LT1507 500kHz Monolithic Buck Mode Switching Regulator FEATURES ■ ■ ■ ■ ■ ■ ■ ■ with all the necessary oscillator, control and logic circuitry. High switching frequency allows a considerable reduction in the size of external components. The topology is current mode for fast transient response and good loop stability. Both fixed output voltage (3.3V) and adjustable parts are available. Constant 500kHz Switching Frequency Uses All Surface Mount Components Operates with Inputs as Low as 4V Saturated Switch Design (0.3Ω) Cycle-by-Cycle Current Limiting Easily Synchronizable Inductor Size as Low as 2µH Shutdown Current: 20µA A special high speed bipolar process and new design techniques allow this regulator to achieve high efficiency at a high switching frequency. Efficiency is maintained over a wide output current range by keeping quiescent supply current to 4mA and by utilizing a supply boost capacitor to allow the NPN power switch to saturate. A shutdown signal will reduce supply current to 20µA. The LT1507 can be externally synchronized from 570kHz to 1MHz with logic level inputs. U APPLICATIONS ■ ■ ■ ■ Portable Computers Battery-Powered Systems Battery Charger Distributed Power U DESCRIPTION The LT®1507 is a 500kHz monolithic buck mode switching regulator, functionally identical to the LT1375 but optimized for lower input voltage applications. It will operate over a 4V to 15V input range, compared with 5.5V to 25V for the LT1375. A 1.5A switch is included on the die along The LT1507 fits into standard 8-pin SO and PDIP packages. Temperature rise is kept to a minimum by the high efficiency design. Full cycle-by-cycle short-circuit protection and thermal shutdown are provided. Standard surface mount external parts are used including the inductor and capacitors. , LTC and LT are registered trademarks of Linear Technology Corporation. U TYPICAL APPLICATION 5V to 3.3V Volt Down Converter 5V to 3.3V Efficiency 100 D2† 1N914 VIN = 5V VOUT = 3.3V C3* 47µF 16V TANTALUM + VSW LT1507-3.3 DEFAULT (OPEN) = ON SHDN GND OUTPUT 3.3V 1.25A L1*** 5µH SENSE VC CC 3.3nF D1 1N5818 * AVX TPSD477M016R0150 OR SPRAGUE 593D EQUIVALENT. RIPPLE CURRENT RATING ≥ 0.6A ** AVX TPSD108M010R0100 OR SPRAGUE 593D EQUIVALENT *** COILTRONICS CTX5-1. SUBSTITUTION UNITS SHOULD BE RATED AT ≥ 1.25A, USING LOW LOSS CORE MATERIAL † SEE BOOST PIN CONSIDERATIONS IN APPLICATIONS INFORMATION SECTION FOR ALTERNATIVE D2 CONNECTION + C1** 100µF 10V TANTALUM EFFICIENCY (%) VIN 5V 90 C2 0.1µF BOOST 80 70 60 50 0 0.25 0.75 1.00 0.50 LOAD CURRENT (A) 1.25 LT1507 • TA02 1 LT1507 U W U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION Input Voltage ........................................................... 16V Boost Pin Voltage .................................................... 25V Shutdown Pin Voltage ............................................... 7V FB Pin Voltage (Adjustable Part) ............................. 3.5V FB Pin Current (Adjustable Part) ............................. 1mA Sense Voltage (Fixed 3.3V Part) ................................ 5V Sync Pin Voltage ....................................................... 7V Operating Ambient Temperature Range LT1507C .................................................. 0°C to 70°C LT1507I .............................................. – 40°C to 85°C Max Operating Junction Temperature................... 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW BOOST 1 8 VC VIN 2 7 FB/SENSE VSW 3 6 GND SHDN 4 5 SYNC N8 PACKAGE 8-LEAD PDIP LT1507CN8 LT1507CN8-3.3 LT1507CS8 LT1507CS8-3.3 LT1507IN8 LT1507IN8-3.3 LT1507IS8 LT1507IS8-3.3 S8 PACKAGE 8-LEAD PLASTIC SO TJMAX = 125°C, θJA = 80°C/W TO 120°C/ W (N) TJMAX = 125°C, θJA = 120°C/W TO 170°C/ W (S) DEPENDING ON PC BOARD LAYOUT S8 PART MARKING 1507 15073 1507I 1507I3 Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS TJ = 25°C, VIN = 5V, VC = 1.5V, boost open, switch open unless otherwise specified. PARAMETER Reference Voltage (Adjustable) CONDITIONS All Conditions ● All Conditions ● Sense Voltage (3.3V) Sense Pin Resistance Reference Voltage Line Regulation FB Input Bias Current Error Amplifier Voltage Gain (Note 8) Error Amplifier Transconductance (Note 8) ● 4.3V ≤ VIN ≤ 15V ● ● (Note 1) ∆I(VC) = ±10µA ● VC Pin to Switch Current Transconductance Error Amplifier Source Current Error Amplifier Sink Current VC Pin Switching Threshold VC Pin High Clamp Switch Current Limit Switch On Resistance (Note 6) MIN 2.39 2.36 3.25 3.23 4.0 VFB = 2.1V or VSENSE = 2.9V VFB = 2.7V or VSENSE = 3.7V Duty Cycle = 0 VFB = 2.1V or VSENSE = 2.9V VC Open, VFB = 2.1V or VSENSE = 2.9V VIN ≥ 5V, VBOOST = VIN + 5V ISW = 1.5A, VBOOST = VIN + 5V DC ≤ 50% DC = 80% 150 1500 1100 ● 150 ● ● 1.50 1.35 TYP 2.42 3.3 6.6 0.01 0.5 400 2000 2 225 2 0.9 2.1 2 0.3 ● Maximum Switch Duty Cycle VFB = 2.1V or VSENSE = 2.9V ● 2 90 86 93 93 MAX 2.45 2.48 3.35 3.37 9.5 0.03 2 UNITS V V V V kΩ %/V µA 2700 3000 µmho µmho 320 3 3 0.4 0.5 A/V µA mA V V A A Ω Ω % % LT1507 ELECTRICAL CHARACTERISTICS TJ = 25°C, VIN = 5V, VC = 1.5V, boost open, switch open unless otherwise specified. PARAMETER Switch Frequency Switch Frequency Line Regulation Frequency Shifting Threshold on FB Pin Minimum Input Voltage (Note 2) Minimum Boost Voltage (Note 3) Boost Current (Note 4) Input Supply Current (Note 5) Shutdown Supply Current CONDITIONS VC Set to Give 50% Duty Cycle – 25°C ≤ TJ ≤ 125°C TJ ≤ –25°C 4.3V ≤ VIN ≤ 15V ∆f = 10kHz MIN 460 440 440 TYP 500 0.05 1.0 4 3 12 ● 0.8 ● ● ISW ≤ 1.5A VBOOST = VIN + 5V ● ISW = 500mA, –25°C ≤ TJ ≤ 125°C TJ ≤ –25°C ISW = 1.5A, –25°C ≤ TJ ≤ 125°C TJ ≤ –25°C 25 3.8 15 ● VSHDN = 0V, VIN ≤ 12V VSW = 0V, VC Open VC Open VC Open Device Shutting Down Device Starting Up Lockout Threshold Shutdown Threshold MAX 540 560 570 0.15 1.3 4.3 3.5 22 25 35 40 5.4 50 75 2.46 0.70 0.70 2.2 1000 ● ● ● ● Minimum Synchronizing Amplitude Synchronizing Frequency Range (Note 7) 2.3 0.15 0.25 2.38 0.37 0.45 1.5 ● 580 The ● denotes specifications which apply over the operating temperature range. Note 1: Gain is measured with a VC swing equal to 200mV above the low clamp level to 200mV below the upper clamp level. Note 2: Minimum input voltage is not measured directly, but is guaranteed by other tests. It is defined as the voltage where internal bias lines are still regulated, so that the reference voltage and oscillator frequency remain constant. Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. See Applications Information. Note 3: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. Note 4: Boost current is the current flowing into the BOOST pin with the pin held 5V above input voltage. It flows only during switch ON time. UNITS kHz kHz kHz %/V V V V mA mA mA mA mA µA µA V V V V kHz Note 5: Input supply current is the bias current drawn by the VIN pin when the SHDN pin is held at 1V (switching disabled). Note 6: Switch ON resistance is calculated by dividing VIN to VSW voltage by the forced current (1.5A). See Typical Performance Characteristics for the graph of switch voltage at other currents. Note 7: For synchronizing frequency above 700kHz, with duty cycles above 50%, external slope compensation may be needed. See Applications Information. Note 8: Transconductance and voltage gain refer to the internal amplifier exclusive of the voltage divider. To calculate gain and transconductance refer to SENSE pin on fixed voltage parts. Divide values shown by the ratio VOUT/2.42. U W TYPICAL PERFORMANCE CHARACTERISTICS VC Pin Shutdown Threshold Switch Peak Current Limit Feedback Pin Voltage and Current 100 1.4 2.44 2.0 2.43 1.5 EFFICIENCY (%) 1.0 0.8 80 70 60 0.6 0.4 –50 FEEDBACK VOLTAGE (V) 90 1.2 50 –25 0 25 50 75 100 JUNCTION TEMPERATURE (°C) 125 LT1507 • TPC01 0 0.25 0.75 1.00 0.50 LOAD CURRENT (A) 1.25 LT1507 • TA02 VOLTAGE 2.42 1.0 CURRENT (µA) THRESHOLD VOLTAGE (V) VIN = 5V VOUT = 3.3V CURRENT 0.5 2.41 2.40 –50 –25 0 25 50 75 100 JUNCTION TEMPERATURE (°C) 0 125 LT1507 • TPC03 3 LT1507 U W TYPICAL PERFORMANCE CHARACTERISTICS Shutdown Pin Bias Current Standby and Shutdown Thresholds CURRENT REQUIRED TO FORCE SHUTDOWN (FLOWS OUT OF PIN). AFTER SHUTDOWN, CURRENT DROPS TO A FEW µA 200 AT 2.38V STANDBY THRESHOLD (CURRENT FLOWS OUT OF PIN) 4 25 INPUT SUPPLY CURRENT (µA) 300 8 VSHDN = 0V STANDBY SHUTDOWN PIN VOLTAGE (V) CURRENT (µA) 400 2.36 2.32 0.8 STARTUP 0.4 20 15 10 5 SHUTDOWN 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 0 125 Shutdown Supply Current VIN = 10V 25 Error Amplifier Transconductance 2500 3000 2000 2500 200 PHASE GAIN (µmho) TRANSCONDUCTANCE (µmho) 75 15 12 1500 1000 GAIN 2000 100 VC 1500 1000 500 150 COUT 12pF ROUT 200k VFB × 2e–3 50 ERROR AMPLIFIER EQUIVALENT CIRCUIT PHASE (DEG) 100 6 9 INPUT VOLTAGE (V) LT1507 • TPC06 Error Amplifier Transconductance 125 3 LT1507 • TPC05 150 50 0 0 50 100 25 75 –50 –25 0 JUNCTION TEMPERATURE (°C) 125 L11507 • TPC04 INPUT SUPPLY CURRENT (µA) Shutdown Supply Current 30 2.40 500 0 RLOAD = 50Ω 0 0 0.1 0.2 0.3 0.4 SHUTDOWN VOLTAGE (V) 0 50 0 75 100 25 –50 –25 JUNCTION TEMPERATURE (°C) 0.5 1k 10k 100k FREQUENCY (Hz) Frequency Foldback –50 10M Minimum Input Voltage with 3.3V Output Switching Frequency 600 500 1M LT1507 • TPC09 LT1507 • TPC08 LT1507 • TPC07 6.5 6.0 400 SWITCHING FREQUENCY 300 200 INPUT VOLTAGE (V) 550 FREQUENCY (kHz) SWITCHING FREQUENCY (kHz) OR CURRENT (µA) 125 500 100 500 450 MINIMUM VOLTAGE TO START WITH STANDARD CIRCUIT 5.5 5.0 4.5 4.0 MINIMUM VOLTAGE TO RUN WITH STANDARD CIRCUIT 100 FEEDBACK PIN CURRENT 3.5 0 0 0.5 1.5 2.0 1.0 FEEDBACK PIN VOLTAGE (V) 2.5 LT1507 • TPC10 400 –50 3.0 –25 0 25 50 75 100 JUNCTION TEMPERATURE (°C) 125 LT1507 • TPC11 1 10 100 LOAD CURRENT (mA) 1000 LT1507 • TPC12 MINIMUM INPUT VOLTAGE CAN BE REDUCED BY ADDING A SMALL EXTERNAL PNP. SEE APPLICATIONS INFORMATION 4 LT1507 U W TYPICAL PERFORMANCE CHARACTERISTICS Maximum Load Current at VOUT = 3.3V Current Limit Foldback 1.50 2.5 L = 20µH 1.25 1.25 L = 10µH CURRENT SOURCE LOAD 1.0 1.00 L = 10µH CURRENT (A) 1.5 1.50 VOUT = 3.3V FOLDBACK CHARACTERISTICS CURRENT (A) *POSSIBLE UNDESIRED STABLE POINT FOR CURRENT SOURCE LOAD 2.0 OUTPUT CURRENT (A) Maximum Load Current at VOUT = 5V L = 5µH 0.75 L = 3µH 0.50 L = 5µH 0.75 0.50 L = 2µH MOS LOAD 0.5 1.00 0.25 0.25 RESISTOR LOAD 0 0 0 20 60 80 40 OUTPUT VOLTAGE (%) 0 4 100 6 8 10 INPUT VOLTAGE (V) 0 14 12 3 6 9 INPUT VOLTAGE (V) LT1507 • TPC14 LT1507 • TPC13 12 15 LT1507 • TPC15 *SEE "MORE THAN JUST VOLTAGE FEEDBACK" IN APPLICATIONS INFORMATION SECTION Inductor Core Loss for 3.3V Output Boost Pin Current VOUT = 3.3V VIN = 5V IOUT = 1A 8 CORE LOSS (W) BOOST PIN CURRENT (mA) 10 6 4 TJ = 25°C 0.1 TYPE 52 POWDERED IRON Kool Mµ® 0.01 PERMALLOY µ = 125 2 0 0.25 0.50 0.75 1.00 SWITCH CURRENT (A) 1.25 SWITCH VOLTAGE (V) TJ = 25°C 0 Switch Voltage Drop 0.8 1.0 12 0.6 0.4 0.2 Metglas® 0 0.001 1 2 4 INDUCTANCE (µH) 6 8 10 LT1507 • TPC17 LT1507 • TPC16 0 0.25 0.50 0.75 1.00 1.25 SWITCH CURRENT (A) 1.50 LT1507 • TPC18 CORE LOSS IS INDEPENDENT OF LOAD CURRENT UNTIL LOAD CURRENT FALLS LOW ENOUGH FOR CIRCUIT TO GO INTO DISCONTINUOUS MODE Kool Mµ is a registered trademark of Magnetics, Incorporated. Metglas is a registered trademark of AlliedSignal Incorporated. U U U PIN FUNCTIONS BOOST (Pin 1): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. Without this added voltage the typical switch voltage loss would be about 1.5V. The additional boost voltage allows the switch to saturate and voltage loss approximates that of a 0.3Ω FET structure, but with a much smaller die area. Efficiency improves from 70% for conventional bipolar designs to greater than 85% for these new parts. VIN (Pin 2): Input Pin. The LT1507 is designed to operate with an input voltage between 4.5V and 15V. Under certain conditions, input voltage may be reduced down to 4V. Actual minimum operating voltage will always be higher than the output voltage. It may be limited by switch 5 LT1507 U U U PIN FUNCTIONS saturation voltage and maximum duty cycle. A typical value for minimum input voltage is 1V above output voltage. Start-up conditions may require more voltage at light loads. See Minimum Input Voltage for details. VSW (Pin 3): The switch pin is driven up to the input voltage in the ON state and is an open circuit in the OFF state. At higher load currents, pin voltage during the off condition will be one diode drop below ground as set by the external catch diode. At lighter loads the pin will assume an intermediate state equal to output voltage during part of the switch OFF time. Maximum negative voltage on the switch pin is 1V with respect to the GND pin, so it must always be clamped with a catch diode to the GND pin. SHDN (Pin 4): The shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. Actually this pin has two separate thresholds, one at 2.38V to disable switching and a second at 0.4V to force complete micropower shutdown. The 2.38V threshold functions as an accurate undervoltage lockout (UVLO). This is sometimes used to prevent the regulator from delivering power until the input voltage has reached a predetermined level. SYNC (Pin 5): The SYNC pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 10% and 90% duty cycle. The synchronizing range is equal to initial operating frequency up to 1MHz. See Sychronizing section for details. FB/SENSE (Pin 7): The feedback pin is used to set output voltage using an external voltage divider that generates 2.42V at the pin with the desired output voltage. The fixed voltage (– 3 .3V) parts have the divider included on the chip and the feedback pin is used as a sense pin connected directly to the 5V output. Two additional functions are performed by the feedback pin. When the pin voltage drops below 1.7V, switch current limit is reduced. Below 1V, switching frequency is also reduced. See More Than Just Voltage Feedback. VC (Pin 8): The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation but can do double duty as a current clamp or control loop override. This pin sets at about 1V for very light loads and 2V at maximum load. It can be driven to ground to shut off the regulator, but if driven high, current must be limited to 4mA. W BLOCK DIAGRAM The LT1507 is a constant frequency, current mode buck converter. This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. In addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS flip-flop to turn the switch on. When switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. This technique means that the error amplifier commands current to be delivered to the output rather than voltage. A voltage fed system will have low phase shift up to the resonant frequency of the inductor 6 and output capacitor, then an abrupt 180° shift will occur. The current fed system will have 90° phase shift at a much lower frequency, but will not have the additional 90° shift until well beyond the LC resonant frequency. This makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. High switch efficiency is attained by using the BOOST pin to provide a voltage to the switch driver which is higher than the input voltage, allowing the switch to be saturated. This boosted voltage is generated with an external capacitor and diode. Two comparators are connected to the shutdown pin. One has a 2.38V threshold for undervoltage lockout and the second has a 0.4V threshold for complete shutdown. LT1507 W BLOCK DIAGRAM 0.1Ω VIN 2 + 2.9V BIAS REGULATOR BIAS – CURRENT SENSE AMPLIFIER VOLTAGE GAIN = 5 INTERNAL VCC BOOST SLOPE COMP Σ 1 0.9V SYNC 500kHz OSCILLATOR 5 S + Q1 POWER SWITCH DRIVER CIRCUITRY RS FLIP-FLOP R SHUTDOWN COMPARATOR – + – CURRENT COMPARATOR 3 VSW 7 FB/SENSE 6 GND 0.4V SHDN FREQUENCY SHIFT CIRCUIT 4 3.5µA FOLDBACK CURRENT LIMIT CLAMP + Q2 – – 2.38V 8 VC ERROR AMPLIFIER gm = 2000µmho + LOCKOUT COMPARATOR 2.42V LT1507 • BD Figure 1. Block Diagram U W U U APPLICATIONS INFORMATION Note: This application section is adapted from the more complete version found in the LT1375/LT1376 data sheet. If more details are desired consult the LT1375/LT1376 Applications Information section, but please acquaint yourself thoroughly with this LT1507 information first so that differences between the LT1375 and the LT1507 do not cause confusion. FEEDBACK PIN FUNCTIONS The feedback pin (FB or SENSE) on the LT1507 is used to set output voltage and also to provide several overload protection features. The first part of this section deals with selecting resistors to set output voltage and the remaining part talks about foldback frequency and current limiting created by the FB pin. Please read both parts before 7 LT1507 U U W U APPLICATIONS INFORMATION committing to a final design. The fixed 3.3V LT1507-3.3 has internal divider resistors and the FB pin is renamed SENSE, connected directly to the output. The suggested value for the output divider resistor from FB to ground (R2) is 5k or less and the formula for R1 is shown below. The output voltage error caused by ignoring the input bias current on the FB pin is less than 0.25% with R2 = 5k. Please read below if R2 is increased above the suggested value. R1 = R2(VOUT – 2.42) 2.42 More Than Just Voltage Feedback The feedback pin is used for more than just output voltage sensing. It also reduces switching frequency and current limit when output voltage is very low (see graph in Typical Performance Characteristics). This is done to control power dissipation in both the IC and in the external diode and inductor during short-circuit conditions. A shorted output requires the switching regulator to operate at very low duty cycles and the average current through the diode and inductor is equal to the short-circuit current limit of the switch (typically 2A of the LT1507, folding back to less than 1A). Minimum switch ON time limitations would prevent the switcher from attaining a sufficiently low duty cycle if switching frequency were maintained at 500kHz, so frequency is reduced by about 5:1 when the feedback pin voltage drops below 1V (see Frequency Foldback graph). This does not affect operation with normal load conditions; one simply sees a gear shift in switching frequency during start-up as the output voltage rises. In addition to lower switching frequency, the LT1507 also operates at lower switch current limit when the feedback pin voltage drops below 1.5V. This foldback current limit greatly reduces power dissipation in the IC, diode and inductor during short-circuit conditions. Again, it is nearly transparent to the user under normal load conditions. The only loads which may be affected are current source loads which maintain full-load current with output voltage less than 50% of final value. In these rare situations, the feedback pin can be clamped above 1.5V with an external diode to defeat foldback current limit. Caution: clamping the feedback pin means that frequency shifting will also be defeated, so a combination of high input voltage and dead shorted output may cause the LT1507 to lose control of current limit. The internal circuitry which forces reduced switching frequency also causes current to flow out of the feedback pin when output voltage is low. If the FB pin falls below 1V, current begins to flow out of the pin and reduces frequency at the rate of approximately 5kHz/µA. To ensure adequate frequency foldback (under worst-case short-circuit conditions) the external divider Thevinin resistance must be low enough to pull 150µA out of the FB pin with 0.6V on the D2 1N914 C2 0.1µF BOOST VIN VIN C3* 33µF 20V TANTALUM + VSW LT1507 DEFAULT (OPEN) = ON SHDN GND L1*** 5µH R1 5.36k D1 1N5818 R2 4.99k OUTPUT 5V FB VC CC 3.3nF + C1** 100µF 10V TANTALUM * AVX TPSD337M020R0200 OR SPRAGUE 593 EQUIVALENT. RIPPLE CURRENT RATING ≥ 0.6A ** AVX TPSD108M010R0100 OR SPRAGUE 593 EQUIVALENT *** COILTRONICS CTX5-1. SUBSTITUTION UNITS SHOULD BE RATED AT ≥ 1.25A, USING LOW LOSS CORE MATERIAL. LOAD CURRENTS ABOVE 0.85A MAY NEED A 10µH OR 20µH INDUCTOR Figure 2. Typical Schematic for LT1507 Adjustable Application 8 LT1507 • F01 LT1507 U W U U APPLICATIONS INFORMATION pin (RDIV = R1/R2 ≤ 4k). The net result is that reductions in frequency and current limit are affected by output voltage divider impedance. Although divider impedance is not critical, caution should be used if resistors are increased beyond the suggested values and short-circuit conditions will occur with high input voltage. High frequency pickup will also increase and the protection accorded by frequency and current foldback will decrease. CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR For most applications the value of the inductor will fall in the range of 2µH to 10µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the LT1507 switch, which has a 1.5A limit. Higher values also reduce output ripple voltage and reduce core loss. Graphs in the Typical Performance Characteristics section show maximum output load current versus inductor size and input voltage. A second graph shows core loss versus inductor size for various core materials. When choosing an inductor you might have to consider maximum load current, core and copper losses, allowable component height, output voltage ripple, EMI, fault current in the inductor, saturation and, of course, cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. Choose a value in microhenries from the graphs of Maximum Load Current and Inductor Core Loss for 3.3V Output. If you want to double check that the chosen inductor value will allow sufficient load current, go to the next section, Maximum Output Load Current. Choosing a small inductor with lighter loads may result in discontinuous mode of operation, but the LT1507 is designed to work well in either mode. Keep in mind that lower core loss means higher cost, at least for closedcore geometries like toroids. Type 52 powdered iron, Kool Mµ and Molypermalloy are old standbys for toroids in ascending order of price. A newcomer, Metglas, gives very low core loss with high saturation current. Assume that the average inductor current is equal to load current and decide whether or not the inductor must withstand continuous fault conditions. If maximum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 1.5A overload condition. Dead shorts (VOUT ≤ 1V) will actually be more gentle on the inductor because the LT1507 has foldback current limiting (see graph in Typical Performance Characteristics). 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall in between somewhere. The following formula assumes a continuous mode of operation, but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions. V (V – V ) IPEAK = IOUT + OUT IN OUT 2(f)(L)(VIN ) VIN = Maximum input voltage f = Switching frequency = 500kHz 3. Decide if the design can tolerate an “open” core geometry like ferrite rods or barrels, which have high magnetic field radiation or whether it needs a closed core like a toroid to prevent EMI problems. One would not want an open core next to a magnetic storage media for instance! This is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. The following is an example of just how subtle the “B” field problems can be with open geometry cores. We had selected an open drum shaped ferrite core for the LTC1376 demonstration board because the inductor was extremely small and inexpensive. It met all the requirements for current and the ferrite core gave low core loss. When the boards came back from assembly, many of them had somewhat higher than expected output ripple voltage. We removed the inductors and output capacitors and found them to be no different than the good boards. After much head scratching and hours of delicate low level ripple measurements on the good and bad boards, I realized that the problem must 9 LT1507 U W U U APPLICATIONS INFORMATION be due to a radiated magnetic field coupling into PC board traces. But why were some boards bad and others good? In a moment of desperation (or divine inspiration) I unsoldered a “bad” inductor, rotated it 180° and resoldered it. Problem fixed!! It turns out that the inductor was symmetrical in all regards except that the polarity of the magnetic field reversed when the unit was rotated 180° because current flowed in the opposite direction in the coil. In one direction, the magnetically induced ripple in the board traces added to output ripple. Rotating the inductor caused the induced field to reduce output ripple. Unfortunately the inductor had no physical package assymmetry to indicate rotation, including part marking, so we had to visually examine the winding in each unit before soldering it to the boards. This little horror story should not preclude the use of open core inductors, but it emphasizes the need to carefully check the effect these seductively small, low cost inductors may have on regulator or system performances. 4. Look for an inductor (see Table 1) which meets the requirements of core shape, peak current (to avoid saturation), average current (to limit heat) and fault current (if the inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts). Keep in mind that all good things like high efficiency, surface mounting, low profile and high temperature operation will increase cost, sometimes dramatically. 5. After making an initial choice, consider secondary things like output voltage ripple, second sourcing, etc. Use the experts in the Linear Technology Applications Department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. 10 Table 1. Representative Surface Mount Units VALUE (µH) DC (A) Coiltronics CTX5-1 CTX10-1 CTX5-1P CTX10-1P 5 10 5 10 2.3 1.9 1.8 1.6 Tor Tor Tor Tor 0.027 0.039 0.021 0.030 KMµ KMµ 52 52 4.2 4.2 4.2 4.2 Sumida CDRH64 CDRH73 CD73 CD104 10 10 10 10 1.7 1.7 1.4 2.4 SC SC Open Open 0.084 0.055 0.062 0.041 Fer Fer Fer Fer 4.5 3.4 3.5 4.0 Gowanda SM20-102K 10 1.3 Open 0.038 Fer 7 Dale IHSM-4825 IHSM-5832 10 10 3.1 4.3 Open Open 0.071 0.053 Fer Fer 5.6 7.1 MANUFACTURER CORE SERIES TYPE (Ω) CORE HEIGHT (mm) SC = Semi-closed geometry Fer = Ferrite core material 52 = Type 52 powdered iron core material KMµ = Kool Mµ OUTPUT CAPACITOR The output capacitor is normally chosen by its effective series resistance (ESR), because that is what determines output ripple voltage. At 500kHz any polarized capacitor is essentially resistive. To get low ESR takes volume ; physically larger capacitors have lower ESR. The ESR range needed for typical LT1507 applications is 0.05Ω to 0.5Ω. A typical output capacitor is an AVX type TPS, 100µF at 10V, with a guaranteed ESR less than 0.1Ω. This is a “D” size surface mount solid tantalum capacitor. TPS capacitors are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. The value in microfarads is not particularly critical and values from 22µF to greater than 500µF work well, but you cannot cheat mother nature on ESR. If you find a tiny 22µF solid tantalum capacitor, it will have high ESR and output ripple voltage will be terrible. The chart in Table 2 shows some typical solid tantalum surface mount capacitors. LT1507 U W U U APPLICATIONS INFORMATION Table 2. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current ESR (MAX Ω) RIPPLE CURRENT (A) AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1 AVX TAJ 0.7 to 0.9 0.4 AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1 AVX TAJ 0.9 to 2.0 0.36 to 0.24 AVX TPS 0.2 (Typ) 0.5 (Typ) AVX TAJ 1.8 to 3.0 0.22 to 0.17 E CASE SIZE D CASE SIZE C CASE SIZE Many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. This is historically true, and type TPS capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. Solid tantalum capacitors fail during very high turn-on surges which do not occur at the output of regulators. High discharge surges, such as when the regulator output is dead shorted, do not harm the capacitors. Unlike the input capacitor, RMS ripple current in the output capacitor is normally low enough that ripple current rating is not an issue. The current waveform is triangular with a typical value of 200mA RMS. The formula to calculate this is: Output Capacitor Ripple Current (RMS) IRIPPLE (RMS ) = 0.29 (VOUT )(VIN – VOUT ) (L)(f)(VIN ) Ceramic Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low ESR. Unfortunately, the ESR is so low that it can cause loop stability problems when ceramic is used for the output capacitor. Solid tantalum capacitor ESR generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective. They are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges. OUTPUT RIPPLE VOLTAGE Ripple voltage is determined by the high frequency impedance of the output capacitor and ripple current through the inductor. Ripple current is triangular (continuous mode) with a peak-to-peak value of: IP-P = ( VOUT )(VIN − VOUT ) ( VIN)(L)(f) Output ripple voltage is also triangular with peak-to-peak amplitude of: VRIPPLE = (IP–P)(ESR) (peak-to-peak) Example: with VIN = 5V, VOUT = 3.3V, L = 5µH, ESR = 0.1Ω; (3.3)(5 − 3.3) = 0.45P-P −6 3 5 5 10 500 10 VRIPPLE = (0.45A)(0.1Ω) = 45mVP-P IP-P = MAXIMUM OUTPUT LOAD CURRENT Maximum load current will be less than the 1.5A rating of the LT1507, especially with lower inductor values. Inductor ripple current must be taken into account as well as reduced switch current at high duty cycles. Maximum switch current rating (IP) of the LT1507 is 1.5A up to 50% duty cycle (DC), decreasing to 1.35A at 80% duty cycle, shown graphically in Typical Performance Characteristics and as a formula below. Current rating decreases with duty cycle because the LT1507 has internal slope compensation to prevent current mode subharmonic switching. For more details on subharmonic oscillation read Application Note 19. Peak guaranteed switch current (IP) is found from: V IP = 1.5A for OUT ≤ 0.5 VIN IP = 1.75A − V 0.5 ( VOUT ) for OUT ≥ 0.5 VIN VIN 11 LT1507 U W U U APPLICATIONS INFORMATION Example: with VOUT = 3.3V, VIN = 5V; Discontinuous mode: (IP )2 (f)(L)(VIN ) IOUT (MAX) = 2(VOUT )(VIN – VOUT ) VOUT/VIN = 3.3/5 = 0.67 IP = 1.75 – (0.5)(0.66) = 1.42A Maximum load current would be equal to maximum switch current for an infinitely large inductor, but with finite inductor size, maximum load current is reduced by one half peak-to-peak inductor current. The following formula assumes continuous mode operation; the term on the right must be less than one half of IP. Example: with L = 2µH, VOUT = 5V and VIN(MAX) = 15V; Continuous mode: The main reason for using such a tiny inductor is that it is physically very small, but keep in mind that peak-to-peak inductor current will be very high. This will increase output ripple voltage. If the output capacitor has to be made larger to reduce ripple voltage, the overall circuit could actually be larger. (VOUT )(VIN – VOUT ) 2(L)( f)(VIN ) For the conditions above, with L = 5µH and f = 500kHz; IOUT (MAX) = IP – (3.3)(5 – 3.3) 2 5 10−6 500 103 = 1.42 – 0.22 = 1.2A IOUT (MAX) = 1.42 – ( ) ( )5 At VIN = 8V, VOUT/ VIN = 0.41, so IP is equal to 1.5A and IOUT(MAX) is equal to; (3.3)(8 – 3.3) 2 5 10−6 500 103 8 = 1.5 – 0.39 = 1.11A 1.5 – ( ) ( ) Note that there is less load current available at the higher input voltage because inductor ripple current increases. This is not always the case. Certain combinations of inductor value and input voltage range may yield lower available load current at the lowest input voltage due to reduced peak switch current at high duty cycles. If load current is close to the maximum available, please check maximum available current at both input voltage extremes. To calculate actual peak switch current with a given set of conditions, use: V (V – V ) ISWITCH(PEAK) = IOUT + OUT IN OUT 2(L)(f)(VIN ) For lighter loads where discontinuous mode operation can be used, maximum load current is equal to: 12 ( ) ( )15 (1.5)2 500 103 2 10−6 IOUT (MAX) = 2(5)(15 – 5) = 338 mA CATCH DIODE The suggested catch diode (D1) is a 1N5818 Schottky or its Motorola equivalent, MBR130. It is rated at 1A average forward current and 30V reverse voltage. Typical forward voltage is 0.42V at 1A. The diode conducts current only during switch OFF time. Peak reverse voltage is equal to regulator input voltage. Average forward current in normal operation can be calculated from: I (V – V ) ID(AVG) = OUT IN OUT VIN This formula will not yield values higher than 1A with maximum load current of 1.25A unless the ratio of input to output voltage exceeds 5:1. The only reason to consider a larger diode is the worst-case condition of a high input voltage and overloaded (not shorted) output. Under shortcircuit conditions, foldback current limit will reduce diode current to less than 1A, but if the output is overloaded and does not fall to less than 1/3 of nominal output voltage, foldback will not take effect. With the overloaded condition, output current will increase to a typical value of 1.8A, determined by peak switch current limit of 2A. With VIN = 10V, VOUT = 2V (3.3V overloaded) and IOUT = 1.8A: ID(AVG) = 1.8 (10 – 2) = 1.44A 10 LT1507 U W U U APPLICATIONS INFORMATION The anode of the diode can be connected to the regulated output voltage or the unregulated input voltage. The “boost voltage” generated across the boost capacitor is then nearly identical to the anode voltage. The input connection minimizes start-up problems and gives plenty of boost voltage, but efficiency is slightly lower, especially with input voltages above 10V. For 5V to 3.3V operation, or any output voltage less than 3.3V, the diode should be connected to the input. With input voltage more than 3V above the output and an output voltage of at least 3.3V the output connection will give better efficiency. Use the BAT85 Schottky diode for 3.3V applications where the anode is connected to the output. This is safe for short periods of time, but it would be prudent to check with the diode manufacturer if continuous operation under these conditions must be tolerated. BOOST PIN CONSIDERATIONS For most applications, the boost components are a 0.22µF capacitor and an MBR0520 or BAT85 Schottky diode. This capacitor value is twice that suggested for the LT1376 because the lower voltages commonly found in LT1507 applications may require lower ripple voltage across the capacitor to ensure adequate boost voltage under worstcase conditions. Efficiency is not affected by the capacitor value, but the capacitor should have an ESR of less than 2Ω to ensure that it can be recharged fully under the worstcase condition of minimum input voltage. Almost any type of film or ceramic capacitor will work fine. LAYOUT CONSIDERATIONS Suggested layout for the LT1507 is shown in Figure 3. The main concern for layout is to minimize the length of the INPUT MINIMIZE AREA OF CONNECTIONS TO THE SWITCH NODE AND BOOST NODE, BUT OBSERVE CURRENT DENSITY LIMITATIONS IN PATH TO L1 CF D2 C2 BOOST VC IN FB CC C3 KEEP INPUT CAPACITOR AND CATCH DIODE CLOSE TO REGULATOR AND TERMINATE THEM TO SAME POINT SW GND RC CF AND RC ARE OPTIONAL. SEE FREQUENCY COMPENSATION R2 D1 SHDN SYNC R1 L1 SYNC C1 OUTPUT GROUND RING NEED NOT BE AS SHOWN. (NORMALLY EXISTS AS INTERNAL PLANE) CONNECT OUTPUT CAPACITOR DIRECTLY TO HEAVY GROUND TAKE OUTPUT DIRECTLY FROM END OF OUTPUT CAPACITOR TO AVOID PARASITIC RESISTANCE AND INDUCTANCE (KELVIN CONNECTION) TERMINATE GND PIN DIRECTLY TO GROUND PLANE WITH VIA TO MINIMIZE EMI. (MINIMIZE DISTANCE TO INPUT CAPACITOR C3). CONNECT FEEDBACK RESISTORS AND COMPENSATION COMPONENTS DIRECTLY TO GROUND PLANE OR TO SWITCHER GND PIN. LT1507 • F03 Figure 3. Suggested Layout 13 LT1507 U W U U APPLICATIONS INFORMATION high speed circulating current path shown in Figure 4 and to make connections to the output capacitor in a manner that minimizes output ripple and noise. For more details, see Applications Information section in the LT1376 data sheet. SWITCH NODE L1 5V VIN C3 HIGH FREQUENCY CIRCULATING PATH C1 LOAD LT1507 • F04 Figure 4. High Speed Switching Path INPUT BYPASSING AND VOLTAGE RANGE Input Bypass Capacitor Stepdown converters draw current from the input supply in pulses. The average height of these pulses is equal to load current and the duty cycle is equal to VOUT/VIN. Rise and fall time of the current is very fast. A local bypass capacitor across the input supply is necessary to ensure proper operation of the regulator and minimize the ripple current fed back into the input supply. The capacitor also forces switching current to flow in a tight local loop, minimizing EMI. Do not cheat on the ripple current rating of the input bypass capacitor, but also don’t get hung up on the value in microfarads. The input capacitor is intended to absorb all the switching current ripple, which can have an RMS value as high as one half of load current. Ripple current ratings on the capacitor must be observed to ensure reliable operation. The actual value of the capacitor in microfarads is not particularly important because at 500kHz, any value above 5µF is essential resistive. Ripple current rating is the critical parameter. RMS ripple current can be calculated from: IRIPPLE (RMS ) = IOUT 14 VOUT (VIN – VOUT ) VIN2 The term inside the radical has a maximum value of 0.5 when input voltage is twice output and stays near 0.5 for a relatively wide range of input voltages. It is common practice, therefore, to simply use the worst-case value and assume that RMS ripple current is one half of load current. At maximum output current of 1.5A for the LT1507, the input bypass capacitor should be rated at 0.75A ripple current. Note however, that there are many secondary considerations in choosing the final ripple current rating. These include ambient temperature, average versus peak load current, equipment operating schedule and required product lifetime. For more details see Application Notes 19 and 46. Input Capacitor Type Some caution must be used when selecting the type of capacitor used at the input of regulators. Aluminum electrolytics are lowest cost, but are physically large to achieve adequate ripple current rating, and size constraints (especially height) may preclude their use. Ceramic capacitors are now available in larger values and their high ripple current and voltage rating make them ideal for input bypassing. Cost is slightly higher and footprint may also be somewhat larger. Solid tantalum capacitors are a good choice except that they have a history of occasional spectacular failures when they are subjected to very large current surges during power-up. The capacitors can short and then burn with a brilliant white light and lots of nasty smoke. This phenomenon occurs in only a small percentage of units, but it has led some OEM companies to forbid their use in high surge applications. The input bypass capacitor of regulators can see such high surges when a battery or high capacitance source is connected. Several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (AVX TPS series for instance, see Table 2). Even these units may fail if the input current surge exceeds a value equal to the voltage rating of the capacitor divided by 1Ω (10A for a 10V capacitor). For this reason, AVX recommends using the highest voltage rating possible for the input capacitor. For equal case size, this means that lower values of capacitance must be used. As stated above, this LT1507 U W U U APPLICATIONS INFORMATION is not a problem, but it should be noted that for equal case size, the ripple current rating and ESR of higher voltage capacitors will be somewhat worse. The lower input operating voltages of the LT1507 allow considerable derating of capacitor voltage. If solid tantalum units are used, it would be wise to use units rated at 25V or more, as long as ripple current requirements are met. Design Note 122 discusses the problem of showing typical input capacitor surges that occur when batteries or adapters are hot plugged to typical regulator systems. A new capacitor type known as OS-CON uses a “semiconductor” dielectric to achieve extremely low ESR and high ripple current rating. These are ideal for input bypassing because they are not surge sensitive. They are not suggested for output capacitors because the very low ESR may present loop stability problems. Price and size (height) are issues to be considered. The original manufacturer is Sanyo but there are now additional sources. Larger capacitors may be necessary when the input voltage is very close to the minimum specified on the data sheet. A 5µF ceramic input capacitor for instance, moves at about 0.1V/µs during switch ON time when load current is 1A, creating a ripple voltage due to reactance. This is in addition to the ripple caused by capacitor ESR. Physically larger input capacitors will have more capacitance (less reactance) and lower ESR. Small voltage dips during switch ON time are not normally a problem, but at very low input voltage they may cause erratic operation because the input voltage drops below the minimum specification. Problems can also occur if the input to output voltage differential is near minimum. to 1.5V higher than the standard running voltage, especially at light loads. An approximate formula to calculate minimum running voltage at load currents above 100mA is: V + (IOUT )(0.3Ω) VIN(MIN) = OUT (IOUT ≥ 100mA) 0.85 With VOUT = 3.3V and IOUT = 0.1A, this formula yields VIN(MIN) = 3.9V. Increasing load current to 1A raises minimum input to 4.2V. For start-up and operation at light loads, see the next section. Minimum Start-Up Voltage and Operation at Light Loads The boost capacitor supplies current to the BOOST pin during switch ON time. This capacitor is recharged only during switch OFF time. Under certain conditions of light load and low input voltage, the capacitor may not be fully recharged during the relatively short OFF time. This causes the boost voltage to collapse and minimum input voltage is increased. Start-up voltage at light loads is higher than normal running voltage for the same reasons. Figure 5 shows minimum input voltage for a 3.3V output, both for start-up and for normal operation. This graph indicates that a 5V to 3.3V converter with 4.7V minimum input voltage, will not start correctly below a 40mA load current and will not run correctly below a 4mA load current. If minimum load current is less than 50mA, a preload should be added or the circuit in Figure 6 can be used. 6.5 VALID ONLY FOR VOUT = 3.3V Minimum Input Voltage (After Start-Up) Minimum input voltage to make the LT1507 “run” correctly is typically 3.6V, but to regulate the output, a buck converter input voltage must always be higher than the output voltage. To calculate minimum operating input voltage, switch voltage loss and maximum duty cycle must be taken into account. With the LT1507 there is the additional consideration of proper operation of the boost circuit. The boost circuit allows the power switch to saturate for high efficiency, but it also sometimes results in a start-up or low current operating voltage that is 0.5V INPUT VOLTAGE (V) 6.0 MINIMUM VOLTAGE TO START WITH PNP ADDED 5.5 MINIMUM VOLTAGE TO START WITH STANDARD CIRCUITS 5.0 4.5 4.0 MINIMUM VOLTAGE TO RUN WITH PNP ADDED 3.5 3.0 1 MINIMUM VOLTAGE TO RUN WITH STANDARD CIRCUIT 10 100 LOAD CURRENT (mA) 1000 LT1400 • GXX MINIMUM VOLTAGE TOFigure RUN WITH 5. PNP ADDED Minimum Input Voltage for VOUT = 3.3V 15 LT1507 U U W U APPLICATIONS INFORMATION The circuit in Figure 6 will allow operation at light loads with low input voltages. It uses a small PNP to charge the boost capacitor (C2) and an extra diode (D3) to complete the power path from VSW to the boost capacitor. Note that the diodes have been changed to Schottky BAT85s to optimize low voltage operation. Figure 5 shows that with the added PNP, minimum load current can be reduced to 6mA and still guarantee proper start-up with 4.7V input. problems. For low input voltage, high sync frequency applications, the circuit shown in Figure 7 can be used to generate an external slope compensation ramp that eliminates subharmonic oscillation. See Frequency Compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. Application Note 19 has more details on the theory of slope compensation. D2 BAT85 VSW C2 0.22µF D3 BAT85 VIN + GND VOUT = 3.3V VSW Q1 2N3906 LT1507-3.3 VC CC D1 1N5818 + VC RC 470Ω CC 2000pF LT1507 • F07 C1 LT1507 • F06 Figure 6. Adding a Small PNP to Reduce Minimum Start-Up Voltage SYNCHRONIZING The LT1507 SYNC pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 10% and 90% duty cycle. The synchronizing range is equal to initial operating frequency up to 1MHz (above 700kHz external slope compensation may be needed). This means that minimum practical sync frequency is equal to the worstcase high self-oscillating frequency (560kHz) not the typical operating frequency of 500kHz. Caution should be used when synchronizing above 700kHz because at higher sync frequencies, the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs at input voltages less than twice the output voltage and shows up as alternating pulse widths at the switch node. It does not cause the regulator to lose regulation, but switch frequency content down to 100kHz may be objectionable. Higher inductor values will tend to eliminate 16 CS 1000pF RS 5.2k SENSE GND VOUT SYNC L1 BOOST INPUT + LT1507 Figure 7. Adding External Slope Compensation for High Sync Frequencies External Slope Compensation Ramp The LT1507 is a current mode switching regulator and therefore, it requires something called “slope compensation” when operated above 50% duty cycle in continuous mode. This condition occurs when input voltage is less than twice output voltage. Slope compensation adds a ramp to the switch current sense signal generated on the chip during switch ON time. Typically the ramp is generated from a portion of the internal oscillator waveform. In the LT1507, the ramp is arranged to be zero until the oscillator waveform reaches about 40% of its final value. This minimizes the total amount of ramp added to switch current. The reason for doing it this way is that the ramp subtracts from switch current limit, so that switch current limit would be considerably lower at high duty cycle compared to low duty cycle if the ramp existed at all duty cycles. By starting the ramp at the 40% point, changes in current limit are minimized. No ramp is needed when operating below 50% duty cycle. Problems can occur with this technique if the regulator is used with a combination of high external sync frequency and more than 50% duty cycle. The basic sync function LT1507 U W U U APPLICATIONS INFORMATION works by prematurely tripping the oscillator before it reaches its normal peak value. For instance, if the oscillator is synchronized at twice its nominal frequency, oscillator amplitude will drop by half. A ramp which previously started at the 40% point now starts at the 80% point! This effectively blocks slope compensation and the regulator may respond with fluctuating pulse widths, a “phase oscillation” if you will. The regulator output stays in regulation but subharmonic frequencies are generated at the switch node. The solution to this problem is to generate an external ramp that replaces the missing internal ramp. As it turns out, this is not difficult if the sync signal can be arranged to have a fairly low duty cycle (< 35%). The ramp is created by AC coupling a resistor from the sync signal to the compensation capacitor as shown in Figure 7. This generates a negative ramp on the VC pin during switch ON time that emulates the missing internally generated ramp. Amplitude of the ramp should be about 100mV to 200mV peak-to-peak. The formulas for calculating the values of RS and CS are shown below. Note that the CS value is unimportant as long as it exceeds the value given. The formula assures that the impedance of CS will be small compared to RS. V ( DCS )(1− DCS ) RS = SYNC VP-P (CC )(f) 20 CS > 2π( f)(RS ) VSYNC = Peak-to-peak value of sync signal DCS = Duty cycle of incoming sync signal VP-P = Desired amplitude of ramp f = Sync frequency Theoretical minimum amplitude for the ramp, assuming no internal ramp, is: (2V − V )(1 − DCS ) VP-P ≥ OUT IN 2(f)(L)(gmP ) gmP = Transconductance from VC pin to switch current (1.8A/V for the LT1507). For VIN = 4.7, VOUT = 3.3V, f = 1MHz, L = 5µH and DCS = 25%: VP-P ≥ (6.6 − 4.7)(1 − 0.25) = 71mV 6 −6 2 1 10 5 10 1.8 To avoid small values of RS, the compensation capacitor (CC) should be made as small as possible. 2000pF will work in most situations. If we increase VPP to 90mV for a little cushion, RS will be: (5)(0.25)(0.75) = 5.2k −9 6 0.09 2 10 1 10 20 C≥ = 612pF 2π 1 106 (5200) RS = ( ) ( ) ( ) THERMAL CALCULATIONS Power dissipation in the LT1507 chip comes from four sources: switch DC loss, switch AC loss, boost circuit current and input quiescent current. The formulas below show how to calculate each of these losses. These formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. Switch loss: RSW (IOUT )2 (VOUT ) PSW = + 16ns(IOUT )(VIN )(f) VIN Boost current loss: V 2 I PBOOST = OUT 0.008 + OUT VIN 75 Quiescent current loss: PQ = VIN (0.003) + VOUT (0.005) RSW = Switch resistance (≈ 0.4Ω) 16ns = Equivalent switch current/voltage overlap time f = Switching frequency 17 LT1507 U U W U APPLICATIONS INFORMATION ( ) GAIN-VC PIN TO INDUCTOR CURRENT (A/V) 2 ( ) (0.4)(1) (3.3) + 16 10−9 (1)(5)500 103 5 = 0.26 + 0.04 = 0.3 W PSW = 2 1 PBOOST = (3.3) 0.008 + = 0.046W 75 5 PQ = 5(0.003) + 3.3(0.005) = 0.032W 2.0 80 GAIN (A/V) 1.5 40 1.0 0 PHASE VOUT = 3.3V IOUT = 250mA VIN = 5V L = 10µH 0.5 –40 0 10 Total power dissipation is 0.3 + 0.046 + 0.032 = 0.38W. 100 –80 100k 1k 10k FREQUENCY (Hz) PHASE-VC PIN TO INDUCTOR CURRENT (C°) Example: with VIN = 5V, VOUT = 3.3V, IOUT = 1A; LT1507 • F08 With the S8 package (θJA = 120°C/W) at an ambient temperature of 70°C; TJ = 70 + 120(0.38) = 116°C FREQUENCY COMPENSATION The LT1507 uses a “current mode” architecture to help alleviate phase shift created by the inductor. The basic connections are shown in Figure 9. Gain of the power stage can be modeled as 1.8A/V transconductance from the VC pin voltage to current delivered to the output. This is shown in Figure 8 where the transconductance from VC pin to inductor current is essentially flat from 50Hz to 50kHz and phase shift is minimal in the important loop unity-gain band of 1kHz to 50kHz. Inductor variation from 3µH to 20µH will have very little effect on these curves. Overall gain from the VC pin to output is then modeled as the product of 1.8A/V transconductance multiplied by the complex impedance of the load in parallel with the output capacitor model. parallel with 12pF. In all practical applications, the compensation network from VC pin to ground has a much lower impedance than the output impedance of the amplifier at frequencies above 500Hz. This means that the error amplifier characteristics themselves do not contribute excess phase shift to the loop and the phase/gain characteristics of the error amplifier section are completely controlled by the external compensation network. The complete small-signal model is shown in Figure 9. R1 and R2 are the divider used to set output voltage. These are internal on the fixed voltage LT1507-3.3 with R1 = 1.8k and R2 = 5k. RC, CC and CF are external compensation POWER STAGE gm = 1.8A/V 12pF LT1507 VSW L1 OUTPUT ERROR AMPLIFIER gm = 2000µho 200k + TJ = TA + θJA(PTOT) Figure 8. Phase and Gain from VC Pin Voltage to Inductor Current R1 FB – Thermal resistance for the LT1507 packages is influenced by the presence of internal or backside planes. With a full plane under the SO package, thermal resistance will be about 120°C/W. No plane will increase resistance to about 150°C/W. To calculate die temperature, use the proper thermal resistance number for the desired package and add in worst-case ambient temperature; ESR + 2.42V C1 R2 GND VC CF RC CC 1507 • F09 The error amplifier can be modeled as a transconductance of 2000µmho, with an output impedance of 200kΩ in 18 Figure 9. Small-Signal Model for Loop Stability Analysis LT1507 U U W U APPLICATIONS INFORMATION components. In many cases only CC is needed. Adding RC will improve phase margin, but this may necessitate the need for CF to limit switching frequency ripple at the VC pin. move around, but at the same time phase moves with it so that adequate phase margin is maintained over a very wide range of ESR (≥ 5:1) Analog experts will note that around 1kHz, phase dips to within 20° of the zero phase margin line. This is typical of switching regulators because of the 2-pole rolloff generated by the output capacitor and the compensation network. This region of low phase is not a problem as long as it does not occur near unity-gain. In practice, the variability of output capacitor ESR tends to dominate all other effects with respect to loop response. Variations in ESR will cause unity-gain to U PACKAGE DESCRIPTION 200 80 VIN = 10V VOUT = 5V, IOUT = 500mA COUT = 100µF, 10V, AVX TPS CC = 3.3nF, RC = 0 L = 10µH LOOP GAIN (dB) 60 GAIN 40 100 PHASE 20 50 0 0 –20 0.01 150 LOOP PHASE (°C) In Figure 10, full loop phase/gain characteristics are shown with a compensation capacitor (CC) of 0.0033µF, giving the error amplifier a pole at 240Hz, with phase rolling off to 90° and staying there. The overall loop has a gain of 77dB at low frequency rolling off to unity gain at 20kHz. Phase shows a 2-pole characteristic until the ESR of the output capacitor brings it back above 10kHz. Phase margin is about 60° at unity-gain. 0.1 1 10 FREQUENCY (kHz) –50 100 100 LT1511 • F10 Figure 10. Overall Loop Phase and Gain Undervoltage Lockout See Application Information in LT1376 data sheet. Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead PDIP (Narrow 0.300) (LTC DWG # 05-08-1510) 0.300 – 0.325 (7.620 – 8.255) 0.009 – 0.015 (0.229 – 0.381) ( +0.025 0.325 –0.015 8.255 +0.635 –0.381 ) 0.045 – 0.065 (1.143 – 1.651) 0.065 (1.651) TYP 0.005 (0.127) MIN 0.100 ± 0.010 (2.540 ± 0.254) 0.400* (10.160) MAX 0.130 ± 0.005 (3.302 ± 0.127) 0.125 (3.175) MIN 0.015 (0.380) MIN 8 7 6 5 1 2 3 4 0.255 ± 0.015* (6.477 ± 0.381) 0.018 ± 0.003 (0.457 ± 0.076) N8 0695 *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm) Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT1507 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 2 3 4 0.053 – 0.069 (1.346 – 1.752) 0.004 – 0.010 (0.101 – 0.254) 0°– 8° TYP 0.016 – 0.050 0.406 – 1.270 0.014 – 0.019 (0.355 – 0.483) *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.050 (1.270) BSC SO8 0695 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1371 3A 500kHz Step-Up Switching Regulator High Current DC/DC Conversion Uses Small Power Components LT1372 1.5A 500kHz Step-Up Switching Regulator Includes Positive and Negative Output Voltage Regulation LT1375 1.5A 500kHz Step-Down Switching Regulator Includes Synchronization Capability LT1376 1.5A 500kHz Step-Down Switching Regulator Output Biasing Yields 90% Efficiency LT1377 1.5A 1MHz Step-Up Switching Regulator Highest Frequency Monolithic Switching Regulator 20 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900 FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com 1507f LT/TP 0697 4K • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 1996