Intersil CA3160AE 4mhz, bimos operational amplifier with mosfet input/cmos output Datasheet

CA3160, CA3160A
Data Sheet
September 1998
4MHz, BiMOS Operational Amplifier with
MOSFET Input/CMOS Output
The CA3160A and CA3160 are integrated circuit operational
amplifiers that combine the advantage of both CMOS and
bipolar transistors on a monolithic chip. The CA3160 series are
frequency compensated versions of the popular CA3130
series.
Gate protected P-Channel MOSFET (PMOS) transistors are
used in the input circuit to provide very high input
impedance, very low input current, and exceptional speed
performance. The use of PMOS field effect transistors in the
input stage results in common-mode input voltage capability
down to 0.5V below the negative supply terminal, an
important attribute in single supply applications.
File Number
Features
• MOSFET Input Stage Provides:
- Very High ZI = 1.5TΩ (1.5 x 1012Ω) (Typ)
- Very Low II . . . . . . . . . . . . . 5pA (Typ) at 15V Operation
. . . . . . . . . . . . . . . . . . . . . . . 2pA (Typ) at 5V Operation
• Common-Mode Input Voltage Range Includes
Negative Supply Rail; Input Terminals Can Be Swung
0.5V Below Negative Supply Rail
• CMOS Output Stage Permits Signal Swing to Either (or
Both) Supply Rails
Applications
• Ground Referenced Single Supply Amplifiers
• Fast Sample Hold Amplifiers
A complementary symmetry MOS (CMOS) transistor-pair,
capable of swinging the output voltage to within 10mV of
either supply voltage terminal (at very high values of load
impedance), is employed as the output circuit.
• Long Duration Timers/Monostables
The CA3160 Series circuits operate at supply voltages
ranging from 5V to 16V, or ±2.5V to ±8V when using split
supplies, and have terminals for adjustment of offset voltage
for applications requiring offset null capability. Terminal
provisions are also made to permit strobing of the output
stage.
• Wien-Bridge Oscillators
• High Input Impedance Wideband Amplifiers
• Voltage Followers (e.g., Follower for Single Supply
D/A Converter)
• Voltage Controlled Oscillators
• Photo Diode Sensor Amplifiers
Pinouts
CA3160
(METAL CAN)
TOP VIEW
The CA3160A offers superior input characteristics over
those of the CA3160.
PART NUMBER
TEMP.
RANGE (oC)
PACKAGE
PKG.
NO.
CA3160AE
-55 to 125
8 Ld PDIP
E8.3
CA3160E
-55 to 125
8 Ld PDIP
E8.3
CA3160T
-55 to 125
TAB
SUPPLEMENTARY
COMPENSATION
Ordering Information
8 Pin Metal Can
976.3
OFFSET
NULL
INV.
INPUT
1
7
-
2
NON-INV.
INPUT
STROBE
8
6
+
5
3
4
T8.C
V+
OUTPUT
OFFSET
NULL
V- AND CASE
CA3160
(PDIP)
TOP VIEW
OFFSET NULL
INV.
INPUT
NON-INV.
INPUT
V-
1
2
3
4
+
8
STROBE
7
V+
6
OUTPUT
5
OFFSET NULL
NOTE: CA3160 Series devices have an on-chip frequency
compensation network. Supplementary phase compensation or
frequency roll-off (if desired) can be connected externally between
Terminals 1 and 8.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 | Copyright © Intersil Corporation 1999
CA3160, CA3160A
Absolute Maximum Ratings
Thermal Information
Supply Voltage (Between V+ and V- Terminals) . . . . . . . . . . . +16V
Differential Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . .8V
Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . (V+ +8V) to (V- -0.5V)
Input Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1mA
Output Short Circuit Duration (Note 2). . . . . . . . . . . . . . . . Indefinite
Thermal Resistance (Typical, Note 1)
θJA (oC/W) θJC (oC/W)
PDIP Package . . . . . . . . . . . . . . . . . . .
110
N/A
Metal Can Package . . . . . . . . . . . . . . .
170
85
Maximum Junction Temperature (Metal Can). . . . . . . . . . . . . . .175oC
Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC
Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC
Operating Conditions
Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . -55oC to 125oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES:
1. θJA is measured with the component mounted on an evaluation PC board in free air.
2. Short Circuit may be applied to ground or to either supply.
TA = 25oC, V+ = 15V, V- = 0V, Unless Otherwise Specified
Electrical Specifications
CA3160
PARAMETER
SYMBOL
TEST CONDITIONS
CA3160A
MIN
TYP
MAX
MIN
TYP
MAX
UNITS
Input Offset Voltage
|VIO|
VS = ±7.5V
-
6
15
-
2
5
mV
Input Offset Current
|IIO|
VS = ±7.5V
-
0.5
30
-
0.5
20
pA
II
VS = ±7.5V
-
5
50
-
5
30
pA
50
320
-
50
320
-
kV/V
94
110
-
94
110
-
dB
CMRR
70
90
-
80
95
-
dB
VlCR
0
-0.5 to 12
10
0
-0.5 to 12
10
V
-
32
320
-
32
150
µV/V
12
13.3
-
12
13.3
-
V
-
0.002
0.01
-
0.002
0.01
V
14.99
15
-
14.99
15
-
V
-
0
0.01
-
0
0.01
V
Input Current
Large-Signal Voltage Gain
AOL
Common-Mode Rejection Ratio
Common-Mode Input-Voltage Range
VO = 10VP-P, RL = 2kΩ
Power-Supply Rejection Ratio
PSRR
∆VIO/∆VS, VS = ±7.5V
Maximum Output Voltage
VOM+
RL = 2kΩ
VOMVOM+
RL = ∞
VOMMaximum Output Current
Supply Current (Note 3)
IOM+
VO = 0V (Source)
12
22
45
12
22
45
mA
IOM-
VO = 15V (Sink)
12
20
45
12
20
45
mA
VO = 7.5V, R L = ∞
-
10
15
-
10
15
mA
VO = 0V, R L = ∞
-
2
3
-
2
3
mA
∆VIO/∆T
-
8
-
-
6
-
µV/oC
I+
Input Offset Voltage Temperature Drift
For Design Guidance, VSUPPLY = ±7.5V, TA = 25oC, Unless Otherwise Specified
Electrical Specifications
PARAMETER
SYMBOL
Input Offset Voltage Adjustment Range
TYP
TYP
UNITS
±22
±22
mV
1.5
1.5
TΩ
4.3
4.3
pF
RS = 1MΩ
40
40
µV
RS = 10MΩ
50
50
µV
10kΩ Across Terminals 4 and 5 or
Terminals 4 and 1
RI
Input Capacitance
CI
f = 1MHz
Equivalent Input Noise Voltage
eN
BW = 0.2MHz
eN
2
CA3160A
TEST CONDITIONS
Input Resistance
Equivalent Input Noise Voltage
CA3160
RS = 100Ω
1kHz
72
72
nV/√Hz
10kHz
30
30
nV/√Hz
CA3160, CA3160A
For Design Guidance, VSUPPLY = ±7.5V, TA = 25oC, Unless Otherwise Specified (Continued)
Electrical Specifications
PARAMETER
SYMBOL
TEST CONDITIONS
CA3160
CA3160A
TYP
TYP
UNITS
Unity Gain Crossover Frequency
fT
4
4
MHz
Slew Rate
SR
10
10
V/µs
CL = 25pF, RL = 2kΩ, (Voltage Follower)
0.09
0.09
µs
10
10
%
CL = 25pF, RL = 2kΩ, (Voltage Follower)
To <0.1%, VIN = 4VP-P
1.8
1.8
µs
Transient Response
Rise and Fall Time
tr
Overshoot
OS
Settling Time
tS
For Design Guidance, V+ = +5V, V- = 0V, TA = 25oC, Unless Otherwise Specified
Electrical Specifications
PARAMETER
SYMBOL
TEST CONDITIONS
CA3160
CA3160A
TYP
TYP
UNITS
Input Offset Voltage
VIO
6
2
mV
Input Offset Current
IIO
0.1
0.1
pA
Il
2
2
pA
CMRR
80
90
dB
100
100
kV/V
100
100
dB
0 to 2.8
0 to 2.8
V
VO = 5V, RL = ∞
300
300
µA
VO = 2.5V, RL = ∞
500
500
µA
∆VIO/∆V+
200
200
µV/V
Input Current
Common-Mode Rejection Ratio
Large Signal Voltage Gain
AOL
Common-Mode Input Voltage Range
VlCR
Supply Current
I+
Power Supply Rejection Ratio
VO = 4VP-P, RL = 5kΩ
PSRR
NOTE:
3. ICC typically increases by 1.5mA/MHz during operation.
Block Diagram
7
200µA
1.35mA
8mA
(NOTE 4)
200µA
NOTES:
0mA
(NOTE 5)
BIAS CKT.
V+
4. Total supply voltage (for indicated voltage
gains) = 15V with input terminals biased so
that Terminal 6 potential is +7.5V above
Terminal 4.
5. Total supply voltage (for indicated voltage
gains) = 15V with output terminal driven to
either supply rail.
+
3
OUTPUT
AV ≈
6000X
AV ≈ 5X
INPUT
AV ≈ 30X
6
2
-
4
CC
5
1
8
COMPENSATION
(WHEN DESIRED)
OFFSET
NULL
3
STROBE
V-
CA3160, CA3160A
Schematic Diagram
BIAS CURRENT
Q1
D1
Z1
8.3V
D2
7
“CURRENT SOURCE
LOAD” FOR Q11
CURRENT SOURCE
FOR Q6 AND Q7
Q2
Q3
Q4
Q5
V+
D3
D4
R1
40kΩ
R2
5kΩ
INPUT STAGE
D5
NON-INV.
INPUT
3
2
D7
D6
SECOND
STAGE
OUTPUT
STAGE
+
Q6
Q7
R3
1kΩ
R5
1kΩ
6
30
pF
R4
1kΩ
Q9
OUTPUT
2kΩ
-
INV. INPUT
Q8
Q12
Q11
Q10
R6
1kΩ
5
1
OFFSET NULL
SUPPLEMENTARY
COMP IF DESIRED
8
4
STROBING
NOTE: Diodes D5 Through D7 Provide Gate Oxide Protection For MOSFET Input Stage.
Application Information
Circuit Description
Refer to the Block Diagram of the CA3160 series CMOS
Operational Amplifiers. The input terminals may be operated
down to 0.5V below the negative supply rail, and the output
can be swung very close to either supply rail in many
applications. Consequently, the CA3160 series circuits are
ideal for single supply operation. Three class A amplifier
stages, having the individual gain capability and current
consumption shown in the Block Diagram provide the total
gain of the CA3160. A biasing circuit provides two potentials
for common use in the first and second stages. Terminals 8
and 1 can be used to supplement the internal phase
compensation network if additional phase compensation or
frequency roll-off is desired. Terminals 8 and 4 can also be
used to strobe the output stage into a low quiescent current
state. When Terminal 8 is tied to the negative supply rail
(Terminal 4) by mechanical or electrical means, the output
potential at Terminal 6 essentially rises to the positive supplyrail potential at Terminal 7. This condition of essentially zero
current drain in the output stage under the strobed “OFF”
condition can only be achieved when the ohmic load
4
resistance presented to the amplifier is very high (e.g., when
the amplifier output is used to drive MOS digital circuits in
comparator applications).
Input Stage - The circuit of the CA3160 is shown in the
Schematic Diagram. It consists of a differential-input stage
using PMOS field-effect transistors (Q6, Q7) working into a
mirror-pair of bipolar transistors (Q9, Q10) functioning as load
resistors together with resistors R3 through R6. The mirrorpair transistors also function as a differential-to-single-ended
converter to provide base drive to the second-stage bipolar
transistor (Q11). Offset nulling, when desired, can be effected
by connecting a 100,000Ω potentiometer across Terminals 1
and 5 and the potentiometer slider arm to Terminal 4.
Cascode-connected PMOS transistors Q2, Q4, are the
constant-current source for the input stage. The biasing circuit
for the constant-current source is subsequently described.
The small diodes D5 through D7 provide gate-oxide protection
against high-voltage transients, including static electricity
during handling for Q6 and Q7.
Second-Stage - Most of the voltage gain in the CA3160 is
provided by the second amplifier stage, consisting of bipolar
CA3160, CA3160A
transistor Q11 and its cascode-connected load resistance
provided by PMOS transistors Q3 and Q5. The source of bias
potentials for these PMOS transistors is described later. Miller
Effect compensation (roll off) is accomplished by means of the
30pF capacitor and 2kΩ resistor connected between the base
and collector of transistor Q11. These internal components
provide sufficient compensation for unity gain operation in
most applications. However, additional compensation, if
desired, may be used between Terminals 1 and 8.
Bias-Source Circuit - At total supply voltages, somewhat
above 8.3V, resistor R2 and zener diode Z1 serve to establish a
voltage of 8.3V across the series-connected circuit, consisting
of resistor R1, diodes D1 through D4, and PMOS transistor Q1.
A tap at the junction of resistor R1 and diode D4 provides a
gate-bias potential of about 4.5V for PMOS transistors Q4 and
Q5 with respect to Terminal 7. A potential of about 2.2V is
developed across diode-connected PMOS transistor Q1 with
respect to Terminal 7 to provide gate bias for PMOS transistors
Q2 and Q3. It should be noted that Q1 is “mirror-connected” to
both Q2 and Q3. Since transistors Q1, Q2, Q3 are designed to
be identical, the approximately 200µA current in Q1 establishes
a similar current in Q2 and Q3 as constant-current sources for
both the first and second amplifier stages, respectively.
At total supply voltages somewhat less than 8.3V, zener diode
Z1 becomes nonconductive and the potential, developed
across series-connected R1, D1 - D4, and Q1, varies directly
with variations in supply voltage. Consequently, the gate bias
for Q4, Q5 and Q2, Q3 varies in accordance with supplyvoltage variations. This variation results in deterioration of the
power-supply-rejection ratio (PSRR) at total supply voltages
below 8.3V. Operation at total supply voltages below about
4.5V results in seriously degraded performance.
Output Stage - The output stage consists of a drain-loaded
inverting amplifier using CMOS transistors operating in the
Class A mode. When operating into very high resistance loads,
the output can be swung within millivolts of either supply rail.
Because the output stage is a drain-loaded amplifier, its gain is
dependent upon the load impedance. The transfer
characteristics of the output stage for a load returned to the
negative supply rail are shown in Figure 17. Typical op amp
loads are readily driven by the output stage. Because largesignal excursions are non-linear, requiring feedback for good
waveform reproduction, transient delays may be encountered.
As a voltage follower, the amplifier can achieve 0.01% accuracy
levels, including the negative supply rail.
Offset Nulling
Offset-voltage nulling is usually accomplished with a
100,000Ω potentiometer connected across Terminals 1 and
5 and with the potentiometer slider arm connected to
Terminal 4. A fine offset-null adjustment usually can be
effected with the slider arm positioned in the mid-point of the
potentiometer's total range.
5
Input Current Variation with Common Mode Input
Voltage
As shown in the Electrical Specifications, the input current for
the CA3160 Series Op Amps is typically 5pA at TA = 25oC
when Terminals 2 and 3 are at a common-mode potential of
+7.5V with respect to negative supply Terminal 4. Figure 23
contains data showing the variation of input current as a
function of common-mode input voltage at TA = 25oC. These
data show that circuit designers can advantageously exploit
these characteristics to design circuits which typically require
an input current of less than 1pA, provided the common-mode
input voltage does not exceed 2V. As previously noted, the
input current is essentially the result of the leakage current
through the gate-protection diodes in the input circuit and,
therefore, a function of the applied voltage. Although the finite
resistance of the glass terminal-to-case insulator of the metal
can package also contributes an increment of leakage current,
there are useful compensating factors. Because the gateprotection network functions as if it is connected to Terminal 4
potential, and the metal can case of the CA3160 is also
internally tied to Terminal 4, input Terminal 3 is essentially
“guarded” from spurious leakage currents.
Input-Current Variation with Temperature
The input current of the CA3160 Series circuits is typically 5pA
at 25oC. The major portion of this input current is due to
leakage current through the gate-protective diodes in the input
circuit. As with any semiconductor junction device, including op
amps with a junction-FET input stage, the leakage current
approximately doubles for every 10oC increase in temperature.
Figure 24 provides data on the typical variation of input bias
current as a function of temperature in the CA3160.
In applications requiring the lowest practical input current and
incremental increases in current because of “warm-up” effects,
it is suggested that an appropriate heat sink be used with the
CA3160. In addition, when “sinking” or “sourcing” significant
output current the chip temperature increases, causing an
increase in the input current. In such cases, heat-sinking can
also very markedly reduce and stabilize input current variations.
Input Offset Voltage (VIO) Variation with DC Bias
vs Device Operating Life
It is well known that the characteristics of a MOSFET device
can change slightly when a DC gate-source bias potential is
applied to the device for extended time periods. The magnitude
of the change is increased at high temperatures. Users of the
CA3160 should be alert to the possible impacts of this effect if
the application of the device involves extended operation at
high temperatures with a significant differential DC bias voltage
applied across Terminals 2 and 3. Figure 25 shows typical data
pertinent to shifts in offset voltage encountered with CA3160
devices in metal can packages during life testing. At lower
temperatures (metal can and plastic) for example at 85oC, this
change in voltage is considerably less. In typical linear
applications where the differential voltage is small and
symmetrical, these incremental changes are of about the same
CA3160, CA3160A
magnitude as those encountered in an operational amplifier
employing a bipolar transistor input stage. The 2V differential
voltage example represents conditions when the amplifier
output state is “toggled”, e.g., as in comparator applications.
Power Supply Considerations
Because the CA3160 is very useful in single supply
applications, it is pertinent to review some considerations
relating to power supply current consumption under both
single and dual supply service. Figures 1A and 1B show the
CA3160 connected for both dual and single supply operation.
Dual-supply operation: When the output voltage at Terminal
6 is 0V, the currents supplied by the two power supplies are
equal. When the gate terminals of Q8 and Q12 are driven
increasingly positive with respect to ground, current flow
through Q12 (from the negative supply) to the load is
increased and current flow through Q8 (from the positive
supply) decreases correspondingly. When the gate terminals
of Q8 and Q12 are driven increasingly negative with respect
to ground, current flow through Q8 is increased and current
flow through Q12 is decreased accordingly.
Single supply operation: Initially, let it be assumed that the
value of RL is very high (or disconnected), and that the inputterminal bias (Terminals 2 and 3) is such that the output
terminal (No. 6) voltage is at V+/2, i.e., the voltage-drops
across Q8 and Q12 are of equal magnitude. Figure 18 shows
typical quiescent supply-current vs supply voltage for the
CA3160 operated under these conditions.
Since the output stage is operating as a Class A amplifier, the
supply current will remain constant under dynamic operating
conditions as long as the transistors are operated in the linear
portion of their voltage-transfer characteristics (see Figure 17).
If either Q8 or Q12 are swung out of their linear regions toward
cutoff (a non-linear region), there will be a corresponding
reduction in supply-current. In the extreme case, e.g., with
Terminal 8 swung down to ground potential (or tied to ground),
NMOS transistor Q12 is completely cut off and the supply
current to series connected transistors Q8, Q12 goes
essentially to zero. The two preceding stages in the CA3160,
however, continue to draw modest supply-current (see the
lower curve in Figure 18) even though the output stage is
strobed off. Figure 1A shows a dual-supply arrangement for the
output stage that can also be strobed off, assuming RL = ∞, by
pulling the potential of Terminal 8 down to that of Terminal 4.
Let it now-be assumed that a load resistance of nominal value
(e.g., 2kΩ) is connected between Terminal 6 and ground in the
circuit of Figure 1B. Let it further be assumed again that the
input-terminal bias (Terminals 2 and 3) is such that the output
terminal (No. 6) voltage is at V+/2. Since PMOS transistor Q8
must now supply quiescent current to both RL and transistor
Q12, it should be apparent that under these conditions the
supply current must increase as an inverse function of the RL
magnitude. Figure 20 shows the voltage-drop across PMOS
transistor Q8 as a function of load current at several supply
6
voltages. Figure 17 shows the voltage transfer characteristics of
the output stage for several values of load resistance.
Wideband Noise
From the standpoint of low-noise performance considerations,
the use of the CA3160 is most advantageous in applications
where in the source resistance of the input signal is on the
order of 1MΩ or more. In this case, the total input-referred
noise voltage is typically only 40µV when the test circuit
amplifier of Figure 2 is operated at a total supply voltage of
15V. This value of total input-referred noise remains
essentially constant, even though the value of source
resistance is raised by an order of magnitude. This
characteristic is due to the fact that reactance of the input
capacitance becomes a significant factor in shunting the
source resistance. It should be noted, however, that for values
of source resistance very much greater than 1MΩ, the total
noise voltage generated can be dominated by the thermal
noise contributions of both the feedback and source resistors.
7
3
V+
+
Q8
CA3160
OUTPUT
Q12
STAGE
6
RL
2
-
4
V-
8
NEGATIVE
SUPPLY
FIGURE 1A. DUAL POWER SUPPLY OPERATION
V+
7
3
+
Q8
CA3160
OUTPUT
Q12
STAGE
6
RL
2
-
4
8
FIGURE 1B. SINGLE POWER SUPPLY OPERATION
FIGURE 1. CA3160 OUTPUT STAGE IN DUAL AND SINGLE
POWER SUPPLY OPERATION
CA3160, CA3160A
+7.5V
0.01µF
RS
3
1MΩ
7
+
2
-
NOISE
VOLTAGE
OUTPUT
6
CA3160
4
30.1kΩ
0.01
µF
-7.5V
BW (3dB) = 200kHz
TOTAL NOISE VOLTAGE
(INPUT REFERRED = 40µV (TYP)
1kΩ
FIGURE 2. TEST CIRCUIT AMPLIFIER (30dB GAIN) USED FOR
WIDEBAND NOISE MEASUREMENTS
Typical Performance Curves
+7.5V
0.01µF
3
10kΩ
7
+
6
CA3160
2
-
4
2kΩ
0.01
µF
-7.5V
2kΩ
BW (-3dB) = 4MHz
SR = 10V/µs
25pF
SIMULATED
LOAD
CAPACITANCE
0.1µF
FIGURE 3A.
Top Trace: Output
Bottom Trace: Input
FIGURE 3B. SMALL SIGNAL RESPONSE
Top Trace: Output Signal
Center Trace: Difference Signal 5mV/Div.
Bottom Trace: Input Signal
FIGURE 3C. INPUT-OUTPUT DIFFERENCE SIGNAL SHOWING
SETTLING TIME
FIGURE 3. DUAL SUPPLY VOLTAGE FOLLOWER WITH ASSOCIATED WAVEFORMS
7
CA3160, CA3160A
Typical Applications
+15V
Voltage Followers
Operational amplifiers with very high input resistances, like
the CA3160, are particularly suited to service as voltage
followers. Figure 3 shows the circuit of a classical voltage
follower, together with pertinent waveforms using the CA3160
in a split-supply configuration.
A voltage follower, operated from a single supply, is shown in
Figure 4 together with related waveforms. This follower circuit
is linear over a wide dynamic range, as illustrated by the
reproduction of the output waveform in Figure 4B with inputsignal ramping. The waveforms in Figure 4C show that the
follower does not lose its input-to-output phase-sense, even
though the input is being swung 7.5V below ground potential.
This unique characteristic is an important attribute in both
operational amplifier and comparator applications. Figure 4C
also shows the manner in which the COS/MOS output stage
permits the output signal to swing down to the negative
supply-rail potential (i.e., ground in the case shown). The
digital-to-analog converter (DAC) circuit, described in the
following section, illustrates the practical use of the CA3160 in
a single supply voltage follower application.
0.01µF
3
10kΩ
7
+
6
CA3160
2
-
4
5
1
OFFSET
ADJUST
2kΩ
BW (-3dB) = 4MHz
SR = 10V/µs
0.1µF
FIGURE 4A.
9-Bit CMOS DAC
A typical circuit of a 9-bit Digital-to-Analog Converter (DAC) (see
Note 6) is shown in Figure 5. This system combines the concepts
of multiple-switch CMOS lCs, a low-cost ladder network of
discrete metal-oxide-film resistors, a CA3160 op amp connected
as a follower, and an inexpensive monolithic regulator in a simple
single power-supply arrangement. An additional feature of the
DAC is that it is readily interfaced with CMOS input logic, e.g.,
10V logic levels are used in the circuit of Figure 5.
The circuit uses an R/2R voltage-ladder network, with the outputpotential obtained directly by terminating the ladder arms at
either the positive or the negative power supply terminal. Each
CD4007A contains three inverters, each inverter functioning as a
single-pole double-throw switch to terminate an arm of the R/2R
network at either the positive or negative power-supply terminal.
The resistor ladder is an assembly of 1% tolerance metal-oxide
film resistors. The five arms requiring the highest accuracy are
assembled with series and parallel combinations of 806,000Ω
resistors from the same manufacturing lot.
A single 15V supply provides a positive bus for the CA3160
follower amplifier and feeds the CA3085 voltage regulator. A
“scale-adjust” function is provided by the regulator output control,
set to a nominal 10V level in this system. The line-voltage
regulation (approximately 0.2%) permits a 9-bit accuracy to be
maintained with variations of several volts in the supply. The
flexibility afforded by the CMOS building blocks simplifies the
design of DAC systems tailored to particular needs.
NOTE:
6. “Digital-to-Analog Conversion Using the Intersil CD4007A
COS/MOS lC”, Application Note AN6080.
8
Top Trace: Output
Bottom Trace: Input
FIGURE 4B. OUTPUT WAVEFORM WITH GROUND REFERENCE
SINE WAVE INPUT
FIGURE 4C. OUTPUT SIGNAL WITH INPUT SIGNAL RAMPING
FIGURE 4. SINGLE SUPPLY VOLTAGE FOLLOWER WITH
ASSOCIATED WAVEFORMS. (e.g., FOR USE IN
SINGLE SUPPLY D/A CONVERTER; SEE FIGURE 9
IN AN6080)
CA3160, CA3160A
Error-Amplifier in Regulated Power Supplies
amplitude (V) and width (T2). Since the output (Terminal 6) of
A1 (a CA3130) can swing within about 10mV of either supplyrail, the output pulse amplitude (V) is essentially equal to V+.
The average output voltage (EAVG = V T2/T1) is applied to the
non-inverting Input terminal of comparator A2 via an integrating
network R3, C2. Comparator A2 operates to establish circuit
conditions such that EAVG = V1. This circuit condition is
accomplished by feeding an output signal from Terminal 6 of A2
through R4, D4 to the inverting terminal (Terminal 2) of A1,
thereby adjusting the multivibrator interval, T3.
The CA3160 is an ideal choice for error-amplifier service in
regulated power supplies since it can function as an erroramplifier when the regulated output voltage is required to
approach zero.
The circuit shown in Figure 6 uses a CA3160 as an error
amplifier in a continuously adjustable 1A power supply. One
of the key features of this circuit is its ability to regulate down
to the vicinity of 0V with only one DC power supply input.
An RC network, connected between the base of the output
drive transistor and the input voltage, prevents “turn-on
overshoot”, a condition typical of many operational
amplifier regulator circuits. As the amplifier becomes
operational, this RC network ceases to have any influence
on the regulator performance.
Voltmeter With High Input Resistance
The voltmeter circuit shown in Figure 8 illustrates an
application in which a number of the CA3160 characteristics
are exploited. Range-switch SW1 is ganged between input
and output circuitry to permit selection of the proper output
voltage for feedback to Terminal 2 via 10kΩ current-limiting
resistor. The circuit is powered by a single 8.4V mercury
battery. With zero input signal, the circuit consumes
somewhat less than 500µA plus the meter current required
to indicate a given voltage. Thus, at full scale input, the total
supply current rises to slightly more than 1500µA.
Precision Voltage-Controlled Oscillator
The circuit diagram of a precision voltage-controlled oscillator is
shown in Figure 7. The oscillator operates with a tracking error
in the order of 0.02% and a temperature coefficient of
0.01%/oC. A multivibrator (A1) generates pulses of constant
10V LOGIC INPUTS
+10.010V
14
LSB
9
8
7
6
3
10
11
6
5
4
3
2
MSB
1
6
3
10
6
3
10
2
CD4007A
“SWITCHES”
9
13
1
7
8
5
4
806K
1%
CD4007A
“SWITCHES”
12
402K
1%
13
1
8
5
200K
1%
VOLTAGE
REGULATOR
+15V
2
62
1
+10.010V
CA3085
8
6
3
22.1K
1%
7
+
-
4
2µF
25V
1K REGULATED
VOLTAGE
ADJUST
3.83K
1%
0.001µF
13
12
806K
1%
806K
1%
100K
1%
806K
1%
806K
1%
CD4007A
“SWITCHES”
1
8
806K
1%
12
5
(2)
806K
1%
(4)
806K
1%
(8)
806K
1%
+15V
750K
1%
BIT
REQUIRED
RATIO-MATCH
1
Standard
2
±0.1%
3
±0.2%
4
±0.4%
5
±0.8%
6-9
±1% ABS.
PARALLELED
RESISTORS
OUTPUT
+
3
VOLTAGE
FOLLOWER
CA3160
6
-
4
5
LOAD
FIGURE 5. 9-BIT DAC USING CMOS DIGITAL SWITCHES AND CA3160
9
10K
7
1
100K
OFFSET
NULL
2K
0.1µF
2
CA3160, CA3160A
2N6385
POWER DARLINGTON
3
SHORT-CIRCUIT CURRENT
LIMIT ADJUSTMENT
40V INPUT
+
1Ω
2
TURN
ON
DELAY
2.4kΩ
1W
1kΩ
1.5kΩ
1W
100kΩ
1
2N2102
1kΩ
56pF
1N914
7
+
+ 5µF
10
9
7
3
5
6
4
-
3
CA3160
6
1
2N2102
8
+
100µF
10kΩ
+
2kΩ
-
CA3086
11
2
43kΩ
8
2.2kΩ
100µF
25V
OUTPUT
0V TO 35V
AT 1A
10kΩ
0.2µF
-
5
2
1
12
10kΩ
14
13
8.2kΩ
4
4.7kΩ
1kΩ
50kΩ
100kΩ
0.01µF
62kΩ
-
-
Hum and Noise Output <250µVRMS; Regulation (No Load to Full Load) <0.005%; Input Regulation <0.01%/V
FIGURE 6. VOLTAGE REGULATOR CIRCUIT (0.1V TO 35V AT 1A)
T2
T3
V
VCO CONTROL VOLTAGE (VI)
(0V - 10V)
(SENSITIVITY = 1kHz/V)
fO
+15V
T1
D1
10K
1M
0.01µF
+15V
R5
100K
100K
3
R6
100K
+15V
D2
0.1
µF
7
+
MULTIVIBRATOR
CA3130
C1
500pF
2
2
EAVG = V T2/T1
6
R3
1M
-
3
4
D4
R1
182K
R2
10K
D1 - D5 = 1N914
D5
COMPARATOR
CA3160
+
4
6
5
C2
0.01µF
D3
7
-
0.01µF
1
R7
100K
R4
3K
FIGURE 7. VOLTAGE CONTROLLED OSCILLATOR
Function Generator
A function generator having a wide tuning range is shown in
Figure 9. The adjustment range, in excess of 1,000,000/1, is
accomplished by a single potentiometer. Three operational
amplifiers are utilized: a CA3160 as a voltage follower, a
CA3080 as a high speed comparator, and a second
10
CA3080A as a programmable current source. Three variable
capacitors C1, C2, and C3 shape the triangular signal
between 500kHz and 1MHz. Capacitors C4, C5, and the
trimmer potentiometer in series with C5 maintain essentially
constant (+10%) amplitude up to 1MHz.
CA3160, CA3160A
300V
300V
100MΩ
100V
100V
30V
30V
1.02
MΩ
10V
BATTERY
TEST
OFF
ON
3 POSITION
SLIDE SWITCH
9.9kΩ
10V
+
500
µF
SW1A 3V
INPUT
SW1B
3V
1V
1V
300V
300V
100V
100V
30V
30V
10V
10V
0-1mA
7
3
+
22MΩ
2.7kΩ
-
2
300V
100V
300V
4
820Ω 200Ω
30V
100V
5
1
30V
100kΩ
ZERO
ADJUST
3V CAL.
500Ω
6
CA3160
0.001µF
M
BATTERY
+9V
BATTERY
10V
1V CAL.
10V
3V
SW1C 3V
1V
300mV
1V
9.1kΩ
9kΩ
100mV
300mV
100mV
10kΩ
SW1D
30mV
900Ω
10mV
30mV
10mV
100Ω
FIGURE 8. HIGH INPUT RESISTANCE DC VOLTMETER
20pF
8.2kΩ
+7.5V
0.9 - 7pF
C1
VOLTAGE-CONTROLLED
CURRENT SOURCE
7
3
+
6.2kΩ
CA3080A
1kΩ
2
-
1kΩ
6
10-80pF
4
5
C2
2MΩ
3
0.1µF
SYMMETRY
-7.5V
100kΩ
+7.5V
5
+
6
3
4
MIN FREQ.SET
-7.5V
+7.5V
10kΩ
6.2kΩ
500Ω
FREQ
ADJUST
500Ω
2
CA3080
-7.5V
0.1µF
C4
4 - 60pF
2kΩ
HIGH FREQ
LEVEL
ADJUST
FIGURE 9A. 1,000,000/1 SINGLE CONTROL FUNCTION GENERATOR: 1Hz to 1MHz
11
6
+
10kΩ
-7.5V
MAX FREQ
SET
7
10kΩ
4 - 60pF CA3160
C3
2
EXTERNAL
SWEEPING INPUT
30kΩ
6.8MΩ
7
-7.5V
4.7kΩ
-7.5V
430pF
+7.5V
HIGH
FREQ.
SHAPE
THRESHOLD
DETECTOR
+7.5V
+7.5V
CENTERING
100kΩ
BUFFER
VOLTAGE FOLLOWER
50kΩ
C5
15 - 115pF
2-1N914
CA3160, CA3160A
NOTE: A square wave signal modulates the external sweeping input
to produce 1Hz and 1MHz, showing the 1,000,000/1 frequency range
of the Function Generator.
NOTE: The bottom trace is the sweeping signal and the top trace is
the actual generator output. The center trace displays the 1MHz signal
via delayed oscilloscope triggering of the upper swept output signal.
FIGURE 9B. TWO-TONE OUTPUT SIGNAL FROM THE
FUNCTION GENERATOR
FIGURE 9C. TRIPLE-TRACE OF THE FUNCTION GENERATOR
SWEEPING TO 1MHz
FIGURE 9. 1,000,000/1 SINGLE CONTROL FUNCTION GENERATOR: 1Hz to 1MHz
5.1kΩ
+15V
1N914
470pF
STAIRCASE
OUTPUT
+15V
100
kΩ
100
kΩ
1MΩ
3
100
kΩ
+
2
15 - 115pF
FREQ
ADJUST
-
7
7
6
CA3130
2
3
4
-
10kΩ
CA3160
1N914
8
+15V
+15V
STEP HEIGHT
ADJUST
4 - 60pF
8.2kΩ
7
6
+
CHARGE
COMMUTATING
NETWORK
1.5
MΩ
+
6
CA3130
2kΩ
4
MULTIVIBRATOR
3
+15V
2
-
8
4
INTEGRATOR
HYSTERESIS SWITCH
MULTIVIBRATOR RETRACE INHIBIT
+15mV TO +10V
51kΩ
100kΩ
FIGURE 10A. STAIRCASE GENERATOR CIRCUIT
Staircase Generator
Figure 10 shows a staircase generator circuit utilizing three
CMOS operational amplifiers. Two CA3130s are used; one
as a multivibrator, the other as a hysteresis switch. The third
amplifier, a CA3160, is used as a linear staircase generator.
Picoammeter Circuit
Figure 11 is a current-to-voltage converter configuration
utilizing a CA3160 and CA3140 to provide a picoampere
12
meter for 13pA full scale meter deflection. By placing
Terminals 2 and 4 of the CA3160 at ground potential, the
CA3160 input is operated in the “guarded mode”. Under this
operating condition, even slight leakage resistance present
between Terminals 3 and 2 or between Terminals 3 and 4
would result in zero voltage across this leakage resistance,
thus substantially reducing the leakage current.
If the CA3160 is operated with the same voltage on input
Terminals 3 and 2 as on Terminal 4, a further reduction in the
CA3160, CA3160A
input current to the less than one picoampere level can be
achieved as shown in Figure 23.
To further enhance the stability of this circuit, the CA3160
can be operated with its output (Terminal 6) near ground,
thus markedly reducing the dissipation by reducing the
supply current to the device.
STAIRCASE
OUTPUT
2V STEPS
The CA3140 stage serves as a X100 gain stage to provide
the required plus and minus output swing for the meter and
feedback network. A 100-to-1 voltage divider network
consisting of a 9.9kΩ resistor in series with a 100Ω resistor
sets the voltage at the 10GΩ resistor (in series with Terminal
3) to ±30mV full-scale deflection. This 30mV signal results
from ±3V appearing at the top of the voltage divider network
which also drives the meter circuitry.
COMPARATOR
OSCILLATOR
Top Trace: Staircase Output 2V Steps
Center Trace: Comparator
Bottom Trace: Oscillator
By utilizing a switching technique in the meter circuit and in
the 9.9kΩ and 100Ω network similar to that used in voltmeter
circuit shown in Figure 8, a current range of 3pA to 1nA full
scale can be handled with the single 10GΩ resistor.
FIGURE 10B. STAIRCASE GENERATOR WAVEFORM
FIGURE 10. STAIRCASE GENERATOR CIRCUIT
10GΩ
+15V
1MΩ
0.1µF
10pF
+15V
7
10MΩ
3
+
7
CA3160
2
6
-
2
10kΩ
-
CA3140
4
5
6
+
3
5.6kΩ
9.9kΩ
1
4
560kΩ
100kΩ
0.1µF
500Ω
9.1kΩ
100Ω
-15V
M
500-0-500µA
-15V
FIGURE 11. CURRENT-TO-VOLTAGE CONVERTER TO PROVIDE A PICOAMMETER WITH ±3pA FULL SCALE DEFLECTION
100kΩ
+15V
+15V
2200pF
30pF
+15V
0.1µF
0.1µF
7
1MΩ
39kΩ
3
+
0.1µF
CA3160
-
2
6
2
1N914
8
4
5
7
3
+
0.1µF
2
6
100kΩ
4
100kΩ
CA3140
1MΩ
3
27kΩ
4
DROOP
ZERO
ADJUST
0.1µF
39kΩ
500µA
STROBE INPUT
SAMPLE = 15V
HOLD = 0V
2kΩ
FIGURE 12A. SINGLE SUPPLY SAMPLE AND HOLD SYSTEM, INPUT 0V TO 10V
13
6
+
5
8.2kΩ
9.1kΩ
7
CA3080A
1
OFFSET
VOLTAGE
ADJUST
8.2Ω
CA3160, CA3160A
SAMPLED
OUTPUT
SAMPLED
OUTPUT
0V-
INPUT
SIGNAL
INPUT 0V-
SAMPLING
PULSES
SAMPLING
PULSES
Top Trace: Sampled Output
Center Trace: Input Signal
Bottom Trace: Sampling Pulses
Top Trace: Sampled Output
Center Trace: Input Signal
Bottom Trace: Sampling Pulses
FIGURE 12B. SAMPLE AND HOLD WAVEFORM
FIGURE 12C. SAMPLE AND HOLD WAVEFORM
FIGURE 12. SINGLE SUPPLY SAMPLE AND HOLD SYSTEM, INPUT 0V TO 10V
Single Supply Sample-and-Hold System
Figure 12 shows a single supply sample-and-hold system
using a CA3160 to provide a high input impedance and an
input voltage range of 0V to 10V. The output from the input
buffer integrator network is coupled to a CA3080A. The
CA3080A functions as a strobeable current source for the
CA3140 output integrator and storage capacitor. The
CA3140 was chosen because of its low output impedance
and constant gain-bandwidth product. Pulse “droop” during
the hold interval can be reduced to zero by adjusting the
100kΩ bias-voltage potentiometer on the positive input of
the CA3140. This zero adjustment sets the CA3080A output
voltage at its zero current position. In this sample-and-hold
circuit it is essential that the amplifier bias current be
reduced to zero to minimize output signal current during the
hold mode. Even with 320mV at the amplifier bias circuit
terminal (5) at least 1100pA of output current will be
available.
Wien Bridge Oscillator
A simple, single supply Wien Bridge oscillator using a
CA3160 is shown in Figure 13. A pair of parallel-connected
1N914 diodes comprise the gain-setting network which
standardizes the output voltage at approximately 1.1V. The
500Ω potentiometer is adjusted so that the oscillator will
always start and the oscillation will be maintained.
Increasing the amplitude of the voltage may lower the
threshold level for starting and for sustaining the oscillation,
but will introduce more distortion.
14
R1
100kΩ
+15V
R3
51kΩ
C2
51pF
7
3
R2
100kΩ
OUTPUT
f = 100kHz
2% THD AT 1.1VP-P
+
CA3160
2
0.1
µF
6
4
2kΩ
C1
10-80
pF
2-1N914
0.01µF
680Ω
f=
1
2 π √(R1 || R2) C1 R3 C2
500Ω
FIGURE 13. SINGLE SUPPLY WEIN BRIDGE OSCILLATOR
Operation with Output Stage Power Booster
The current sourcing and sinking capability of the CA3160
output stage is easily supplemented to provide power-boost
capability. In the circuit of Figure 14, three CMOS transistorpairs in a single CA3600 lC array are shown parallel-connected
with the output stage in the CA3160. In the Class A mode of
CA3600E shown, a typical device consumes 20mA of supply
current at 15V operation. This arrangement boosts the currenthandling capability of the CA3160 output stage by about 2.5X.
The amplifier circuit in Figure 14 employs feedback to
establish a closed-loop gain of 20dB. The typical largesignal-bandwidth (-3dB) is 190kHz.
CA3160, CA3160A
+15V
2
14
0.01µF
11
1MΩ
-
1µF
CA3600
+
QP1
QP2
QP3
7
680kΩ
3
+
CA3160
INPUT
2
1µF
6
13
1
3
10
-
2kΩ
500µF
8
6
12
50Ω
100mW
AT 10%
THD
4
8
A = 20dB
LARGE SIGNAL
BW (-3dB) = 190kHz
5
QN1
QN2
7
QN3
4
9
20kΩ
FIGURE 14. CMOS TRANSISTOR ARRAY (CA3600E) CONNECTED AS POWER BOOSTER IN THE OUTPUT STAGE OF THE CA3160
VS = ±7.5V
TA = 25oC
100
0
50
100
φ OL
80
150
200
60
40
CL = 30pF
RL = 2kΩ
20
0
101
102
103
104
105
106
FREQUENCY (Hz)
107
108
FIGURE 15. OPEN LOOP VOLTAGE GAIN AND PHASE SHIFT
vs FREQUENCY
15
150
RL = 2kΩ
OPEN LOOP VOLTAGE GAIN (dB)
OPEN LOOP VOLTAGE GAIN (dB)
120
OPEN LOOP PHASE (DEGREES)
Typical Performance Curves
140
130
120
110
100
90
80
-100
-50
0
50
100
TEMPERATURE (oC)
FIGURE 16. OPEN LOOP GAIN vs TEMPERATURE
CA3160, CA3160A
(Continued)
15.0
17.5
15
QUIESCENT SUPPLY CURRENT (mA)
V+ = 15V, V- = 0V
TA = 25oC
RL = 5kΩ
2kΩ
1kΩ
500Ω
12.5
10
7.5
5
2.5
10.0
5.0
2.5
5
7.5
10 12.5
15 17.5
20
GATE VOLTAGE [TERMINALS 4 AND 8] (V)
2.5
14
TA = -55oC
25oC
125oC
6
4
2
0
0
2
4
6
8
10
12
14
POSITIVE SUPPLY VOLTAGE (V)
16
10
12
14
18
FIGURE 19. QUIESCENT SUPPLY CURRENT vs SUPPLY
VOLTAGE
50
10
V- = 0V
TA = 25oC
18
V+ = 15V
10V
5V
1
0.1
0.01
0.001
0.001
0.01
0.1
1
10
MAGNITUDE OF LOAD CURRENT (mA)
100
FIGURE 20. VOLTAGE ACROSS PMOS OUTPUT TRANSISTOR
(Q8) vs LOAD CURRENT
1000
50
V- = 0V
TA = 25oC
16
FIGURE 18. QUIESCENT SUPPLY CURRENT vs SUPPLY
VOLTAGE
10
8
8
POSITIVE SUPPLY VOLTAGE (V)
VO = V+ / 2
V- = 0
12
6
22.5
FIGURE 17. VOLTAGE TRANSFER CHARACTERISTICS OF
CMOS OUTPUT STAGE
QUIESCENT SUPPLY CURRENT (mA)
HIGH VO = V+
OR LOW VO = V-
0
0
TA = 25oC
VS = ±7.5V
V+ = 15V
10V
5V
100
1
EN (nV/√Hz)
VOLTAGE DROP ACROSS NMOS OUTPUT STAGE
TRANSISTOR (Q12) (V)
BALANCED
VO = V+/2
7.5
0
10
TA = 25oC
RL = ∞
V- = 0
12.5
VOLTAGE DROP ACROSS PMOS OUTPUT STAGE
TRANSISTOR (Q8) (V)
OUTPUT VOLTAGE [TERMS. 4 AND 6] (V)
Typical Performance Curves
0.1
10
0.01
0.001
0.001
1
0.01
0.1
1
10
100
MAGNITUDE OF LOAD CURRENT (mA)
FIGURE 21. VOLTAGE ACROSS NMOS OUTPUT TRANSISTOR
(Q12) vs LOAD CURRENT
16
1
101
102
103
FREQUENCY (Hz)
104
105
FIGURE 22. EQUIVALENT NOISE VOLTAGE vs FREQUENCY
CA3160, CA3160A
Typical Performance Curves
10
(Continued)
4000
TA = 25oC
VS = ±7.5V
1000
5
INPUT CURRENT (pA)
INPUT VOLTAGE (V)
7.5
V+ 15V
TO
5V
7
2
CA3160
PA
6
3
2.5
0V
TO
V- -10V
0
-1
0
1
2
3
4
5
6
INPUT CURRENT (pA)
10
8
4
VIN
100
1
-80 -60 -40 -20
7
FIGURE 23. INPUT CURRENT vs COMMON MODE VOLTAGE
0
20 40 60 80
TEMPERATURE (oC)
100 120 140
FIGURE 24. INPUT CURRENT vs TEMPERATURE
OFFSET VOLTAGE SHIFT (mV)
7
DIFFERENTIAL DC VOLTAGE
(ACROSS TERMINALS 2 AND 3) = 2V
OUTPUT STAGE TOGGLED
6
5
TA = 125oC FOR
METAL CAN PACKAGES
4
3
2
DIFFERENTIAL DC VOLTAGE
(ACROSS TERMINALS 2 AND 3) = 0V
OUTPUT VOLTAGE = V+ / 2
1
0
0
500
1000 1500 2000 2500 3000 3500 4000
TIME (HOURS)
FIGURE 25. TYPICAL INCREMENTAL OFFSET VOLTAGE SHIFT vs OPERATING LIFE
All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification.
Intersil semiconductor products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see web site www.intersil.com
Sales Office Headquarters
NORTH AMERICA
Intersil Corporation
P. O. Box 883, Mail Stop 53-204
Melbourne, FL 32902
TEL: (321) 724-7000
FAX: (321) 724-7240
17
EUROPE
Intersil SA
Mercure Center
100, Rue de la Fusee
1130 Brussels, Belgium
TEL: (32) 2.724.2111
FAX: (32) 2.724.22.05
ASIA
Intersil (Taiwan) Ltd.
7F-6, No. 101 Fu Hsing North Road
Taipei, Taiwan
Republic of China
TEL: (886) 2 2716 9310
FAX: (886) 2 2715 3029