ON MC33067P High performance zero voltage switch resonant mode controller Datasheet

Order this document by MC34067/D
The MC34067/MC33067 are high performance zero voltage switch
resonant mode controllers designed for off–line and dc–to–dc converter
applications that utilize frequency modulated constant off–time or constant
deadtime control. These integrated circuits feature a variable frequency
oscillator, a precise retriggerable one–shot timer, temperature compensated
reference, high gain wide bandwidth error amplifier, steering flip–flop, and
dual high current totem pole outputs ideally suited for driving power
MOSFETs.
Also included are protective features consisting of a high speed fault
comparator, programmable soft–start circuitry, input undervoltage lockout
with selectable thresholds, and reference undervoltage lockout.
These devices are available in dual–in–line and surface mount packages.
• Zero Voltage Switch Resonant Mode Operation
•
•
•
•
•
•
•
•
•
Variable Frequency Oscillator with a Control Range Exceeding 1000:1
Precision One–Shot Timer for Controlled Off–Time
HIGH PERFORMANCE
ZERO VOLTAGE SWITCH
RESONANT MODE
CONTROLLERS
SEMICONDUCTOR
TECHNICAL DATA
P SUFFIX
PLASTIC PACKAGE
CASE 648
16
Internally Trimmed Bandgap Reference
1
4.0 MHz Error Amplifier
Dual High Current Totem Pole Outputs
Selectable Undervoltage Lockout Thresholds with Hysteresis
Enable Input
DW SUFFIX
PLASTIC PACKAGE
CASE 751G
(SO–16L)
Programmable Soft–Start Circuitry
Low Startup Current for Off–Line Operation
16
1
PIN CONNECTIONS
16 One–Shot RC
Osc Charge 1
15 VCC
Osc RC 2
Osc Control Current 3
Simplified Block Diagram
VCC
15
Enable / 9
UVLO Adjust
1
Osc Charge
2
Osc RC
Oscillator 3
Control Current
One–Shot
11
5.0 V
Reference
Vref
Vref UVLO
Variable
Frequency
Oscillator
Steering
Flip–Flop
One–Shot
Gnd 4
13 Power Gnd
Vref 5
12 Drive Output B
Error Amp Out 6
11 CSoft–Start
Inverting Input 7
10 Fault Input
Enable/UVLO
9
Adjust
Noninverting Input 8
14
16
Error Amp 6
Output
Noninverting 8
Input
Inverting Input 7
Soft–Start
VCC UVLO /
Enable
5
14 Drive Output A
(Top View)
Output A
12
Output B
2.5 V
Clamp
13
Pwr Gnd
ORDERING INFORMATION
Error
Amp
10
Soft–Start
Fault Detector
Fault Input
Device
Operating
Temperature Range
MC34067DW
4
Ground
MC34067P
MC33067DW
MC33067P
TA = 0 to + 70°C
TA = – 40° to + 85°C
 Motorola, Inc. 1999
MOTOROLA ANALOG IC DEVICE DATA
Package
SO–16L
Plastic DIP
SO–16L
Plastic DIP
Rev 1, 05/99
1
MC34067 MC33067
MAXIMUM RATINGS
Rating
Symbol
Value
Unit
VCC
20
V
Drive Output Current, Source or Sink (Note 1)
Continuous
Pulsed (0.5 µs, 25% Duty Cycle
IO
0.3
1.5
Error Amplifier, Fault, One–Shot, Oscillator and
Soft–Start Inputs
Vin
– 1.0 to + 6.0
V
Vin(UVLO)
– 1.0 to VCC
V
PD
RθJA
862
145
mW
°C/W
PD
RθJA
1.25
100
W
°C/W
Operating Junction Temperature
TJ
+ 150
°C
Operating Ambient Temperature
MC34067
MC33067
TA
Power Supply Voltage
UVLO Adjust Input
A
Power Dissipation and Thermal Characteristics
DW Suffix, Plastic Package, Case 751G
TA = 25°C
Thermal Resistance, Junction–to–Air
P Suffix, Plastic Package, Case 648
TA = 25°C
Thermal Resistance, Junction–to–Air
°C
0 to + 70
– 40 to + 85
Storage Temperature
Tstg
°C
– 55 to + 150
ELECTRICAL CHARACTERISTICS (VCC = 12 V [Note 2], ROSC= 18.2 k, RVFO = 2940, COSC = 300 pF, RT = 2370 k, CT = 300 pF,
CL = 1.0 nF. For typical values TA = 25°C, for min/max values TA is the operating ambient temperature range that applies [Note 3], unless
otherwise noted.)
Characteristic
Symbol
Min
Typ
Max
Unit
Vref
5.0
5.1
5.2
V
Line Regulation (VCC = 10 TO 18 V)
Regline
–
1.0
20
mV
Load Regulation (IO = 0 mA to 10 mA)
Regload
–
1.0
20
mV
Vref
4.9
–
5.3
V
REFERENCE SECTION
Reference Output Voltage (IO = 0 mA, TJ = 25°C)
Total Output Variation Over Line, Load, and Temperature
Output Short Circuit Current
IO
25
100
190
mA
Reference Undervoltage Lockout Threshold
Vth
3.8
4.3
4.8
V
VIO
–
1.0
10
mV
Input Bias Current (VCM = 1.5 V)
IIB
–
0.2
1.0
µA
Input Offset Current (VCM = 1.5 V)
IIO
–
0
0.5
µA
Open Loop Voltage Gain (VCM = 1.5 V, VO = 2.0 V)
AVOL
70
100
–
dB
Gain Bandwidth Product (f = 100 kHz)
GBW
3.0
5.0
–
MHz
Input Common Mode Rejection Ratio (VCM = 1.5 to 5.0 V)
CMR
70
95
–
dB
Power Supply Rejection Ratio (VCC = 10 to 18 V, f = 120 Hz)
PSR
80
100
–
dB
Output Voltage Swing
High State
Low State
VOH
VOL
2.8
–
3.2
0.6
–
0.8
ERROR AMPLIFIER
Input Offset Voltage (VCM = 1.5 V)
V
NOTES: 1. Maximum package power dissipation limits must be observed.
2. Adjust VCC above the Startup threshold before setting to 12 V.
3. Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.
Tlow = 0°C for the MC34067
Thigh = + 70°C for MC34067
= – 40°C for the MC33067
Thigh = + 85°C for MC33067
2
MOTOROLA ANALOG IC DEVICE DATA
MC34067 MC33067
ELECTRICAL CHARACTERISTICS (VCC = 12 V [Note 2], ROSC= 18.2 k, RVFO = 2940, COSC = 300 pF, RT = 2370 k, CT = 300 pF,
CL = 1.0 nF. For typical values TA = 25°C, for min/max values TA is the operating ambient temperature range that applies [Note 3], unless
otherwise noted.)
Characteristic
Symbol
Min
Typ
Max
Unit
Frequency (Error Amp Output Low)
TA = 25°C
Total Variation (VCC = 10 to 18 V, TA = TLow to THigh
fOSC(low)
500
490
525
–
540
550
Frequency (Error Amp Output High)
TA = 25°C
Total Variation (VCC = 10 to 18 V, TA = TLow to THigh
fOSC(high)
1900
1850
2050
–
2150
2200
Vin
–
2.5
–
tBlank
235
225
250
–
270
280
VOL
–
–
9.5
9.0
0.8
1.5
10.3
9.7
1.2
2.0
–
–
VOL(UVLO)
–
0.8
1.2
V
Output Voltage Rise Time (CL = 1.0 nF)
tr
–
20
50
ns
Output Voltage Fall Time (CL = 1.0 nF)
tf
–
15
50
ns
Input Threshold
Vth
0.93
1.0
1.07
V
Input Bias Current (VPin 10 = 0 V)
IIB
–
– 2.0
– 10
µA
tPLH(In/Out)
–
60
100
ns
Ichg
4.5
9.0
14
µA
Idischg
3.0
8.0
–
mA
Startup Threshold, VCC Increasing
Enable/UVLO Adjust Pin Open
Enable/UVLO Adjust Pin Connected to VCC
Vth(UVLO)
14.8
8.0
16
9.0
17.2
10
Minimum Operating Voltage After Turn–On
Enable/UVLO Adjust Pin Open
Enable/UVLO Adjust Pin Connected to VCC
VCC(min)
8.0
7.6
9.0
8.6
10
9.6
Enable/UVLO Adjust Shutdown Threshold Voltage
Vth(Enable)
6.0
7.0
–
Enable/UVLO Adjust Input Current (Pin 9 = 0 V)
Iin(Enable)
–
– 0.2
– 1.0
ICC
–
–
0.5
27
0.8
35
OSCILLATOR
Oscillator Control Input Voltage, Pin 3 @ 25°C
kHz
kHz
V
ONE–SHOT
Drive Output Off–Time
TA = 25°C
Total Variation (VCC = 10 to 18 V, TA = TLow to THigh
ns
DRIVE OUTPUTS
Output Voltage
Low State (ISink = 20 mA)
Low State (ISink = 200 mA)
High State (ISource = 20 mA)
High State (ISource = 200 mA)
Output Voltage with UVLO Activated (VCC = 6.0 V, ISink = 1.0 mA)
V
VOH
FAULT COMPARATOR
Propagation Delay to Drive Outputs (100 mV Overdrive)
SOFT–START
Capacitor Charge Current (VPin 11 = 2.5 V)
Capacitor Discharge Current (VPin 11 = 2.5 V)
UNDERVOLTAGE LOCKOUT
V
V
V
mA
TOTAL DEVICE
Power Supply Current (Enable/UVLO Adjust Pin Open)
Startup (VCC = 13.5 V)
Operating (fOSC = 500 kHz) (Note 2)
mA
NOTES: 1. Maximum package power dissipation limits must be observed.
2. Adjust VCC above the Startup threshold before setting to 12 V.
3. Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.
Tlow = 0°C for the MC34067
Thigh = + 70°C for MC34067
= – 40°C for the MC33067
Thigh = + 85°C for MC33067
MOTOROLA ANALOG IC DEVICE DATA
3
MC34067 MC33067
500
3500
COSC = 300 pF
COSC = 200 pF
400
COSC = 500 pF
300
VCC = 12 V
RVFO = ∞
RT = ∞
CT = 500 pF
TA = 25°C
200
100
0
Figure 2. Oscillator Frequency versus
Oscillator Control Current
f OSC , OSCILLATOR FREQUENCY (kHz)
ROSC, OSCILLATOR TIMING RESISTOR (kΩ )
Figure 1. Oscillator Timing Resistor
versus Discharge Time
Oscillator Discharge Time is Measured at the Drive Outputs.
0
20
40
60
80
tdischg, OSCILLATOR DISCHARGE TIME (µs)
VCC = 12 V
TA = 25°C
ROSC = 18.2 k
3000
2500
2000
COSC = 300 pF
1500
1000
100
500
0
0
400
800
1200
1600
IOSC, OSCILLATOR CONTROL CURRENT (mA)
Figure 4. One–Shot Timing Resistor
versus Period
60
0.35
RT, TIMING RESISTOR (k Ω )
0.30
0.25
0.20
0.15
0.10
0.05
0
0.5
1.0
1.5
2.0
2.5
IOSC, OSCILLATOR CONTROL CURRENT (mA)
3.0
50
40
VCC = 12 V
VO = 2.0 V
RL = 100 k
TA = 25°C
Gain
30
70
20
80
10
90
0
–10
– 20
10 k
Phase
Phase
Margin
= 64°
100 k
1.0M
f, FREQUENCY (Hz)
100
110
120
10M
∇
4
60
0, EXCESS PHASE (DEGREES)
A VOL, OPEN LOOP VOLTAGE GAIN (dB)
Figure 5. Open Loop Voltage Gain and Phase
versus Frequency
50
VCC = 12 V
COSC = 500 pF
ROSC = 100 k
TA = 25°C
30
CT = 200 pF
20
CT = 300 pF
CT = 500 pF
10
400
One–Shot Period is Measured
at the Drive Outputs.
3.0
0.1
V ref , REFERENCE OUTPUT VOLTAGE CHANGE (mV)
Vsat, OUTPUT SATURATION VOLTAGE (V)
Figure 3. Error Amp Output Saturation
Voltage versus Oscillator Control Current
2000
0.3
0.6
1.0
3.0
tOS, ONE–SHOT PERIOD (µs)
6.0
10
Figure 6. Reference Output Voltage Change
versus Temperature
*Vref = 5.0 V
0
– 10
– 20
*Vref = 5.0 V
VCC = 12 V
RL = ∞
*Vref at TA = 25°C
– 30
– 40
– 50
– 55
*Vref = 5.0 V
– 25
0
25
50
75
TA, AMBIENT TEMPERATURE (°C)
100
125
MOTOROLA ANALOG IC DEVICE DATA
Figure 7. Reference Voltage Change
versus Source Current
Figure 8. Drive Output Saturation Voltage
versus Load Current
0
V sat , OUTPUT SATURATION VOLTAGE (V)
V ref , REFERENCE OUTPUT VOLTAGE CHANGE (mV)
MC34067 MC33067
TA = – 40°C
–10
TA = – 20°C
– 20
TA = –125°C
– 30
– 40
– 50
VCC = 12 V
0
20
∇
40
60
80
Iref, REFERENCE SOURCE CURRENT (mA)
100
0
TA = 25°C
– 2.0
3.0
TA = – 40°C
2.0
TA = 25°C
Source Saturation
(Load to VCC)
1.0
0
0
V OL , SOFT–START SATURATION VOLTAGE (V)
CL = 1.0 nF
TA = 25 °C
1.6
0.8
0
0
VCC = 12 V
CL = 1.0 nF
TA = 25 °C
1200
800
400
50
60
70
ICC, INPUT SUPPLY CURRENT (mA)
MOTOROLA ANALOG IC DEVICE DATA
80
2.0
4.0
6.0
8.0
Idchg, CAPACITOR DISCHARGE CURRENT (mA)
10
Figure 12. Supply Current versus Supply Voltage
I CC, SUPPLY CURRENT (mA)
f, FREQUENCY (kHz)
VCC = 12 V
Pin 10 = Vref
TA = 25 °C
24
2000
40
1.0
2.4
Figure 11. Operating Frequency
versus Supply Current
30
0.4
0.6
0.8
IO, OUTPUT LOAD CURRENT (A)
3.2
20 ns/DIV
0
0.2
Gnd
Figure 10. Soft–Start Saturation Voltage
versus Capacitor Discharge Current
10%
1600
VCC = 12 V
80 µs Pulsed Load
120 Hz Rate
TA = – 40°C
– 3.0
Figure 9. Drive Output Waveform
90%
Source Saturation
(Load to Ground)
VCC
–1.0
90
TA = 25 °C
20
Enable/UVLO
Adjust Pin
Open
16
12
8.0
Enable/UVLO
Adjust Pin
to VCC
4.0
0
0
4.0
8.0
12
VCC, SUPPLY VOLTAGE (V)
16
20
5
MC34067 MC33067
Figure 13. MC34067 Representative Block Diagram
VCC
15
50k
Enable /
UVLO Adjust
7.0k
7.0k
9
50k
5.1V
Reference
VCC UVLO
8.0V
Q1
4.2/4.0V
Q2
2
IOSC
4.9V/3.6V
One–Shot RC
CT
Oscillator 16
Control Current
RT
IOSO
14
13
One–Shot
3.1V
3
RVFO
Error Amp Output 6
8
Noninverting Input
Inverting Input
7
Soft–Start
Output A
Steering
Flip–Flop
Q
T
RQ
Oscillator
OSC RC
COSC
Vref
D1
1
ROSC
5
Vref UVLO
Vref
OSC Charge
Vref
Power Ground
Output B
12
4.9V/3.6V
Fault Comparator
Error Amp
Clamp
Fault Input
10
1.0V
9.0µA
Error Amp
11
4
Ground
Timing Diagram
5.1 V
COSC
3. 6 V
One–Shot
5.1 V
3.6 V
Output A
Output B
tOS
tOS
tOS
Error Amp output high, minimum IOSC current
occurring at minimum input voltage, maximum load.
6
tOS
tOS
tOS
Error Amp output low, maximum IOSC current
occurring at maximum input voltage, minimum load.
MOTOROLA ANALOG IC DEVICE DATA
MC34067 MC33067
OPERATING DESCRIPTION
Introduction
As power supply designers have strived to increase power
conversion efficiency and reduce passive component size,
high frequency resonant mode power converters have
emerged as attractive alternatives to conventional
pulse–width modulated control. When compared to
pulse–width modulated converters, resonant mode control
offers several benefits including lower switching losses,
higher efficiency, lower EMI emission, and smaller size. A
new integrated circuit has been developed to support this
trend in power supply design. The MC34067 Resonant Mode
Controller is a high performance bipolar IC dedicated to
variable frequency power control at frequencies exceeding
1.0 MHz. This integrated circuit provides the features and
performance specifically for zero voltage switching resonant
mode power supply applications.
The primary purpose of the control chip is to provide a
fixed off–time to the gates of external power MOSFETs at a
repetition rate regulated by a feedback control loop.
Additional features of the IC ensure that system startup and
fault conditions are administered in a safe, controlled manner.
A simplified block diagram of the IC is shown on the front
page, which identifies the main functional blocks and the
block–to–block interconnects. Figure 13 is a detailed
functional diagram which accurately represents the internal
circuitry. The various functions can be divided into two
sections. The first section includes the primary control path
which produces precise output pulses at the desired
frequency. Included in this section are a variable frequency
Oscillator, a One–Shot, a pulse Steering Flip–Flop, a pair of
power MOSFET Drivers, and a wide bandwidth Error
Amplifier. The second section provides several peripheral
support functions including a voltage reference, undervoltage
lockout, Soft–Start circuit, and a fault detector.
Primary Control Path
The output pulse width and repetition rate are regulated
through the interaction of the variable frequency Oscillator,
One–Shot timer and Error Amplifier. The Oscillator triggers
the One–Shot which generates a pulse that is alternately
steered to a pair of totem pole output drivers by a toggle
Flip–Flop. The Error Amplifier monitors the output of the
regulator and modulates the frequency of the Oscillator. High
speed Schottky logic is used throughout the primary
control channel to minimize delays and enhance high
frequency characteristics.
Oscillator
The characteristics of the variable frequency Oscillator are
crucial for precise controller performance at high operating
frequencies. In addition to triggering the One–Shot timer and
initiating the output deadtime, the oscillator also determines
the initial voltage for the one–shot capacitor. The Oscillator is
designed to operate at frequencies exceeding 1.0 MHz. The
Error Amplifier can control the oscillator frequency over a
1000:1 frequency range, and both the minimum and
maximum frequencies are easily and accurately
programmed by the proper selection of external components.
MOTOROLA ANALOG IC DEVICE DATA
The functional diagram of the Oscillator and One–Shot
timer is shown in Figure 14. The oscillator capacitor (COSC) is
initially charged by transistor Q1. When COSC exceeds the
4.9 V upper threshold of the oscillator comparator, the base
of Q1 is pulled low allowing COSC to discharge through the
external resistor, (ROSC), and the oscillator control current,
(IOSC). When the voltage on COSC falls below the
comparator’s 3.6 V lower threshold, Q1 turns on and again
charges COSC.
COSC charges from 3.6 V to 5.1 V in less than 50 ns. The
high slew rate of COSC and the propagation delay of the
comparator make it difficult to control the peak voltage. This
accuracy issue is overcome by clamping the base of Q1
through a diode to a voltage reference. The peak voltage of
the oscillator waveform is thereby precisely set at 5.1 V.
Figure 14. Oscillator and One–Shot Timer
+
OSC Charge
VOE
Vref
+ VOE
Q1
D1
1
OSC RC
ROSC
COSC
2
Oscillator
IOSC
4.9V/3.6V
One–Shot RC
CT
RT
Oscillator 10
Control Current
IOSO
3
RVFO
6
Error Amp Output
One–Shot
4.9V/3.6V
3.1V
Error Amp
Charge
The frequency of the Oscillator is modulated by varying
the current flowing out of the Oscillator Control Current (IOSC)
pin. The IOSC pin is the output of a voltage regulator. The
input of the voltage regulator is tied to the variable frequency
oscillator. The discharge current of the Oscillator increases
by increasing the current out of the IOSC pin. Resistor RVFO is
used in conjunction with the Error Amp output to change the
IOSC current. Maximum frequency occurs when the Error
Amplifier output is at its low state with a saturation voltage of
0.1 V at 1.0 mA.
The minimum oscillator frequency will result when the
IOSC current is zero, and COSC is discharged through the
external resistor (ROSC). This occurs when the Error
Amplifier output is at its high state of 2.5 V. The minimum and
maximum oscillator frequencies are programmed by the
proper selection of resistor ROSC and RVFO. The minimum
frequency is programmed by ROSC using Equation 1:
1
– t PD
t (max) – 70 ns
ƒ(min)
R OSC =
=
0.348 C OSC
C OSC ȏ n ǒ 5.1 Ǔ
3.6
where tPD is the internal propagation delay.
(1)
7
MC34067 MC33067
The maximum oscillator frequency is set by the current
through resistor RVFO. The current required to discharge
COSC at the maximum oscillator frequency can be calculated
by Equation 2:
I (max) = C OSC
5.1 – 3.6
1
ƒ(max)
= 1.5COSC ƒ(max)
5.1 – 3.6
=
IR
ε
OSC
ROSC
=
1.5
R OSC
ε
ǒ–
Oscillator
Control Current
(2)
1
ƒ (min)
R OSC COSC
3.1V
3
IOSC
The discharge current through ROSC must also be known
and can be calculated by Equation 3:
ǒ–
Figure 15. Error Amplifier and Clamp
RVFO
6
Error Amp
Charge
Error Amp Output
8
Noninverting Input
Ǔ
Inverting Input
Ǔ
(3)
7
Error
Amp
1
ƒ (min) R OSC COSC
Resistor RVFO can now be calculated by Equation 4:
2.5 – V EAsat
RVFO =
(4)
I(max) – I R
OSC
One–Shot Timer
The One–Shot is designed to disable both outputs
simultaneously providing a deadtime before either output is
enabled. The One–Shot capacitor (C T ) is charged
concurrently with the oscillator capacitor by transistor Q1, as
shown in Figure 14. The one–shot period begins when the
oscillator comparator turns off Q1, allowing CT to discharge.
The period ends when resistor RT discharges CT to the
threshold of the One–Shot comparator. The lower threshold
of the One–Shot is 3.6 V. By choosing CT, RT can by solved
by Equation 5:
t OS
t OS
RT =
=
(5)
0.348 C T
C T ȏ n ǒ 5.1 Ǔ
3.6
When the Error Amplifier output is coupled to the IOSC pin
by RVFO, as illustrated in Figure 15, it provides the Oscillator
Control Current, IOSC. The output swing of the Error Amplifier
is restricted by a clamp circuit to improve its transient
recovery time.
Output Section
The pulse(tOS), generated by the Oscillator and One–Shot
timer is gated to dual totem–pole output drives by the
Steering Flip–Flop shown in Figure 16. Positive transitions of
tOS toggle the Flip–Flop, which causes the pulses to alternate
between Output A and Output B. The flip–flop is reset by the
undervoltage lockout circuit during startup to guarantee that
the first pulse appears at Output A.
Figure 16. Steering Flip–Flop and Output Drivers
VOE
±
Output A
Errors in the threshold voltage and propagation delays
through the output drivers will affect the One–Shot period. To
guarantee accuracy, the output pulse of the control chip is
trimmed to within 5% of 250 ns with nominal values of RT and
CT.
The outputs of the Oscillator and One–Shot comparators
are OR’d together to produce the pulse tOS, which drives the
Flip–Flop and output drivers. The output pulse (tOS) is
initiated by the Oscillator and terminated by the One–Shot
comparator. With zero–voltage resonant mode converters,
the oscillator discharge time should never be set less than
the one–shot period.
Error Amplifier
A fully accessible high performance Error Amplifier is
provided for feedback control of the power supply system.
The Error Amplifier is internally compensated and features dc
open loop gain greater than 70 dB, input offset voltage of less
than 10 mV and a guaranteed minimum gain–bandwidth
product of 2.5 MHz. The input common mode range extends
from 1.5 V to 5.1 V, which includes the reference voltage.
8
Steering
Flip–Flop
Q
T
14
Power Ground
Pwr
Gnd
RQ
13
VOE
±
Output B
12
Pwr
Gnd
The totem–pole output drivers are ideally suited for driving
power MOSFETs and are capable of sourcing and sinking
1.5 A. Rise and fall times are typically 20 ns when driving a
1.0 nF load. High source/sink capability in a totem–pole
driver normally increases the risk of high cross conduction
current during output transitions. The MC34067 utilizes a
unique design that virtually eliminates cross conduction, thus
controlling the chip power dissipation at high frequencies. A
separate power ground pin is provided to isolate the sensitive
analog circuitry from large transient currents.
MOTOROLA ANALOG IC DEVICE DATA
MC34067 MC33067
Figure 17. Undervoltage Lockout and Reference
VCC
15
50k
Enable /
UVLO Adjust
7.0k
Vref
7.0k
5.1V
Reference
9
50k
8.0V
Vref
5
Vref UVLO
VCC UVLO
4.2/4.0V
UVLO
PERIPHERAL SUPPORT FUNCTIONS
The MC34067 Resonant Controller provides a number of
support and protection functions including a precision voltage
reference, undervoltage lockout comparators, soft–start
circuitry, and a fault detector. These peripheral circuits
ensure that the power supply can be turned on and off in a
controlled manner and that the system will be quickly
disabled when a fault condition occurs.
Undervoltage Lockout and Voltage Reference
Separate undervoltage lockout comparators sense the
input VCC voltage and the regulated reference voltage as
illustrated in Figure 17. When VCC increases to the upper
threshold voltage, the VCC UVLO comparator enables the
Reference Regulator. After the Vref output of the Reference
Regulator rises to 4.2 V, the Vref UVLO comparator switches
the UVLO signal to a logic zero state enabling the primary
control path. Reducing VCC to the lower threshold voltage
causes the VCC UVLO comparator to disable the Reference
Regulator. The Vref UVLO comparator then switches the
UVLO output to a logic one state disabling the controller.
The Enable/UVLO Adjust pin allows the power supply
designer to select the VCC UVLO threshold voltages. When
this pin is open, the comparator switches the controller on at
16 V and off at 9.0 V. If this pin is connected to the VCC
terminal, the upper and lower thresholds are reduced to
9.0 V and 8.6 V, respectively. Forcing the Enable/UVLO
Adjust pin low will pull the VCC UVLO comparator input low
(through an internal diode) turning off the controller.
The Reference Regulator provides a precise 5.1 V
reference to internal circuitry and can deliver up to 10 mA to
external loads. The reference is trimmed to better than 2%
initial accuracy and includes active short circuit protection.
Fault Detector
The high speed Fault Comparator illustrated in Figure 18
can protect a power supply from destruction under fault
conditions. The Fault Input pin connects to the input of the
Fault Comparator. The Fault Comparator output connects to
the output drivers. This direct path reduces the propagation
MOTOROLA ANALOG IC DEVICE DATA
delay from the Fault Input to the A and B outputs to typically
70 ns. The Fault Comparator output is also OR’d with the
UVLO output from the Vref UVLO comparator to produce the
logic output labeled “UVLO+Fault”. This signal disables the
Oscillator and One–Shot by forcing both the COSC and CT
capacitors to be continually charged.
Figure 18. Fault Detector and Soft–Start
UVLO
UVLO + Fault
Fault
Fault
Comparator Input
10
9.0µA
1.0V
Ea Clamp
CSoft–Start
Soft–Start
Buffer
11
6
Ground
Soft–Start Circuit
The Soft–Start circuit shown in Figure 18 forces the
variable frequency Oscillator to start at the maximum
frequency and ramp downward until regulated by the
feedback control loop. The external capacitor at the
CSoft–Start terminal is initially discharged by the UVLO+Fault
signal. The low voltage on the capacitor passes through the
Soft–Start Buffer to hold the Error Amplifier output low. After
UVLO+Fault switches to a logic zero, the soft–start
capacitor is charged by a 9.0 µA current source. The buffer
allows the Error Amplifier output to follow the soft–start
capacitor until it is regulated by the Error Amplifier inputs. The
soft–start function is generally applicable to controllers
operating below resonance and can be disabled by simply
opening the CSoft–Start terminal.
9
MC34067 MC33067
APPLICATIONS INFORMATION
The MC34067 is specifically designed for zero voltage
switching (ZVS) quasi–resonant converter (QRC)
applications. The IC is optimized for double–ended push–pull
or bridge type converters operating in continuous conduction
mode. Operation of this type of ZVS with resonant properties
is similar to standard push–pull or bridge circuits in that the
energy is transferred during the transistor on–time. The
difference is that a series resonant tank is usually introduced
to shape the voltage across the power transistor prior to
turn–on. The resonant tank in this topology is not used to
deliver energy to the output as is the case with zero current
switch topologies. When the power transistor is enabled the
voltage across it should already be zero, yielding minimal
switching loss. Figure 19 shows a timing diagram for a
half–bridge ZVS QRC. An application circuit is shown in
Figure 20. The circuit built is a dc to dc half–bridge converter
delivering 75 W to the output from a 48 V source.
When building a zero voltage switch (ZVS) circuit, the
objective is to waveshape the power transistor’s voltage
waveform so that the voltage across the transistor is zero
when the device is turned on. The purpose of the control IC is
to allow a resonant tank to waveshape the voltage across the
power transistor while still maintaining regulation. This is
accomplished by maintaining a fixed deadtime and by
varying the frequency; thus the effective duty cycle is
changed.
Primary side resonance can be used with ZVS circuits. In
the application circuit, the elements that make the resonant
tank are the primary leakage inductance of the transformer
(LL) and the average output capacitance (COSS) of a power
MOSFET (CR). The desired resonant frequency for the
application circuit is calculated by Equation 6:
ƒr =
10
1
2π
L L 2C R
In the application circuit, the operating voltage is low and
the value of COSS versus Drain Voltage is known. Because
the COSS of a MOSFET changes with drain voltage, the value
of the CR is approximated as the average COSS of the
MOSFET. For the application circuit the average COSS can be
calculated by Equation 7:
CR =
2 * C OSS measured at
1
V
2 in
(7)
The MOSFET chosen fixes CR and that LL is adjusted to
achieve the desired resonant frequency.
However, the desired resonant frequency is less critical
than the leakage inductance. Figure 19 shows the primary
current ramping toward its peak value during the resonant
transition. During this time, there is circulating current
flowing through the secondary inductance, which effectively
makes the primary inductance appear shorted. Therefore,
the current through the primary will ramp to its peak value at
a rate controlled by the leakage inductance and the applied
voltage. Energy is not transferred to the secondary during
this stage, because the primary current has not overcome the
circulating current in the secondary. The larger the leakage
inductance, the longer it takes for the primary current to slew.
The practical effect of this is to lower the duty cycle, thus
reducing the operating range.
The maximum duty cycle is controlled by the leakage
inductance, not by the MC34067. The One–Shot in the
MC34067 only assures that the power switch is turned on
under a zero voltage condition. Adjust the one–shot period so
that the output switch is activated while the primary current is
slewing but before the current changes polarity. The resonant
stage should then be designed to be as long as the time for
the primary current to go to zero amps.
(6)
MOTOROLA ANALOG IC DEVICE DATA
MC34067 MC33067
Figure 19. Application Timing Diagram
5.1 V
COSC
3.6 V
One–Shot
5.1 V
3.6 V
Output A
Output B
Vin
1/2 Vin
0V
+ Iprimary
0A
– Iprimary
Vin/Turns Ratio
Output Diode
Voltage
MOTOROLA ANALOG IC DEVICE DATA
11
12
VFB
2.7k
18k
6
4.0 mV = ±0.039%
25 mVp–p
83.5%
84.2%
V in = 48 V, I O = 15 A, fswitch = 1.0 MHz
V in = 48 V, I O = 10 A, fswitch = 1.7 MHz
V in = 48 V, I O = 15 A, fswitch = 1.0 MHz
Output Ripple
Efficiency
20 mV = ±0.198%
Results
V in = 48 V, IO = 10 A to 15 A
Conditions
4
V in = 40 V to 56 V, IO =15 A
11
7
8
220pF
Reference
Regulator
Load Regulation
Test
0.01
1500pF
1.1k
10k
3
10
2
1
9
15
Line Regulation
16k
1.6k 330pF
100pF
330pF
10
VCC
Figure 20. Application Circuit
1N5819
470
T2
3.9k
1.0k
1.0k
MTP33N10E
100
1.0
1.0
100
T3
1N5819 x 4
CTL
MBR2535
500pF 51, 0.5W
30
L1 =
1.8µ
VFB
2
L2 =
100ns
Insulators = Berquist Sil–Pad 1500
Heatsinks = AAVID Engineering Inc. 533402B02552 with clip
MC34067–5803
L2 = 5 turns #48 AWG (1300 strands litz wire)
Core: 0.5″ diameter air code
Inductance = 100 nH
L1 = 2 turns #48 AWG (1300 strands litz wire)
Core: Philips 3F3 EP10–3F3
Bobbin: Philips EP10PCB1–8
Inductance = 1.8 µH
T3 = Coilcraft D1870 (100 turns)
T2 = All windings: 8 turns #36 AWG
Core: Philips 3F3 EP7–3F3
Bobbin: Philips EP7PCB1–6
T1 = Primary: 12 turns #48 AWG (1300 strands litz wire)
Secondary: 6 turns center tapped #48 AWG (1300 strands litz wire)
Core: Philips 3F3 4312 020 4124
Bobbin: Philips 4322 021 3525
Primary Leakage Inductance = 1.0 µH
470pF
10
12
13
14
5
Vin = 36 – 56V
Vout = 5.0V
MC34067 MC33067
MOTOROLA ANALOG IC DEVICE DATA
MC34067 MC33067
Figure 21. Printed Circuit Board and Component Layout
3.875″
5.0″
(Bottom View)
(Top View)
MOTOROLA ANALOG IC DEVICE DATA
13
MC34067 MC33067
OUTLINE DIMENSIONS
P SUFFIX
PLASTIC PACKAGE
CASE 648–08
ISSUE R
–A–
16
9
1
8
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION L TO CENTER OF LEADS WHEN
FORMED PARALLEL.
4. DIMENSION B DOES NOT INCLUDE MOLD FLASH.
5. ROUNDED CORNERS OPTIONAL.
B
F
C
L
S
–T–
SEATING
PLANE
K
H
G
D
M
J
16 PL
0.25 (0.010)
M
T A
M
DIM
A
B
C
D
F
G
H
J
K
L
M
S
INCHES
MIN
MAX
0.740
0.770
0.250
0.270
0.145
0.175
0.015
0.021
0.040
0.70
0.100 BSC
0.050 BSC
0.008
0.015
0.110
0.130
0.295
0.305
0_
10 _
0.020
0.040
MILLIMETERS
MIN
MAX
18.80
19.55
6.35
6.85
3.69
4.44
0.39
0.53
1.02
1.77
2.54 BSC
1.27 BSC
0.21
0.38
2.80
3.30
7.50
7.74
0_
10 _
0.51
1.01
DW SUFFIX
PLASTIC PACKAGE
CASE 751G–03
(SO–16L)
ISSUE B
A
D
9
1
8
h X 45 _
E
M
0.25
8X
H
B
M
16
q
16X
M
14X
e
T A
S
B
S
14
A1
L
A
0.25
DIM
A
A1
B
C
D
E
e
H
h
L
B
B
NOTES:
1. DIMENSIONS ARE IN MILLIMETERS.
2. INTERPRET DIMENSIONS AND TOLERANCES
PER ASME Y14.5M, 1994.
3. DIMENSIONS D AND E DO NOT INLCUDE MOLD
PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 PER SIDE.
5. DIMENSION B DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.13 TOTAL IN EXCESS
OF THE B DIMENSION AT MAXIMUM MATERIAL
CONDITION.
SEATING
PLANE
T
q
MILLIMETERS
MIN
MAX
2.35
2.65
0.10
0.25
0.35
0.49
0.23
0.32
10.15
10.45
7.40
7.60
1.27 BSC
10.05
10.55
0.25
0.75
0.50
0.90
0_
7_
C
MOTOROLA ANALOG IC DEVICE DATA
MC34067 MC33067
Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals”
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arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that
Motorola was negligent regarding the design or manufacture of the part. Motorola and
are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal
Opportunity/Affirmative Action Employer.
MOTOROLA ANALOG IC DEVICE DATA
15
MC34067 MC33067
Mfax is a trademark of Motorola, Inc.
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16
◊
MC34067/D
MOTOROLA ANALOG IC DEVICE
DATA