Intersil ISL55211IRTZ-EVAL1Z Wideband, low noise, low distortion, fixed gain differential amplifier Datasheet

Wideband, Low Noise, Low Distortion, Fixed Gain,
Differential Amplifier
ISL55211
Features
The ISL55211 is a wideband, differential input to differential
output amplifier offering 3 possible internal gain settings.
Using fixed 500Ω internal feedback resistors, the amplifier
may be configured for a differential gain of 2, 4 or 5V/V
depending on which combination of input pins are connected
to the signal source. Internal feedback capacitors controls the
signal bandwidth to be a constant 1.4GHz in all gain settings.
• 3 Fixed Gain Options . . . . . . . . . . . . . . . . . . . . . . . 2, 4, or 5V/V
• Constant Bandwidth Over Gain . . . . . . . . . . . . . . . . . . 1.4GHz
• Differential Slew Rate . . . . . . . . . . . . . . . . . . . . . . . 5,600V/µs
• 2VP-P, 2-tone IM3 (200Ω) 100MHz . . . . . . . . . . . . . . -103dBc
• Low Differential Output Noise (Gain 5V/V) . . . . . . <12nV/√Hz
• Supply Voltage Range . . . . . . . . . . . . . . . . . . . . . . 3.0V to 4.2V
Ideally suited for AC-coupled data acquisition applications, the
output DC common mode voltage is controlled through an
external VCM pin or left to default to 1.2V above the negative
supply pin. Where the differential signal source is AC-coupled,
the input common mode voltage will equal the output
common mode voltage.
• Quiescent Power (3.3V Supply) . . . . . . . . . . . . . . . . . .115mW
Applications
• Low Power, High Dynamic Range ADC Interface
• Differential Mixer Output Amplifier
Intended for very high dynamic range ADC interface
applications, the ISL55211 offers 5600V/µs differential slew
rate, <12nV/√Hz output noise, and >100dBc SFDR to
>100MHz for 2VP-P 2-tone 3rd order intermodulation. Its
balanced architecture effectively suppresses even order
distortion terms - an important issue for very wide band 1st
Nyquist zone ADC interface applications. Minimum gain
operation of 2V/V (6dB) with <1dB peaking ensures stable
performance over-temperature. It's ultra high differential slew
rate of 5600V/µs provides adequate performance margin for
large signal application through 500MHz.
• SAW Filter Pre/Post Driver
• Fixed Gain Coax Receiver
Related Devices
• ISL55210 - External Gain Set Version
• ISLA112P50 - 12-bit, 500MSPS ADC (<500mW)
• ISLA214P50 - 14-bit, 500MSPS ADC (<850mW)
Related Literature
The ISL55211 requires only a single 3.3V (max. 4.2V) power
supply and 35mA quiescent current, providing a very low
power solution (115mW). Further power savings are possible
using the optional power shutdown control - where the
quiescent current can be reduced to <0.4mA. A companion
device, the ISL55210, offers similar performance where the
feedback and gain resistors are external. Both are available in
a 16 Ld TQFN (Pb-free) package and are specified for
operation over the -40°C to +85°C ambient temperature
range.
• AN1649 - “Designer’s guide to the ISL55210 and ISL55211
Evaluation Boards”
+3.3V
ISLA214P50
(850mW)
35mA
(115mW)
10
+
120nH
1:2
ISL55211
VCM
50
ADT4-1T
180mVP-P
For ADC -1dBFS
1.2V
1:1.4
22pF
VADC
300
G = 5V/V
120nH
-
10
20 log
50
VADC
= 20dB
Vi
VCM
8pF
GAIN (dB)
Vi
14Bit 500MSPS
ADT2-1T 50
23
20
17
14
11
8
5
2
-1
-4
-7
-10
-13
1M
HIGH GAIN, VERY LOWPOWER, ADC INTERFACE WITH 3RD ORDER OUTPUT FILTER
MEASURED FREQUENCY RESPONSE
10M
100M
1G
FREQUENCY (Hz)
FIGURE 1. TYPICAL APPLICATION CIRCUIT
June 21, 2011
FN7868.0
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2011. All Rights Reserved
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All other trademarks mentioned are the property of their respective owners.
ISL55211
Pin Configuration
ISL55211
(3x3 16 LD TQFN)
TOP VIEW
VIN2-
1
VIN1-
2
GND
VS+
VCM
GND
16
15
14
13
348
500
750
0.2pF
-
140
12
VO+
11
NC
10
NC
9
VO-
VCM
VIN1+
3
140
+
0.2pF
750
VIN2+
4
500
348
5
GND
Pin Descriptions
6
VS+
7
8
GND
GND
Pd
PIN NUMBER
SYMBOL
DESCRIPTION
1
VIN2-
Balanced Differential Input for Av = 6dB, strapped to VIN1- for Av = 14dB
2
VIN1-
Balanced Differential Input for Av = 12dB, strapped to VIN2- for Av = 14dB
3
VIN1+
Balanced Differential Input for Av = 12dB, strapped to VIN2+ for Av = 14dB
4
VIN2+
Balanced Differential Input for Av = 6dB, strapped to VIN1+ for Av = 14dB
5, 8, 13, 16
GND
Supply Ground (thermal pad electrically connected)
6, 15
VS+
Positive Power Supply (3.0V~4.2V)
7
Pd
Power-down: Pd = logic low. Puts part into low power mode; Pd = logic high or open for normal operation
9
VO-
Inverting Amplifier Output
10, 11
NC
No Internal Connection
12
VO+
Non-Inverting Amplifier Output
14
VCM
Common-mode Voltage Input
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART MARKING
ISL55211IRTZ
5211
ISL55211IRTZ-EVAL1Z
Evaluation Board
TEMP RANGE
(°C)
-40 to +85
PACKAGE
(Pb-free)
16 Ld 3x3 TQFN
PKG.
DWG. #
L16.3x3D
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL55211. For more information on MSL please see techbrief TB363.
2
FN7868.0
June 21, 2011
ISL55211
Absolute Maximum Ratings (TA = +25°C)
Thermal Information
Supply Voltage from VS+ to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .4.5V
Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .VS+ +0.3V to GND-0.3V
Power Dissipation. . . . . . . . . . . . . . . . . . . . See Thermal Conditions Section
ESD Rating
ESD Rating
Human Body Model (Per MIL-STD-883 Method 3015.7) . . . . . . . . 3500V
Machine Model (Per EIAJ ED-4701 Method C-111) . . . . . . . . . . . . . 250V
Charged Device Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1500V
Latch up (Per JESD-78; Class II; Level A) . . . . . . . . . . . . . . . . . . . . . . 100mA
Thermal Resistance (Typical)
θJA (°C/W) θJC (°C/W)
16 Ld TQFN Package (Notes 4, 5) . . . . . . .
63
16.5
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +125°C
Max. Continuous Operating Junction Temperature . . . . . . . . . . . . .+135°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Ambient Operating Temperature . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications VS+ = +3.3V Test Conditions: G = 12dB, VCM = open, VO = 2VP-P, RL = 200Ω differential, TA = +25°C,
differential input, differential output, input and output referenced to internal default VCM (1.2V nominal) unless otherwise specified.
PARAMETER
CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNIT
TESTED
AC PERFORMANCE
Small-Signal Bandwidth (4-port S
parameter, Test Circuit 2)
G = 6dB, VO = 100mVP-P
1.6
GHz
G = 12dB, VO = 100mVP-P
1.4
GHz
G = 14dB, VO = 100mVP-P
1.4
GHz
Bandwidth for 0.1-dB Flatness
G = 12dB, VO = 2VP-P (Figure 17)
150
MHz
Large-Signal Bandwidth
G = 12dB, VO = 2VP-P
1.2
GHz
Gain Accuracy
G = 6dB, RL = Open
1.96
2
2.04
V/V
*
G = 12dB, RL = Open
3.88
4
4.12
V/V
*
G = 14dB, RL = Open
4.8
5
5.2
V/V
Slew Rate (Differential)
5,600
V/µs
Differential Rise/Fall Time
2-V step (simulated)
0.22
ns
2nd-order Harmonic Distortion,
Test Circuit 1, 15dB Gain
f = 20MHz, VO = 2VP-P
-110
dBc
f = 50MHz, VO = 2VP-P
-98
dBc
f = 100MHz, VO = 2VP-P
-85
dBc
f = 20MHz, VO = 2VP-P
-120
dBc
f = 50MHz, VO = 2VP-P
-110
dBc
f = 100MHz, VO = 2VP-P
-100
dBc
2nd-order Intermodulation Distortion,
Test Circuit 1, 15dB Gain
fc = 70MHz, 200kHz spacing (2VP-P envelope)
-89
dBc
fc = 140MHz, 200kHz spacing (2VP-P envelope)
-78
dBc
3rd-order Intermodulation Distortion,
Test Circuit 1, 15dB Gain
fc = 70MHz, 200kHz spacing (2VP-P envelope)
-104
dBc
fc = 140MHz, 200kHz spacing (2VP-P envelope)
-92
dBc
11.2
nV/√Hz
3rd-order Harmonic Distortion,
Test Circuit 1, 15dB Gain
Output Voltage Noise
Test Circuit 1, total gain 15dB, ADT2-1T
DC PERFORMANCE (Internal Nodes)
Input Offset Voltage
3
TA = +25°C
-1.4
±0.1
+1.4
mV
TA = -40°C to +85°C
-1.6
±0.1
+1.6
mV
*
FN7868.0
June 21, 2011
ISL55211
Electrical Specifications VS+ = +3.3V Test Conditions: G = 12dB, VCM = open, VO = 2VP-P, RL = 200Ω differential, TA = +25°C,
differential input, differential output, input and output referenced to internal default VCM (1.2V nominal) unless otherwise specified. (Continued)
PARAMETER
CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6)
UNIT
µV/°C
Average Offset Voltage Drift
TA = -40°C to +85°C
±3
Input Bias Current
TA = +25°C, positive current into the pin
+50
+120
µA
TA = -40°C to +85°C
+50
+140
µA
Average Bias Current Drift
TA = -40°C to +85°C
200
Input Offset Current
TA = +25°C
-5
TA = -40°C to +85°C
-6
Average Offset Current Drift
TA = -40°C to +85°C
±1
TESTED
*
nA/°C
+5
µA
+6
µA
*
nA/°C
±8
INPUT
Common-mode Input Range High
Internal Nodes
1.7
Common-mode Input Range Low
Internal Nodes
1.1
Common-mode Rejection Ratio
f < 10MHz, common mode to differential
output
56
Differential Input Impedance
VIN1- Connected to VIN2VIN1+ Connected to VIN2+
V
*
V
*
75
dB
*
200
Ω
2.35
V
*
V
*
VP-P
*
OUTPUT (Pins 9 AND 12)
Maximum Output Voltage
Minimum Output Voltage
Differential Output Voltage Swing
Each output (with 200Ω differential load)
Linear Operation
2.15
TA = +25°C
3.04
TA = -40°C to +85°C
2.95
Differential Output Current Drive
RL = 10Ω [sourcing or sinking]
Closed-loop Output Impedance
f < 10MHz, differential
0.45
40
0.63
3.8
V
45
mA
0.6
Ω
*
OUTPUT COMMON-MODE VOLTAGE CONTROL (Pin 14)
Small-signal Bandwidth
From VCM pin to Output VCM
30
MHz
Slew Rate
Rising/Falling
150
V/µs
Gain
VCM input pin 1.0V to 1.4V
0.995
0.999
V/V
*
-8
±1
+8
mV
*
1.18
1.2
1.22
V
*
Output Common-Mode Offset from CM Input
CM Default Voltage
Output VCM with VCM pin floating
CM Input Bias Current
At control pin
CM Input Voltage Range
At control pin
CM Input Impedance
At control pin
2
0.9
µA
1.9
15 || 50
V
*
kΩ || pF
POWER SUPPLY
Specified Operation Voltage
Quiescent Current
TA = +25°, VS+ = 3.3V, VS- = 0V
TA = -40°C to +85°C
Power-supply Rejection (PSRR) VS+
3.0V to 4.5V range
f < 10MHz [PSRR to differential output]
POWER-DOWN (Pin 7)
Referenced to GND
Enable Voltage Threshold
Assured on above 1.55V
Disable Voltage Threshold
Assured off below 0.54V
4
3
3.3
4.2
V
*
33
35
37
mA
*
30.5
35
39.5
mA
50
67
1.3
0.54
0.7
1.55
dB
*
V
*
V
*
FN7868.0
June 21, 2011
ISL55211
Electrical Specifications VS+ = +3.3V Test Conditions: G = 12dB, VCM = open, VO = 2VP-P, RL = 200Ω differential, TA = +25°C,
differential input, differential output, input and output referenced to internal default VCM (1.2V nominal) unless otherwise specified. (Continued)
MIN
(Note 6)
TYP
TA = +25°C
0.2
0.3
TA = -40°C to +85°C
0.15
PARAMETER
CONDITIONS
Power-down Quiescent Current
Input Bias Current
PD = 0V, current positive into pin
-5
1
MAX
(Note 6)
UNIT
TESTED
*
0.4
mA
0.45
mA
+5
µA
2 || 5
Input Impedance
MΩ || pF
Turn-on Time Delay
Measured to output on
200
ns
Turn-off Time Delay
Measured to output off
400
ns
NOTE:
6. Compliance to datasheet limits is assured by one or more methods: production test, characterization, and/or design.
TABLE 1. ISL55211 INTENDED TRANSFORMER + INTERNAL GAIN
SETTINGS
ISL55211
500
RG
+
VI
1:n
50
INPUT
VO
RT
RG
-
INPUT
XFMR
TURNS
RATIO
INTERNAL
RG VALUE
(Ω)
GAIN (V/V)
VO/VI
GAIN (dB)
VO/VI
RT VALUE (Ω)
TO GET 50Ω
MATCH
1:1.4
250
2.8
9
122
1:1.4
125
5.6
15
162
1:1.4
100
7
17
192
1:2
250
4
12
333
1:2
125
8
18
1020
1:2
100
10
20
Open
500
FIGURE 2. INTENDED CONFIGURATION
5
FN7868.0
June 21, 2011
ISL55211
Typical Performance Curves Vs+ = 3.3V, TA ≈ +25°C, unless otherwise noted.
3
3
2
2
1
Av = 2
NORMALIZED GAIN (dB)
NORMALIZED GAIN (dB)
1
0
-1
-2
-3
Av = 5
-4
Av = 4
-5
-6
-7
-8
TEST CIRCUIT #1
VO = 500mVP-P
100M
FREQUENCY (Hz)
-3
-5
-6
3
2
NORMALIZED GAIN (dB)
-1
-2
Av = 5
-5
Av = 4
-6
-7
-9
Av = 2
0
-1
-2
-3
Av = 5
-4
-5
-6
-7
TEST CIRCUIT #1
VO(P-P) = 3VP-P
-8
100M
FREQUENCY (Hz)
Av = 4
TEST CIRCUIT #1
VO(P-P) = 3VP-P
-8
10M
1G
1
Av = 2
-4
100M
FREQUENCY (Hz)
FIGURE 4. SMALL SIGNAL FREQUENCY RESPONSE WITH
ADT4-1WT INPUT TRANSFORMER
2
-3
Av = 4
TEST CIRCUIT #1
VO = 500mVP-P
-9
10M
3
0
Av = 5
-4
1G
1
NORMALIZED GAIN (dB)
-2
-8
FIGURE 3. SMALL SIGNAL FREQUENCY RESPONSE WITH ADT2-1T
INPUT TRANSFORMER
-9
10M
1G
100M
FREQUENCY (Hz)
1G
FIGURE 5. LARGE SIGNAL FREQUENCY RESPONSE WITH ADT2-1T
INPUT TRANSFORMER
FIGURE 6. LARGE SIGNAL FREQUENCY RESPONSE WITH
ADT4-1WT INPUT TRANSFORMER
25
20
20
NOISE FIGURE (dB)
16
15
10
GAIN = 17dB
GAIN = 15dB
5
14
12
10
8
6
GAIN = 20dB
GAIN = 18dB
4
2
TEST CIRCUIT #1
0
GAIN = 12dB
18
GAIN = 9dB
NOISE FIGURE (dB)
-1
-7
-9
10M
Av = 2
0
50
100
150
200
250
300
350
400
450
500
FREQUENCY (MHz)
FIGURE 7. NOISE FIGURE WITH ADT2-1T INPUT TRANSFORMER
6
0
TEST CIRCUIT #1
50
100
150
200
250
300
350
400
450
500
FREQUENCY (MHz)
FIGURE 8. NOISE FIGURE WITH ADT4-1WT INPUT TRANSFORMER
FN7868.0
June 21, 2011
ISL55211
Typical Performance Curves Vs+ = 3.3V, TA ≈ +25°C, unless otherwise noted. (Continued)
HD2, HD3 SPURIOUS (dBc)
-65
-70
-60
HD2 of 3VP-P
TEST CIRCUIT 1
RL = 200Ω
GAIN = 15dB
TEST CIRCUIT 1
RL = 200Ω
-70 GAIN = 15dB
HD2 of 2VP-P
IM2, IM3 SPURIOUS (dBc)
-60
-75
HD3 of 3VP-P
-80
-85
-90
-95
-100
HD3 of 2VP-P
-105
HD2 of 1VP-P
-110
50M
100M
150M
FREQUENCY (Hz)
IM2 of 2VP-P
-90
-100
IM3 of 2VP-P
100
150
FREQUENCY (MHz)
-60
-80
-70
HD2 of 9dB
HD2 of 15dB
HD2 of 17dB
-85
-90
HD3 of 17dB
-95
-100
IM2, IM3 SPURIOUS (dBc)
HD2, HD3 SPURIOUS (dBc)
-60
-75
HD3 of 15dB
-105
TEST CIRCUIT 1
VO = 1VP-P EACH TONE
IM2 of 15dB
-80
-90
IM3 of 9dB
-100
IM3 of 15dB
-110
100M
150M
FREQUENCY (Hz)
-120
50M
200M
100M
150M
FREQUENCY (MHz)
HD2 of 200Ω
HD2 of 500Ω
-70
-80
-90
HD3 of 500Ω
-100
-110
50M
HD3 of 100Ω
100M
150M
FREQUENCY (Hz)
FIGURE 13. HD2, HD3 vs DIFFERENTIAL LOAD
7
IM2 of 100Ω
IM2 of 50Ω
-70
IM2 of 200Ω
-80
-90
-100
IM2 of 500Ω
-110
IM3 of 50Ω
-120
HD3 of 200Ω
HD3 of 50Ω
TEST CIRCUIT 1
-60
IM2, IM3 SPURIOUS (dBc)
HD2, HD3 SPURIOUS (dBc)
-60 HD2 of 100Ω
-50
HD2 of 50Ω
TEST CIRCUIT 1
GAIN = 15dB
200M
FIGURE 12. IM2 AND IM3 vs GAIN
FIGURE 11. HD2, HD3 vs GAIN
-50
IM2 of 9dB
IM2 of 17dB
IM3 of 17dB
HD3 of 9dB
-110
50M
200
FIGURE 10. IM2 AND IM3 vs OUTPUT SWING
FIGURE 9. HD2, HD3 vs OUTPUT SWING
TEST CIRCUIT 1
-65 RL = 200Ω
VO = 2VP-P
-70
IM3 of 1VP-P
IM3 of 3VP-P
-120
50
200M
IM2 of 1VP-P
-80
-110
HD3 of 1VP-P
IM2 of 3VP-P
IM3 of 500Ω
IM3 of 100Ω
IM3 of 200Ω
200M
-130
50M
100M
150M
FREQUENCY (Hz)
200M
FIGURE 14. IM2, IM3 vs DIFFERENTIAL LOAD
FN7868.0
June 21, 2011
ISL55211
Typical Performance Curves Vs+ = 3.3V, TA ≈ +25°C, unless otherwise noted. (Continued)
180
TEST CIRCUIT 1
WITH ADT2-1T INPUT
PHASE OF 15dB
PHASE (°)
1.4
140
1.3
120
15
14
1.6
1.5
PHASE OF 9dB
160
1.7
1.2
GROUP DELAY OF 17dB
100
1.1
GROUP DELAY OF 15dB
80
10
GROUP DELAY OF 9dB
30
50
70
90
110
130
150
170
190
OUTPUT SPOT NOISE (nV/√Hz)
PHASE OF 17dB
GROUP DELAY (ns)
200
GAIN = 17dB
13
GAIN = 15dB
12
11
10
9
8
7
1.0
5
1M
10M
FREQUENCY (Hz)
FREQUENCY (MHz)
3
12
10
0
OUTPUT IMPEDANCE (Ω)
NORMALIZED GAIN (dB)
6dB
-3
12dB
-6
14dB
-9
10
TEST CIRCUIT 2
SIMULATED
8
6
GAIN = 2
4
100
1000
0
1M
5000
10M
100M
1000M
FREQUENCY (Hz)
FREQUENCY (MHz)
FIGURE 17. SMALL SIGNAL FREQUENCY RESPONSE
FIGURE 18. DIFFERENTIAL OUTPUT IMPEDANCE
-50
DIFFERENTIAL TO COMMON MODE
CONVERSION (dBc)
3
0
-3
GAIN (dB)
GAIN = 5
GAIN = 4
2
TEST CIRCUIT 2
NO TRANSFORMERS
-6
100M
FIGURE 16. DIFFERENTIAL OUTPUT NOISE vs GAIN
FIGURE 15. PHASE AND GROUP DELAY vs GAIN
-12
TEST CIRCUIT 1 ADT2-1T
OUTPUT NOISE INCLUDING
50Ω SOURCE NOISE
GAIN = 9dB
6
200mVP-P
10mVP-P
-9
-12
-15
-18 TEST CIRCUIT 3
COMMON MODE AC OUTPUT
-21
1
10
FREQUENCY (MHz)
100
200
FIGURE 19. VCM PIN INPUT FREQUENCY RESPONSE TO OUTPUT
COMMON MODE
8
-55
TEST CIRCUIT 3
VO(P-P) DIFFERENTIAL IS 2VP-P
9dB
-60
15dB
-65
-70
17dB
-75
-80
2M
20M
FREQUENCY (Hz)
200M
FIGURE 20. OUTPUT BALANCE ERROR
FN7868.0
June 21, 2011
ISL55211
Typical Performance Curves Vs+ = 3.3V, TA ≈ +25°C, unless otherwise noted. (Continued)
0.15
1.5
TEST CIRCUIT #1 WITH ADT2-1T
100MHz SQUARE WAVE INPUT
TEST CIRCUIT #1 WITH ADT2-1T
100MHz SQUARE WAVE INPUT
OUTPUT
INPUT
0.05
0
-0.05
INPUT
0.5
0
-0.5
-0.10
-0.15
OUTPUT
1.0
AMPLITUDE (V)
AMPLITUDE (V)
0.10
-1
0
2
4
6
8
10
12
14
16
18
-1.5
20
0
2
4
6
8
10
12
14
16
18
20
TIMEBASE (ns)
TIMEBASE (ns)
FIGURE 21. SMALL SIGNAL STEP RESPONSE
FIGURE 22. LARGE SIGNAL RESPONSE
-14.4
100MHz
100MHz OUTPUT
OUTPUT
OUTPUT vs INPUT (dBc)
-14.6
ENABLED
ENABLED
PD
DISABLED
DISABLED
2VP-P
-14.8
-15.0
-15.2
100mVP-P
-15.4
-15.6
-15.8
-16.0
1M
TEST CIRCUIT 1
2µs/DIV
2µs/DIV
FIGURE 23. ENABLE/DISABLE TIMES (2µs/DIV)
95
OUTPUT
100M
PSRR TO VO (DIFFERENTIAL)
2.0
85
1.5
1.0
INPUT
0.5
0
-0.5
-1.0
-1.5
75
CMRR TO VO (DIFFERENTIAL)
65
55
45
-2.0
-2.5
10M
FREQUENCY (Hz)
FIGURE 24. SHUTDOWN FEED-THROUGH
PSRR/CMRR (dB)
INPUT AND OUTPUT WAVEFORMS (V)
2.5
TEST CIRCUIT 1 WITH ADT2-1T INPUT
OUTPUT VP-P RELATIVE TO INPUT VP-P
TEST CIRCUIT 1
0
20
40
60
80
100 120
TIME (ns)
140
FIGURE 25. OVERDRIVE RECOVERY
9
160
180
200
35
TEST CIRCUIT 1 SIMULATED
EXACT EXTERNAL R’s
1
10
100
1000
FREQUENCY (MHz)
FIGURE 26. PSRR/CMRR TO DIFFERENTIAL VO
FN7868.0
June 21, 2011
ISL55211
Typical Performance Curves Vs+ = 3.3V, TA ≈ +25°C, unless otherwise noted. (Continued)
6
5
4
MAXIMUM DIFFERENTIAL VP-P
OUTPUT USING DEFAULT VCM
3
2
INTERNALLY SET VCM
1
3.0 3.1 3.2 3.3 3.4 3.5 3.6 3.7 3.8 3.9 4.0 4.1 4.2 4.3 4.4 4.5
SUPPLY VOLTAGE (V)
FIGURE 27. DEFAULT VCM AND MAX VOPP vs SUPPLY VOLTAGE
Applications
SUPPLY CURRENT (mA)
OUTPUT DEFAULT VCM AND
MAX DIFFERENTIAL VOPP (V)
TEST CIRCUIT 1
45
44 TEST CIRCUIT 1
43
TA = +85°C
42
41
40
39
38
TA = +25°C
37
36
35
TA = -40°C
34
33
32
31
30
3.0 3.1 3.2 3.3 3.4 3.5 3.6 3.7 3.8 3.9 4.0 4.1 4.2 4.3 4.4 4.5
SINGLE SUPPLY VOLTAGE (V)
FIGURE 28. SUPPLY CURRENT vs SUPPLY VOLTAGE
shown in Table 1, gives a 9dB to 20dB operating gain range in
approximately 3dB steps.
Basic Operation
The ISL55211 is a very wideband, voltage feedback based,
differential amplifier including an output common mode control
loop and optional power shutdown feature. Intended for very low
distortion differential signal driving, this internally fixed gain
device provides 3 possible gain settings by simply picking the
input side connections as shown in Table 1. Including internal
compensation, the ISL55211 holds a constant bandwidth over
gain settings. Most applications are intended for AC-coupled I/O
using a single 3.3V supply and an input transformer. The internal
resistor values have been scaled up slightly to require an external
termination element along with the two internal resistors where
a 50Ω differential input match is desired. This does increase the
output noise slightly but narrows up the input VSWR tolerance
and lowers the added loading of the feedback resistors
improving SFDR.
Where DC-coupled differential I/O operation is desired, the
ISL55211 can be connected directly to the source as long as the
internal input common mode range limits are observed (1.1V to
1.7V for a 3.3V single supply operation). For a DC-coupled, single
to differential requirement, consider the ISL55210. This device is
an external resistor version of the ISL55211 where the flexibility
in the external resistors will enable single to differential
operation. For a ground referenced input signal, this will require a
negative supply when using the ISL55210.
Most applications behave as a differential inverting op amp
design. There is therefore an input gain resistor on each side of
the inputs that must be driven. The 3 possible connections to the
two pairs of input pins will give a 100Ω, 125Ω, or 250Ω input
resistor on each side. Combined with the two input turns ratio's
10
The device can be powered down to < 400µA supply current
using the optional disable pin. To operate normally, this pin
should be asserted high using a simple logic gate to +VCC or tied
high through a 10kΩ resistor to +VCC. When disabled, the power
dissipation drops to < 1mW but, due to the inverting op amp type
architecture, the input signal will feed-forward through the
feedback and gain resistors giving limited isolation.
Application and Characterization Circuits
Test Circuit 1 of Figure 29 forms a starting point for many of the
characterization curves for the ISL55211. Since most lab sources
and measurement devices are single-ended, this circuit converts
to differential at the input through a wideband transformer and
would also be a typical application circuit coming from a
single-ended source. Assuming the source is a 50Ω impedance,
the internal RG resistors and external RT are set to provide both
the input termination and the gain. Since the inverting summing
nodes act as virtual ground points for AC signal analysis, the total
termination impedance across the input transformer secondary
will be (2*RG)||RT. Setting this equal to n2*RS will give a
matched input impedance inside the bandwidth of the
transformer (where "n" is the turns ratio). The amplifier gain is
fixed by the selected input RG element and the internal 500Ω
feedback resistors. While the ISL55211 is internally a Voltage
Feedback Design (VFA) to give the lowest possible noise, internal
compensation caps hold the bandwidth over gain setting
approximately constant at 1.4GHz. For wider small signal
bandwidth at lower gains, consider the ISL55210, which provides
>2.2GHz at a gain of 12dB.
FN7868.0
June 21, 2011
ISL55211
+3.3V
115mW 35mA
10k
500
RG
1:1.4
RT
Vi 1µF
ADT21T
or
ADT41Wt
85
0.2pF
+
50
PD
1µF
200
35
VO
VCM
1µF
0.1µF
35
-
0.2pF
RG
50
1:1
Vm
ADT11WT
Where just the amplifier is tested, a 4-port network analyzer is
used and the very simple test circuit of Figure 30 is
implemented. This is used to measure the differential S21 curves
vs gain of Figure 17 and as a simulation circuit for the differential
output impedance vs gain of Figure 18. Changing the gain is a
simple matter of adjusting the connections to the four input RG
connections resistors, as shown in Table 1. This circuit depends
on the two AC-coupled source 50Ω of the 4 port network analyzer
and presents an AC-coupled differential 100Ω load to the
amplifier as the input impedance of the remaining two ports of
the network analyzer.
+3.3V
1µF 85
RF
500
ISL55211
RG
50
10k
PD
50
+
FIGURE 29. TEST CIRCUIT 1
Working with a transformer coupled input as shown in Figure 29,
or with two DC blocking caps from a differential source, means
the output common mode voltage set by either the default
internal VCM setting, or a voltage applied to the VCM control pin,
will also appear as the input common mode voltage. This
provides a very easy way to control the ISL55211 I/O common
mode operating voltages for an AC-coupled signal path. The
internal common mode loop holds the output pins to VCM and,
since there is no DC path for an ICM current back towards the
input in Figure 29, that VCM setting will also appear as the input
common mode voltage. It is useful, for this reason, to leave any
input transformer secondary centertap unconnected. The
internally set VCM voltage is referenced from the negative supply
pin. With a single 3.3V supply, it is very close to 1.2V but will
change with total supply voltage across the device as shown in
Figure 27.
Most of the characterization curves starting with Figure 29 then
get different gains by changing the connections to the two pairs
of input RG connections, as shown on the pin configuration
drawing on page 2. Two input turns ratios are intended for Test
Circuit 1; either a 1:1.4 turns ratio (ohms ratio of 2) or a 1:2 turns
ratio (ohm ratio of 4). The specific transformers shown in
Figure 29 are representative of broadband RF transformers but
alternate devices and manufacturers of these turns ratio devices
are certainly applicable. The output side of this test circuit
presents a differential 200Ω load while converting the
differential to single-ended through a resistive attenuator and a
1:1 transformer. This inserts approximately a 17dB insertion loss
that is removed to report the characteristic curves. For load tests
below the 200Ω shown in Figure 29, a simple added shunt
resistor is placed across the output pins. For loads > 200Ω, the
series and shunt load R's are adjusted to show that total load
(including the 50Ω measurement load reflected through the 1:1
output measurement port transformer) and provide an apparent
50Ω differential source to that transformer. This output side
transformer is for measurement purposes only and is not
necessary for final applications circuits. There are output
interface designs that do benefit from a transformer as part of
the signal path as shown in Figure 1. In that case, the 1:1:4
output side transformer becomes part of a filter design and
recovers the filter insertion loss from the amplifier output pins to
the ADC inputs.
11
1/2 of a 4-port
S-parameter
1/2 of a 4-port
S-parameter
VCM
RT
50
50
RG
RF
ISL55211
FIGURE 30. TEST CIRCUIT 2 4-PORT S-PARAMETER
MEASUREMENTS
Using this measurement allows the small single bandwidth of
just the ISL55211 to be exposed. Many of the other
measurements are using I/O transformers that are limiting the
apparent bandwidth to a reduced level. Figure 17 shows the 3
normalized differential S21 curves for the possible internal gains
of 9dB, 14dB and 15dB. The small signal bandwidth is remaining
nearly constant at 1.4GHz due to the internal capacitive
feedback network.
The closed loop differential output impedance of Figure 18 is
simulated using Figure 30 in ADS. This shows a relatively low
output impedance (< 1Ω through 100MHz) constant with signal
gain setting. Typical FDA outputs show a closed loop output
impedance that increases with signal gain setting. The ISL55211
holds a more constant response due to internal design elements
unique to this device.
Common mode output measurements are made using the circuit
in Figure 31. Here, the outputs are summed together through two
100Ω resistors (still a 200Ω differential load) to a center point
where the average, or common mode, output voltage may be
sensed. This is coupled through a 1µF DC blocking capacitor and
measured using 50Ω test equipment. The common mode source
impedance for this circuit is the parallel combination of the
2-100Ω elements, or 50Ω. Figure 19 uses this circuit to measure
the small and large signal response from the VCM control pin to
the output common mode. This pin includes an internal 50pF
capacitor on the default bias network (to filter supply noise when
there is no connection to this pin), which bandlimits the response
to approximately 30MHz. This is far lower than the actual
bandwidth of the common mode loop. Figure 20 uses this output
FN7868.0
June 21, 2011
ISL55211
CM measurement circuit with a large signal (2VP-P) differential
output voltage (generated through the Vi path of Figure 31) to
measure the differential to common mode conversion - often
called the "Output Balance Error" for an FDA.
ISL55211
500
RS
+3.3V
PD
Vi
50
ADT2-1T
1:1.4
500
+
75
INPUT
1:n
1µF
V CM
500
V CM Input
RS
1:1.4 -> CX2045NL
1:2 -> CX2032
50
RG
VO
RT
100
Output
V CM
1µF
1µF
+
Vi
10k
RG
RG
-
RG
500
100
ISL55211
50
FIGURE 31. TEST CIRCUIT #3 COMMON MODE AC OUTPUT
MEASUREMENTS
Single Supply, Input Transformer Coupled,
Design Considerations
The characterization circuit of Figure 29 shows one possible
input stage interface that offers several advantages. Where AC
coupling is adequate, the circuit of Figure 29 simplifies the input
common mode voltage control. If the source coming into this
stage is single-ended, the input transformer provides a zero
power conversion to differential. The two gain resistors (RG in
Figure 29) provide both a portion of the input termination
impedance and the gain element for the amplifier. For 50Ω
systems, these RG resistors are too high with the turns ratios
shown in Figure 29 to provide the full match and an external RT
resistor is required. This RT element goes away at the highest
gain setting using a 1:2 input turns ratio transformer.
It is also possible to adapt this circuit to other input characteristic
impedances. Figure 32 shows a 75Ω example similar to Figure 2
while Table 2 shows the necessary external R values and
resulting gains.
FIGURE 32. 75Ω IMPEDANCE IMPLEMENTATIONS
Here, the sum of the two internal RG resistors at the higher two
gain settings is too low to retain a match for the 1:2 input step up
case. There, a pair of external series resistors are added to get
the total differential input impedance up to 300Ω on the
secondary side of the transformer and the RT element goes to
infinity. These two conditions are not particularly useful but
Figure 32 shows how to implement the full range of internal
conditions with the two turns ratios considered in Table 2.
Figure 32 also shows a pair of alternate input transformer types
from Pulse Engineering particularly suitable to the 75Ω case.
TABLE 2. EXTERNAL RESISTORS FOR A 75Ω INPUT
IMPEDANCE DESIGN
ISL55211 INTENDED TRANSFORMER + INTERNAL GAIN SETTINGS
INPUT
XFMR
TURNS
RATIO
INTERNAL
RG VALUE
(Ω)
GAIN
(V/V)
VO/VI
GAIN
(dB)
VO/VI
1:1.4
250
2.8
9
214
0
1:1.4
125
5.6
15
375
0
1:1.4
100
7
17
600
0
1:2
250
4
12
750
0
1:2
125
6.7
16.5
Open
25
1:2
100
6.7
16.5
Open
50
EXTERNAL EXTERNAL
RT VALUE RS VALUE
(Ω)
(Ω)
This input interface also simplifies the input common mode
control. The VCM pin controls the output common mode voltage.
In most DC-coupled FDA applications, the input common mode
voltage is determined by both this output common mode and the
source signal. In a configuration like Figure 29, there is no path
for a common mode current to flow from output to input, so the
input common mode voltage equals the output. A similar effect
could be achieved with just two blocking caps on the two RG
resistors. A DC-coupled, single to differential, configuration will
also have a common mode input that is moving with the input
signal. Converting to just a differential signal at the amplifier, as
in Figure 29, removes any input signal related artifacts from the
input common mode making the ISL55211 behave as a
differential only VFA amplifier. There is only a very small
differential error signal at the inputs set by the loop gain, as in a
12
FN7868.0
June 21, 2011
ISL55211
normal single-ended VFA application, but no common mode
signal related terms.
Amplifier I/O Range Limits
maximum differential VP-P swing will be 4x this 0.6V
single-ended limit or 2.4VP-P. Where +Vs is increased, the limit
then becomes the 0.9V below VCM, but then the absolute
maximum differential VP-P is then 4 x 0.9V to 3.6VP-P. So for
instance, to get this maximum output swing, increase the supply
voltage until +Vs - 1.5V > VCM + 0.9V. If we assume a VCM voltage
of 1.3V for instance, then 1.3V + 0.9V + 1.5V = 3.7V will give an
unclipped 3.6VP-P output capability. The VP-P reported in
Figure 27 is an asymmetrically clipped maximum swing. Going
10% above this 3.7V target to 4.1V will be within the
recommended operating range and give some tolerancing
headroom that would also suggest the VCM voltage be moved up
to approximately 1.5V, which coincides with the default output
VCM from Figure 27. Operating at +4.1V single supply in a
Figure 29 type configuration will give the maximum linear
differential output swing of 3.6VP-P.
The ISL55211 is intended principally to give the lowest IM3
performance on the lowest power for a differential I/O
application. The amplifier will work DC coupled and over a
relatively wide supply range of 3.0V to 4.2V supplies. The outputs
have both a differential and common mode operating range
limits while the input pins internal to the ISL55211 have a
common mode voltage operating range. For single supply
operation, the -Vs pins are at ground as is the exposed metal pad
on the underside of the package. The ISL55211 can operate split
supply where then -Vs will be a negative supply voltage and the
exposed metal pad is either connected to this negative supply or
left unconnected on an insulating board layer.
The differential inputs internal to the ISL55211 also have
operating range limits relative to the supply voltages. Operating
in an AC-coupled circuit like Figure 29 will produce an input
common mode voltage equal to the outputs. The inputs can
operate with full linearity with this VCM voltage down to 1.1V
above the -Vs supply. On the default 1.2V output VCM on +3.3V
supplies this gives a 100mV guardband on the input VCM
voltages. Overriding the default VCM by applying a control voltage
to the VCM pin should be done with care in going towards the
negative supply due to this limit. On the + side, the maximum
input VCM above the -Vs supply is 2V so there is more room to
move the output VCM up than down from the default value.
Briefly, the I/O and VCM limits are as follows:
Power Supply, Shutdown, and Thermal
Considerations
The examples shown are using the transformer to convert from
single to differential. However, if the source is already
differential, these same transformer input circuits can drive the
transformer differentially still providing impedance scaling if
needed and common mode rejection for both DC and AC
common mode issues. A good example would be differential
mixer outputs or SAW filter outputs. Those differential sources
could also be connected into the ISL55211 RG resistors through
blocking caps as well eliminating the input transformer. The AC
termination impedance for the differential source will then be
the sum of the two RG resistors when simple blocking caps are
used.
1. Maximum VCM setting = -Vs +2V
2. Input common mode operating range (internal summing
junction pints of the ISL55211) of -Vs + 1.1V or to output VCM
+ 0.5V
3. Output VO minimum (on each side) is either -Vs + 0.3V or
output VCM - 0.9V
4. Output VO maximum (on each side) is +Vs - 1.5V
The output swing limits are often asymmetrical around the VCM
voltage. The maximum single-ended swings are set by these two
limits - VO(MIN) is either -Vs + 0.3V or VCM - 0.9V, whichever is
less. So for instance, on a single 3.3V supply with the default VCM
voltage of 1.2V, these two limits give the same result and the
output pins can swing down to 0.3V above -Vs (= 0V). If, however,
the VCM pin is raised to 1.5V, then the minimum output voltage
will become 1.5V - 0.9V = 0.6V.
VO(MAX) is set by a headroom limit to the positive supply to be
-VO(MAX) = +Vs - 1.5V. Again, on a 3.3V single supply and the
default 1.2V VCM setting, this means the maximum referenced to
ground output pin voltages can be 3.3V - 1.5V = +1.8V or 0.6V
above the default VCM voltage.
Using these default conditions, and the maximum positive
excursion of 0.6V above the 1.2V output VCM setting, the
13
The ISL55211 is intended for single supply operation from 3.0V
to 4.2V with an absolute maximum setting of 4.5V. The 3.3V
supply current is trimmed to be nominally 35mA at +25°C
ambient. Figure 28 shows the supply current for nominal +25°C
and -40°C to +85°C operation over the specified maximum
supply range. The input stage is biased from an internal voltage
reference from the negative supply giving the exceptional 90dB
low frequency PSRR shown in Figure 26.
Since the input stage bias is from a re-regulated internal supply,
a simple approach to single +5V operation can be supported as
shown in Figure 33. Here, a simple IR drop from the +5V supply
will bring the operating supply voltage for the ISL55211 into its
allowed range. Figure 33 shows example calculations for the
voltage range at the ISL55211 +Vs pin assuming a ±5%
tolerance on the +5V supply and a 35mA to 55mA range on the
total supply current. Considering the 34mA to 44mA quiescent
current range from Figure 28 over the -40°C to +85°C ambient,
and the 3.4V to 4.4V supply voltage range assumed here, this is
designing for a 1mA to 11mA average load current, which should
be adequate for most intended application loads. Good supply
decoupling at the device pins is required for this simple solution
to still provide exceptional HD performance.
FN7868.0
June 21, 2011
ISL55211
+85°C maximum operating ambient from Figure 27, we get
4.5V*45mA*+120°C/W = +24°C rise above +85°C or
approximately +109°C operating TJ maximum - still well below the
specified Absolute Maximum operating junction temperature of
+135°C.
+5V ±5%
35
24.3
3.4
55mA
4.4V
+ 2.2µF
10nF
Noise Analysis
10k
RF
PD
+
Vi
RO
RG
C IN 1:n
V CM
VO
RG
RO
RF
The decompensated voltage feedback design of the ISL55211
provides very low input voltage and current noise. Based on the
ISL55210, these internal noise terms are 0.85nV/√Hz differential
voltage noise and a 5pA/√Hz current noise term on each side. Since
the ISL55211 is an internally fixed gain version, these internal noise
terms will produce only a few set of output noise values. Figure 34
shows the analysis model for just the ISL55211 with no input
transformer while Table 3 shows the resulting output and input
referred differential spot noise voltages using Equation 1.
ISL55211
4kTR F
*
RF
FIGURE 33. OPERATING FROM A SINGLE +5V SUPPLY
500
The ISL55211 includes a power shutdown feature that can be used
to reduce system power dissipation when signal path operation is
not required. This pin (Pd) is referenced to -Vs and must be asserted
low to activate the shutdown feature. When not used, a 10kΩ
external resistor to +Vs should be used to assert a high level at this
pin. Digital control on this pin can be either an open collector output
(using that 10kΩ pull-up) or a CMOS logic line running off the same
+Vs as the amplifier. For split supply operation, the Pd pins must be
pulled to below -Vs + 0.54V to disable.
4kTR G
The very low internal power dissipation of the ISL55211, along with
the excellent thermal conductivity of the TQFN package when the
exposed metal pad is tied to a conductive plate, reduces the TJ rise
above ambient to very modest levels. Assuming a nominal 115mW
dissipation and using the 63°C/W measured thermal impedance
from Junction to ambient, gives a rise of only 0.115*63 = 7.2°C.
Operation at elevated ambient temperatures is easily supported
given this very low internal rise to junction.
The maximum internal junction temperatures would occur at
maximum supply voltage, +85°C maximum ambient operating,
and where the TQFN exposed pad is not tied to a conductive layer.
Where the TQFN must be mounted with an insulating layer to the
exposed metal plate, such as in a split supply application, device
measurements show an increased thermal impedance junction to
ambient of +120°C/W. Using this, and a maximum quiescent
internal power on 4.5V absolute maximum, which shows 45mA for
14
+
RG
en *
4kTR G R G
Since the ISL55211 operates as a differential inverting op amp,
there is only modest signal path isolation when disabled, as shown
in Figure 24. The inputs include 2 pairs of back to back low
capacitance diodes intended to protect any subsequent devices
from large input signals during shutdown. Those diodes limit the
maximum overdrive voltage across the input to approximately 1.0V
in each polarity. The internal RG resistors of Test Circuit 1 limit the
current into those diodes under this condition.
The supply current in shutdown does not reduce to zero as internal
circuitry is still active to hold the output common mode voltage at
the VCM voltage even during shutdown. This is intended to hold the
ISL55211 outputs near the desired common mode output level
during shutdown. This improves the turn on characteristic and keeps
those output voltages in a safe range for downstream circuitry.
iN *
eO
ISL55211
iN
*
500
*
4kTR F
RF
FIGURE 34. AMPLIFIER ONLY NOISE MODEL
With equal feedback and gain resistors, the total output noise
expression becomes very simple. This is shown as Equation 1.
(EQ. 1)
e 0 ( e N∗ NG ) 2 + 2 ( i N R F ) 2 + 2 ( 4kTR F NG )
The NG term in this equation is the Noise Gain = 1 + RF/RG. The
last term in Equation 1 captures both the RF and RG resistor
noise terms. Table 3 evaluates this expression for the 3 possible
internal gains with a fixed 500Ω internal feedback. nV/√HZ
TABLE 3. OUTPUT AND INPUT SPOT NOISE FROM EQUATION 1 FOR
THE 3 GAINS OF THE ISL55211
INPUT REFERRED
RG
(Ω)
GAIN
V/V
NOISE GAIN
V/V
EO
nV/√Hz
ENI
nV/√Hz
250
2
3
8.19
4.09
125
4
5
10.51
2.63
100
5
6
11.60
2.32
FN7868.0
June 21, 2011
ISL55211
TABLE 4. OUTPUT NOISE AND INPUT REFERRED EQUIVALENT NOISE FOR THE TRANSFORMER COUPLED INPUT
INPUT REFERRED
ISL55211 INTENDED TRANSFORMER + INTERNAL GAIN SETTINGS
INPUT XFMR
TURNS RATIO
INTERNAL
RG VALUE (Ω)
GAIN (V/V)
VO/VI
GAIN (dB)
VO/VI
EXTERNAL
RT VALUE (Ω)
TOTAL GAIN
RESISTOR FOR
NG (Ω)
NOISE GAIN
V/V
EO
nV/√Hz
ENI
nV/√Hz
1:1.4
250
2.8
9
122
277.48
2.80
7.94
2.834811
1:1.4
125
5.6
15
162
155.92
4.21
9.62
1.718338
1:1.4
100
7
17
192
132.88
4.76
10.25
1.46452
1:2
250
4
12
333
312.48
2.60
7.68
1.920066
1:2
125
8
18
1020
208.61
3.40
8.67
1.083876
1:2
100
10
20
1008
200.00
3.50
8.79
0.879492
Adding an input transformer can improve the input referred noise
by adding a noiseless voltage gain. Starting from Test Circuit 1 of
Figure 29, and assuming the source shows a matched
broadband source RS that will be matched by the input referred
parallel combination of 2*RG||RT, a noise gain analysis circuit
can be developed as shown in Figure 35.
RF
500
RT/2
RG
+
n2RS/2
ISL55211
n2RS/2
-
Driving Cap and Filter Loads
Most applications will drive a resistive or filter load. The
ISL55211 is robust to direct capacitive load on the outputs up to
approximately 10pF. For frequency response flatness, it is best to
avoid any output pin capacitance as much as possible - as the
capacitance increases, the high frequency portion of the
ISL55211 (>1GHz) response will start to show considerable
peaking. No oscillations were observed up through 10pF load on
each output.
For AC-coupled applications, an output network that is a small
series resistor (10 to 50Ω) into a blocking capacitor is preferred.
This series resistor will isolate parasitic capacitance to ground
from the internally closed loop output stage of the amplifier and
de-que the self resonance of the blocking capacitors. Once the
output stage sees this resistive element first, the remaining part
of a passive filter design can be done without fear of amplifier
instability.
Driving ADC's
RG
500
RT/2
RF
FIGURE 35. NOISE GAIN MODEL FOR THE TRANSFORMER
COUPLED INPUT CIRCUIT OF FIGURE 29
Stepping through the 3 gain settings with two input transformers
will allow the noise gain to be calculated for the circuit of
Figure 35, which is all that is needed in Equation 1 to arrive at an
output differential noise (since RF is fixed at 500Ω). Doing this
gives Table 4.
The signal gain is taken from the input of the transformer for this
analysis and shows the total input referred noise going below
0.9nV at the highest gain setting here. While this analysis is
including the approximate 0.9nV noise of a 50Ω source R, that
noise is assumed to be divided down by 2 to the input of the
transformer, which explains the total input referred noise
showing up as less than just a 50Ω resistor. The total output
differential noise goes below 9nV/√Hz at the higher gains
settings using this input transformer technique. For even lower
noise, consider the ISL55210 where the input RT element is
generally not required. In that case, simply setting RG to the
desired input Z and adjusting RF to the desired gain will give an
output noise that is slightly lower than shown previously for the
same input transformer due to the removal of the RT element.
15
Many of the intended applications for the ISL55211 are as a low
power, very high dynamic range, last stage interface to high
performance ADC's. The lowest power ADC's, such as the
ISLA214P50 shown on the front page, include an innovative
"Femto-Charge™" internal architecture that eliminates op amps
from the ADC design and only passes signal charge from stage to
stage. This greatly reduces the required quiescent power for
these ADC's but then that signal charge has to be provided by the
external circuit at the two input pins. This appears on an ADC like
the ISLA112P50 as a clock rate dependent common mode input
current that must be supplied by the interface circuit. At
500MHz, this DC current is 1.3mA on each input for the 14-bit
ISLA214P50.
Most interfaces will also include an interstage noise power
bandlimiting filter between the amplifier and the ADC. This filter
needs to be designed considering the loading of the amplifier,
any VCM level shifting that needs to take place, the filter shape,
and this Icm issue into the ADC input pins. Here are 4 example
topologies suitable for different situations.
1. AC-coupled, broadband RLC interstage filter design. This
approach lets the amplifier operate at its desired output
common mode, then provides the ADC common mode
voltage and current through a bias path as part of the filters
designs last stage R values. The Vb is set to include the IR loss
from that voltage to the ADC inputs due to the ICM current.
FN7868.0
June 21, 2011
ISL55211
ADC
LS
+3.3V
RT
VCM1
IN+
CB
RS
ISL55211
ICM
CT
RIN
VB
1.2V
CB
RS
LS
RT
CIN
CT
3. AC-coupled with output side transformer. This design includes
an output side transformer, very similar to ADC
characterization circuits. This approach allows a slightly lower
amplifier output swing (if N>1 is used) and very easy 2nd or
3rd order low pass responses to be implemented. It also
provides the ICM and VCM bias to the ADC through the
transformer centertap. This approach would be attractive for
higher ADC input swing targets and more aggressive noise
power bandwidth control needs. Figure 1 on page 1 is an
example showing this approach.
IN-
Rt >Rs
Vb −Icm ×Rt =Vcm2
ICM
ADC
+3.3V
VCM2
ICM
CB
RS
ISL55211
IN+
RT
1:N
CT
FIGURE 36. AC- COUPLED BROADBAND RLC INTERSTAGE FILTER
DESIGN
2. AC-coupled, higher frequency range interstage filter design.
This design replaces the RT resistors in Figure 35 with large
valued inductors and implements the filter just using shunt
resistors at the end of the RLC filter. In this case, the ADC VCM
can be tied to the centerpoint of the bias path inductors (very
much like a Bias-T) to provide the common mode voltage and
current to the ADC inputs. These bias inductors do limit the
low frequency end of the operation where, with 1µH values,
operation from 10MHz to 200MHz is supported using the
approach of Figure 37.
ADC
LS
+3.3V
RS
ISL55211
VCM1
ICM
CB
LP
IN+
RS
CB
LS
RIN
1.2V
RIN
CT
RT
INICM
RT<30
VCM2
2ICM
FIGURE 38. AC-COUPLED WITH OUTPUT SIDE TRANSFORMER
4. DC-coupled with ADC VCM and ICM provided from the
amplifier. Here, DC to very high frequency interstage low pass
filter can be provided. Again, the RS element must be low to
reduce the IR drop from the VCM of the converter, which now
shows up on the output of the ISL55211, to the ADC input
pins.
ADC
CIN
LS
+3.3V
LP
CT
ICM
Lp >>Ls
IN+
RS
ISL55211
INICM
CIN
CB
RS
CT
RT
1.2V
VCM1
CT
VCM2
FIGURE 37. AC-COUPLED, HIGHER FREQUENCY RLC INTERSTAGE
FILTER DESIGN
RT
RIN
VCM
RS
LS
CIN
CT
IN-
Rs ≤30Ω
ICM
VCM2
FIGURE 39. DC-COUPLED WITH A VCM VOLTAGE FROM THE ADC
16
FN7868.0
June 21, 2011
ISL55211
Layout Considerations
The ISL55211 pinout is organized to isolate signal I/O along one
axis of the package with ground, power and control pins on the
other axis. Ground and power should be planes coming into the
upper and lower sides of the package (see “Pin Configuration” on
page 2). The signal I/O should be laid out as tight as possible.
The ground pins and package backside metal contact should be
connected into a good ground plane. The power supply should
have both a large values electrolytic cap to ground, then a high
frequency ferrite beads, then 0.01µF SMD ceramic caps at the
supply pins. Some improvement in HD2 performance may be
experienced by placing and X2Y cap between the two Vs+ pins
and ground underneath the package on the board back side. This
is 3 terminal device that is included in the Evaluation board
layout.
Evaluation Board (Rev. C)
Test circuit 1 (Figure 29) is implemented on an Evaluation Board
available from Intersil. This board includes a number of optional
features that not populated as the board is delivered. The full
Evaluation circuit is shown in Figure 40 where unloaded
(optional) elements are shown in green.
The nominal supply voltage for the board and device is a single
3.3V supply. From this, the ISL55210, ISL55211 generates an
internal common mode voltage of approximately 1.2V. That
voltage can be overridden by populating the two resistors and
potentiometer shown as R19 to R21 above.
17
The primary test purpose for this board is to implement different
interstage differential passive filters intended for the ADC
interface along with the ADC input impedances. The board is
delivered with only the output R's loaded to give a 200Ω
differential load. This is done using the two 85Ω resistors as R9
and R10, then the 4 0Ω elements (R10, R12, R24, and R25) and
finally the two shunt elements R13 and R14 set to 35.5Ω.
Including the 50Ω measurement load on the output side of the
1:1 transformer reflecting in parallel with the two 35Ω resistors
takes the nominal AC shunt impedance to 71Ω||50Ω = 29.3Ω.
This adds to the two 85Ω series output elements to give a total
load across the amplifier outputs of 170Ω + 29.3Ω = 199.3Ω.
To test a particular ADC interface RLC filter and converter input
impedance, replace R11 and R12 with RF chip inductors, load
C10 and C11 with the specified ADC input capacitance and R26
with the specified ADC differential input R. With these loaded,
the remaining resistive elements (R24, R25, R13, R14) are set to
hit a desired total parallel impedance to implement the desired
filter (must be < than the ADC input differential R since that sits
in parallel with any "external" elements) and achieve a 250Ω
source looking into each side of the tap point transformer.
This Evaluation board includes a user's manual showing a
number of example circuits and tested results and is available on
the Intersil web site on the ISL55211 Product Information Page.
FN7868.0
June 21, 2011
+Vs
C1001 + BEAD
4.7uF
GND
R21
VCC
L1
200ohm/DNP
C3
C1002
1.0uF
C2
100nF
R19
1k/DNP
100nf
R17
R18
200ohm
R1001
R1002
R1003
R1004
R1005
0ohm/DNP 0ohm/DNP 0ohm/DNP 0ohm/DNP 0ohm/DNP
50ohm
C9
100nf
TEST POINT
R20
200ohm/DNP
C10
DNP
13
15
14
R26
DNP
TP1
18
16
C11
DNP
R70ohm
0ohm
R8DNP
Cterm2
2.2pF
4
GND
Vs+
Vcm
NC
Vi+
NC
Fb-
Vo-
11
1uf
C6
9
1uf
85ohm
R11
0ohm
R24
0ohm
0ohm/DNP
R10
R12
R25
85ohm
0ohm
0ohm
R15
50ohm
C5
100nf
TP2
DIFPROBE
R16
100nf
50ohm
Pd
1
R22
50ohm
2
3
NC
VCC
5
A
GND
Y
R13
35.5ohm
C8
4
FN7868.0
June 21, 2011
74AHC1G04
FIGURE 40. ISL55210, ISL55211 SINGLE INPUT TRANSFORMER EVALUATION BOARD REV C
R14
35.5ohm
OUT
ADT1-1WT
R28
ISL55210/11
C4
R1006
R1007
R1008
R1009
R1010
R1011
0ohm/DNP 0ohm/DNP 0ohm/DNP 0ohm/DNP 0ohm/DNP 0ohm/DNP
R9
10
GND
R4
Vi-
C7
12
8
3
Vo+
Pd
R2
DNP
2
Fb+
1uf
R27
0ohm
ISL55211
R23
0ohm
R60ohm
1
Vs+
0ohm
R0
162ohm
1uF
R5DNP
7
R3
GND
ADT2-1T
6
C1
5
IN
Cterm1
2.2pF
GND
U1
R1
DNP
ISL55211
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make
sure you have the latest revision.
DATE
REVISION
June 21, 2011
FN7868.0
CHANGE
Initial Release
Products
Intersil Corporation is a leader in the design and manufacture of high-performance analog semiconductors. The Company's products
address some of the industry's fastest growing markets, such as, flat panel displays, cell phones, handheld products, and notebooks.
Intersil's product families address power management and analog signal processing functions. Go to www.intersil.com/products for a
complete list of Intersil product families.
*For a complete listing of Applications, Related Documentation and Related Parts, please see the respective device information page
on intersil.com: ISL55211
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For additional products, see www.intersil.com/product_tree
Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted
in the quality certifications found at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time
without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be
accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
19
FN7868.0
June 21, 2011
ISL55211
Package Outline Drawing
L16.3x3D
16 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 0, 3/10
4X 1.50
3.00
A
12X 0.50
B
13
6
PIN 1
INDEX AREA
16
6
PIN #1
INDEX AREA
12
3.00
1
1.60 SQ
4
9
(4X)
0.15
0.10 M C A B
5
8
16X 0.40±0.10
TOP VIEW
4 16X 0.23 ±0.05
BOTTOM VIEW
SEE DETAIL “X”
0.10 C
0.75 ±0.05
C
0.08 C
SIDE VIEW
(12X 0.50)
(2.80 TYP) (
1.60)
(16X 0.23)
C
0 . 2 REF
5
0 . 02 NOM.
0 . 05 MAX.
(16X 0.60)
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES:
1.
Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2.
Dimensioning and tolerancing conform to ASME Y14.5m-1994.
3.
Unless otherwise specified, tolerance : Decimal ± 0.05
4.
Dimension applies to the metallized terminal and is measured
between 0.15mm and 0.25mm from the terminal tip.
5.
Tiebar shown (if present) is a non-functional feature.
6.
The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
7.
JEDEC reference drawing: MO-220 WEED.
either a mold or mark feature.
20
FN7868.0
June 21, 2011