LINER LT1511 Constant-current/ constant-voltage 3a battery charger with input current limiting Datasheet

LT1511
Constant-Current/
Constant-Voltage 3A Battery
Charger with Input Current Limiting
DESCRIPTIO
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FEATURES
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The LT®1511 current mode PWM battery charger is the
simplest, most efficient solution to fast charge modern
rechargeable batteries including lithium-ion (Li-Ion), nickelmetal-hydride (NiMH) and nickel-cadmium (NiCd) that
require constant-current and/or constant-voltage charging. The internal switch is capable of delivering 3A* DC
current (4A peak current). Full-charging current can be
programmed by resistors or a DAC to within 5%. With 0.5%
reference voltage accuracy, the LT1511 meets the critical
constant-voltage charging requirement for Li-Ion cells.
Simple Design to Charge NiCd, NiMH and Lithium
Rechargeable Batteries—Charging Current
Programmed by Resistors or DAC
Adapter Current Loop Allows Maximum Possible
Charging Current During Computer Use
Precision 0.5% Accuracy for Voltage Mode Charging
High Efficiency Current Mode PWM with 4A Internal
Switch
5% Charging Current Accuracy
Adjustable Undervoltage Lockout
Automatic Shutdown When AC Adapter is Removed
Low Reverse Battery Drain Current: 3µA
Current Sensing Can Be at Either Terminal of the Battery
Charging Current Soft-Start
Shutdown Control
A third control loop is provided to regulate the current
drawn from the AC adapter. This allows simultaneous
operation of the equipment and battery charging without
overloading the adapter. Charging current is reduced to
keep the adapter current within specified levels.
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APPLICATIO S
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The LT1511 can charge batteries ranging from 1V to 20V.
Ground sensing of current is not required and the battery’s
negative terminal can be tied directly to ground. A saturating switch running at 200kHz gives high charging efficiency and small inductor size. A blocking diode is not
required between the chip and the battery because the
chip goes into sleep mode and drains only 3µA when the
wall adapter is unplugged.
Chargers for NiCd, NiMH, Lead-Acid, Lithium
Rechargeable Batteries
Switching Regulators with Precision Current Limit
, LTC and LT are registered trademarks of Linear Technology Corporation.
*See LT1510 for 1.5A Charger
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TYPICAL APPLICATIO
R7†
500Ω
BOOST
LT1511
COMP1
L1**
20µH
D2
MBR0540T
CLN
VCC
SW
C2
0.47µF
D1
MBRD340
CLP
GND
200pF
+
+
10µF
RS4†
ADAPTER
CURRENT SENSE
CIN*
10µF
UV
VC
SPIN
RS3
200Ω
1%
BAT
RS2
200Ω
1%
RS1
0.033Ω
BATTERY CURRENT
SENSE
50pF
VIN (ADAPTER INPUT)
11V TO 28V
TO MAIN
SYSTEM POWER
R5†
UNDERVOLTAGE
LOCKOUT
R6
5k
PROG
OVP SENSE
NOTE: COMPLETE LITHIUM-ION CHARGER,
NO TERMINATION REQUIRED. RS4, R7
AND C1 ARE OPTIONAL FOR IIN LIMITING
*TOKIN OR UNITED CHEMI-CON/MARCON
CERAMIC SURFACE MOUNT
**20µH COILTRONICS CTX20-4
†SEE APPLICATIONS INFORMATION FOR
INPUT CURRENT LIMIT AND UNDERVOLTAGE LOCKOUT
C1
1µF
D3
MBRD340
300Ω
1k
CPROG
1µF
RPROG
4.93k
1%
0.33µF
R3
390k
0.25%
BATTERY
VOLTAGE SENSE
+
COUT
22µF
TANT
+
4.2V
+
VBAT
2 Li-Ion
4.2V
1511 • F01
R4
162k
0.25%
Figure 1. 3A Lithium-Ion Battery Charger
1
LT1511
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
Supply Voltage
(VMAX, CLP and CLN Pin Voltage) ...................... 30V
Switch Voltage with Respect to GND ...................... – 3V
Boost Pin Voltage with Respect to VCC ................... 25V
Boost Pin Voltage with Respect to GND ................. 57V
Boost Pin Voltage with Respect to SW Pin .............. 30V
VC, PROG, OVP Pin Voltage ...................................... 8V
IBAT (Average)........................................................... 3A
Switch Current (Peak) .............................................. 4A
Operating Junction Temperature Range
Commercial ........................................... 0°C to 125°C
Industrial ......................................... – 40°C to 125°C
Operating Ambient Temperature
Commercial ............................................ 0°C to 70°C
Industrial ........................................... – 40°C to 85°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
GND** 1
24 GND**
SW 2
23 GND**
BOOST 3
22 VCC1*
GND** 4
21 VCC2*
GND** 5
20 VCC3*
UV 6
19 PROG
GND** 7
LT1511CSW
LT1511ISW
*ALL VCC PINS SHOULD
BE CONNECTED
TOGETHER CLOSE TO
THE PINS
** ALL GND PINS ARE
FUSED TO INTERNAL DIE
ATTACH PADDLE FOR
HEAT SINKING. CONNECT
THESE PINS TO
EXPANDED PC LANDS
FOR PROPER HEAT
SINKING. 30°C/W
THERMAL RESISTANCE
ASSUMES AN INTERNAL
GROUND PLANE
DOUBLING AS A HEAT
SPREADER
18 VC
OVP 8
17 UVOUT
CLP 9
16 GND**
CLN 10
15 COMP2
COMP1 11
14 BAT
SENSE 12
13 SPIN
SW PACKAGE
24-LEAD PLASTIC SO WIDE
TJMAX = 125°C, θJA = 30°C/ W**
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, VMAX (maximum operating VCC) = 28V,
RS2 = RS3 = 200Ω (see Block Diagram), VCLN = VCC. No load on any outputs unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
4.5
4.6
6.8
7.0
mA
mA
100
10
105
12
13
mV
mV
mV
110
13
14
mV
mV
mV
Overall
Supply Current
VPROG = 2.7V, VCC ≤ 20V
VPROG = 2.7V, 20V < VCC ≤ 25V
Sense Amplifier CA1 Gain and Input Offset Voltage
(With RS2 = 200Ω, RS3 = 200Ω)
(Measured across RS1)(Note 2)
8V ≤ VCC ≤ 25V , 0V ≤ VBAT ≤ 20V
RPROG = 4.93k
RPROG = 49.3k
TJ < 0°C
VCC = 28V, VBAT = 20V
RPROG = 4.93k
RPROG = 49.3k
TJ < 0°C
●
●
●
●
95
8
7
●
●
90
7
6
6
VCC Undervoltage Lockout (Switch OFF) Threshold
Measured at UV Pin
●
7
8
V
UV Pin Input Current
0.2V ≤ VUV ≤ 8V
●
0.1
5
µA
UV Output Voltage at UVOUT Pin
In Undervoltage State, IUVOUT = 70µA
●
0.1
0.5
V
UV Output Leakage Current at UVOUT Pin
8V ≤ VUV, VUVOUT = 5V
●
0.1
3
µA
Reverse Current from Battery (When VCC Is
Not Connected, VSW Is Floating)
VBAT ≤ 20V, VUV ≤ 0.4V
3
15
µA
2
LT1511
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, VMAX (maximum operating VCC) = 28V,
RS2 = RS3 = 200Ω (see Block Diagram), VCLN = VCC. No load on any outputs unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0.1
0.25
6
8
10
20
9
12
µA
µA
mA
mA
0.15
0.25
Ω
25
35
mA/A
2
4
100
200
µA
µA
20
µA
Overall
Boost Pin Current
VCC = 20V, VBOOST = 0V
VCC = 28V, VBOOST = 0V
2V ≤ VBOOST – VCC < 8V (Switch ON)
8V ≤ VBOOST – VCC ≤ 25V (Switch ON)
Switch
Switch ON Resistance
8V ≤ VCC ≤ VMAX, ISW = 3A,
VBOOST – VSW ≥ 2V
∆IBOOST/∆ISW During Switch ON
VBOOST = 24V, ISW ≤ 3A
Switch OFF Leakage Current
VSW = 0V, VCC ≤ 20V
20V < VCC ≤ 28V
●
●
●
Minimum IPROG for Switch ON
●
2
4
Minimum IPROG for Switch OFF at VPROG ≤ 1V
●
1
2.4
Maximum VBAT for Switch ON
●
mA
VCC – 2
V
– 125
µA
Current Sense Amplifier CA1 Inputs (Sense, BAT)
Input Bias Current
●
Input Common Mode Low
●
Input Common Mode High
●
– 50
– 0.25
SPIN Input Current
V
VCC – 2
V
– 100
– 200
µA
2.465
2.477
V
2.489
2.489
V
V
Reference
Reference Voltage (Note 3)
Reference Voltage
RPROG = 4.93k, Measured at OVP with
VA Supplying IPROG and Switch OFF
All Conditions of VCC, TJ > 0°C
TJ < 0°C (Note 4)
2.453
●
●
2.441
2.43
Oscillator
Switching Frequency
Switching Frequency
180
200
220
kHz
●
●
170
160
200
230
230
kHz
kHz
●
TA = 25°C
85
90
93
VC = 1V, IVC = ±1µA
150
250
All Conditions of VCC, TJ > 0°C
TJ < 0°C
Maximum Duty Cycle
%
%
Current Amplifier CA2
Transconductance
Maximum VC for Switch OFF
IVC Current (Out of Pin)
●
VC ≥ 0.6V
VC < 0.45V
550
µmho
0.6
V
100
3
µA
mA
3
LT1511
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, VMAX (maximum operating VCC) = 28V.
No load on any outputs unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Transconductance (Note 3)
Output Current from 50µA to 500µA
0.25
0.6
1.3
mho
Output Source Current
VOVP = VREF + 10mV, VPROG = VREF + 10mV
1.1
OVP Input Bias Current
At 0.75mA VA Output Current
At 0.75mA VA Output Current, TJ > 90°C
– 15
Turn-On Threshold
0.75mA Output Current
Transconductance
Output Current from 50µA to 500µA
CLP Input Current
CLN Input Current
Voltage Amplifier VA
mA
±3
±10
25
nA
nA
93
100
107
mV
0.5
1
2
mho
0.75mA Output Current, VUV ≥ 0.4V
0.3
1
µA
0.75mA Output Current VUV ≥ 0.4V
0.8
2
mA
Current Limit Amplifier CL1, 8V ≤ Input Common Mode
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: Tested with Test Circuit 1.
Note 3: Tested with Test Circuit 2.
Note 4: A linear interpolation can be used for reference voltage
specification between 0°C and – 40°C.
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TYPICAL PERFORMANCE CHARACTERISTICS
Thermally Limited Maximum
Charging Current
8
2.8
96
16.8V BATTERY
92
CHARGER EFFICIENCY
90
88
(θJA =30°C/W)
TAMAX =60°C
TJMAX =125°C
5
10
20
15
INPUT VOLTAGE (V)
INCLUDES LOSS
IN DIODE D3
84
25
30
1511 • TPC01
NOTE: FOR 4.2V AND 8.4V BATTERIES MAXIMUM
CHARGING CURRENT IS 3A FOR VIN – VBAT ≥ 3V
0°C
125°C
4
25°C
2
1
82
2.0
5
3
86
2.2
ICC (mA)
94
4.2V BATTERY
VIN ≥ 8V
2.4
7
6
8.4V BATTERY
VIN ≥ 11V
2.6
VCC = 16V
VIN = 16.5
VBAT = 8.4V
98
EFFICIENCY (%)
MAXIMUM CHARGING CURRENT (A)
100
12.6V BATTERY
4
ICC vs Duty Cycle
Efficiency of Figure 1 Circuit
3.0
0
80
0.2
0.6
1.0
1.4 1.8
IBAT (A)
2.2
2.6
3.0
1511 • TPC02
0
10
20
30 40 50 60
DUTY CYCLE (%)
70
80
1511 • TPC03
LT1511
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TYPICAL PERFORMANCE CHARACTERISTICS
Switching Frequency vs
Temperature
VREF Line Regulation
ICC vs VCC
7.0
210
0.003
MAXIMUM DUTY CYCLE
0.002
0°C
6.5
25°C
200
195
0.001
6.0
∆VREF (V)
ICC (mA)
FREQUENCY (kHz)
205
125°C
5.5
ALL TEMPERATURES
0
–0.001
190
5.0
185
180
–20
–0.002
4.5
0
20
40 60 80 100 120 140
TEMPERATURE (°C)
0
10
5
15
VCC (V)
20
25
30
–0.003
0
5
10
15
VCC (V)
20
1511 • TPC05
1511 • TPC04
IVA vs ∆VOVP (Voltage Amplifier)
30
1511 • TPC06
Maximum Duty Cycle
4
25
VC Pin Characteristics
98
–1.20
–1.08
97
–0.96
2
125°C
1
96
–0.84
95
–0.72
IVC (mA)
DUTY CYCLE (%)
94
–0.48
–0.36
92
–0.24
–0.12
25°C
91
0
90
0
0
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
IVA (mA)
20
40
60
80
100
120
0.12
140
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
VC (V)
1511 • TPC08
1511 • TPC09
Reference Voltage
vs Temperature
Switch Current vs Boost Current
vs Boost Voltage
PROG Pin Characteristics
50
6
45
BOOST CURRENT (mA)
40
125°C
25°C
0
0
TEMPERATURE (°C)
1511• TPC07
IPROG (mA)
–0.60
93
35
2.470
VCC = 16V
2.468
VBOOST = 38V
28V
18V
REFERENCE VOLTAGE (V)
∆VOVP (mV)
3
30
25
20
15
10
2.466
2.464
2.462
2.460
5
–6
0
1
2
3
VPROG (V)
4
5
1511 • TPC10
0
2.458
0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
SWITCH CURRENT (A)
1511 • TPC11
0
25
50
75
100
TEMPERATURE
125
150
LT1511 • TPC12
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LT1511
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PIN FUNCTIONS
GND (Pins 1, 4, 5, 7, 16, 23, 24): Ground Pin.
SW (Pin 2): Switch Output. The Schottky catch diode must
be placed with very short lead length in close proximity to
SW pin and GND.
BOOST (Pin 3): This pin is used to bootstrap and drive the
switch power NPN transistor to a low on-voltage for low
power dissipation. In normal operation, VBOOST = VCC +
VBAT when switch is on. Maximum allowable VBOOST is
55V.
UV (Pin 6): Undervoltage Lockout Input. The rising threshold is at 6.7V with a hysteresis of 0.5V. Switching stops in
undervoltage lockout. When the supply (normally the wall
adapter output) to the chip is removed, the UV pin has to
be pulled down to below 0.7V (a 5k resistor from adapter
output to GND is required) otherwise the reverse battery
current drained by the chip will be approximately 200µA
instead of 3µA. Do not leave UV pin floating. If it is
connected to VIN with no resistor divider, the built-in 6.7V
undervoltage lockout will be effective.
OVP (Pin 8): This is the input to the amplifier VA with a
threshold of 2.465V. Typical input current is about 3nA out
of pin. For charging lithium-ion batteries, VA monitors the
battery voltage and reduces charging when battery voltage
reaches the preset value. If it is not used, the OVP pin
should be grounded.
CLP (Pin 9): This is the positive input to the supply current
limit amplifier CL1. The threshold is set at 100mV. When
used to limit supply current, a filter is needed to filter out
the 200kHz switching noise.
CLN (Pin 10): This is the negative input to the amplifier
CL1.
COMP1 (Pin 11): This is the compensation node for the
amplifier CL1. A 200pF capacitor is required from this pin
to GND if input current amplifier CL1 is used. At input
adapter current limit, this node rises to 1V. By forcing
COMP1 low with an external transistor, amplifier CL1 will
be defeated (no adapter current limit). COMP1 can source
200µA.
6
SENSE (Pin 12): Current Amplifier CA1 Input. Sensing can
be at either terminal of the battery.
SPIN (Pin 13): This pin is for the internal amplifier CA1
bias. It has to be connected to RS1 as shown in the 3A
Lithium Battery Charger (Figure 1).
BAT (Pin 14): Current Amplifier CA1 Input.
COMP2 (Pin 15): This is also a compensation node for the
amplifier CL1. It gets up to 2.8V at input adapter current
limit and/or at constant-voltage charging.
UVOUT (Pin 17): This is an open collector output for
undervoltage lockout status. It stays low in undervoltage
state. With an external pull-up resistor , it goes high at valid
VCC. Note that the base drive of the open collector NPN
comes from CLN pin. UVOUT stays low only when CLN is
higher than 2V. Pull-up current should be kept under
100µA.
VC (Pin 18): This is the control signal of the inner loop of
the current mode PWM. Switching starts at 0.7V. Higher
VC corresponds to higher charging current in normal
operation. A capacitor of at least 0.33µF to GND filters out
noise and controls the rate of soft-start. To shut down
switching, pull this pin low. Typical output current is 30µA.
PROG (Pin 19): This pin is for programming the charging
current and for system loop compensation. During normal
operation, VPROG stays close to 2.465V. If it is shorted to
GND the switching will stop. When a microprocessor
controlled DAC is used to program charging current, it
must be capable of sinking current at a compliance up to
2.465V.
VCC (Pins 20, 21, 22): This is the supply of the chip. For
good bypass, a low ESR capacitor of 20µF or higher is
required, with the lead length kept to a minimum. VCC
should be between 8V and 28V and at least 3V higher than
VBAT. Undervoltage lockout starts and switching stops
when VCC goes below 7V. Note that there is a parasitic
diode inside from SW pin to VCC pin. Do not force VCC
below SW by more than 0.7V with battery present. All three
VCC pins should be shorted together close to the pins.
LT1511
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BLOCK DIAGRAM
–
UV
UVOUT
+
+
6.7V
200kHz
OSCILLATOR
+
SHUTDOWN
0.7V
+
VCC
–
VSW
S
BOOST
–
VCC
R
R
+
+
SLOPE COMPENSATION
SW
1.5V
VCC
SPIN
VBAT
–
PWM
C1
–
+
B1
+
GND
QSW
R
R2
+
SENSE
RS3
–
BAT
RS2
CA1
IPROG
R1
1k
R3
0VP
–
VA
CA2
+
75k
RS1
BAT
+
VC
IBAT
gm = 0.64
VREF
Ω
–
VREF
2.465V
100mV
+
CLP
+
CL1
–
CLN
COMP1
COMP2
PROG
CPROG
IPROG
RPROG
(I
)(R )
IBAT = PROG S2
RS1
RS2
= 2.465V
RPROG RS1
1511 BD
( )( )
(RS3 = RS2)
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LT1511
TEST CIRCUITS
Test Circuit 1
SPIN
LT1511
CA1
CA2
1k
+
0.047µF
+
–
VC
60k
–
SENSE
RS3
200Ω
BAT
RS2
200Ω
RS1
10Ω
+
VBAT
VREF
PROG
1µF
RPROG
300Ω
+
LT1006
1k
+
1511 • TC01
–
≈ 0.65V
20k
Test Circuit 2
LT1511
OVP
+
VA
–
VREF
PROG
10k
IPROG
10k
0.47µF
RPROG
–
+
+
LT1013
2.465V
1511 • TC02
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OPERATION
The LT1511 is a current mode PWM step-down (buck)
switcher. The battery DC charging current is programmed
by a resistor RPROG (or a DAC output current) at the PROG
pin (see Block Diagram). Amplifier CA1 converts the
charging current through RS1 to a much lower current
IPROG fed into the PROG pin. Amplifier CA2 compares the
output of CA1 with the programmed current and drives the
PWM loop to force them to be equal. High DC accuracy is
achieved with averaging capacitor CPROG. Note that IPROG
has both AC and DC components. IPROG goes through R1
and generates a ramp signal that is fed to the PWM control
comparator C1 through buffer B1 and level shift resistors
8
R2 and R3, forming the current mode inner loop. The
Boost pin drives the switch NPN QSW into saturation and
reduces power loss. For batteries like lithium-ion that
require both constant-current and constant-voltage charging, the 0.5%, 2.465V reference and the amplifier VA
reduce the charging current when battery voltage reaches
the preset level. For NiMH and NiCd, VA can be used for
overvoltage protection. When input voltage is not present,
the charger goes into low current (3µA typically) sleep
mode as input drops down to 0.7V below battery voltage.
To shut down the charger, simply pull the VC pin low with
a transistor.
LT1511
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APPLICATIONS INFORMATION
Input and Output Capacitors
Soft-Start
In the 3A Lithium Battery Charger (Figure 1), the input
capacitor (CIN) is assumed to absorb all input switching
ripple current in the converter, so it must have adequate
ripple current rating. Worst-case RMS ripple current will
be equal to one half of output charging current. Actual
capacitance value is not critical. Solid tantalum capacitors
such as the AVX TPS and Sprague 593D series have high
ripple current rating in a relatively small surface mount
package, but caution must be used when tantalum capacitors are used for input bypass. High input surge currents
can be created when the adapter is hot-plugged to the
charger and solid tantalum capacitors have a known
failure mechanism when subjected to very high turn-on
surge currents. Highest possible voltage rating on the
capacitor will minimize problems. Consult with the manufacturer before use. Alternatives include new high capacity
ceramic (5µF to 20µF) from Tokin or United Chemi-Con/
Marcon, et al., and the old standby, aluminum electrolytic,
which will require more microfarads to achieve adequate
ripple rating. Sanyo OS-CON can also be used.
The LT1511 is soft started by the 0.33µF capacitor on the
VC pin. On start-up, VC pin voltage will rise quickly to 0.5V,
then ramp at a rate set by the internal 45µA pull-up current
and the external capacitor. Battery charging current starts
ramping up when VC voltage reaches 0.7V and full current
is achieved with VC at 1.1V. With a 0.33µF capacitor, time
to reach full charge current is about 10ms and it is
assumed that input voltage to the charger will reach full
value in less than 10ms. The capacitor can be increased up
to 1µF if longer input start-up times are needed.
The output capacitor (COUT) is also assumed to absorb
output switching current ripple. The general formula for
capacitor current is:
(
)
V
0.29 (VBAT) 1 – BAT
VCC
IRMS =
(L1)(f)
For example, VCC = 16V, VBAT = 8.4V, L1 = 20µH,
and f = 200kHz, IRMS = 0.3A.
EMI considerations usually make it desirable to minimize
ripple current in the battery leads, and beads or inductors
may be added to increase battery impedance at the 200kHz
switching frequency. Switching ripple current splits between the battery and the output capacitor depending on
the ESR of the output capacitor and the battery impedance. If the ESR of COUT is 0.2Ω and the battery impedance
is rased to 4Ω with a bead or inductor, only 5% of the
current ripple will flow in the battery.
In any switching regulator, conventional timer-based soft
starting can be defeated if the input voltage rises much
slower than the time out period. This happens because the
switching regulators in the battery charger and the computer power supply are typically supplying a fixed amount
of power to the load. If input voltage comes up slowly
compared to the soft start time, the regulators will try to
deliver full power to the load when the input voltage is still
well below its final value. If the adapter is current limited,
it cannot deliver full power at reduced output voltages and
the possibility exists for a quasi “latch” state where the
adapter output stays in a current limited state at reduced
output voltage. For instance, if maximum charger plus
computer load power is 30W, a 15V adapter might be
current limited at 2.5A. If adapter voltage is less than
(30W/2.5A = 12V) when full power is drawn, the adapter
voltage will be sucked down by the constant 30W load until
it reaches a lower stable state where the switching regulators can no longer supply full load. This situation can be
prevented by utilizing undervoltage lockout, set higher
than the minimum adapter voltage where full power can be
achieved.
A fixed undervoltage lockout of 7V is built into the VCC pin,
but an additional adjustable lockout is also available on the
UV pin. Internal lockout is performed by clamping the VC
pin low. The VC pin is released from its clamped state when
the UV pin rises above 6.7V and is pulled low when the UV
pin drops below 6.2V (0.5V hysteresis). At the same time
UVOUT goes high with an external pull-up resistor. This
signal can be used to alert the system that charging is
about to start. The charger will start delivering current
about 4ms after VC is released, as set by the 0.33µF
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capacitor. A resistor divider is used to set the desired VCC
lockout voltage as shown in Figure 2. A typical value for R6
is 5k and R5 is found from:
R5 =
R6(VIN – VUV )
VUV
VUV = Rising lockout threshold on the UV pin
VIN = Charger input voltage that will sustain full load power
Example: With R6 = 5k, VUV = 6.7V and setting VIN at 12V;
R5 = 5k (12V – 6.7V)/6.7V = 4k
The resistor divider should be connected directly to the
adapter output as shown, not to the VCC pin to prevent
battery drain with no adapter voltage. If the UV pin is not
used, connect it to the adapter output (not VCC) and
connect a resistor no greater than 5k to ground. Floating
the pin will cause reverse battery current to increase from
3µA to 200µA.
If connecting the unused UV pin to the adapter output is
not possible for some reason, it can be grounded. Although it would seem that grounding the pin creates a
permanent lockout state, the UV circuitry is arranged for
phase reversal with low voltages on the UV pin to allow the
grounding technique to work.
100mV
+
+
CLP
1µF
CL1
CLN
–
500Ω
AC ADAPTER
OUTPUT
RS4*
VCC
VIN
+
LT1511
R5
being charged without complex load management algorithms. Additionally, batteries will automatically be charged at
the maximum possible rate of which the adapter is capable.
This feature is created by sensing total adapter output
current and adjusting charging current downward if a
preset adapter current limit is exceeded. True analog
control is used, with closed loop feedback ensuring that
adapter load current remains within limits. Amplifier CL1
in Figure 2 senses the voltage across RS4, connected
between the CLP and CLN pins. When this voltage exceeds
100mV, the amplifier will override programmed charging
current to limit adapter current to 100mV/RS4. A lowpass
filter formed by 500Ω and 1µF is required to eliminate
switching noise. If the current limit is not used, both CLP
and CLN pins should be connected to VCC.
Charging Current Programming
The basic formula for charging current is (see Block
Diagram):
IBAT = IPROG
( )(
RS2
2.465V
=
RPROG
RS1
)( )
RS2
RS1
where RPROG is the total resistance from PROG pin to ground.
For the sense amplifier CA1 biasing purpose, RS3 should
have the same value as RS2 and SPIN should be connected
directly to the sense resistor (RS1) as shown in the Block
Diagram.
For example, 3A charging current is needed. To have low
power dissipation on RS1 and enough signal to drive the
amplifier CA1, let RS1 = 100mV/3A = 0.033Ω. This limits
RS1 power to 0.3W. Let RPROG = 5k, then:
UV
*RS4 =
100mV
ADAPTER CURRENT LIMIT
R6
1511 • F02
Figure 2. Adapter Current Limiting
Adapter Limiting
An important feature of the LT1511 is the ability to
automatically adjust charging current to a level which
avoids overloading the wall adapter. This allows the
product to operate at the same time that batteries are
10
)(R )
(I )(R
RS2 = RS3 = BAT PROG S1
2.465V
(3A)(5k)(0.033)
=
= 200Ω
2.465V
Charging current can also be programmed by pulse width
modulating IPROG with a switch Q1 to RPROG at a frequency
higher than a few kHz (Figure 3). Charging current will be
proportional to the duty cycle of the switch with full current
at 100% duty cycle.
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LT1511
PROG
300Ω
RPROG
4.7k
5V
0V
Q1
VN2222
PWM
CPROG
1µF
1511 • F03
IBAT = (DC)(3A)
Figure 3. PWM Current Programming
When power is on, there is about 200µA of current flowing
out of the BAT and Sense pins. If the battery is removed
during charging, and total load including R3 and R4 is less
than the 200µA, VBAT could float up to VCC even though the
loop has turned switching off. To keep VBAT regulated to
the battery voltage in this condition, R3 and R4 can be
chosen to draw 0.5mA and Q3 can be added to disconnect
them when power is off (Figure 4). R5 isolates the OVP pin
from any high frequency noise on VIN. An alternative way is
to use a Zener diode with a breakdown voltage two or three
volts higher than battery voltage to clamp the VBAT voltage.
Lithium-Ion Charging
The 3A Lithium Battery Charger (Figure 1) charges lithiumion batteries at a constant 3A until battery voltage reaches
a limit set by R3 and R4. The charger will then automatically go into a constant-voltage mode with current decreasing to zero over time as the battery reaches full
charge. This is the normal regimen for lithium-ion charging, with the charger holding the battery at “float” voltage
indefinitely. In this case no external sensing of full charge
is needed.
+
R3
12k
0.25%
LT1511
OVP
Q3
VN2222
–
+
VIN
R5
220k
–
VBAT
4.2V
4.2V
R4
4.99k
0.25%
LT1511 • F04
Figure 4. Disconnecting Voltage Divider
Battery Voltage Sense Resistors Selection
To minimize battery drain when the charger is off, current
through the R3/R4 divider is set at 15µA. The input current
to the OVP pin is 3nA and the error can be neglected.
With divider current set at 15µA, R4 = 2.465/15µA = 162k
and,
R3 =
(R4)(VBAT − 2.465) = 162k (8.4 − 2.465)
2.465
2.465
= 390k
Li-Ion batteries typically require float voltage accuracy of
1% to 2%. Accuracy of the LT1511 OVP voltage is ±0.5%
at 25°C and ±1% over full temperature. This leads to the
possibility that very accurate (0.1%) resistors might be
needed for R3 and R4. Actually, the temperature of the
LT1511 will rarely exceed 50°C in float mode because
charging currents have tapered off to a low level, so 0.25%
resistors will normally provide the required level of overall
accuracy.
Some battery manufacturers recommend termination of
constant-voltage float mode after charging current has
dropped below a specified level (typically around 10% of
the full current) and a further time out period of 30 minutes
to 90 minutes has elapsed. This may extend the life of the
battery, so check with manufacturers for details. The
circuit in Figure 5 will detect when charging current has
dropped below 400mA. This logic signal is used to initiate
a timeout period, after which the LT1511 can be shut down
by pulling the VC pin low with an open collector or drain.
Some external means must be used to detect the need for
additional charging or the charger may be turned on
periodically to complete a short float-voltage cycle.
Current trip level is determined by the battery voltage, R1
through R3 and the sense resistor (RS1). D2 generates
hysteresis in the trip level to avoid multiple comparator
transitions.
11
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IBAT
R1 =
RS3
200Ω
SENSE
RS1
0.033Ω
LT1511
RS2
200Ω
BAT
VBAT
ADAPTER
OUTPUT
BAT
R1* C1
1.6k 0.1µF
3.3V OR 5V
D1
1N4148
3
8
–
7
LT1011
2
R2
560k
+
R4
470k
NEGATIVE EDGE
TO TIMER
4
1
D2
1N4148
R3
430k
* TRIP CURRENT =
=
R1(VBAT)
(R2 + R3)(RS1)
(1.6k)(8.4V)
≈ 400mA
(560k + 430k)(0.033Ω)
1511 • F04
Figure 5. Current Comparator for Initiating Float Time Out
Nickel-Cadmium and Nickel-Metal-Hydride Charging
The circuit in the 3A Lithium Battery Charger (Figure 1) can
be modified to charge NiCd or NiMH batteries. For example, 2-level charging is needed; 2A when Q1 is on and
200mA when Q1 is off.
PROG
0.33µF
R1
49.3k
R2
5.49k
Q1
1511 • F05
Figure 6. 2-Level Charging
For 2A full current, the current sense resistor (RS1) should
be increased to 0.05Ω so that enough signal (10mV) will
be across RS1 at 0.2A trickle charge to keep charging
current accurate.
For a 2-level charger, R1 and R2 are found from;
12
ILOW
R2 =
(2.465)(4000 )
IHI − ILOW
All battery chargers with fast charge rates require some
means to detect full charge state in the battery to terminate
the high charging current. NiCd batteries are typically
charged at high current until temperature rise or battery
voltage decrease is detected as an indication of near full
charge. The charging current is then reduced to a much
lower value and maintained as a constant trickle charge.
An intermediate “top off” current may be used for a fixed
time period to reduce 100% charge time.
NiMH batteries are similar in chemistry to NiCd but have
two differences related to charging. First, the inflection
characteristic in battery voltage as full charge is approached is not nearly as pronounced. This makes it more
difficult to use dV/dt as an indicator of full charge, and
change of temperature is more often used with a temperature sensor in the battery pack. Secondly, constant trickle
charge may not be recommended. Instead, a moderate
level of current is used on a pulse basis (≈ 1% to 5% duty
cycle) with the time-averaged value substituting for a
constant low trickle. Please contact the Linear Technology
Applications Department about charge termination circuits.
If overvoltage protection is needed, R3 and R4 should be
calculated according to the procedure described in LithiumIon Charging section. The OVP pin should be grounded if
not used.
LT1511
1k
(2.465)(4000)
When a microprocessor DAC output is used to control
charging current, it must be capable of sinking current at a
compliance up to 2.5V if connected directly to the PROG pin.
Thermal Calculations
If the LT1511 is used for charging currents above 1.5A, a
thermal calculation should be done to ensure that junction
temperature will not exceed 125°C. Power dissipation in
the IC is caused by bias and driver current, switch resistance and switch transition losses. The SO wide package,
with a thermal resistance of 30°C/W, can provide a full 3A
charging current in many situations. A graph is shown in
the Typical Performance Characteristics section.
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PBIAS = (3.5mA )(VIN) + 1.5mA(VBAT )
2
VBAT )
(
+
7.5mA + (0.012)(I )
VIN
[
SW
BAT ]

(IBAT )(VBAT )  1+ VBAT

30 
PDRIVER =
55(VIN)
C2
BOOST
L1
D2
2
PSW
2
IBAT ) (RSW )(VBAT )
(
=
+
VIN
( tOL)(VIN)(IBAT )( f)
RSW = Switch ON resistance ≈ 0.16Ω
tOL = Effective switch overlap time ≈ 10ns
f = 200kHz
Example: VIN = 15V, VBAT = 8.4V, IBAT = 3A;
PBIAS = (3.5mA )(15) + 1.5mA(8.4)
+
(8.4)2
15
[7.5mA + (0.012)(3)] = 0.27W
(3)(8.4)2  1+ 830.4 
PDRIVER =
= 0.33W
55(15)
2
3) (0.16)(8.4)
(
PSW =
+ 10−9 (15)(3)(200kHz )
15
= 0.81 + 0.09 = 0.9W
Total Power in the IC is: 0.27 + 0.33 + 0.9 = 1.5W
Temperature rise will be (1.5W)(30°C/W) = 45°C. This
assumes that the LT1511 is properly heat sunk by connecting the seven fused ground pins to expanded traces
and that the PC board has a backside or internal plane for
heat spreading.
The PDRIVER term can be reduced by connecting the boost
diode D2 (see Figure 1) to a lower system voltage (lower
than VBAT) instead of VBAT.
(IBAT )(VBAT )(VX ) 1+ V30X 
Then PDRIVER =
55(VIN )
For example, VX = 3.3V then:
LT1511
SPIN
VX
IVX
+
1511 • F07
10µF
Figure 7. Lower VBOOST
.3V 
(3A)(8.4V)(3.3V) 1+ 330


PDRIVER =
= 0.11W
55(15V )
The average IVX required is:
PDRIVER 0.11W
=
= 34mA
3.3V
VX
Fused-lead packages conduct most of their heat out the
leads. This makes it very important to provide as much PC
board copper around the leads as is practical. Total
thermal resistance of the package-board combination is
dominated by the characteristics of the board in the
immediate area of the package. This means both lateral
thermal resistance across the board and vertical thermal
resistance through the board to other copper layers. Each
layer acts as a thermal heat spreader that increases the
heat sinking effectiveness of extended areas of the board.
Total board area becomes an important factor when the
area of the board drops below about 20 square inches. The
graph in Figure 8 shows thermal resistance vs board area
for 2-layer and 4-layer boards with continuous copper
planes. Note that 4-layer boards have significantly lower
thermal resistance, but both types show a rapid increase
for reduced board areas. Figure 9 shows actual measured
lead temperatures for chargers operating at full current.
Battery voltage and input voltage will affect device power
dissipation, so the data sheet power calculations must be
used to extrapolate these readings to other situations.
Vias should be used to connect board layers together.
Planes under the charger area can be cut away from the
rest of the board and connected with vias to form both a
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low thermal resistance system and to act as a ground
plane for reduced EMI.
Glue-on, chip-mounted heat sinks are effective only in
moderate power applications where the PC board copper
cannot be used, or where the board size is small. They
offer very little improvement in a properly laid out multilayer board of reasonable size.
Higher Duty Cycle for the LT1511 Battery Charger
Maximum duty cycle for the LT1511 is typically 90%, but
this may be too low for some applications. For example, if
an 18V ±3% adapter is used to charge ten NiMH cells, the
charger must put out 15V maximum. A total of 1.6V is lost
in the input diode, switch resistance, inductor resistance
and parasitics, so the required duty cycle is 15/16.4 =
91.4%. As it turn out, duty cycle can be extended to 93%
THERMAL RESISTANCE (°C/W)
45
by restricting boost voltage to 5V instead of using VBAT as
is normally done. This lower boost voltage also reduces
power dissipation in the LT1511, so it is a win-win decision. Connect an external source of 3V to 6V at VX node in
Figure 10 with a 10µF CX bypass capacitor.
Even Lower Dropout
For even lower dropout and/or reducing heat on the board,
the input diode D3 should be replaced with a FET (see
Figure 11). It is pretty straightforward to connect a
P-channel FET across the input diode and connect its gate
to the battery so that the FET commutates off when the
input goes low. The problem is that the gate must be
pumped low so that the FET is fully turned on even when
the input is only a volt or two above the battery voltage.
Also there is a turn-off speed issue. The FET should turn
STANDARD CONNECTION
40
35
SW
C3
0.47µF
2-LAYER BOARD
SW
C3
0.47µF
BOOST
LT1511
D2
30
HIGH DUTY CYCLE CONNECTION
D2
SPIN
4-LAYER BOARD
25
BOOST
LT1511
SPIN
SENSE
VX
3V TO 6V
BAT
SENSE
BAT
CX
10µF
20
VBAT
MEASURED FROM AIR AMBIENT
TO DIE USING COPPER LANDS
AS SHOWN ON DATA SHEET
15
VBAT
+
+
10
0
5
20
15
25
10
BOARD AREA (IN2)
30
35
1511 F10
LT1511 • F08
Figure 10. High Duty Cycle
Figure 8. LT1511 Thermal Resistance
VIN
110
+
Q1
NOTE: PEAK DIE TEMPERATURE WILL BE
ABOUT 10°C HIGHER THAN LEAD TEMPERATURE AT 3A CHARGING CURRENT
100
LEAD TEMPERATURE (°C)
HIGH DUTY CYCLE CONNECTION
Q2
90
2-LAYER BOARD
RX
50k
80
D1
C2
0.47µF
D2
4-LAYER BOARD
70
60
VIN = 16V
VBAT = 8.4V
ICHRG = 3A
TA = 25°C
50
40
0
5
VCC
BOOST
LT1511
SPIN
Q1 = Si4435DY
Q2 = TP0610L
4-LAYER BOARD
WITH VBOOST = 3.3V
20
15
25
10
BOARD AREA (IN2)
SW
VX
3V TO 6V
SENSE
BAT
CX
10µF
VBAT
30
+
35
LT1511 • F09
1511 F11
Figure 9. LT1511 Lead Temperature
14
Figure 11. Replacing the Input Diode
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off instantly when the input is dead shorted to avoid large
current surges from the battery back through the charger
into the FET. Gate capacitance slows turn-off, so a small
P-channel (Q2) is to discharge the gate capacitance quickly
in the event of an input short. The body diode of Q2 creates
the necessary pumping action to keep the gate of Q1 low
during normal operation. Note that Q1 and Q2 have a VGS
spec limit of 20V. This restricts VIN to a maximum of 20V.
For low dropout operation with VIN > 20V consult factory.
Optional Connection of Input Diode and
Current Sense Resistor
D3
ADAPTER
IN
CLP
+
LT1511
C1
1µF
CLN
L1
RS4
TO
SYSTEM
POWER
VCC
SW
+
PARASITIC
INTERNAL
DIODE
The circuit in Figure 12b allows system power to go to 0V
without drawing battery current by adding an additional
diode, D4. To ensure proper operation, the LT1511 current
sense amplifier inputs (CLP and CLN) were designed to
work above VCC and not to draw current from VCC when the
inputs are pulled to ground by a powered-down adapter.
Layout Considerations
The typical application shown in Figure 1 on the first page
of this data sheet shows a single diode to isolate the VCC
pin from the adapter input. This simple connection may be
unacceptable in situations where the main system power
must be disconnected from both the battery and the
adapter under some conditons. In particular, if the adapter
is disconnected or turned off and it is desired to also
R7
500Ω
disconnect the system load from the battery, the system
will remain powered through the parasitic diode from the
SW pin to the VCC pin.
CIN
RS1
1511 F12a
Switch rise and fall times are under 10ns for maximum
efficiency. To prevent radiation, the catch diode, SW pin
and input bypass capacitor leads should be kept as short
as possible. A ground plane should be used under the
switching circuitry to prevent interplane coupling and to
act as a thermal spreading path. All ground pins should be
connected to expanded traces for low thermal resistance.
The fast-switching high current ground path, including the
switch, catch diode and input capacitor, should be kept
very short. Catch diode and input capacitor should be
close to the chip and terminated to the same point. This
path contains nanosecond rise and fall times with several
amps of current. The other paths contain only DC and/or
200kHz tri-wave and are less critical. Figure 13 indicates
the high speed, high current switching path. Figure 14
shows critical path layout. Contact Linear Technology for
an actual LT1511 circuit PCB layout or Gerber file.
Figure 12a. Standard Connection
SWITCH NODE
L1
R7
500Ω
CLP
LT1511
SW
L1
+
CLN
VCC
PARASITIC
INTERNAL
DIODE
C1
1µF
+
CIN
VBAT
ADAPTER
IN
RS4
VIN
D3
D4
TO
SYSTEM
POWER
CIN
HIGH
FREQUENCY
CIRCULATING
PATH
D1
COUT
BAT
LT1511 • F13
RS1
1511 F12b
Figure 13. High Speed Switching Path
Figure 12b. Modified Input Diode Connection
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
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GND
D1
GND
SW
GND
GND
GND
GND
VCC1
CIN
CIN
GND
L1
GND
TO
GND
TO
GND
RS1
COUT
GND
NOTE: CONNECT ALL GND PINS TO EXPANDED PC LANDS FOR PROPER HEAT SINKING
LT1511 • F14
Figure 14. Critical Electrical and Thermal Path Layout
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
SW Package
24-Lead Plastic Small Outline (Wide 0.300)
(LTC DWG # 05-08-1620)
0.598 – 0.614*
(15.190 – 15.600)
0.291 – 0.299**
(7.391 – 7.595)
0.010 – 0.029 × 45°
(0.254 – 0.737)
0.037 – 0.045
(0.940 – 1.143)
0.093 – 0.104
(2.362 – 2.642)
24
23
22
21
20
19
18
17
16
15
14
13
0° – 8° TYP
0.009 – 0.013
(0.229 – 0.330)
NOTE 1
0.050
(1.270)
TYP
0.394 – 0.419
(10.007 – 10.643)
NOTE 1
0.004 – 0.012
(0.102 – 0.305)
0.014 – 0.019
0.016 – 0.050
(0.356 – 0.482)
(0.406 – 1.270)
TYP
NOTE:
1. PIN 1 IDENT, NOTCH ON TOP AND CAVITIES ON THE BOTTOM OF PACKAGES ARE THE MANUFACTURING OPTIONS.
THE PART MAY BE SUPPLIED WITH OR WITHOUT ANY OF THE OPTIONS
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
1
2
3
4
5
6
7
8
9
10
11
12
S24 (WIDE) 0996
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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High Frequency, Small Inductor, High Efficiency Switchers, 1.5A Switch
LT1376
500kHz Step-Down Switching Regulator
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LT1505
High Current, High Efficiency Battery Charger
94% Efficiency, Synchronous Current Mode PWM
LT1510
Constant-Voltage/Constant-Current Battery Charger
Up to 1.5A Charge Current for Lithium-Ion, NiCd and NiMH Batteries
LT1512
SEPIC Battery Charger
VIN Can Be Higher or Lower Than Battery Voltage
LT1769
Constant-Voltage/Constant-Current Battery Charger
Up to 2A Charge Current for Lithium-Ion, NiCd and NiMH Batteries
®
16
Linear Technology Corporation
1511fb LT/TP 0399 REV B 2K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
 LINEAR TECHNOLOGY CORPORATION 1995