Maxim MAX746ESE High-efficiency, pwm, step-down, n-channel dc-dc controller Datasheet

19-0192; Rev 1; 11/93
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
____________________________Features
♦ 93% to 96% Efficiency for 50mA to 3A
Output Currents
♦ 4V to 15V Input Voltage Range
♦ Low 950µA Supply Current
♦ 1.4µA Shutdown Current
♦ Drives External N-Channel FETs
♦ Fixed-Frequency Current-Mode PWM (Heavy Loads)
♦ Idle-Mode PFM (Light Loads)
♦ Cycle-by-Cycle Current Limiting
♦ 2V ±1.5% Accurate Reference Output
♦ Adjustable Soft-Start
♦ Undervoltage Lockout
♦ Precision Comparator for Power-Fail or
Low-Battery Warning
________________________Applications
5V-to-3.3V Green PC Applications
Notebook/Laptop Computers
Personal Digital Assistants
Battery-Operated Equipment
Cellular Phones
______________Ordering Information
PART
TEMP. RANGE
MAX746CPE
0°C to +70°C
16 Plastic DIP
PIN-PACKAGE
MAX746CSE
MAX746C/D
MAX746EPE
0°C to +70°C
0°C to +70°C
-40°C to +85°C
16 Narrow SO
Dice*
16 Plastic DIP
MAX746ESE
MAX746MJE
-40°C to +85°C
-55°C to +125°C
16 Narrow SO
16 CERDIP
* Contact factory for dice specifications.
__________Typical Operating Circuit
__________________Pin Configuration
INPUT 6V TO 15V
V+
TOP VIEW
AV+
40mΩ
CP
HIGH
SHDN
MAX746
LBO 1
CS
LBI 2
EXT
SS 3
ON/OFF
39µH
OUTPUT
5V
REF 4
16 GND
15 V+
14 CP
MAX746
SHDN 5
LOW-BATTERY
DETECTOR INPUT
440µF
LBI
REF
SS
OUT
LBO
CC FB AGND GND
LOW-BATTERY
DETECTOR OUTPUT
13 HIGH
12 EXT
FB 6
11 AGND
CC 7
10 CS
AV+ 8
9
OUT
DIP/SO
™Dual-Mode and Idle-Mode are trademarks of Maxim Integrated Products.
________________________________________________________________ Maxim Integrated Products
Call toll free 1-800-998-8800 for free samples or literature.
1
MAX746
_______________General Description
The MAX746 is a high-efficiency, high-current, step-down
DC-DC power-supply controller that drives external N-channel FETs. It provides 93% to 96% efficiency from a 6V supply
voltage with load currents ranging from 50mA up to 3A. It
uses a pulse-width-modulating (PWM) current-mode control
scheme to provide precise output regulation and low output
noise. The MAX746's 4V to 15V input voltage range, fixed
5V/adjustable (Dual-ModeTM) output, and adjustable current
limit make this device ideal for a wide range of applications.
High efficiency is maintained with light loads due to a proprietary automatic pulse-skipping control (Idle-ModeTM) scheme
that minimizes switching losses by reducing the switching frequency at light loads. The low 950µA quiescent current and
ultra-low 1.4µA shutdown current further extend battery life.
External components are protected by the MAX746's cycleby-cycle current limit. The MAX746 also features a 2V ±1.5%
reference, a comparator for low-battery detection or level
translating, and soft-start and shutdown capability.
The MAX747—discussed in a separate data sheet—
functions similarly to the MAX746, but drives P-channel logic
level FETs.
MAX746
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
ABSOLUTE MAXIMUM RATINGS
Supply Voltage V+, AV+ to GND ..............................-0.3V to 17V
HIGH, EXT to GND....................................................-0.3V to 21V
AGND to GND..........................................................-0.3V to 0.3V
All Other Pins................................................-0.3V to (V+ + 0.3V)
Reference Current (IREF) ....................................................±2mA
Continuous Power Dissipation (TA = +70°C)
Plastic DIP (derate 10.53mW/°C above +70°C) ..........842mW
Narrow SO (derate 8.70mW/°C above +70°C) ............696mW
CERDIP (derate 10.00mW/°C above +70°C) ...............800mW
Operating Temperature Ranges:
MAX746C_E ........................................................0°C to +70°C
MAX746E_E .....................................................-40°C to +85°C
MAX746MJE ..................................................-55°C to +125°C
Junction Temperatures:
MAX746C_E/E_E..........................................................+150°C
MAX746MJE.................................................................+175°C
Storage Temperature Range .............................-65°C to +160°C
Lead Temperature (soldering, 10sec) .............................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V+ = 10V, ILOAD = 0A, IREF = 0µA, TA = TMIN to TMAX, unless otherwise noted.)
PARAMETER
Input Voltage
SYMBOL
CONDITIONS
MIN
V+
Output Voltage
VOUT
V+ = 6V to 15V, 0V < (V+ - CS) < 0.125V,
FB = 0V (includes line and load regulation)
Feedback Voltage
VFB
(V+ - CS) = 0V,
external feedback mode
MAX
UNITS
15
V
5.25
V
4.85
5.08
MAX746C
1.96
2.00
2.04
MAX746E/M
1.95
2.00
2.05
V+ = 6V to 15V, FB = 0V
Line Regulation
TYP
4
0.05
V+ = 4V to 15V, external feedback mode
0.1
%/V
Load Regulation
0V < (V+ - CS) < 0.125V
1.3
Efficiency
Circuit of Figure 1, ILOAD = 0.5A to 2.5A,
V+ = 6V
94
OUT Leakage Current
VOUT = 5V
50
80
µA
FB Input Logic Low
For dual-mode switchover
40
mV
FB Input Leakage Current
FB = 2V
1
100
nA
Reference Voltage
VREF
IREF = 0µA
2.5
V
%
%
MAX746C
1.97
2.00
2.03
MAX746E/M
1.96
2.00
2.04
V
Reference Load Regulation
IREF = 0µA to 100µA
9
20
mV
Soft-Start Source Current
SS = 0V
0.5
1.0
1.5
µA
Soft-Start Fault Current (Note 1)
SS = 2V
100
500
Operating, V+ = 15V
Supply Current (Note 2)
ISUPP
MAX746C
1.1
MAX746E/M
Operating, V+ = 10V
2
fOSC
1.4
1.7
mA
µA
0.95
Shutdown mode
Oscillator Frequency
µA
1.4
20
MAX746C
85
100
115
MAX746E/M
80
100
120
_______________________________________________________________________________________
kHz
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
(V+ = 10V, ILOAD = 0A, IREF = 0µA, TA = TMIN to TMAX, unless otherwise noted.)
PARAMETER
SYMBOL
Maximum Duty Cycle
Charge-Pump Output Voltage
Current-Sense Amplifier
Current-Limit Threshold
CONDITIONS
V+ = 6V
VHIGH
VLIMIT
IHIGH = 0mA to 10mA
MIN
TYP
91
96
V+ + 4
V+ – CS
MAX
%
V+ + 5 V+ + 6
125
150
UNITS
175
VHIGH - 0.1
V
mV
EXT Output High
VHIGH forced to 15V, IEXT = -1mA
EXT Output Low
VHIGH forced to 15V, IEXT = 1mA
EXT Sink Current
VHIGH = 15V, VEXT = 12.5V
160
mA
EXT Source Current
VHIGH = 15V, VEXT = 2.5V
270
mA
24
kΩ
Compensation Pin Impedance
LBI Threshold Voltage
LBO Output Voltage Low
VOL
LBO Output Leakage Current
1.97
2.00
2.03
MAX746E/M
1.96
2.00
2.04
VIL
SHDN Input Voltage High
VIH
V
V
ISINK = 0.5mA
0.4
V
LBI = 2.5V
100
nA
1
µA
0.4
V
V+ = 15V, LBO = 15V, LBI = 2.5V
SHDN Input Voltage Low
Note 1:
Note 2:
MAX746C
LBI falling
LBI Input Leakage Current
SHDN Input Leakage Current
V
0.25
2.0
SHDN = 10V
V
0.1
100
nA
The soft-start fault current is the current sink capability of SS when VREF < 1V or when the device is in shutdown.
ISUPP is the supply current drawn by V+, which includes the current drawn by the charge pump. The charge pump
doubles the current drawn by HIGH from the V+ input, so ISUPP = IV+ + 2IHIGH.
_______________________________________________________________________________________
3
MAX746
ELECTRICAL CHARACTERISTICS (continued)
__________________________________________Typical Operating Characteristics
(Circuit of Figure 1a, TA = +25°C, unless otherwise noted.)
0.9
2
ENTIRE
CIRCUIT
1
7
9
11
13
SUPPLY VOLTAGE (V)
-75 -50 -25
15
0
25
50
75
5
0.7
100 125
MAX746-07
VIN = 6V
VIN = 9V
VIN = 12V
80
1.1
1.3
1.5
1.7
EFFICIENCY vs. OUTPUT CURRENT
100
CIRCUIT OF FIGURE 1c
VOUT = 5V
VIN = 6V
EFFICIENCY (%)
CIRCUIT OF FIGURE 1b
VOUT = 3.3V
V+ = 5V
0.9
OUTPUT CURRENT (A)
EFFICIENCY vs. OUTPUT CURRENT
100
EFFICIENCY (%)
EFFICIENCY (%)
90
CONTINUOUSCONDUCTION
REGION
TEMPERATURE (°C)
EFFICIENCY vs. OUTPUT CURRENT
100
9
7
MAX746-08
5
PEAK
INDUCTOR
CURRENT
11
SCHOTTKY DIODE
LEAKAGE EXCLUDED
0
0.8
DISCONTINUOUSCONDUCTION REGION
13
3
MAX746-09
V+ = 9V
VOUT = 5V
SUPPLY VOLTAGE (V)
1.0
15
MAX746-02
MAX746-01
1.1
4
NO-LOAD SUPPLY CURRENT (mA)
NO-LOAD SUPPLY CURRENT (mA)
1.2
CONTINUOUS-CONDUCTION MODE
BOUNDARY AND CORRESPONDING
PEAK INDUCTOR CURRENT
NO-LOAD SUPPLY CURRENT
vs. TEMPERATURE
90
80
MAX746-06
N0-LOAD SUPPLY CURRENT
vs. SUPPLY VOLTAGE
90
VIN = 12V
80
CIRCUIT OF FIGURE 1a
VOUT = 5V
70
70
0.1
0.01
1
0.1
1
10
0.1
1
OUTPUT CURRENT (A)
PEAK INDUCTOR CURRENT
vs. OUTPUT CURRENT
PEAK INDUCTOR CURRENT
vs. OUTPUT CURRENT
PEAK INDUCTOR CURRENT
vs. OUTPUT CURRENT
2
VIN = 12V
1
VIN = 9V
3
2
1
1.5
CIRCUIT OF FIGURE 1c
VOUT = 5V
1.0
VIN = 12V
0.5
VIN = 6V
VIN = 6V
0
0
0.1
10
MAX746-03
MAX746-05
CIRCUIT OF FIGURE 1b
VOUT = 3.3V
V+ = 5V
PEAK INDUCTOR CURRENT (A)
3
4
PEAK INDUCTOR CURRENT (A)
MAX746-03
CIRCUIT OF FIGURE 1a
VOUT = 5V
0.01
0.01
OUTPUT CURRENT (A)
0
1
OUTPUT CURRENT (A)
4
70
0.01
10
OUTPUT CURRENT (A)
4
PEAK INDUCTOR CURRENT (A)
MAX746
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
10
0.01
0.1
1
OUTPUT CURRENT (A)
10
0.01
0.1
1
OUTPUT CURRENT (A)
_______________________________________________________________________________________
10
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
LOAD-TRANSIENT RESPONSE
LINE-TRANSIENT RESPONSE
LOAD-TRANSIENT RESPONSE
10V
A
8V
A
A
B
200µs/div
B
500ms/div
1ms/div
A: LOAD CURRENT, 0.1A TO 1.5A, 1A/div
B: VOUT RIPPLE, 50mV/div, AC-COUPLED
V+ = 10V
A: V+ = 8V TO 10V, 2V/div
B: VOUT RIPPLE, 100mV/div
IOUT = 3A
A: LOAD CURRENT, 0.1A TO 1.5A, 1A/div
B: VOUT RIPPLE, 50mV/div, AC COUPLED
V+ = 10V
CONTINUOUS-CONDUCTION MODE
WAVEFORMS
B
MODERATE-LOAD, IDLE-MODE
WAVEFORMS
DISCONTINUOUS-CONDUCTION
IDLE-MODE WAVEFORMS
A
A
A
B
B
B
C
C
0V
C
5µs/div
A : EXT VOLTAGE, 20V/div
B : INDUCTOR CURRENT 1A/div
C : VOUT RIPPLE, 50mV/div
V+ = 10V, IOUT = 3A
20µs/div
A: EXT VOLTAGE, 10V/div
B: INDUCTOR CURRENT, 500mA/div
C: VOUT RIPPLE, 50mV/div, AC-COUPLED
V+ = 10V, IOUT = 75mA
20µs/div
A: EXT VOLTAGE, 10V/div
B: INDUCTOR CURRENT, 500mA/div
C: VOUT RIPPLE, 50mV/div, AC-COUPLED
V+ = 6V, IOUT = 480mA
_______________________________________________________________________________________
5
MAX746
____________________________Typical Operating Characteristics (continued)
(Circuit of Figure 1a, TA = +25°C, unless otherwise noted.)
MAX746
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
______________________________________________________________Pin Description
PIN
NAME
FUNCTION
1
LBO
Low-battery output is an open-drain output that goes low when LBI is less than 2V. Connect to V+ through a
pull-up resistor. Leave floating if not used. LBO is disabled in shutdown mode.
2
LBI
Input to the low-battery comparator. Tie to V+ or GND if not used.
3
SS
Soft-start limits start-up surge currents. On power-up, it charges the soft-start capacitor, slowly raising the peak
current limit to the level set by the sense resistor.
4
REF
2V reference output can source 100µA for external loads. Bypass with 1µF. The reference is disabled in shutdown mode.
5
SHDN
Active-high logic input. In shutdown mode, VOUT = 0V and the supply current is reduced to less than 20µA.
Connect to GND for normal operation.
6
FB
Feedback input for adjustable-output operation. Connect to GND for fixed 5V output. Use a resistor-divider network to adjust the output voltage (see Setting the Output Voltage section).
7
CC
AC compensation input for the error amplifier. Connect a capacitor between CC and GND for fixed 5V-output
operation (see Compensation Capacitor section).
8
AV+
Quiet supply voltage for sensitive analog circuitry. Also the noninverting input to the current-sense amplifier. A
separate bypass capacitor is not recommended for AV+.
9
OUT
Output voltage sense that connects to the internal resistor divider. Bypass with 0.1µF to AGND, close to the IC
for fixed output operation. Leave unconnected for adjustable-output operation.
10
CS
11
AGND
12
EXT
13
HIGH
14
CP
Charge-pump output that generates a 0V to V+, 50kHz square wave (see Charge Pump section).
15
V+
High-current supply voltage for the charge pump.
16
GND
Inverting input to the current-sense amplifier. Connect the current-sense resistor (RSENSE) from AV+ to CS.
Quiet analog ground.
Power MOSFET gate-drive output that swings between HIGH and GND. EXT is not protected against short circuits to V+ or AGND.
Regulated high-side voltage, 5V above the V+ supply voltage.
High-current ground return for the output driver and charge pump.
____________________Getting Started
Figure 1a shows the 5V-output 3A standard application
circuit, Figure 1b shows the 3.3V-output 3A standard
application circuit, and Figure 1c shows the 5V-output
1.5A standard application circuit. Most applications will
be served by these circuits. To learn more about component selection for particular applications, refer to the
Design Procedure section. To learn more about the operation of the MAX746, refer to the Detailed Description.
_______________Detailed Description
The MAX746 monolithic, CMOS, step-down, switchmode power-supply controller provides high-side drive
for external logic-level N-channel FETs. A charge pump
generates a voltage 5V above the supply voltage for
high-side drive capability. The MAX746 uses a unique
6
current-mode pulse-width-modulating (PWM) control
scheme that results in tight output-voltage regulation,
excellent load- and line-transient response, low noise,
and high efficiency over a wide range of load currents.
Efficiency at light loads is further enhanced by a proprietary idle-mode switching control scheme that skips
oscillator cycles in order to reduce switching losses.
Other features include undervoltage lockout, shutdown,
and a low-battery detection comparator.
Operating Principle
Figure 2 is the MAX746 block diagram. The MAX746
regulates using an inner current-feedback loop and an
outer voltage-feedback loop. A slope-compensation
scheme stabilizes the current loop; the dominant pole,
formed by the output filter capacitor and the load,
stabilizes the voltage loop.
_______________________________________________________________________________________
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
MAX746
V IN
6V TO 15V
C3
0.1µF
C2
100µF
R2
2
R1
D2
1N914 *
15
V+
LBI
CP
HIGH
C5
0.1µF
3
C6
1.0µF
4
6
5
11
AV+
SS
REF
C9
4.7µF
D4
1N5817
MAX746
CS
14
D3
1N914 *
C8
0.1µF
13
8
R SENSE
40mΩ
10
Q1
Si9410DY
EXT 12
FB
CC
SHDN
AGND
OUT
LB0
7
C7
2.7nF
N
D1
NSQ03A03
L1
39µH
5V
AT 3A
C1
430µF
9
C4
0.1µF
1
R3
100k
GND
16
* SEE TABLE 2 FOR DIODE SELECTION.
Figure 1a. 5V Standard Application Circuit (15W)
Discontinuous-/ContinuousConduction Modes
Under these conditions, the inductor must be scaled to the
current-sense resistor value.
The MAX746 is designed to operate in continuous-conduction mode (CCM) but can also operate in discontinuous-conduction mode (DCM), making it ideal for variableload applications. In DCM, the current starts at zero and
returns to zero on each cycle. In CCM, the inductor current
never returns to zero; it consists of a small AC component
superimposed on a DC offset. This results in higher current
capability because the AC component in the inductor current waveform is small. It also results in lower output noise,
since the inductor does not exhibit the ringing that would
occur if the current reached zero (see inductor waveforms
in the Typical Operating Characteristics). To transfer equal
amounts of energy to the load in one cycle, the peak current level for the discontinuous waveform must be much
larger than the peak current for the continuous waveform.
Overcompensation adds a pole to the outer voltage feedback-loop response, degrading loop stability. This may cause
voltage-mode pulse-frequency-modulation instead of PWM
operation. Undercompensation results in inner current feedback-loop instability, and may cause the inductor current to
staircase. Ideal matching between the sense resistor and
inductor is not required; it can differ by ±30% or more.
Slope Compensation
Slope compensation stabilizes the inner current-feedback
loop by adding a ramp signal to the current-sense amplifier
output. Ideal slope compensation can be achieved by
adding a linear ramp, with the same slope as the declining
inductor current, to the rising inductor current-sense voltage.
Oscillator and EXT Control
The oscillator frequency is nominally 100kHz, and the duty
cycle varies from 5% to 96%, depending on the input/output voltage ratio. EXT, which provides the gate drive for the
external logic-level N-FET, is switched between HIGH and
GND at the switching frequency. EXT is controlled by a
unique two-comparator control scheme consisting of a PWM
comparator and an idle-mode comparator (Figure 2). The
PWM comparator determines the cycle-by-cycle peak current with heavy loads, and the idle-mode comparator sets
the light-load peak current. As VOUT begins to drop, EXT
goes high and remains high until both comparators trip.
With heavy loads, the idle-mode comparator trips first and
the PWM control comparator determines the EXT on-time;
_______________________________________________________________________________________
7
MAX746
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
V IN
4.5V TO 6V
C3
0.1µF
C2
100µF
R2
2
R1
D2
D3
D5
1N914 1N914 1N914
15
V+
LBI
CP
HIGH
C5
0.1µF
3
C6
1µF
4
AV+
SS
REF
MAX746
CS
14
C8
0.1µF
C9
1µF
D6
1N914
11
CC
SHDN
13
8
R SENSE
40mΩ
10
Q1
Si9410DY
FB
LB0
N
7
OUT 9
AGND
C11
1µF
C10
0.1µF
EXT 12
5
D4
1N5817
R5
13k (1%)
6
C7
2nF
1
R4
20k (1%)
L1 *
22µH
D1
NSQ03A03
3.3V
AT 3A
C4
0.1µF
C3
660µF
R3
100k
GND
16
*
SUMIDA CDR125 22µH SURFACE-MOUNT INDUCTOR
Figure 1b. 3.3V Standard Application Circuit (9.9W)
with light loads, the PWM comparator trips quickly and the
idle-mode comparator sets the EXT on-time.
Traditional PWM converters continue to switch on every
cycle, even when the inductor current is discontinuous
due to smaller loads, decreasing light-load efficiency.
In contrast, the MAX746’s idle-mode comparator increases the switch on-time, allowing more energy to be transferred per cycle. Since fewer cycles are required, the
switching frequency is reduced, resulting in minimal
switching losses and increased efficiency.
The light-load output noise spectrum widens due to the
variable switching frequency in idle-mode, but output
ripple remains low. Using the Typical Operating Circuit,
with a 9V input and a 125mA load current, output ripple
is less than 40mV.
Charge Pump
The MAX746 contains all the control circuitry required
to provide a regulated charge-pump voltage 5V
above V+ for high-side driving N-channel logic FETs.
The charge pump operates with a nominal 50kHz fre-
8
quency. When the voltage at HIGH exceeds AV+ by
5V, the charge-pump oscillator is inhibited (Figure 2).
When the voltage at HIGH is less than 4.3V below V+,
undervoltage lockout occurs. Use the voltage tripler
(Figure 3b) when V+ ≤ 6V; otherwise, use the voltage
doubler (Figure 3a).
Soft-Start and Current Limiting
The MAX746 draws its highest current at power-up. If
the power source to the MAX746 cannot provide this
initial elevated current, the circuit may not function correctly. For example, after prolonged use the increased
series resistance of a battery may prevent it from providing adequate initial surge currents when the
MAX746 is brought out of shutdown. Using soft-start
(SS) minimizes the possibility of overloading the incoming supply at power-up by gradually increasing the
peak current limit. Connect an external capacitor from
SS to AGND to reduce the initial peak currents drawn
from the supply.
_______________________________________________________________________________________
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
MAX746
VIN
6V TO 15V
C3
0.1µF
C2
47µF
R2
2
R1
D2
1N914 *
15
V+
LBI
CP
HIGH
C5
0.1µF
3
C6
1µF
4
6
5
11
AV+
SS
REF
C9
4.7µF
D4
1N5817
MAX746
CS
14
D3
1N914 *
C8
0.1µF
13
8
R SENSE
75mΩ
10
Q1
Si9410DY
EXT 12
FB
CC
SHDN
AGND
OUT
LB0
7
C7
1nF
N
D1
NSQ03A03
L1 **
82µH
5V
AT 1.5A
C1
220µF
9
C4
0.1µF
1
R3
100k
GND
16
* SEE TABLE 2 FOR DIODE SELECTION.
** SUMIDA CDR125 SURFACE-MOUNT INDUCTOR.
Figure 1c. 5V Standard Application Circuit (7.5W)
The steady-state SS pin voltage is typically 3.8V. On
power-up, SS sources 1µA until its voltage reaches
3.8V. The current-limit comparator inhibits EXT switching until the SS voltage reaches 1.8V. The peak current
limit is set by:
VLIMIT
150mV (typ)
IPK = _________ = ___________
RSENSE
RSENSE
where VLIMIT is the differential voltage across the currentsense amplifier inputs. Figure 4 shows how the SS peak
current limit increases as the voltage on SS rises for two
RSENSE values.
Undervoltage Lockout
Undervoltage lockout inhibits operation of EXT until the
charge pump is capable of generating a voltage greater
than 4.3V above the supply voltage (Figure 2). When
the undervoltage-lockout comparator detects an undervoltage condition, the switching action at EXT is halted.
Shutdown Mode
When SHDN is high, the MAX746 is shut down. In this
mode, the internal biasing circuitry (including EXT) is
turned off, VOUT drops to 0V, and the supply current
drops to 1.4µA (20µA max). This excludes external
component leakage, which may add several
microamps to the shutdown supply current for the
entire circuit. SHDN is a logic input. Connect SHDN to
GND for normal operation.
Low-Battery Detector
The MAX746 provides a low-battery comparator that
compares the voltage on LBI to the reference voltage.
LBO, an open-drain output, goes low when the LBI voltage is below V REF. Use a resistor-divider network, as
shown in the Input Voltage Monitor Circuit (Figure 5),
to set the trip voltage (VTRIP) at the desired level. In
this circuit, LBO goes low when V+ ≤ VTRIP. LBO is high
impedance in shutdown mode.
_______________________________________________________________________________________
9
MAX746
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
LBO
HIGH
EXT
V+
PUMP
FROM AV+
LBI
LOW-BATTERY
COMPARATOR
CHARGE-PUMP CONTROL
COMPARATOR
4.3V
N
T T FLIPFLOP Q
5V
+2V
REFERENCE
UNDERVOLTAGELOCKOUT
COMPARATOR
REF
100kHz
OSCILLATOR
OUT
EXT
CONTROL
SHDN
CC
ERROR
AMPLIFIER
FB
PWM
COMPARATOR
DUAL-MODE
COMPARATOR
100mV
AV+
CS
CURRENT-SENSE
AMPLIFIER
SLOPECOMPENSATION
RAMP
VRAMP
LIGHT-LOAD
COMPARATOR
Σ
50mV
SS
SOFT-START
CIRCUITRY
CURRENT-LIMIT
COMPARATOR
AGND
GND
Figure 2. Block Diagram
10
______________________________________________________________________________________
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
C8
0.1µF
D4
1N5817
RSENSE = 50mΩ
PEAK CURRENT LIMIT (A)
15
V+
D3
1N914
C9
1µF
MAX746
T FLIPT FLOP Q
CLK
MAX746-FG03
3
D2
1N914
V IN
MAX746
PEAK CURRENT LIMIT
vs. SOFT-START VOLTAGE
3a. CHARGE-PUMP
VOLTAGE DOUBLER
CP 14
2
V+ - VCS = 150mV
1
HIGH 13
RSENSE = 100mΩ
100kHz
OSCILLATOR
0
0
1
2
3
SOFT-START VOLTAGE (V)
5V
Figure 4. Peak Current Limit vs. Soft-Start Voltage
3b. CHARGE-PUMP
VOLTAGE TRIPLER
AV+
GND
D4
1N5817
16
D3
D2
1N914 1N914
V IN
VIN
D5
1N914
D6
1N914
15
…TO VOUT OR VIN
V+
C8
0.1µF
15
V+
CP
MAX746
HIGH
4
C9
1µF
C10
0.1µF
C11
1µF
R2
14
2
13
R3
100k
MAX746
1
LBO
LBI
LOW-BATTERY
OUTPUT
R1
GND
GND
16
16
R2 = R1
( VVTRIP
-1 )
REF
VREF = 2.0V
Figure 3. Charge-Pump Configurations
Figure 5. Input Voltage Monitor Circuit
__________________Design Procedure
Setting the Output Voltage
The MAX746’s dual-mode output voltage can be set
to 5V by grounding FB, or it can be adjusted from
2V to 14V using external resistors R4 and R5 configured as shown in Figure 6. Select feedback resistor
R4 in the 10kΩ to 60kΩ range. R5 is given by:
VOUT
R5 = (R4) _______ – 1
2V
The MAX746 is designed to use either internal or external feedback mode, but should not be toggled between
(
)
the two modes while operating. If two different output
voltages are required, use external feedback mode
with a resistor network similar to the 3.3V/5V adjustable
output circuit shown in Figure 7.
Selecting RSENSE
To select the sense-resistor value (R SENSE ), first
approximate the peak current assuming I PK is
(1.1) (ILOAD), where ILOAD is the maximum load current. Once all component values have been determined, the actual peak current is given by:
VOUT
IPK = ILOAD + ___________
(2L) (fOSC)
(
V
OUT
)(1– _______
)
VIN
______________________________________________________________________________________
11
MAX746
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
VIN
15
EXT
V+
12
N
L
R5
FB
6
VOUT
C7*
MAX746
R4
OUT
OUT
9
9
16
R4 = 10kΩ TO 60kΩ
VOUT
R5 = R4
-1
VREF
VREF = 2.0V NOMINAL
(
C1
FB 6
MAX746
GND
V OUT
D1
R5
26.1k (1%)
)
* SEE COMPENSATION CAPACITOR SECTION.
SELECT WITH FET OFF:
VOUT = VREF
SELECT WITH FET OFF:
VOUT = VREF
R5
+1 )
( R4a
R4a
17.4k (1%)
5V/3.3V
N
C7
R4b
22.6k (1%)
( R4a R5+ R4b +1 )
VREF = 2.0V NOMINAL
Figure 6. Adjustable Output Circuit
Figure 7. 3.3V/5V Ajustable Output Circuit
Next, determine the value of RSENSE such that:
where VRAMP(max) is the 50mV peak value of the slopecompensation linear ramp signal.
VLIMIT(min)
125mV
RSENSE = _____________ = ________
IPK
IPK
For example, to obtain 5V at 3A, I PK = 3.3A and
RSENSE = 125mV/3.3A = 38mΩ.
The sense resistor should have a power rating greater
than (I PK2) (RSENSE) with an adequate safety margin.
With a 3A load current, IPK = 3.3A and RSENSE = 38mΩ.
The power dissipated by the resistor (assuming an 80%
duty cycle) is 331mW. Metal-film resistors are recommended. Do not use wire-wound resistors because
their inductance will adversely affect circuit operation.
The duty cycle (for continuous conduction) is determined
from the following equation:
VOUT + VDIODE
Duty Cycle (%) = _____________________ x 100%
V+ - VSW + VDIODE
where V SW is the voltage drop across the external
N-FET and sense resistor. VSW can be approximated
as [ILOAD x (rDS(ON) + RSENSE)].
Inductor Selection
Once the sense-resistor value is determined, calculate
the inductor value (L) using the following equation. The
correct inductor value ensures proper slope compensation. Continuing from the equations above:
(RSENSE) (VOUT)
L = ______________________
(VRAMP(max)) (fOSC)
(38mΩ) (5V)
= _____________________ = 38µH
(50mV) (100kHz)
12
Although 38µH is the calculated value, the component
used may have a tolerance of ±30% or more.
Inductors with molypermalloy powder (MPP), Kool Mµ,
or ferrite are recommended. Inexpensive iron-powder
core inductors are not suitable, due to their increased
core losses, especially at switching frequencies in the
100kHz range. MPP and Kool Mµ cores have low permeability, allowing larger currents.
For highest efficiency, use a coil with low DC resistance. To minimize radiated noise, use a toroid, a pot
core, or a shielded coil.
It is customary to select an inductor with a saturation
rating that exceeds the peak current set by R SENSE,
but inductors are often specified very conservatively.
If the inductor’s core losses do not cause excessive
temperature rise (inductor wire insulation is usually
rated for +125°C) and the associated efficiency losses are minimal, inductors with lower current ratings
are acceptable.
In the 3.3V Standard Application Circuit (Figure 1b), the
inductor selected has a 2.2A current rating even
though the peak current is 3.3A. This inductor was
selected for two reasons: it is the highest-rated readily
available surface-mount inductor of its size, and lab
tests have verified that the core-loss increase is minimal. With a 3A load current, the inductor current does
not begin showing significant losses due to saturation
until the supply voltage increases to 10V (the maximum
supply for this circuit is 6V).
______________________________________________________________________________________
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
To ensure the external N-FET is turned on hard, use
logic-level or low-threshold N-FETs. Three important
parameters to note when selecting the N-FET are the
total gate charge (Qg), on resistance (rDS(ON)), and
reverse transfer capacitance (CRSS).
Qg includes all capacitances associated with charging
the gate. Use the typical Qg value for best results; the
maximum value is usually grossly overspecified, since
it is a guaranteed limit and not the measured value.
The typical total gate charge should be 50nC or less;
with larger numbers, EXT may not be able to adequately drive the gate. EXT sink/source capability
(IEXT) is typically 210mA.
The two most significant losses contributing to the
N-FET’s power dissipation are I2R losses and switching
losses. CCM power dissipation (PD), is approximated by:
PD = (Duty Cycle) (IPK2) (rDS(ON)) +
(V+2) (CRSS) (IPK) (fOSC)
__________________________
(IEXT)
where the duty cycle is approximately V OUT /V+,
fOSC = 100kHz, and rDS(ON) and CRSS are given in the
data sheet of the chosen N-FET. In the equation,
rDS(ON) is assumed constant, but is actually a function
of temperature. The equation given does not account
for losses incurred by charging and discharging the
gate capacitance, because that energy is dissipated
by the gate-drive circuitry, not the N-FET.
The Standard Application Circuits (Figure 1) use an
8-pin, Si9410DY, surface-mount N-FET that has 0.05Ω
on resistance with a 4.5V V GS. Optimum efficiency is
obtained when the voltage at the source swings between
the supply rails (within a few hundred millivolts).
Diode Selection
The MAX746’s high switching frequency demands a
high-speed rectifier. Schottky diodes are recommended. Ensure that the Schottky diode average current
rating exceeds the maximum load current.
Capacitor Selection
Output Filter Capacitor
The output filter capacitor C1 should have a low effective series resistance (ESR), and its capacitance should
remain fairly constant over temperature. This is especially true when in CCM, since the output filter capacitor and the load form the dominant pole that
stabilizes the voltage loop.
To ensure stability, the minimum capacitance and maximum ESR values are:
(5) (VREF)
C1(min) > ______________________________
(2π) (GBW) (VOUT) (RSENSE)
and,
(VOUT ) (RSENSE)
ESRC1 < ___________________
(VREF)
where GBW = the loop gain-bandwidth product, 15kHz.
Sprague 595D surface-mount solid tantalum capacitors
and Sanyo OS-CON through-hole capacitors are recommended due to their extremely low ESR. OS-CON
capacitors are particularly useful at low temperatures.
For best results when using other capacitors, increase
the output filter capacitor’s size or use capacitors in
parallel to reduce the ESR.
Bypass OUT with a 0.1µF (C4) capacitor to GND when using
a fixed 5V output (Figures 1a and 1c). With adjustable-output
operation, place C4 between the output voltage and AGND
as close to the IC as possible (Figure 1b).
The circuit load-step response is improved by using a
larger output filter capacitor or by placing a low-cost
bulk capacitor in parallel with the required low-ESR
output filter capacitor. The output voltage sag under a
load step (ISTEP) is approximated by:
(ISTEP2) (L)
VSAG = _____________________________________
(2) (C1) (VIN(MIN) (DMAX - VOUT)
where DMAX is the maximum duty cycle (91% worst
case). The equation assumes an input/output voltage
differential of 2V or more. Table 1 gives measured values of output voltage sag with a 30mA to 3A load step
for various input voltages and output filter capacitors.
Refer also to the AC Stability with Low Input/Output
Differentials section.
Input Bypass Capacitor
The input bypass capacitor C2 reduces peak currents
drawn from the voltage source, and also reduces the
amount of noise at the voltage source caused by the
MAX746’s fast switching action (this is especially
important when other circuitry is operated from the
same source). The input capacitor ripple current rating
must exceed the RMS input ripple current.
IRMS = RMS AC input current
= ILOAD
√(V ) (V - V )
(_______________________
)
V
OUT
IN
OUT
IN
______________________________________________________________________________________
13
MAX746
External Logic-Level N-FET Selection
MAX746
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
Table 1. Measured Output Voltage Sag
with 30mA to 3A Load Step*
OUTPUT
FILTER
CAPACITOR
C1 (mF)
VIN =6V
VIN=6.5V
VIN=7V
VIN=9V
VIN =10V
440
400
250
210
140
90
660
260
190
160
70
50
880
200
100
90
40
25
OUTPUT VOLTAGE SAG (mV)
FOR VARIOUS INPUT VOLTAGES
*Circuit of Figure 1a.
For load currents up to 3A, 100µF (C2) in parallel with
0.1µF (C3) is adequate. Smaller bypass capacitors may
also be acceptable for lighter loads. The input voltage
source impedance determines the size of the capacitor
required at the V+ input. As with the output filter capacitor, a low-ESR capacitor (Sanyo OS-CON, Sprague 595D
or equivalent) is recommended for input bypassing.
Charge-Pump Capacitors
Figure 3a shows the charge-pump doubler circuit configured with a 0.1µF charge-pump capacitor C8 and a
1.0µF reservoir capacitor C9. The ratio of the capacitors, along with the input voltage, determines the
amount of ripple on HIGH. If the input supply range
exceeds 12V, increase C9 to 4.7µF to reduce the
charge-pump ripple. C9 should be 10µF for less.
Figure 3b shows the charge-pump tripler circuit.
Refer to Table 2 to determine the proper charge-pump
configuration (which is based on the minimum expected supply voltage at V+).
Some interaction occurs between the switch oscillator
and the charge-pump oscillator. This interaction modulates the inductor-current waveform, but has negligible
impact on the output.
Soft-Start and Reference Capacitors
Soft-start provides a ramp to the full current limit. A typical value for the soft-start capacitor (C5) is 0.1µF,
which provides a 380ms soft-start time. Use values in
the 0.001µF to 1µF range. The nominal time for C5 to
reach its steady-state value is given by:
Table 2. Charge-Pump Configuration
V+
CHARGE-PUMP CONFIGURATION
V+ ≤ 6V
Voltage tripler with 1N914 diodes for D2,
D3, D5, and D6
6V < V+ < 6.5V*
Voltage doubler with 1N5817 Schottky
diodes for D2 and D3
V+ ≥ 6.5V*
Voltage doubler with 1N914 diodes for
D2 and D3
* When using the voltage-doubler circuit over the military
temperature range, increase the 6.5V limit to 7V.
voltage and load current. With a 3A load current, a 10V
input voltage, and a 0.1µF soft-start capacitor, it typically takes 240ms for the MAX746 to power up. A
0.47µF soft-start capacitor increases the start-up time
to approximately 2.3sec.
Bypass REF with a 1µF capacitor (C6).
Compensation Capacitor
With a fixed 5V output, connect a compensation capacitor (C7) between CC and AGND to optimize transient
response. Appropriate compensation is determined by
the size and ESR of the output filter capacitor (C1), and
by the load current.
In the standard 5V application circuit, 2.7nF is appropriate for load currents up to 3A; for lighter loads,
C7’s value can be reduced. If 2.7nF does not compensate adequately, use the following equations to
determine C7.
For fixed 5V-output operation:
(C1) (ESRC1)
C7 = _____________
12kΩ
For adjustable-output operation, FB becomes the
compensation input pin, and CC and OUT are left
unconnected. Connect C7 between FB and GND in
parallel with R4 (Figure 6). C7 is determined by:
(2) (C1) (ESRC1)
C7 = ___________________
R4  R5
tSS (sec) = (C5) (3.8 x 106)
For example, with a fixed 5V output with C1 = 470µF
and an ESRC1 of 0.04Ω (at a frequency of 100kHz):
Note that tSS does NOT equal the time it takes for the
MAX746 to power-up, although it does affect the startup time. The start-up time is also a function of the input
(C1) (ESRC1)
C7 = _____________ = 1560pF
12kΩ
14
______________________________________________________________________________________
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
Setting the Low-Battery
Detector Voltage
Select R1 between 10kΩ and 1MΩ. Determine R2 using
the following equation:
R2 = R1
(VTRIP - VREF)
________________
VREF
(
)
MAX746
Increasing C7 by up to 50% enhances outer-loop
stability by adding stability to the inductor current
waveform. But increasing C7 too much causes
FB’s response time to decrease (due to the larger
RC time constant caused by the feedback resistors
and the compensation capacitor), which reduces
load-transient stability.
VIN
V+ AV+
KELVIN SENSE
CONNECTION
RSENSE
MAX746
CS
EXT
N
L1
VOUT
where VREF is typically 2.0V. Connect a pull-up resistor
(e.g., 100kΩ) between LBO and VOUT (Figure 5).
Using a Second Supply in
Place of the Charge Pump
If a secondary power supply (a minimum of 5V above
the main supply) is available, it can be substituted for
the charge-pump high-side supply. In this case,
bypass HIGH with a 1µF capacitor and leave CP
unconnected. Since this secondary supply voltage
is applied to the gate, V GS must not exceed the
gate-source breakdown voltage of the external N-FET.
Also, the voltage at HIGH must not exceed 20V. If
a secondary supply is used, the shutdown function
cannot be used because HIGH is internally tied to
V+ in shutdown mode. In this case, SHDN must be
tied low. With the main supply off and HIGH at 12V,
HIGH will typically sink 130µA.
Layout Considerations
Because high current levels and fast switching waveforms radiate noise, proper PC board layout is essential. Use a ground plane, and minimize ground noise by
connecting GND, the anode of the steering Schottky
diode, the input bypass-capacitor ground lead, and the
output filter capacitor ground lead to a single point (star
ground configuration). Also minimize lead lengths to
reduce stray capacitance, trace resistance, and radiated noise. Place bypass capacitor C3 as close to V+
and GND as possible.
AV+ and CS are the inputs to the differential-input
current-sense amplifier. Use a Kelvin connection
across the sense resistor, as shown in Figure 8.
Although AV+ also functions as the supply voltage
for sensitive analog circuitry, a separate AV+ bypass
capacitor should not be used. By not using a capaci-
Figure 8. Kelvin Connection for Current-Sense Amplifier
tor, any noise at the CS input will also appear at the
AV+ input, and will be interpreted by the currentsense amplifier as a common-mode signal . A separate AV+ capacitor causes the noise to appear on
only one input, and this differential noise will be
amplified, adversely affecting circuit operation.
Additional Notes
When probing the MAX746 circuit, avoid shorting
V+ to GND (the two pins are adjacent) as this may
cause the IC to malfunction because of large ground
currents. Because of its fast switching and high drivecapability requirements, EXT is a low-impedance point
that is not short-circuit protected. Therefore, do not
short EXT to any node (including AGND and V+, which
are adjacent to EXT).
Similarly, CC (or FB in adjustable-output operation) is a
sensitive input that should not be shorted to any node.
Avoid shorting CC when probing the circuit, as this may
damage the device.
The MAX746 may continue to operate with AV+ disconnected, but erratic switching waveforms will appear at EXT.
Switching Waveforms
There is a region between CCM and DCM where the
inductor current operates in both modes, as shown
in the Idle-Mode Moderate Current EXT waveform in
the Typical Operating Characteristics . As the output voltage varies, it is fed back into CC and the
duty cycle adjusts to compensate for this change.
The switch is considered off when VEXT is less than
______________________________________________________________________________________
15
MAX746
High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller
or equal to the N-FET’s V GS threshold voltage. Once
the switch is off, the voltage at EXT is pulled to GND
and the N-FET source voltage is a Schottky diode
drop below GND. However, this is not always the case
in the “in-between” mode, due to the changing duty
cycle inherent with DCM. When the device is at maximum duty cycle, EXT turns off at V GS, but the switch
sometimes turns on again after the minimum off-time
before EXT can be pulled to GND. This results in short
spikes, which can be seen on the EXT waveform in the
Typical Operating Characteristics .
___________________Chip Topography
Table 3. Component Suppliers
SUPPLIER
AC Stability with Low Input/Output Differentials
At low input/output differentials, the inductor current
cannot slew quickly enough to respond to load
changes, so the output filter capacitor must hold up the
voltage as the load transient is applied. In Figure 1a’s
circuit, for V+ = 6V, increase the output filter capacitor
to 900µF (Sprague 595D low-ESR capacitors) to obtain
a transient response less than 250mV with a load step
from 0.1A to 3A. As V+ increases, the inductor current
slews faster, so the size of the output filter capacitor can
be reduced (see Table 1).
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INDUCTORS
SS
CP
CAPACITORS
HIGH
EXT
REF
SHDN
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POWER TRANSISTORS
FB CC
AV+ OUT
CS
0.080"
(2.03mm)
TRANSISTOR COUNT: 508;
SUBSTRATE CONNECTED TO HIGH.
RESISTORS
IRC
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