AD AD9864BCPZRL If digitizing subsystem Datasheet

IF Digitizing Subsystem
AD9864
Data Sheet
FEATURES
GENERAL DESCRIPTION
10 MHz to 300 MHz input frequency
6.8 kHz to 270 kHz output signal bandwidth
7.5 dB single sideband noise figure (SSB NF)
−7.0 dBm input third-order intercept (IIP3)
AGC free range up to −34 dBm
12 dB continuous AGC range
16 dB front-end attenuator
Baseband I/Q 16-bit (or 24-bit) serial digital output
LO and sampling clock synthesizers
Programmable decimation factor, output format, AGC, and
synthesizer settings
370 Ω input impedance
2.7 V to 3.6 V supply voltage
Low current consumption: 17 mA
48-lead LFCSP package
The AD98641 is a general-purpose IF subsystem that digitizes a
low level, 10 MHz to 300 MHz IF input with a signal bandwidth
ranging from 6.8 kHz to 270 kHz. The signal chain of the AD9864
consists of a low noise amplifier (LNA), a mixer, a band-pass Σ-∆
analog-to-digital converter (ADC), and a decimation filter with
programmable decimation factor. An automatic gain control
(AGC) circuit gives the AD9864 12 dB of continuous gain
adjustment. Auxiliary blocks include both clock and local
oscillator (LO) synthesizers.
The high dynamic range of the AD9864 and inherent antialiasing
provided by the band-pass Σ-∆ converter allow the device to cope
with blocking signals up to 95 dB stronger than the desired signal.
This attribute often reduces the cost of a radio by reducing IF
filtering requirements. Also, it enables multimode radios of varying
channel bandwidths, allowing the IF filter to be specified for the
largest channel bandwidth.
APPLICATIONS
The SPI port programs numerous parameters of the AD9864,
allowing the device to be optimized for any given application.
Programmable parameters include synthesizer divide ratios, AGC
attenuation and attack/decay time, received signal strength level,
decimation factor, output data format, 16 dB attenuator, and the
selected bias currents.
Multimode narrow-band radio products
Analog/digital UHF/VHF FDMA receivers
TETRA, APCO25, GSM/EDGE
Portable and mobile radio products
SATCOM terminals
The AD9864 is available in a 48-lead LFCSP package and operates
from a single 2.7 V to 3.6 V supply. The total power consumption
is typically 56 mW and a power-down mode is provided via
serial interfacing.
FUNCTIONAL BLOCK DIAGRAM
MXOP MXON IF2P IF2N
GCP GCN
DAC
AD9864
AGC
–16dB
IFIN
LNA
DECIMATION
FILTER
Σ-Δ ADC
FORMATTING/SSI
DOUTA
DOUTB
FS
CLKOUT
FREF
CONTROL LOGIC
LO
SYN
LOP LON
LO VCO AND
LOOP FILTER
IOUTC
CLKP
CLKN
VREFP VCM VREFN
SPI
PC
PD
PE
SYNCB
04319-0-001
IOUTL
VOLTAGE
REFERENCE
CLK SYN
LOOP FILTER
Figure 1.
1
Protected by U.S. Patent No. 5,969,657; other patents pending.
Rev. A
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AD9864
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
SSI Control Registers ................................................................. 21
Applications ....................................................................................... 1
Synchronization Using SYNCB ................................................ 24
General Description ......................................................................... 1
Interfacing to DSPs .................................................................... 24
Functional Block Diagram .............................................................. 1
Power Control ............................................................................. 24
Revision History ............................................................................... 2
LO Synthesizer ............................................................................ 25
Specifications..................................................................................... 3
Clock Synthesizer ....................................................................... 26
Digital Specifications ................................................................... 5
IF LNA/Mixer ............................................................................. 28
Absolute Maximum Ratings............................................................ 6
Band-Pass Σ-Δ ADC .................................................................. 29
Thermal Resistance ...................................................................... 6
Decimation Filter ....................................................................... 32
ESD Caution .................................................................................. 6
Pin Configuration and Functional Descriptions .......................... 7
Variable Gain Amplifier Operation with Automatic
Gain Control ............................................................................... 33
Typical Performance Characteristics ............................................. 9
Applications Considerations ..................................................... 38
Terminology .................................................................................... 14
External Passive Component Requirements .......................... 40
Serial Peripheral Interface (SPI) ................................................... 15
Applications ................................................................................ 40
Theory of Operation ...................................................................... 17
Layout Example, Evaluation Board, and Software ................. 45
Introduction ................................................................................ 17
SPI Initialization Example ......................................................... 45
Serial Port Interface (SPI) .......................................................... 18
Device SPI Initialization ............................................................ 46
Power-On Reset .......................................................................... 19
Outline Dimensions ....................................................................... 47
Synchronous Serial Interface (SSI) ........................................... 19
Ordering Guide .......................................................................... 47
REVISION HISTORY
2/16—Rev. 0 to Rev. A
Changes to Figure 2 .......................................................................... 7
Changes to Typical Performance Characteristics Section ........... 9
Changes to Figure 19 ...................................................................... 11
Changes to Table 6 .......................................................................... 16
Changed General Description Section to Introduction Section ... 17
Changes to Serial Port Interface (SPI) Section ........................... 18
Added Figure 31; Renumbered Sequentially .............................. 19
Added Power-On Reset Section ................................................... 19
Deleted Table 9; Renumbered Sequentially ................................ 19
Added SSI Control Registers Section and Table 8 to Table 13 .... 21
Changes to Synchronization Using SYNCB Section and
Figure 38 .......................................................................................... 24
Changes to Clock Synthesizer Section ......................................... 26
Changes to Band-Pass Σ-Δ ADC Section and Table 20 ............ 30
Changes to Table 21 ....................................................................... 31
Changes to Variable Gain Control Section ................................. 34
Deleted Table 17 ............................................................................. 34
Added Figure 64 ............................................................................. 36
Changes to Figure 72...................................................................... 40
Changes to Layout Example, Evaluation Board, and
Software Section ............................................................................. 45
Added Figure 77 and SPI Initialization Example Section ......... 45
Added Device SPI Initialization Section and Table 24 .............. 46
Updated Outline Dimensions ....................................................... 47
Changes to Ordering Guide .......................................................... 47
8/03—Revision 0: Initial Version
Rev. A | Page 2 of 47
Data Sheet
AD9864
SPECIFICATIONS
VDDI = VDDF = VDDA = VDDC = VDDL = VDDH = 2.7 V to 3.6 V, VDDQ = VDDP = 2.7 V to 5.5 V, fCLK = 18 MSPS, fIF = 109.65 MHz,
fLO = 107.4 MHz, fREF = 16.8 MHz, unless otherwise noted. Standard operating mode: VGA at minimum attenuation setting, synthesizers
in normal (not fast acquire) mode, decimation factor = 900, 16-bit digital output, and 10 pF load on SSI output pins.
Table 1.
Parameter
SYSTEM DYNAMIC PERFORMANCE1
SSB Noise Figure at Minimum VGA Attenuation2, 3
SSB Noise Figure at Maximum VGA Attenuation2, 3
Dynamic Range with AGC Enabled2, 3
IF Input Clip Point at Maximum VGA Attenuation3
IF Input Clip Point at Minimum VGA Attenuation3
Input Third-Order Intercept (IIP3)
Gain Variation over Temperature
LNA + MIXER
Maximum RF and LO Frequency Range
LNA Input Impedance
Mixer LO Input Resistance
LO SYNTHESIZER
LO Input Frequency
LO Input Amplitude
FREF Frequency (for Sinusoidal Input Only)
FREF Input Amplitude
FREF Slew Rate
Minimum Charge Pump Current at 5 V4
Maximum Charge Pump Current at 5 V4
Charge Pump Output Compliance5
Synthesizer Resolution
CLOCK SYNTHESIZER
CLK Input Frequency
CLK Input Amplitude
Minimum Charge Pump Output Current4
Maximum Charge Pump Output Current4
Charge Pump Output Compliance5
Synthesizer Resolution
Σ-∆ ADC
Resolution
Clock Frequency (fCLK)
Center Frequency
Pass-Band Gain Variation
Alias Attenuation
GAIN CONTROL
Programmable Gain Step
AGC Gain Range
GCP Output Resistance
Temperature
Test Level
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
Full
25°C
25°C
IV
V
V
300
Full
Full
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
VI
VI
VI
IV
7.75
0.3
8
0.3
7.5
0.4
6.25
VDDP − 0.4
Full
Full
Full
Full
Full
Full
IV
IV
VI
VI
VI
VI
13
0.3
26
VDDC
0.4
2.2
VDDQ − 0.4
Full
Full
Full
Full
Full
IV
IV
V
IV
IV
16
13
24
26
Full
Full
Full
V
V
IV
Rev. A | Page 3 of 47
Min
91
−20
−32
−12
Typ
Max
Unit
7.5
13
95
−19
−31
−7.0
0.7
9.5
dB
dB
dB
dBm
dBm
dBm
dB
2
500
370||1.4
1
MHz
Ω||pF
kΩ
300
2.0
26
3
0.67
5.3
0.67
5.3
fCLK/8
1.0
80
50
16
12
72.5
95
MHz
V p-p
MHz
V p-p
V/µs
mA
mA
V
kHz
MHz
V p-p
mA
mA
V
kHz
Bits
MHz
MHz
dB
dB
dB
dB
kΩ
AD9864
Parameter
OVERALL
Analog Supply Voltage (VDDA, VDDF, VDDI)
Digital Supply Voltage (VDDD, VDDC, VDDL)
Interface Supply Voltage (VDDH)6
Charge Pump Supply Voltage (VDDP, VDDQ)
Total Current
Operation Mode7
Standby
OPERATING TEMPERATURE RANGE
Data Sheet
Temperature
Test Level
Min
Typ
Max
Unit
Full
Full
Full
Full
VI
VI
VI
VI
2.7
2.7
1.8
2.7
3.0
3.0
3.6
3.6
3.6
5.5
V
V
V
V
Full
Full
VI
VI
+85
mA
mA
°C
5.0
17
0.01
−40
This includes 0.9 dB loss of matching network.
AGC with DVGA enabled.
3
Measured in 10 kHz bandwidth.
4
Programmable in 0.67 mA steps.
5
Voltage span in which LO (or CLK) charge pump output current is maintained within 5% of nominal value of VDDP/2 (or VDDQ/2).
6
VDDH must be less than VDDD + 0.5 V.
7
Clock VCO off and additional 0.7 mA with VGA at maximum attenuation.
1
2
Rev. A | Page 4 of 47
Data Sheet
AD9864
DIGITAL SPECIFICATIONS
VDDI = VDDF = VDDA = VDDC = VDDL = VDDH = 2.7 V to 3.6 V, VDDQ = VDDP = 2.7 V to 5.5 V, fCLK = 18 MSPS, fIF = 109.65 MHz,
fLO = 107.4 MHz, fREF = 16.8 MHz, unless otherwise noted. Standard operating mode: VGA at minimum attenuation setting, synthesizers
in normal (not fast acquire) mode, decimation factor = 900, 16-bit digital output, and 10 pF load on SSI output pins.
Table 2.
Parameter
DECIMATOR
Decimation Factor1
Pass-Band Width
Pass-Band Gain Variation
Alias Attenuation
SPI READ OPERATION (See Figure 30)
PC Clock Frequency
PC Clock Period (tCLK)
PC Clock High (tHI)
PC Clock Low (tLOW)
PC to PD Setup Time (tDS)
PC to PD Hold Time (tDH)
PE to PC Setup Time (tS)
PC to PE Hold Time (tH)
SPI WRITE OPERATION2 (See Figure 29)
PC Clock Frequency
PC Clock Period (tCLK)
PC Clock High (tHI)
PC Clock Low (tLOW)
PC to PD Setup Time (tDS)
PC to PD Hold Time (tDH)
PC to PD (or DOUTB) Data Valid Time (tDV)
PE to PD Output Valid to High-Z (tEZ)
SSI2 (See Figure 33)
CLKOUT Frequency
CLKOUT Period (tCLK)
CLKOUT Duty Cycle (tHI, tLOW)
CLKOUT to FS Valid Time (tV)
CLKOUT to DOUT Data Valid Time (tDV)
CMOS LOGIC INPUTS3
Logic 1 Voltage (VIH)
Logic 0 Voltage (VIL)
Logic 1 Current (IIH)
Logic 0 Current (IIL)
Input Capacitance
CMOS LOGIC OUTPUTS2, 3, 4
Logic 1 Voltage (VOH)
Logic 0 Voltage (VOL)
Temperature
Test Level
Min
Full
Full
Full
Full
IV
V
IV
IV
48
Full
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
IV
Full
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
IV
100
45
45
2
2
3
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
0.867
38.4
33
−1
−1
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
0.7 × VDDH
Full
Full
IV
IV
VDDH − 0.2
Programmable in steps of 48 or 60.
CMOS output mode with CLOAD = 10 pF and drive strength = 7.
3
Absolute maximum and minimum input/output levels are VDDH + 0.3 V and −0.3 V.
4
IOL = 1 mA; specification is also dependent on drive strength setting.
1
2
Rev. A | Page 5 of 47
Typ
Max
Unit
960
50%
1.2
88
10
MHz
ns
ns
ns
ns
ns
ns
ns
10
MHz
ns
ns
ns
ns
ns
ns
ns
26
1153
67
+1
+1
MHz
ns
ns
ns
ns
100
45
45
2
2
5
5
8
50
fCLKOUT
dB
dBm
0.3 × VDDH
10
10
3
0.2
V
V
µA
µA
pF
V
V
AD9864
Data Sheet
ABSOLUTE MAXIMUM RATINGS
Table 3. AD9864 Absolute Maximum Ratings
Parameter
VDDF, VDDA, VDDC,
VDDD, VDDH, VDDL,
VDDI
VDDF, VDDA, VDDC,
VDDD, VDDH, VDDL,
VDDI
VDDP, VDDQ
GNDF, GNDA, GNDC,
GNDD, GNDH, GNDL,
GNDI, GNDQ, GNDP,
GNDS
MXOP, MXON, LOP, LON,
IFIN, CXIF, CXVL, CXVM
PC, PD, PE, CLKOUT,
DOUTA, DOUTB, FS,
SYNCB
IF2N, IF2P, GCP, GCN
With Respect To
GNDF, GNDA, GNDC,
GNDD, GNDH, GNDL,
GNDI, GNDS
VDDR, VDDA, VDDC,
VDDD, VDDH, VDDL,
VDDI
GNDP, GNDQ
GNDF, GNDA, GNDC,
GNDD, GNDH, GNDL,
GNDI, GNDQ, GNDP,
GNDS
GNDH
VFEFP, VREGN, RREF
GNDA
IOUTC
GNDQ
IOUTL
GNDP
CLKP, CLKN
GNDC
FREF
GNDL
Maximum Junction
Temperature
Storage Temperature
Maximum Lead
Temperature
GNDH
GNDF
Rating
−0.3 to +4.0
−4.0 V to +4.0 V
−0.3 V to +6.0 V
−0.3 V to +0.3 V
−0.3 V to
VDDI + 0.3 V
−0.3 V to
VDDH + 0.3 V
Stresses at or above those listed under Absolute Maximum
Ratings may cause permanent damage to the product. This is a
stress rating only; functional operation of the product at these
or any other conditions above those indicated in the operational
section of this specification is not implied. Operation beyond
the maximum operating conditions for extended periods may
affect product reliability.
THERMAL RESISTANCE
θJA is specified for the worst-case conditions, that is, θJA is
specified for device soldered in circuit board for surface-mount
packages.
Table 4. Thermal Resistance
Package Type
48-Lead LFCSP
−0.3 V to
VDDF + 0.3 V
−0.3 V to
VDDA + 0.3 V
−0.3 V to
VDDQ + 0.3 V
−0.3 V to
VDDP + 0.3 V
−0.3 V to
VDDC + 0.3 V
−0.3 V to
VDDL + 0.3 V
150°C
ESD CAUTION
−65°C to +150°C
300°C
Rev. A | Page 6 of 47
θJA
29.5
Unit
°C/W
Data Sheet
AD9864
GNDP
IOUTL
VDDP
VDDL
CXVM
LOP
LON
GNDI
CXVL
IFIN
CXIF
VDDI
PIN CONFIGURATION AND FUNCTIONAL DESCRIPTIONS
48 47 46 45 44 43 42 41 40 39 38 37
MXOP 1
36 GNDL
PIN 1
IDENTIFIER
MXON 2
35 FREF
GNDF 3
34 GNDS
IF2N 4
33 SYNCB
IF2P 5
VDDF 6
32 GNDH
AD9864
31 FS
TOP VIEW
(Not to Scale)
GCP 7
GCN 8
30 DOUTB
29 DOUTA
VDDA 9
GNDA 10
28 CLKOUT
27 VDDH
VREFP 11
VREFN 12
26 VDDD
25 PE
NOTES
1. EXPOSED PAD. THE BACKSIDE PADDLE CONTACT IS NOT CONNECTED TO GROUND.
A PCB GROUND PAD IS OPTIONAL.
Figure 2. 48-Lead LFCSP Pin Configuration
Table 5. 48-Lead Lead Frame Chip Scale Package (LFCSP) Pin Function Descriptions
Pin No.
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
Mnemonic
MXOP
MXON
GNDF
IF2N
IF2P
VDDF
GCP
GCN
VDDA
GNDA
VREFP
VREFN
RREF
VDDQ
IOUTC
GNDQ
VDDC
GNDC
CLKP
CLKN
GNDS
GNDD
PC
PD
PE
VDDD
VDDH
CLKOUT
DOUTA
DOUTB
FS
Description
Mixer Output, Positive.
Mixer Output, Negative.
Ground for Front End of ADC.
Second IF Input (to ADC), Negative.
Second IF Input (to ADC), Positive.
Positive Supply for Front End of ADC.
Filter Capacitor for ADC Full-Scale Control.
Full-Scale Control Ground.
Positive Supply for ADC Back End.
Ground for ADC Back End.
Voltage Reference, Positive.
Voltage Reference, Negative.
Reference Resistor: Requires 100 kΩ to GNDA.
Positive Supply for Clock Synthesizer.
Clock Synth Charge Pump Out Current.
Ground for Clock Synthesizer Charge Pump.
Positive Supply for Clock Synthesizer.
Ground for Clock Synthesizer.
Sampling Clock Input/Clock VCO Tank, Positive.
Sampling Clock Input/Clock VCO Tank, Negative.
Substrate Ground.
Ground for Digital Functions.
Clock Input for SPI Port.
Data I/O for SPI Port.
Enable Input for SPI Port.
Positive Supply for Internal Digital.
Positive Supply for Digital Interface.
Clock Output for SSI Port.
Data Output for SSI Port.
Data Output for SSI Port (Inverted) or SPI Port.
Frame Sync for SSI Port.
Rev. A | Page 7 of 47
04319-0-002
PD
PC
GNDD
CLKN
GNDS
CLKP
GNDC
VDDC
GNDQ
IOUTC
RREF
VDDQ
13 14 15 16 17 18 19 20 21 22 23 24
AD9864
Pin No.
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
Data Sheet
Mnemonic
GNDH
SYNCB
GNDS
FREF
GNDL
GNDP
IOUTL
VDDP
VDDL
CXVM
LON
LOP
CXVL
GNDI
CXIF
IFIN
VDDI
EPAD
Description
Ground for Digital Interface.
Resets SSI and Decimator Counters; Active Low. Connect to VDDH if unused.
Substrate Ground.
Reference Frequency Input for Both Synthesizers.
Ground for LO Synthesizer.
Ground for LO Synthesizer Charge Pump.
LO Synthesizer Charge Pump Out Current.
Positive Supply for LO Synthesizer Charge Pump.
Positive Supply for LO Synthesizer.
External Filter Capacitor; DC Output of LNA.
LO Input to Mixer and LO Synthesizer, Negative.
LO Input to Mixer and LO Synthesizer, Positive.
External Bypass Capacitor for LNA Power Supply.
Ground for Mixer and LNA.
External Capacitor for Mixer V-I Converter Bias.
First IF Input (to LNA).
Positive Supply for LNA and Mixer.
Exposed Pad. The backside paddle contact is not connected to ground. A PCB ground
pad is optional.
Rev. A | Page 8 of 47
Data Sheet
AD9864
TYPICAL PERFORMANCE CHARACTERISTICS
VDDI = VDDF = VDDA = VDDC = VDDL = VDDH = VDDx , VDDQ = VDDP = 2.7 V to 5.5 V, fCLK = 18 MSPS, fIF = 109.65 MHz,
fLO = 107.4 MHz, fREF = 16.8 MHz, TA=25 Co, LO and CLK synthesizer disabled, 16-bit data with AGC and DVGA enabled, unless
otherwise noted.
0
9.5
9.0
–2
+85°C
–4
IIP3 (dBm)
8.0
+25°C
7.5
+85°C
–6
+25°C
–8
–40°C
–40°C
–10
6.5
3.0
3.3
3.6
VDDx (V)
–12
04319-0-003
6.0
2.7
3.0
2.7
Figure 6. IIP3 vs. Supply
98
–17.5
97
–18.0
DR (dB)
INPUT CLIP POINT (dBm)
+25°C
96
–40°C
95
+85°C
–18.5
+85°C
–19.0
+25°C
–19.5
–40°C
–20.0
93
3.0
3.3
3.6
VDDx (V)
–20.5
2.7
04319-0-004
92
2.7
3.6
VDDx (V)
Figure 3. SSB Noise Figure vs. Supply
94
3.3
04319-0-006
7.0
3.0
3.3
3.6
VDDx (V)
Figure 4. Dynamic Range (DR) vs. Supply
Figure 7. Maximum VGA Attenuation Clip Point vs. Supply
–29.5
0.1
0
GAIN VARIATION (dB)
–0.1
–30.5
+85°C
+25°C
3.3
VDDx (V)
–0.4
–0.5
–0.7
–40°C
3.0
–0.3
–0.6
–31.5
–32.0
2.7
–0.2
3.6
Figure 5. Minimum VGA Attenuation Clip Point vs. Supply
–0.8
–20
–17
–14
–11
–8
–5
LO DRIVE (dBm)
Figure 8. Normalized Gain Variation vs. LO Drive (VDDx = 3.0 V)
Rev. A | Page 9 of 47
04319-0-008
–31.0
04319-0-005
INPUT CLIP POINT (dBm)
–30.0
04319-0-007
NF (dB)
8.5
AD9864
Data Sheet
VDDI = VDDF = VDDA = VDDC = VDDL = VDDH = 3.0 V , VDDQ = VDDP = 2.7 V to 5.5 V, fCLK = 18 MSPS, fIF = 109.65 MHz, fLO =
107.4 MHz, fREF = 16.8 MHz, TA=25 Co, LO and CLK synthesizer disabled, unless otherwise noted.
–10
–15
–40
7.8
–50
IMD
7.6
–60
7.4
7.2
–10
–15
0
–5
5
–33
–80
–36
–36 –33 –30 –27 –24 –21 –18 –15 –12
Figure 9. Noise Figure and IMD vs. LO Drive (VDDx = 3.0 V)
–61
2.7V
2.7V
–27
3.0V
–85
–30
3.3V
–91
–97
–33
–36
3.6V
–39
–42
–28
–26
–24
–22
–20
–18
–16
–14
04319-0-010
–109
IFIN (dBm)
–115
–51
–48
–45
–42
–39
–36
–33
–45
–30
IFIN (dBm)
Figure 10. Gain Compression vs. IFIN
Figure 13. IMD vs. IFIN
10.0
10.0
16-BIT
I/Q DATA
9.5
NOISE FIGURE (dB)
16-BIT
I/Q DATA WITH
DVGA ENABLED
9.0
8.5
16-BIT
DATA
9.0
16-BIT DATA
WITH DVGA ENABLED
8.5
8.0
24-BIT
I/Q DATA
24-BIT
DATA
100
CHANNEL BANDWIDTH (kHz)
1000
04319-0-011
7.5
10
–24
–103
–12
8.0
–21
–79
IMD (dBm)
3.0V
–18
PIN
–73
–10
9.5
–0
–15
–67
–6
–14
–30
–3
–55
3.3V
–8
–6
Figure 12. Gain Compression vs. IFIN with 16 dB LNA Attenuator Enabled
3.6V
–4
–9
IFIN (dBm)
ADC DOES NOT GO INTO
HARD COMPRESSION
–2
dBFS
–30
–70
LO DRIVE (dBm)
0
–27
Figure 11. Noise Figure vs. Bandwidth
(Minimum Attenuation, fCLK = 13 MSPS)
7.5
10
100
CHANNEL BANDWIDTH (kHz)
Figure 14. Noise Figure vs. Bandwidth
(Minimum Attenuation, fCLK = 18 MSPS)
Rev. A | Page 10 of 47
1000
04319-0-014
7.0
–20
–24
04319 0 013
8.0
–21
04319-0-012
NF
PIN (dBFS)
–30
8.2
–18
dBm
–20
8.4
NOISE FIGURE (dB)
NOISE FIGURE (dBc)
8.6
–12
IMD WITH IFIN = –36 dBm (dBc)
8.8
0
04319-0-009
9.0
Data Sheet
AD9864
VDDI = VDDF = VDDA = VDDC = VDDL = VDDH = 3.0 V , VDDQ = VDDP = 2.7 V to 5.5 V, fCLK = 18 MSPS, fIF = 109.65 MHz, fLO =
107.4 MHz, fREF = 16.8 MHz, TA=25 Co, LO and CLK Synthesizer Disabled, unless otherwise noted.
11.5
10.0
16-BIT DATA
WITH DVGA ENABLED
11.0
10.5
24-BIT
DATA
16-BIT
DATA
9.0
NOISE FIGURE (dB)
NOISE FIGURE (dB)
9.5
8.5
BW = 27.08kHz
(K = 0, M = 3)
10.0
BW = 12.04kHz
(K = 0, M = 8)
9.5
9.0
BW = 6.78kHz
(K = 0, M = 15)
8.5
8.0
8.0
100
1000
CHANNEL BANDWIDTH (kHz)
7.0
04319-0-015
7.5
10
14
13
BW = 75kHz
(K = 0, M = 1)
12
BW = 50kHz
(K = 0, M = 2)
11
NOISE FIGURE (dB)
BW = 15kHz
(K = 0, M = 9)
10
9
BW = 90.28kHz
(K = 1, M = 2)
12
BW = 135.42kHz
(K = 1, M = 1)
11
10
BW = 27.08kHz
(K = 1, M = 9)
9
8
6
12
9
VGA ATTENUATION (dB)
7
–40
–50
–5
–30
–10
–40
–25
–90
–30
–100
IMD (dBFS)
–80
–39
–36
–33
–30
–27
IFIN (dB)
–24
–45
IMD
–80
–25
–90
–30
–35
–40
–120
04319-0-017
–42
–20
–70
–110
–40
–120
–15
PIN
–100
–35
–110
–10
–60
PIN (dBFS)
IMD
–5
–50
–20
–70
12
Figure 19. Noise Figure vs. VGA Attenuation (fCLK = 26 MSPS)
–15
PIN
–60
9
6
VGA ATTENUATION (dB)
Figure 16. Noise Figure vs. VGA Attenuation (fCLK = 18 MSPS)
–30
3
0
–130
–45
Figure 17. IMD vs. IFIN (fCLK = 13 MSPS)
–42
–39
–36
–33
–30
–27
IFIN (dBm)
Figure 20. IMD vs. IFIN (fCLK = 18 MSPS)
Rev. A | Page 11 of 47
PIN (dBFS)
3
04319-0-016
0
04319-0-019
8
–24
–45
04319-0-020
NOISE FIGURE (dB)
13
IMD (dBFS)
12
Figure 18. Noise Figure vs. VGA Attenuation (fCLK = 13 MSPS)
14
–130
–45
9
6
VGA ATTENUATION (dB)
Figure 15. Noise Figure vs. Bandwidth
(Minimum Attenuation, fCLK = 26 MSPS)
7
3
0
04319-0-018
7.5
AD9864
Data Sheet
VDDI = VDDF = VDDA = VDDC = VDDL = VDDH = 3.0 V , VDDQ = VDDP = 2.7 V to 5.5 V, fCLK = 18 MSPS, fIF = 109.65 MHz, fLO =
107.4 MHz, fREF = 16.8 MHz, TA=25 Co, LO and CLK synthesizer disabled, 16-bit data with AGC and DVGA enabled, unless otherwise noted.
–40
PIN
–10
12
16-BIT WITH DVGA
–15
IMD
–80
–25
–90
–30
–100
PIN (dBFS)
–20
–70
11
10
9
–130
–45
–42
–39
–36
–33
–30
–27
–24
–40
7
–45
6
04319-0-021
–120
IFIN (dBm)
24-BIT
8
–35
–110
0
50
100
150
200
250
300
350
400
450
500
FREQUENCY (MHz)
Figure 21. IMD vs. IFIN (fCLK = 26 MSPS)
Figure 24. Noise Figure vs. Frequency
(Minimum Attenuation, fCLK = 18 MSPS, BW = 10 kHz)
13
0
16-BIT WITH DVGA
12
11
10
9
–4
IIP3 (dBm)
24-BIT
–6
8
0
50
100
150
200
250
300
350
400
450
500
FREQUENCY (MHz)
–10
04319-0-022
50
0
100
150
200
250
300
350
400
450
Figure 25. Input IIP3 vs. Frequency (fCLK = 18 MSPS)
Figure 22. Noise Figure vs. Frequency
(Minimum Attenuation, fCLK = 26 MSPS, BW = 10 kHz)
0
20.0
128
18.5
112
AGC
NOISE FIGURE (dBc)
–2
–4
–6
17.0
96
15.5
80
14.0
64
12.5
48
NOISE FIGURE
11.0
–8
32
0
50
100
150
200
250
300
350
400
450
FREQUENCY (MHz)
500
04319-0-023
9.5
–10
500
FREQUENCY (MHz)
Figure 23. Input IIP3 vs. Frequency (fCLK = 26 MSPS)
8.0
–55
16
–50
–45
–40
–35
–30
–25
–20
–15
–10
–5
INTERFERER LEVEL (dBm)
Figure 26. Noise Figure vs. Interferer Level (16-Bit Data,
BW = 12.5 kHz, AGCR = 1, fINTERFERER = fIF + 110 kHz)
Rev. A | Page 12 of 47
MEAN AGC ATTN VALUE
6
04319-0-025
–8
7
0
04319-0-026
NOISE FIGURE (dB)
–2
IIP3 (dBm)
IMD (dBFS)
–60
13
NOISE FIGURE (dB)
–50
–5
04319-0-024
–30
Data Sheet
AD9864
VDDI = VDDF = VDDA = VDDC = VDDL = VDDH = 3.0 V , VDDQ = VDDP = 2.7 V to 5.5 V, fCLK = 18 MSPS, fIF = 109.65 MHz, fLO =
107.4 MHz, fREF = 16.8 MHz, TA=25 Co, LO and CLK Synthesizer Disabled, AGC enabled, unless otherwise noted.
224
15
14
192
14
13
160
NOISE FIGURE
11
96
10
64
9
32
8
–50
–45
–40
–35
–30
–25
–20
–15
0
–10
INTERFERER LEVEL (dBm)
AGC ATTN
96
13
64
12
11
NOISE FIGURE
10
32
9
8
–65
–55
–45
–35
–25
–15
–5
0
INTERFERER LEVEL (dBm)
Figure 28. Noise Figure vs. Interferer Level (24-Bit Data,
BW = 12.5 kHz, AGCR = 1, fINTERFERER = fIF + 110 kHz)
Figure 27. Noise Figure vs. Interferer Level (16-Bit Data with DVGA,
BW = 12.5 kHz, AGCR = 1, fINTERFERER = fIF + 110 kHz)
Rev. A | Page 13 of 47
04319-0-028
128
12
NOISE FIGURE (dBc)
AGC ATTN
04319-0-027
NOISE FIGURE (dBc)
15
128
MEAN AGC ATTN VALUE
16
MEAN AGC ATTN VALUE
256
16
AD9864
Data Sheet
TERMINOLOGY
Single Sideband Noise Figure (SSB NF)
Noise figure (NF) is defined as the degradation in SNR
performance (in dB) of an IF input signal after it passes through
a component or system. It can be expressed with the equation
Noise Figure = 10 × log(SNRIN/SNROUT)
The term SSB is applicable for heterodyne systems containing a
mixer. It indicates that the desired signal spectrum resides on only
one side of the LO frequency (that is, single sideband); therefore,
a noiseless mixer has a noise figure of 3 dB.
The SSB noise figure of the AD9864 is determined by the equation
SSB NF = PIN − [10 × log(BW)] − (−174 dBm/Hz) − SNR
where:
PIN is the input power of an unmodulated carrier.
BW is the noise measurement bandwidth.
−174 dBm/Hz is the thermal noise floor at 293 K.
SNR is the measured signal-to-noise ratio in dB of the AD9864.
Note that PIN is set to −85 dBm to minimize any degradation in
measured SNR due to phase noise from the RF and LO signal
generators. The IF frequency, CLK frequency, and decimation
factors are selected to minimize any spurious components
falling within the measurement bandwidth. Note also that a
bandwidth of 10 kHz is used for the data sheet specification. All
references to noise figures within this data sheet imply single
sideband noise figure.
Input Third-Order Intercept (IIP3)
IIP3 is a figure of merit used to determine the susceptibility of a
component or system to intermodulation distortion (IMD) from
its third-order nonlinearities. Two unmodulated carriers at a
specified frequency relationship (f1 and f2) are injected into a
nonlinear system exhibiting third-order nonlinearities producing
IMD components at 2f1 − f2 and 2f2 − f1. IIP3 graphically
represents the extrapolated intersection of the carrier’s input
power with the third-order IMD component when plotted in dB.
The difference in power (D in dBc) between the two carriers, and
the resulting third-order IMD components can be determined
from the equation
Dynamic Range (DR)
Dynamic range is the measure of a small target input signal
(PTARGET) in the presence of a large unwanted interferer signal
(PINTER). Typically, the large signal causes some unwanted
characteristic of the component or system to degrade, thus
making it unable to detect the smaller target signal correctly.
For the AD9864, it is often a degradation in noise figure at
increased VGA attenuation settings that limits its dynamic
range.
The test method for the AD9864 is as follows. The small target
signal (an unmodulated carrier) is input at the center of the IF
frequency, and its power level (PTARGET) is adjusted to achieve an
SNRTARGET of 6 dB. The power of the signal is then increased by
3 dB prior to injecting the interferer signal. The offset frequency
of the interferer signal is selected so that aliases produced by
the response of the decimation filter, as well as phase noise from
the LO (due to reciprocal mixing), do not fall back within the
measurement bandwidth. For this reason, an offset of 110 kHz
was selected. The interferer signal (also an unmodulated carrier)
is then injected into the input and its power level is increased to
the point (PINTER) where the target signal SNR is reduced to 6 dB.
The dynamic range is determined with the equation
DR = PINTER − PTARGET + SNRTARGET
Note that the AGC of the AD9864 is enabled for this test.
IF Input Clip Point
The IF input clip point is defined as the input power that results
in a digital output level 2 dB below full scale. Unlike other linear
components that typically exhibit a soft compression
(characterized by its 1 dB compression point), an ADC exhibits
a hard compression when its input signal exceeds its rated
maximum input signal range. For the AD9864, which contains
a Σ-∆ ADC, hard compression must be avoided because it
causes severe SNR degradation.
D = 2 × (IIP3 − PIN)
Rev. A | Page 14 of 47
Data Sheet
AD9864
SERIAL PERIPHERAL INTERFACE (SPI)
The SPI is a bidirectional serial port. It is used to load the configuration information into the registers listed in Table 6, as well as to read
back their contents. Table 6 provides a list of the registers that can be programmed through the SPI port. Addresses and default values are
given in hexadecimal form.
Table 6. SPI Address Map
Address
(Hex)
Bit(s)
Power Control Registers
0x00
[7:0]
0x01
0x02
AGC
0x03
0x04
0x05
0x06
Width
Default
Value
Name
Description
8
0xFF
STBY
[3:2]
[1:0]
[7:0]
2
2
8
0x00
0x00
0x00
CKOB
ADCB
TEST
Standby control bits (REF, LO, CKO, CK, GC, LNAMX, unused, and ADC).
Default is power-up condition of standby.
CK oscillator bias (0 = 0.25 mA, 1 = 0.35 mA, 2 = 0.40 mA, 3 = 0.65 mA).
Do not use.
Factory test mode. Do not use.
7
[6:0]
[7:0]
[7:4]
[3:0]
7
[6:4]
3
[2:0]
1
7
8
4
4
1
3
1
3
0
0x00
0x00
0x00
0x00
0
0x00
0
0x00
ATTEN
AGCG [14:8]
AGCG [7:0]
AGCA
AGCD
AGCV
AGCO
AGCF
AGCR
3
1
4
0
0x04
Unused
K
M
6
8
0x00
0x38
LOR [13:8]
LOR [7:0]
Decimation Factor
0x07
[7:5]
4
[3:0]
LO Synthesizer
0x08
[5:0]
0x09
[7:0]
Decimation factor = 60 × (M + 1), if K = 0; 48 × (M + 1), if K = 1.
Default is decimate-by-300.
0x0A
[7:5]
[4:0]
3
5
0x05
0x00
LOA
LOB [12:8]
0x0B
0x0C
[7:0]
6
5
[4:2]
[1:0]
8
1
1
3
2
0x1D
0
0
0x00
0x03
LOB [7:0]
LOF
LOINV
LOI
LOTM
0x0D
[5:0]
0x0E
[7:0]
Clock Synthesizer
0x10
[5:0]
0x11
[7:0]
6
8
0x00
0x04
LOFA [13:8]
LOFA [7:0]
6
8
0x00
0x38
CKR [13:8]
CKR [7:0]
0x12
0x13
5
8
0x00
0x3C
CKN [12:8]
CKN [7:0]
[4:0]
[7:0]
Apply 16 dB attenuation in the front end.
AGC attenuation setting (7 MSBs of a 15-bit unsigned word).
AGC attenuation setting (8 LSBs of a 15-bit unsigned word).
AGC attack bandwidth setting. Default yields 50 Hz loop bandwidth.
AGC decay time setting. Default is decay time = attack time.
Enable digital VGA to increase AGC range by 12 dB.
AGC overload update setting. Default is slowest update.
Fast AGC (minimizes resistance seen between GCP and GCN).
AGC enable/reference level (disabled, 3 dB, 6 dB, 9 dB, 12 dB, 15 dB
below clip).
Reference frequency divider (6 MSBs of a 14-bit word).
Reference frequency divisor (8 LSBs of a 14-bit word). Default (56) yields
300 kHz from fREF = 16.8 MHz.
A counter (prescaler control counter).
B counter MSB (5 MSB of a 13-bit word). Default LOA and LOB values
yield 300 kHz from 73.35 MHz to 2.25 MHz.
B counter LSB (8 LSB of a 13-bit word).
Enable fast acquire.
Invert charge pump (0 = source current to increase VCO frequency).
Charge pump current in normal operation. IPUMP = (LOI +1) × 0.625 mA.
Manual control of LO charge pump (0 = off, 1 = up, 2 = down, and
3 = normal).
LO fast acquire time unit (6 MSBs of a 14-bit word).
LO fast acquire time unit (8 LSBs of a 14-bit word).
Reference frequency divisor (6 MSBs of a 14-bit word).
Reference frequency divisor (8 LSBs of a 14-bit word). Default yields
300 kHz from fREF = 16.8 MHz; minimum = 3, maximum = 16383.
Synthesized frequency divisor (5 MSBs of a 13-bit word).
Synthesized frequency divisor (8 LSBs of a 13-bit word). Default yields
300 kHz from 18 MHz; minimum = 3, maximum = 8191.
Rev. A | Page 15 of 47
AD9864
Address
(Hex)
0x14
Data Sheet
Width
1
1
3
2
Default
Value
0
0
0x00
0x03
Name
CKF
CKINV
CKI
CKTM
0x15
[5:0]
0x16
[7:0]
SSI Control
0x18
[7:0]
6
8
0x00
0x04
CKFA [13:8]
CKFA [7:0]
8
0x12
SSICRA
0x19
8
0x07
SSICRB
Bit(s)
6
5
[4:2]
[1:0]
[7:0]
0x1A
[3:0]
4
0x01
ADC Tuning
0x1C
1
1
0
0
1
0
0x1D
[3:0]
3
0x00
0x1E
[5:0]
6
0x00
0x1F
[7:0]
8
0x00
Test Registers and SPI Port Read Enable
0x37
[7:0]
8
0x00
0x38
[7:1]
7
0x00
0
1
0
0x39
[7:0]
8
0x00
0x3A
[7:4]
4
0x00
3
1
0
[2:0]
3
0x00
0x3B
[7:4]
4
0x00
3
1
0
[2:0]
3
0x00
0x3C to
[7:0]
8
0x00
0x3D
0x3E
7
1
0
6
1
0
[5:3]
3
0
2
2
0
1
1
0
0
1
0
0x3F
[7:0]
8
Subject
to change
Description
Enable fast acquire.
Invert charge pump (0 = source current to increase VCO frequency).
Charge pump current in normal operation. IPUMP = (CKI + 1) × 0.625 mA.
Manual control of CLK charge pump (0 = off, 1 = up, 2 = down, and
3 = normal).
CK fast acquire time unit (6 LSBs of a 14-bit word).
CK fast acquire time unit (8 LSBs of a 14-bit word).
SSIORD
SSI Control Register A. See the SSI Control Registers section. Default is
FS and CLKOUT three-stated.
SSI Control Register B. See the SSI Control Registers section (16-bit data,
maximum drive strength).
Output rate divisor. fCLKOUT = fCLK/SSIORD.
TUNE_LC
TUNE_RC
CAPL1 [2:0]
CAPL0 [5:0]
CAPR
Perform tuning on LC portion of the ADC (cleared when done).
Perform tuning on RC portion of the ADC (cleared when done).
Coarse capacitance setting of LC tank (LSB is 25 pF, differential).
Fine capacitance setting of LC tank (LSB is 0.4 pF, differential).
Capacitance setting for RC resonator (64 LSB of fixed capacitance).
TEST
TEST
DACCR
DACDATA
TEST
SPIREN
TEST
TEST
TRI
TEST
TEST
Factory test mode. Do not use.
Factory test mode. Do not use.
Manual feedback DAC control
Feedback DAC data setting in manual mode.
Factory test mode. Do not use.
Enable read from SPI port.
Factory test mode. Do not use.
Factory test mode. Do not use.
Three-state DOUTB.
Factory test mode. Do not use.
Factory test mode. Do not use.
TEST
OVL
TEST
RC_Q
RC_BYP
SC_BYP
ID
Factory test mode. Do not use.
ADC overload detector.
Factory test mode. Do not use.
RC Q enhancement.
Bypass RC resonator.
Bypass SC resonators.
Revision ID (read-only). A write of 0x99 to this register is equivalent to a
power-on reset.
Rev. A | Page 16 of 47
Data Sheet
AD9864
THEORY OF OPERATION
INTRODUCTION
The AD9864 is a general-purpose, narrow-band IF subsystem
that digitizes a low level, 10 MHz to 300 MHz IF input with a
signal bandwidth ranging from 6.8 kHz to 270 kHz. The signal
chain of the AD9864 consists of an LNA, a mixer, a band-pass
Σ-∆ ADC, and a decimation filter with programmable
decimation factor.
The input LNA is a fixed gain block with an input impedance of
approximately 370 Ω||1.4 pF. The LNA input is single-ended
and self biasing, allowing the input IF to be ac-coupled. The
LNA can be disabled through the serial interface, providing a
fixed 16 dB attenuation to the input signal.
The LNA drives the input port of a Gilbert-type active mixer.
The mixer LO port is driven by the on-chip LO buffer, which can
be driven externally, single-ended, or differential. The LO buffer
inputs are self biasing and allow the LO input to be ac-coupled.
The open-collector outputs of the mixer drive an external
resonant tank consisting of a differential LC network tuned to
the IF of the band-pass Σ-∆ ADC.
The external differential LC tank forms the resonator for the
first stage of the band-pass Σ-∆ ADC. The tank LC values must
be selected for a center frequency of fCLK/8, where fCLK is the
sample rate of the ADC. The fCLK/8 frequency is the IF digitized
by the band-pass Σ-∆ ADC. On-chip calibration allows
standard tolerance inductor and capacitor values. The
calibration is typically performed once at power-up.
The ADC contains a sixth-order, multibit band-pass Σ-∆
modulator that achieves very high instantaneous dynamic range
over a narrow frequency band centered at fCLK/8. The modulator
output is quadrature mixed to baseband and filtered by three
cascaded linear phase FIR filters to remove out-of-band noise.
The first FIR filter is a fixed, decimate by 12, using a fourthorder comb filter. The second FIR filter also uses a fourth-order
comb filter with programmable decimation from 1 to 16. The
third FIR stage is programmable for decimation of either 4 or 5.
The cascaded decimation factor is programmable from 48 to 960.
The decimation filter data is output via the synchronous serial
interface (SSI) of the chip.
Additional functionality built into the AD9864 includes LO and
clock synthesizers, programmable AGC, and a flexible synchronous
serial interface for output data.
The LO synthesizer is a programmable phase-locked loop (PLL)
consisting of a low noise phase frequency detector (PFD), a
variable output current charge pump (CP), a 14-bit reference
divider, A and B counters, and a dual modulus prescaler. The
user only needs to add an appropriate loop filter and VCO for
complete operation.
The clock synthesizer is equivalent to the LO synthesizer with
the following differences:
•
•
It does not include the prescaler or A counter.
It includes a negative resistance core used for VCO
generation.
The AD9864 contains both a variable gain amplifier (VGA) and
a digital VGA (DVGA). Both of these can operate manually or
automatically. In manual mode, the gain for each is programmed
through the SPI. In automatic gain control mode, the gains are
adjusted automatically to ensure that the ADC does not clip and
that the rms output level of the ADC is equal to a programmable
reference level.
The VGA has 12 dB of attenuation range and is implemented by
adjusting the ADC full-scale reference level. The DVGA gain is
implemented by scaling the output of the decimation filter. The
DVGA is most useful in extending the dynamic range in narrowband applications requiring 16-bit I and Q data format.
The SSI provides a programmable frame structure, allowing 24-bit
or 16-bit I and Q data and flexibility by including attenuation and
RSSI data if required.
Rev. A | Page 17 of 47
AD9864
Data Sheet
SERIAL PORT INTERFACE (SPI)
register are shifted into the data pin (PD) on the rising edges of
the next eight clock cycles. PE stays low during the operation
and goes high at the end of the transfer. If PE rises before the
eight clock cycles have passed, the operation is aborted. If PE
stays low for an additional eight clock cycles, the destination
address is incremented and another eight bits of data are shifted
in. Again, if PE rises early, the current byte is ignored. By using
this implicit addressing mode, the chip can be configured with
a single write operation. Registers identified as being subject to
frequent updates, namely those associated with power control
and AGC operation, have been assigned adjacent addresses to
minimize the time required to update them. Note that multibyte
registers are big endian (the most significant byte has the lower
address) and are updated when a write to the least significant
byte occurs.
The serial port of the AD9864 has 3-wire or 4-wire SPI capability,
allowing read/write access to all registers that configure the internal
parameters of the device. The default 3-wire serial communication
port consists of a clock (PC), peripheral enable (PE), and
bidirectional data (PD) signal. The inputs to PC, PE, and PD
contain a Schmitt trigger with a nominal hysteresis of 0.4 V
centered about the digital interface supply, that is, VDDH/2.
A 4-wire SPI interface can be enabled by setting the MSB of the
SSICRB register (Register 0x19, Bit 7) and setting Register 0x3A
to 0x00, resulting in the output data appearing on only the
DOUTB pin with the PD pin functioning as an input pin only.
Note that because the default power-up state sets DOUTB low,
bus contention is possible for systems sharing the SPI output
line. To avoid any bus contention, the DOUTB pin can be threestated by setting the fourth control bit in the three-state bit
(Register 0x3B, Bit 3). This bit can then be toggled to gain
access to the shared SPI output line.
Figure 30 and Figure 31 illustrates the timing for 3-wire and
4-wire SPI read operations. Although the AD9864 does not
require read access for proper operation, it is often useful in the
product development phase or for system authentication. Note
that the readback enable bit (Register 0x3A, Bit 3) must be set
for a read operation with a 3-wire SPI interface. For 4-wire SPI
operation, this bit remains low (Register 0x3A = 0x00) but
DOUTB is enabled via the SSICRB register (Register 0x19, Bit 7).
Note that for the 4-wire SPI interface, the eight data bits appear
on the DOUTB pin with the same timing relationship as those
appearing at PD for 3-wire SPI interface case.
An 8-bit instruction header must accompany each read and
write SPI operation. Only the write operation supports an autoincrement mode, which allows the entire chip to be configured
in a single write operation. The instruction header is shown in
Table 7. It includes a read/not-write indicator bit, six address
bits, and a Don’t Care bit. The data bits immediately follow the
instruction header for both read and write operations. Note that
the address and data are always given MSB first.
After the peripheral enable (PE) signal goes low, data (PD)
pertaining to the instruction header is read on the rising edges
of the clock (PC). A read operation occurs if the read/not-write
indicator is set high. After the address bits of the instruction
header are read, the eight data bits pertaining to the specified
register are shifted out of the data pin (PD) on the falling edges
of the next eight clock cycles. After the last data bit is shifted
out, the user must return PE high, causing PD to become threestated (for 3-wire case) and return to its normal status as an
input pin. Since the auto-increment mode is not supported for
read operations, an instruction header is required for each
register read operation and PE must return high before
initiating the next read operation.
Table 7. Instruction Header Information
1
I6
A5
I5
A4
I4
A3
I3
A2
I2
A1
LSB
I0
X1
I1
A0
X = don’t care.
Figure 29 illustrates the timing requirements for a write
operation to the SPI port. After the peripheral enable (PE)
signal goes low, data (PD) pertaining to the instruction header
is read on the rising edges of the clock (PC). To initiate a write
operation, the read/not-write bit is set low. After the instruction
header is read, the eight data bits pertaining to the specified
tS
PE
tCLK
tHI
tH
tLOW
PC
tDS
PD
tDH
R/W
A5
A4
A0
DON'T
CARE
D7
Figure 29. SPI Write Operation Timing
Rev. A | Page 18 of 47
D6
D1
D0
04319-0-029
MSB
I7
R/W
Data Sheet
AD9864
tCLK
tS
PE
tHI
tLOW
PC
tDV
tDH
R/W
PD
A5
A1
A0
tEZ
DON'T
CARE
D7
D6
D1
04319-0-130
tDS
D0
Figure 30. 3-Wire SPI Read Operation Timing
tS
PE
tCLK
tHI
tLOW
PC
PD
DOUTB
DON'T
T
CARE
tDV
tDH
R/W
A5
DON'T
CARE
DON'T
CARE
A1
A0
DON'T
CARE
DON'T
CARE
DON'T
CARE
D7
D6
D1
D0
DON'T
CARE
04319-0-030
tDS
Figure 31. 4-Wire SPI Read Operation Timing
POWER-ON RESET
The SPI registers are automatically set to their default settings
upon power-up when the VDDD supply crosses a threshold. This
ensures that the AD9864 is in a known state and placed in standby
for minimal power consumption. In the unlikely event that the
SPI registers were not reset to their default settings, an equivalent
software reset by writing 0x99 to Register 0x3F can be used as the
first SPI write command to provide additional assurance.
SYNCHRONOUS SERIAL INTERFACE (SSI)
The AD9864 provides a high degree of programmability of its
SSI output data format, control signals, and timing parameters
to accommodate various digital interfaces. In a 3-wire digital
interface, the AD9864 provides a frame sync signal (FS), a clock
output (CLKOUT), and a serial data stream (DOUTA) signal to
the host device. In a 2-wire interface, the frame sync information
is embedded into the data stream, thus only CLKOUT and
DOUTA output signals are provided to the host device. The SSI
control registers are SSICRA, SSICRB, and SSIORD. Table 8 to
Table 13 show the bit fields associated with these registers.
The primary output of the AD9864 is the converted I and Q
demodulated signal available from the SSI port as a serial bit
stream contained within a frame. The output frame rate is equal
to the modulator clock frequency (fCLK) divided by the digital
filter’s decimation factor that is programmed in the Decimator
Register (0x07). The bit stream consists of an I word followed
by a Q word, where each word is either 24 bits or 16 bits long
and is given MSB first in twos complement form. Two optional
bytes may also be included within the SSI frame following the Q
word. One byte contains the AGC attenuation and the other
byte contains both a count of modulator reset events and an
estimate of the received signal amplitude (relative to full scale of
the AD9864 ADC). Figure 32 illustrates the structure of the SSI
data frames in a number of SSI modes.
The two optional bytes are output if the EAGC bit of SSICRA is
set. The first byte contains the 8-bit attenuation setting (0 = no
attenuation, 255 = 24 dB of attenuation), whereas the second byte
contains a 2-bit reset field and 6-bit received signal strength
field. The reset field contains the number of modulator reset
events since the last report, saturating at 3. The received signal
strength (RSSI) field is a linear estimate of the signal strength at
the output of the first decimation stage; 60 corresponds to a
full-scale signal.
The two optional bytes follow the I and Q data as a 16-bit word
provided that the AAGC bit of SSICRA is not set. If the AAGC
bit is set, the two bytes follow the I and Q data in an alternating
fashion. In this alternate AGC data mode, the LSB of the byte
containing the AGC attenuation is a 0, whereas the LSB of the
byte containing reset and RSSI information is always a 1.
In a 2-wire interface, the embedded frame sync bit (EFS) within
the SSICRA register is set to 1. In this mode, the framing
information is embedded in the data stream, with each eight
bits of data surrounded by a start bit (low) and a stop bit (high),
and each frame ends with at least 10 high bits. FS remains either
low or three-stated (default), depending on the state of the SFST
bit. Other control bits can be used to invert the frame sync
(SFSI), to delay the frame sync pulse by one clock period
(SLFS), to invert the clock (SCKI), or to three-state the clock
(SCKT). Note that if EFS is set, SLFS is a don’t care bit.
Rev. A | Page 19 of 47
AD9864
Data Sheet
the pass band of the target signal. Users must verify that the
output bit rate is sufficient to accommodate the required number
of bits per frame for a selected word size and decimation factor.
Idle (high) bits are used to fill out each frame.
The SSIORD register controls the output bit rate (fCLKOUT) of the
serial bit stream. fCLKOUT can be set equal to the modulator clock
frequency (fCLK) or an integer fraction of it. It is equal to fCLK divided
by the contents of the SSIORD register. Note that fCLKOUT must
be chosen such that it does not introduce harmful spurs within
24-BIT I AND Q, EAGC = 0, AAGC = X:48 DATA BITS
I(23:0)
Q(23:0)
24-BIT I AND Q, EAGC = 1, AAGC = 0:64 DATA BITS
I(23:0)
ATTN(7:0)
Q(23:0)
SSI(5:0)
RESET COUNT
16-BIT I AND Q, EAGC = 0, AAGC = X:32 DATA BITS
I(15:0)
Q(15:0)
16-BIT I AND Q, EAGC = 0, AAGC = 0:32 DATA BITS
I(15:0)
Q(15:0)
ATTN(7:0)
SSI(5:0)
I(15:0)
Q(15:0)
ATTN(7:1) 0
I(15:0)
Q(15:0)
SSI(5:1) 1
RESET COUNT
Figure 32. SSI Frame Structure
Rev. A | Page 20 of 47
04319-0-031
16-BIT I AND Q, EAGC = 1, AAGC = 1:40 DATA BITS
Data Sheet
AD9864
SSI CONTROL REGISTERS
SSICRA (Address 0x18)
Table 8. SSICRA Bitmap
Bit 7
AAGC
Bit 6
EAGC
Bit 5
EFS
Bit 4
SFST
Bit 3
SFSI
Bit 2
SLFS
Bit 1
SCKT
Bit 0
SCKI
Bit 1
DS_1
Bit 0
DS_0
Table 9. SSICRA Bit Descriptions
Bit(s)
7
6
5
4
3
2
1
0
Name
AAGC
EAGC
EFS
SFST
SFSI
SLFS
SCKT
SCKI
Width
1
1
1
1
1
1
1
1
Default
0
0
0
1
0
0
1
0
Description
Alternate AGC data bytes.
Embed AGC data.
Embed frame sync.
Three-state frame sync.
Invert frame sync.
Late frame sync (1 = late, 0 = early).
Three-state CLKOUT.
Invert CLKOUT.
SSICRB (Address 0x19)
Table 10. SSICRB Bitmap
Bit 7
4_SPI
Bit 6
Reserved
Bit 5
Reserved
Bit 4
Reserved
Bit 3
DW
Bit 2
DS_2
Table 11. SSICRB Bit Descriptions
Bit(s)
7
[6:4]
3
[2:0]
Name
4_SPI
Reserved
DW
DS
Width
1
3
1
3
Default
0
0
0
7
Description
Enable 4-wire SPI interface for SPI read operation via DOUTB.
Reserved.
I/Q data-word width (0 = 16 bit, 1 = 24 bit). Automatically 16-bit when AGCV = 1.
FS, CLKOUT, and DOUT drive strength Level 0 to Level 7, with 7 being the highest level.
SSIORD (Address 0x1A)
Table 12. SSIORD Bitmap
Bit 7
Reserved
Bit 6
Reserved
Bit 5
Reserved
Bit 4
Reserved
Bit 3
DIV_3
Bit 2
DIV_2
Bit 1
DIV_1
Table 13. SSIORD Bit Descriptions
Bit(s)
[7:4]
[3:0]
Name
Reserved
SSIORD
Width
4
4
Default
0
1
Description
Reserved.
Output bit rate divisor setting fCLKOUT = fCLK/SSIORD where SSIORD = 1 to 15.
Rev. A | Page 21 of 47
Bit 0
DIV_0
AD9864
Data Sheet
CLKOUT
FS
DOUT
I15
I0
Q15
Q14
Q0
SCKI = 0, SCKT = 0, SLFS = 0, SFSI = 0, EFS = 0, SFST = 0, EAGC = 0
CLKOUT
FS
DOUT
I15
I0
Q15
Q14
Q0
SCKI = 0, SCKT = 0, SLFS = 1, SFSI = 0, EFS = 0, SFST = 0, EAGC = 0
CLKOUT
FS
DOUT
I15
I0
Q15
Q14
Q0
ATTN7
ATTEN6
RSSI0
SCKI = 0, SCKT = 0, SLFS = 0, SFSI = 0, EFS = 0, SFST = 0, EAGC = 1, AAGC
CLKOUT
FS
START
START
STOP
BIT
I7
I0
STOP
BIT
BIT
BIT
SCKI = 0, SCKT = 0, SLFS = X, SFSI = X, EFS = 1, SFST = 1, EAGC = 0
SCKI = 0, SCKT = 0, SLFS = X, SFSI = X, EFS = 1, SFST = 0, EAGC = 0; AS ABOVE, BUT FS IS LOW
I15
IDLE (HIGH) BITS
I8
Q15
04319-0-032
HI-Z
DOUT
Figure 33. SSI Timing for Several SSICRA Settings with 16-Bit I/Q Data
Table 14. Number of Bits per Frame for Different SSICR
Settings
DW
0 (16 Bits)
1 (24 Bits)
EAGC
0
0
1
1
1
1
0
0
1
1
1
1
EFS
0
1
0
0
1
1
0
1
0
0
1
1
AAGC
N/A
N/A
0
1
0
1
N/A
N/A
0
1
0
1
Number of Bits per Frame
32
491
48
40
691
591
48
691
64
56
891
791
The number of bits per frame with embedded frame sync (EFS = 1); assume
at least 10 idle bits are desired.
2
N/A means not applicable.
1
The maximum SSIORD setting can be determined by the
following equation:
SSIORD < TRUNC[(Decimation Factor)/(Number of
Bits per Frame)]
where TRUNC is the truncated integer value.
(1)
If SSIORD = (decimation factor)/(number of bits per frame),
the last bit in the SSI frame is not clocked out prior to FS
returning high.
Table 14 lists the number of bits within a frame for 16-bit and 24-bit
output data formats for all of the different SSICR settings. The
decimation factor is determined by the contents of Register 0x07.
An example helps illustrate how the maximum SSIORD setting
is determined. Suppose a user selects a decimation factor of 600
(Register 0x07, K = 0, M = 9) and prefers a 3-wire interface with a
dedicated frame sync (EFS = 0) containing 24-bit data (DW = 1)
with nonalternating embedded AGC data included (EAGC = 1,
AAGC = 0). Referring to Table 14, each frame consists of 64
data bits. Using Equation 1, the maximum SSIORD setting is
9 (= TRUNC(600/64)). Therefore, the user can select any
SSIORD setting between 1 and 9.
Figure 33 illustrates the output timing of the SSI port for several
SSI control register settings with 16-bit I/Q data, and Figure 34
shows the associated timing parameters. Note that the same
timing relationship holds for 24-bit I/Q data, with the exception
that I and Q word lengths now become 24 bits. In the default mode
of operation, data is shifted out on rising edges of CLKOUT
after a pulse equal to a clock period is output from the frame
sync (FS) pin. As described above, the output data consists of a
16-bit or 24-bit I sample followed by a 16-bit or 24-bit Q sample,
plus two optional bytes containing AGC and status information.
Rev. A | Page 22 of 47
Data Sheet
AD9864
tCLK
tHI tLOW
14
tV
12
13
DOUT
I15
I14
Figure 34. SSI Timing Parameters for SSI Timing
In Figure 34, the timing parameters also apply to inverted
CLKOUT or FS modes, with tDV relative to the falling edge of
the CLK and/or FS.
The AD9864 also provides the means for controlling the
switching characteristics of the digital output signals via the drive
strength (DS) field of the SSICRB. This feature is useful in
limiting switching transients and noise from the digital output
that may ultimately couple back into the analog signal path,
potentially degrading the sensitivity performance of the AD9864.
Figure 35 and Figure 36 show how the NF can vary as a function
of the SSI setting for an IF frequency of 109.65 MHz. The
following two observations can be made from these figures:
10.0
9.8
9.6
16-BIT I/O DATA
DS
0
1
2
3
4
5
6
7
8.6
8.4
1
2
4
3
5
6
SSI OUTPUT DRIVE STRENGTH SETTING
7
1
3
5
6
2
4
SSI OUTPUT DRIVE STRENGTH SETTING
Table 15. Typical Rise/Fall Times (±25%) with a 10 pF
Capacitive Load for Each DS Setting
24-BIT I/O DATA
16-BIT I/0 DATA
w/ DVGA ENABLED
9
7
Table 15 lists the typical output rise/fall times as a function of
DS for a 10 pF load. Rise/fall times for other capacitor loads can
be determined by multiplying the typical values presented by a
scaling factor equal to the desired capacitive load divided by 10 pF.
9.0
8.0
16-BIT I/O DATA
Figure 36. NF vs. SSI Output Drive Strength
(VDDx = 3.0 V, fCLK = 18 MSPS, BW = 75 kHz)
9.2
8.2
10
7
9.4
8.8
16-BIT I/O DATA
w/DVGA ENABLED
11
8
04319-0-034
2.
The NF becomes more sensitive to the SSI output drive
strength level at higher signal bandwidth settings.
The NF is dependent on the number of bits within an SSI
frame that become more sensitive to the SSI output drive
strength level as the number of bits is increased. Therefore,
select the lowest possible SSI drive strength setting that still
meets the SSI timing requirements.
NOISE FIGURE (dB)
1.
24-BIT I/O DATA
04319-0-035
tDV
04319-0-033
FS
NOISE FIGURE (dB)
CLKOUT
Figure 35. NF vs. SSI Output Drive Strength
(VDDx = 3.0 V, fCLK = 18 MSPS, BW = 10 kHz)
Rev. A | Page 23 of 47
Typ (ns)
13.5
7.2
50
3.7
3.2
2.8
2.3
2.0
AD9864
Data Sheet
SYNCHRONIZATION USING SYNCB
INTERFACING TO DSPS
Many applications require the ability to synchronize one or
more AD9864 devices in a way that causes the output data to
be precisely aligned to an external asynchronous signal. For
example, receiver applications employing diversity often require
synchronization of the digital outputs of multiple AD9864
devices. Satellite communication applications using TDMA
methods may require synchronization between payload bursts
to compensate for reference frequency drift and Doppler effects.
The AD9864 connects directly to an Analog Devices
programmable digital signal processor (DSP). Figure 38
illustrates an example with the Blackfin® series processors, such
as the ADSP-BF609. The Blackfin DSP series of 16-bit products
is optimized for low power telecommunications applications
with its dynamic power management feature, making it well
suited for portable radio products. The code compatible family
members share the fundamental core attributes of high
performance, low power consumption, and the ease-of-use
advantages of a microcontroller instruction set.
Figure 37 shows the timing relationship between SYNCB and
the CLKOUT and FS signals of the SSI port. When the clock
synthesizer is enabled to generate the input ADC clock, SYNCB
is considered an asynchronous active-low signal that must remain
low for at least half an input clock period, that is, 1/(2 × fCLK).
CLKOUT remains high while FS remains low upon SYNCB
going low. CLKOUT becomes active within one to two output
clock periods upon SYNCB returning high. If an external ADC
clock input is supplied along with a synchronous SYNCB signal,
it is recommended that SYNCB go low and returns high on the
falling edges of the CLKIN signal to ensure consistent CLKOUT
delay relative to rising edge of SYNCB. FS reappears several
output cycles later, depending on the decimation factor of the
digital filter and the SSIORD setting. Note that for any
decimation factor and SSIORD setting, this delay is fixed and
repeatable. To verify proper synchronization, monitor the FS
signals of the multiple AD9864 devices.
As shown in Figure 38, the synchronous serial interface (SSI)
of the AD9864 links the receive data stream to the serial port
(SPORT) of the DSP. For AD9864 setup and register
programming, the device connects directly to the SPI port of
the DSP. Dedicated select lines (SEL) allow the DSP to program
and read back registers of multiple devices using only one SPI
port. The DSP driver code pertaining to this interface is
available on the AD9864 product page.
SSI
CLKOUT
FS
DOUTA
SPI1_CLK
SPI1_MOSI
SPI1_MISO
SPI1_SSEL7
SPI
SPORT0_ACLK
SPORT0_AFS
SPORT0_AD0
SPORT
SYNCB
PE0
DGND
GND
DGND
GND
POWER CONTROL
To allow power consumption to be minimized, the AD9864
possesses numerous SPI programmable power-down and bias
control bits. The AD9864 powers up with all of its functional
blocks placed into a standby state, that is, STBY register default
is 0xFF. Each major block can then be powered up by writing a
0 to the appropriate bit of the STBY register. This scheme
provides the greatest flexibility for configuring the IC to a
specific application as well as for tailoring the power-down and
wake-up characteristics of the IC. Table 16 summarizes the
function of each of the STBY bits. Note that when all the blocks
are in standby, the master reference circuit is also put into
standby, and therefore the current is reduced further by 0.4 mA.
04319-0-036
FS
SPI
PC
PD
DOUTB
PE
Figure 38. Example of AD9864 and ADSP-BF609 Interface
SYNCB
CLKOUT
ADSP-BF609
AD9864
04319-0-037
SYNCB can be used for this purpose. It is an active-low signal
that clears the clock counters in both the decimation filter and
the SSI port. The counters in the clock synthesizers are not reset
because it is presumed that the CLK signals of multiple chips
would be connected. SYNCB also resets the modulator,
resulting in a large-scale impulse that must propagate through
the digital filter and SSI data formatting circuitry of the
AD9864 before recovering valid output data. As a result, data
samples unaffected by this SYNCB induced impulse can be
recovered 12 output data samples after SYNCB goes high
(independent of the decimation factor). Because SYNCB also
resets the modulator, apply SYNCB only after the tuning of the
band-pass Σ-Δ ADC has been completed during the
initialization phase. For applications that may be performing a
periodic SYNCB signal that is synchronous to FS, it is
recommended that SYNCB assertion be applied after the rising
edge of FS and three CLKOUT cycles before the arrival of the
next FS pulse to avoid a possible runt FS pulse that could
disrupt the host DSP/FPGA. Lastly, SYNCB must be tied high if
unused because it does not include an internal pull-up resistor.
Figure 37. SYNCB Timing
Rev. A | Page 24 of 47
Data Sheet
AD9864
STBY Bit
7: REF
6: LO
5: CKO
4: CK
3: GC
2: LNAMX
1: Unused
0: ADC
Effect
Voltage reference off; all
biasing shut down.
LO synthesizer off,
IOUTL three-state.
Clock oscillator off.
Clock synthesizer off,
IOUTC three-state. Clock
buffer off if ADC is off.
Gain control DAC OFF.
GCP and GCN threestate.
LNA and mixer off.
CXVM, CXVL, and CXIF
three-state.
ADC off; clock buffer off
if CLK synthesizer off;
VCM three-state; clock
to the digital filter
halted; digital outputs
static.
Current
Reduction
(mA)1
0.6
1.2
fREF
Wake-Up
Time (ms)
<0.1
(CREF = 4.7 nF)
See Note 2
1.1
1.3
See Note 2
See Note 2
0.2
Depends on
CGC
8.2
<2.2
9.2
<0.1
When all blocks are in standby, the master reference circuit is also put into
standby, and thus the current is further reduced by 0.4 mA.
2
Wake-up time is dependent on programming and/or external components.
1
REF
BUFFER
fREF
÷R
LOR
PHASEFREQUENCY
DETECTOR
fLO
÷8/9
The LO synthesizer shown in Figure 39 is a fully programmable
phase-locked loop (PLL) capable of 6.25 kHz resolution at input
frequencies up to 300 MHz and reference clocks of up to 26 MHz.
It consists of a low noise, digital, phase-frequency detector (PFD),
a variable output current charge pump (CP), a 14-bit reference
divider, programmable A and B counters, and a dual-modulus
8/9 prescaler.
The A (3-bit) and B (13-bit) counters, in conjunction with the
dual 8/9 modulus prescaler, implement an N divider with N =
8 × B + A. In addition, the 14-bit reference counter (R counter)
allows selectable input reference frequencies, fREF, at the PFD
input. A complete PLL can be implemented if the synthesizer is
used with an external loop filter and voltage controlled
oscillator (VCO).
The A, B, and R counters can be programmed via the following
registers: LOA, LOB, and LOR. The charge pump output current
is programmable via the LOI register from 0.625 mA to 5.0 mA
using the equation
IPUMP = (LOI + 1) × 0.625 mA
(2)
LO
BUFFER
fLO
FROM
VCO
Figure 39. LO Synthesizer
The LO (and CLK) synthesizer works in the following manner.
The externally supplied reference frequency, fREF, is buffered and
divided by the value held in the R counter. The internal fREF is
then compared to a divided version of the VCO frequency, fLO.
The phase/frequency detector provides up and down pulses
whose widths vary, depending upon the difference in phase and
frequency of the input signals of the detector. The up/down
pulses control the charge pump, making current available to
charge the external low-pass loop filter when there is a
discrepancy between the inputs of the PFD. The output of the
low-pass filter feeds an external VCO whose output frequency,
fLO, is driven such that its divided down version, fLO, matches
that of fREF, thus closing the feedback loop.
The synthesized frequency is related to the reference frequency
and the LO register contents as follows:
fLO = (8 × LOB + LOA)/LOR × fREF
LO SYNTHESIZER
PUMP
FAST
ACQUIRE
LOA, LOB
A. B
COUNTERS
TO EXTERNAL
LOOP
CHARGE FILTER
04319-0-038
Table 16. Standby Control Bits
(3)
Note that the minimum allowable value in the LOB register is 3,
and its value must always be greater than that loaded into LOA.
An example helps to illustrate how the values of LOA, LOB, and
LOR can be selected. Consider an application employing a
13 MHz crystal oscillator (fREF = 13 MHz), with the requirement
that fREF = 100 kHz and fLO = 143 MHz, that is, high side injection
with fIF = 140.75 MHz and fCLK = 18 MSPS. LOR is selected to be
130 so that fREF = 100 kHz. The N-divider factor is 1430, which
can be realized by selecting LOB = 178 and LOA = 6.
The stability, phase noise, spur performance, and transient
response of the AD9864 LO (and CLK) synthesizers are
determined by the external loop filter, the VCO, the N-divide
factor, and the reference frequency, fREF. A good overview of the
theory and practical implementation of PLL synthesizers
(featured as a 3-part series in Analog Dialogue) can be found on
the Analog Devices website. A free software copy of the Analog
Devices ADIsimPLL, a PLL synthesizer simulation tool, is also
available at www.analog.com. Note that the ADF4112 can be
used as a close approximation to the LO synthesizer of the
AD9864 when using this software tool.
An on-chip fast acquire function (enabled by the LOF bit)
automatically increases the output current for faster settling
during channel changes. The synthesizer can also be disabled
using the LO standby bit located in the STBY register.
Rev. A | Page 25 of 47
AD9864
Data Sheet
LOP
84kΩ
LO
BUFFER
~VDDL/2
LON
TO MIXER
LO PORT
500Ω
FREF
500Ω
NOTES
1. ESD DIODE STRUCTURES OMITTED FOR CLARITY.
2. FREF STBY SWITCHES SHOWN WITH LO SYNTHESIZER ON.
04319-0-039
1.75V
BIAS
Figure 40. Equivalent Input of LO and REF Buffers
Figure 40 shows the equivalent input structures of the LO and
REF buffers of the synthesizers (excluding the ESD structures).
The LO input is fed to the buffer of the LO synthesizer as well as
the LO port of the AD9864 mixer. Both inputs are self biasing and
thus tolerate ac-coupled inputs. The LO input can be driven with
a single-ended or differential signal. Single-ended, dc-coupled
inputs ensure sufficient signal swing above and below the
common-mode bias of the LO and REF buffers (that is, 1.75 V
and VDDL/2). Note that the fREF input is slew rate dependent
and must be driven with input signals exceeding 7.5 V/µs to
ensure proper synthesizer operation. If this condition cannot be
met, an external logic gate can be inserted prior to the fREF input
to square up the signal, thus allowing an fREF input frequency
approaching dc.
Fast Acquire Mode
The fast acquire circuit attempts to boost the output current when
the phase difference between the divided-down LO (that is, fLO)
and the divided-down reference frequency (that is, fREF) exceeds
the threshold determined by the LOFA register. The LOFA
register specifies a divisor for the fREF signal that determines the
period (T) of this divided-down clock. This period defines the
time interval used in the fast acquire algorithm to control the
charge pump current.
Assume that the nominal charge pump current is at its lowest
setting (LOI = 0), and denote this minimum current by I0.
When the output pulse from the phase comparator exceeds T,
the output current for the next pulse is 2I0. When the pulse is
wider than 2T, the output current for the next pulse is 3I0, and
so forth, up to eight times the minimum output current. If the
nominal charge pump current is more than the minimum value
(LOI > 0), the preceding rule is only applied if it results in an
increase in the instantaneous charge pump current. If the
charge pump current is set to its lowest value (LOI = 0) and the
fast acquire circuit is enabled, the instantaneous charge pump
current never falls below 2I0 when the pulse width is less than T.
Thus, the charge pump current when fast acquire is enabled is
given by
IPUMP-FA = I0 × [1 = Max(1, LOI, Pulse Width/T)]
(4)
instantaneous charge pump current is less than that available for
a LOFA value of LOR/16. Similarly, a smaller value for LOFA
decreases T, making more current available for the same phase
difference. In other words, a smaller value of LOFA enables the
synthesizer to settle faster in response to a frequency hop than a
large LOFA value. Take care to choose a value for LOFA that is
large enough (values greater than 4 are recommended) to
prevent the loop from oscillating back and forth in response to
a frequency hop.
Table 17. SPI Registers Associated with LO Synthesizer
Address (Hex)
0x00
0x08
0x09
0x0A
0x0B
0x0C
0x0D
0x0E
Bit(s)
[7:0]
[5:0]
[7:0]
[7:5]
[4:0]
[7:0]
6
5
[4:2]
[1:0]
[3:0]
[7:0]
Width
1
6
8
3
5
8
1
1
3
2
4
8
Default Value
0xFF
0x00
0x38
0x5
0x00
0xiD
0
0
0x00
0x00
0x00
0x04
Name
STBY
LOR [13:8]
LOR [7:0]
LOA
LOB [12:8]
LOB [7:0]
LOF
LOINV
LOI
LOTM
LOFA [13:8]
LOFA [7:0]
CLOCK SYNTHESIZER
The clock synthesizer is a fully programmable integer-N PLL
capable of supporting input clock and reference frequencies up
to 26 MHz. It is similar to the LO synthesizer described in
Figure 39 with the following exceptions:
•
•
It does not include an 8/9 prescaler nor an A counter.
It includes a negative-resistance core that, when used in
conjunction with an external LC tank and varactor, serves
as the VCO.
The 14-bit reference counter and 13-bit N-divider counter can
be programmed via the CKR and CKN registers. The clock
frequency, fCLK, is related to the reference frequency by the
equation
fCLK = (CKN/CKR × fREF)
(5)
The charge pump current is programmable via the CKI register
from 0.625 mA to 5.0 mA using the equation
IPUMP = (CKI + 1) × 0.625 mA
(6)
The fast acquire subcircuit of the charge pump is controlled
by the CKFA register in the same manner the LO synthesizer
is controlled by the LOFA register. An on-chip lock detect
function (enabled by the CKF bit) automatically increases the
output current for faster settling during channel changes. The
synthesizer may also be disabled using the CK standby bit
located in the STBY register.
The recommended setting for LOFA is LOR/16. Choosing a
larger value for LOFA increases T. Thus, for a given phase
difference between the LO input and the fREF input, the
Rev. A | Page 26 of 47
Data Sheet
AD9864
VDDC = 3.0V
LOOP
FILTER
RBIAS
RD
RF
values were selected for the synthesizer: RF = 390 Ω, RD = 2 kΩ,
CZ = 0.68 μF, CP = 0.1 μF, COSC = 91 pF, LOSC = 1.2 μH, and
CVAR = Toshiba 1SV228 varactor.
COSC
LOSC
0
0.1µF
CP
CVAR
–10
CZ
–20
CLKP
CLKN
CLK OSC. BIAS
2
IBIAS = 0.15mA, 0.25mA,
0.40mA, OR 0.65mA
04319-0-040
VCM = VDDC – RBIAS × IBIAS > 1.6V
fOSC > 1/{2 × (LOSC × (CVARACTOR ||COSC))1/2}
The bias, IBIAS, of the negative-resistance core has four
programmable settings. The lower equivalent Q of the LC tank
circuit may require a higher bias setting of the negative
resistance core to ensure proper oscillation. Select RBIAS so that the
common-mode voltage at CLKP and CLKN is approximately 1.6 V.
The synthesizer may be disabled via the CK standby bit to allow the
user to employ an external synthesizer and/or VCO in place of
those resident on the IC. Note that if an external CLK source or
VCO is used, the clock oscillator must be disabled via the CKO
standby bit.
The phase noise performance of the clock synthesizer is
dependent on several factors, including the CLK oscillator IBIAS
setting, charge pump setting, loop filter component values, and
internal fREF setting. Figure 42 and Figure 43 show how the
measured phase noise attributed to the clock synthesizer varies
(relative to an external fCLK) as a function of the IBIAS setting and
charge pump setting for a −31 dBm IFIN signal at 73.35 MHz
with an external LO signal at 71.1 MHz. Figure 42 shows that
the optimum phase noise is achieved with the highest IBIAS
(CKO) setting, while Figure 43 shows that the higher charge
pump values provide the optimum performance for the given
loop filter configuration. The AD9864 clock synthesizer and
oscillator were set up to provide an fCLK of 18 MHz from an
external fREF of 16.8 MHz. The following external component
–50
–60
–70
–80
CKO = 2
–90
CK0 = 0
–100
CKO = 3
CKO = 1
–110
EXT CLK
–120
–130
Figure 41. External Loop Filter, Varactor, and LC Tank Required to Realize a
Complete Clock Synthesizer
–140
–25
–20
–15
–10
–5
0
5
10
15
20
25
FREQUENCY OFFSET (kHz)
Figure 42. CLK Phase Noise vs. IBIAS Setting (CKO);
IF = 73.35 MHz, IF = 71.1 MHz, IFIN = −31 dBm, fCLK = 18 MHz, fREF = 16.8 MHz;
CLK SYN Settings: CKI = 7, CLR = 56, and CLN = 60 with fREF = 300 kHz
0
–10
–20
–30
PHASE NOISE (dBc/Hz)
The AD9864 clock synthesizer circuitry includes a negative
resistance core so that only an external LC tank circuit with a
varactor is needed to realize a voltage controlled clock oscillator
(VCO). Figure 41 shows the external components required to
complete the clock synthesizer along with the equivalent input
circuitry of the CLK input. The resonant frequency of the VCO
is approximately determined by LOSC and the series equivalent
capacitance of COSC and CVAR. As a result, LOSC, COSC, and CVAR
must be selected to provide a sufficient tuning range to ensure
both proper start-up oscillation and locking of the clock
synthesizer. Both the COSC and LOSC values must have ±5%
tolerance along with LOSC having a Q > 20 at the desired clock
frequency.
–40
04319-0-041
AD9864
20
–40
–50
–60
–70
–80
CP = 0
–90
CP = 2
–100
–110
CP = 4
CP = 6
EXT CLK
–120
–130
–140
–25
–20
–15
–10
5
0
5
10
FREQUENCY OFFSET (kHz)
15
20
25
04319-0-042
IOUTC
19
PHASE NOISE (dBc/Hz)
–30
15
Figure 43. CLK Phase Noise vs. IBIAS Setting (CKO);
IF = 73.35 MHz, IF = 71.1 MHz, IFIN = −31 dBm, fCLK = 18 MHz, fREF = 16.8 MHz;
CLK SYN Settings: CKO Bias = 3, CKR = 56, and CKN = 60 with fREF = 300 kHz
Table 18. SPI Registers Associated with CLK Synthesizer
Address (Hex)
0x00
0x01
0x10
0x11
0x12
0x13
0x14
0x15
0x16
Rev. A | Page 27 of 47
Bit(s)
[7:0]
[3:2]
[5:0]
[7:0]
[4:0]
[7:0]
6
5
[4:2]
[1:0]
[3:0]
[7:0]
Width
8
2
6
8
5
8
1
1
3
1
4
8
Default Value
0xFF
0x00
0x00
0x38
0x00
0x3C
0
0
0x00
0x00
0x00
0x04
Name
STBY
CKOB
CKR [13:8]
CKR [7:0]
CKN [12:8]
CKN [7:0]
CKF
CKINV
CKI
CKTM
CKFA [13:8]
CKFA [7:0]
AD9864
Data Sheet
2.5
IF LNA/MIXER
2.0
CAPACITANCE (pF)
0
RBIAS
L
MXON
CXVL
LO INPUT =
0.3V p-p TO
1.0V p-p
RGAIN
MULTI-TANH
V–I STAGE
RF
CXIF
CXVM
04319-0-043
IFIN
DC SERVO
LOOP
Figure 44. Simplified Schematic of AD9864 LNA/Mixer
600
RESISTANCE (Ω)
100
200
150
250
FREQUENCY (MHz)
300
350
The external differential LC tank forms the resonant element for
the first resonator of the band-pass Σ-∆ modulator, and so must
be tuned to the fCLK/8 center frequency of the modulator. The
inductors must be chosen such that their impedance at fCLK/8 is
approximately 140, that is, L = 180/fCLK. An accuracy of 20% is
considered to be adequate. For example, at fCLK = 18 MHz, L =
10 µH is a good choice. Once the inductors have been selected,
the required tank capacitance may be calculated using the
relation
fCLK / 8 = 1/[2 × π × (2L × C ) ]
550
500
450
400
0
50
100
150
200
250
FREQUENCY (MHz)
300
350
04319-0-044
350
300
50
The differential LO port of the mixer is driven by the LO buffer
stage shown in Figure 44, which can be driven single-ended or
differential. Because it is self biasing, the LO signal level can be
ac-coupled and range from 0.3 V p-p to 1.0 V p-p with negligible
effect on performance. The open-collector outputs of the mixer,
MXOP and MXON, drive an external resonant tank consisting of
a differential LC network tuned to the IF of the band-pass Σ-∆
ADC, that is, fIF2_ADC = fCLK/8. The two inductors provide a dc
bias path for the mixer core via a series resistor of 50 Ω, which
is included to dampen the common-mode response. The output
of the mixer must be ac-coupled to the input of the band-pass
Σ-∆ ADC, IF2P, and IF2N via two 100 pF capacitors to ensure
proper tuning of the LC center frequency.
C
MXOP
0
Figure 46. Shunt Capacitance vs. Frequency of AD9864 IF1 Input
50Ω
VDDI
1.0
0.5
2.7V TO 3.6V
L
1.5
04319-0-045
The AD9864 contains a single-ended LNA followed by a Gilbert
type active mixer, shown in Figure 44 with the required external
components. The LNA uses negative shunt feedback to set its
input impedance at the IFIN pin, thus making it dependent on
the input frequency. It can be modeled as approximately
370 Ω||1.4 pF (±20%) below 100 MHz. Figure 45 and Figure 46
show the equivalent input impedance vs. frequency characteristics
of the AD9864. The increase in shunt resistance vs. frequency
can be attributed to the reduction in bandwidth, thus the amount
of negative feedback of the LNA. Note that the input signal into
IFIN must be ac-coupled via a 10 nF capacitor because the LNA
input is self biasing.
Figure 45. Shunt Input Resistance vs. Frequency of AD9864 IF1 Input
For example, at fCLK = 18 MHz and L = 10 µH, a capacitance of
250 pF is needed. However, to accommodate an inductor tolerance
of ±10%, the tank capacitance must be adjustable from 227 pF
to 278 pF. Selecting an external capacitor of 180 pF ensures that
even with a 10% tolerance and stray capacitances as high as 30 pF,
the total capacitance is less than the minimum value needed by
the tank. Extra capacitance is supplied by the AD9864 on-chip
programmable capacitor array. Because the programming range
of the capacitor array is at least 160 pF, the AD9864 has plenty of
range to make up for the tolerances of low cost external
components. Note that if fCLK is increased by a factor of 1.44 MHz
to 26 MHz so that fCLK/8 becomes 3.25 MHz, reducing L and C
by approximately the same factor (L = 6.9 µH and C = 120 pF)
satisfies the requirements stated previously.
Rev. A | Page 28 of 47
Data Sheet
AD9864
Table 19. SPI Registers Associated with LNA/Mixer
Address (Hex)
0x00
0x03
Bit(s)
[7:0]
7
Width
8
1
Default Value
0xFF
0
Name
STBY
ATTEN
Figure 48 shows the measured power spectral density measured
at the output of the undecimated band-pass Σ-Δ modulator.
Note that the wide dynamic range achieved at the center
frequency, fCLK/8, is achieved once the LC and RC resonators of
the Σ-Δ modulator have been successfully tuned. The out-ofband noise is removed by the decimation filters following
quadrature mixer.
0
–2dBFS OUTPUT
–10
fCLK = 18MHz
NBW = 3.3kHz
–20
FFT OUTPUT (dBFS/NBW)
A 16 dB step attenuator is also included within the LNA/mixer
circuitry to prevent large signals (that is, > −18 dBm) from
overdriving the Σ-Δ modulator. In such instances, the Σ-Δ
modulator becomes unstable, thus severely desensitizing the
receiver. The 16 dB step attenuator can be invoked by setting
the ATTEN bit (Register 0x03, Bit 7), causing the mixer gain to
be reduced by 16 dB. The 16 dB step attenuator can be used in
applications where a potential target or blocker signal could
exceed the IF input clip point. Although the LNA is driven into
compression, it may still be possible to recover the desired signal
if it is FM. See Table 19 for the gain compression characteristics
of the LNA and mixer with the 16 dB attenuator enabled.
–30
–40
–50
–60
–70
–80
BAND-PASS Σ-∆ ADC
The ADC of the AD9864 is shown in Figure 47. The ADC
contains a sixth-order, multibit band-pass Σ-Δ modulator that
achieves very high instantaneous dynamic range over a narrow
frequency band. The loop filter of the band-pass Σ-Δ modulator
consists of two continuous-time resonators followed by a discrete
time resonator, with each resonator stage contributing a pair of
complex poles. The first resonator is an external LC tank, while
the second is an on-chip active RC filter. The output of the LC
resonator is ac-coupled to the second resonator input via 100 pF
capacitors. The center frequencies of these two continuous-time
resonators must be tuned to fCLK/8 for the ADC to function
properly. The center frequency of the discrete time resonator
automatically scales with fCLK, thus no tuning is required.
EXTERNAL
LC
IF2P
fCLK = 13 MSPS TO 26 MSPS
RC
RESONATOR
IF2N
MXOP
SC
RESONATOR
NINELEVEL
FLASH
–100
1
2
3
4
5
6
7
8
9
FREQUENCY (MHz)
Figure 48. Measured Undecimated Spectral Output of Σ-∆
Modulator ADC with fCLK = 18 MSPS and Noise Bandwidth of 3.3 kHz
The signal transfer function of the AD9864 possesses inherent
anti-alias filtering by virtue of the continuous time portions of
the loop filter in the band-pass Σ-Δ modulator. Figure 49
illustrates this property by plotting the nominal signal transfer
function of the ADC for frequencies up to 2fCLK. The notches
that naturally occur for all frequencies that alias to the fCLK/8
pass band are clearly visible. Even at the widest bandwidth
setting, the notches are deep enough to provide greater than
80 dB of alias protection. Thus, the wideband IF filtering
requirements preceding the AD9864 are determined mostly by
the image band of the mixer, which is offset from the desired IF
input frequency by fCLK/4 (that is, 2 × fCLK/8) rather than any
aliasing associated with the ADC.
0
MXON
GAIN
CONTROL
Figure 47. Equivalent Circuit of Sixth-Order Band-Pass Σ-∆ Modulator
–20
–30
NOTCH AT ALL ALIAS FREQUENCIES
–40
–50
–60
–70
–80
0
0.5
1.0
1.5
NORMALIZED FREQUENCY (RELATIVE TO fOUT)
Figure 49. Signal Transfer Function of the Band-Pass Σ-∆
Modulator from 0 fCLK to 2fCLK
Rev. A | Page 29 of 47
2.0
04319-0-049
TO DIGITAL
FILTER
04319-0-047
ESL
SIGNAL TRANSFER FUNCTION (dB)
–10
DAC1
MIXER
OUTPUT
0
04319-0-048
–90
AD9864
Data Sheet
Figure 50 shows the nominal signal transfer function
magnitude for frequencies near the fCLK/8 pass band. The width
of the pass band determines the transfer function droop, but
even at the lowest oversampling ratio (48) where the pass band
edges are at ±fCLK/192 (±0.005 fCLK), the gain variation is less
than 0.5 dB. Also consider the amount of attenuation offered by
the signal transfer function near fCLK/8 when determining the
narrow-band IF filtering requirements preceding the AD9864.
The following mechanisms prevent the tuning procedure from
finishing (that is, Register 0x1C does not clear):
•
0
–5
–10
•
•
–15
–20
–0.10
–0.05
0
0.05
NORMALIZED FREQUENCY (RELATIVE TO fCLK)
0.10
04319-0-050
SIGNAL TRANSFER FUNCTION (dB)
•
Figure 50. Magnitude of the Signal Transfer Function of the ADC near fCLK/8
Tuning of the Σ-∆ modulator’s two continuous-time resonators
is essential in realizing the full dynamic range of the ADC, and
must be performed upon system startup. To facilitate tuning of
the LC tank, a capacitor array is internally connected to the MXOP
and MXON pins. The capacitance of this array is programmable
from 0 pF to 200 pF ± 20% and can be programmed either
automatically or manually via the SPI port. The capacitors of
the active RC resonator are similarly programmable. Note that
the AD9864 can be placed in and out of its standby mode without
retuning since the tuning codes are stored in the SPI registers.
When tuning the LC tank, the sampling clock frequency must
be stable and the LNA/mixer, LO synthesizer, and ADC must all
be placed in standby. Because large LO and IF signals
(>−40 dBm) present at the inputs of the AD9864 can corrupt
the calibration, these signals must be minimized or disabled
during the calibration sequence. Tuning is triggered when the
ADC is taken out of standby if the TUNE_LC bit of Register
0x1C has been set. This bit clears when the tuning operation is
complete (less than 6 ms). The tuning codes can be read from
the 3-bit CAPL1 (0x1D) and the 6-bit CAPL0 (0x1E) registers.
In a similar manner, tuning of the RC resonator is activated if
the TUNE_RC bit of Register 0x1C is set when the ADC is
taken out of standby. This bit clears when tuning is complete.
The tuning code can be read from the CAPR (0x1F) register.
Setting both the TUNE_LC and TUNE_RC bits tunes the LC
tank and the active RC resonator in succession. During tuning,
the ADC is not operational and neither data nor a clock is
available from the SSI port.
CLK signal is not present or scaled/biased properly into
CLKP (and/or CLKN) pin such that internal clock receiver
does not square-up the input clock signal. To determine if
the CLK input signal is being received correctly, a clock
signal appears at the CLKOUT when the ADC is not in
standby mode (Register 0x00), and the CLKOUT buffer is
not three-stated (Register 0x18).
LC resonator fails to resonate during tune operation.
Check to see proper values are used for LC tank and that it
is connected to MXOP/MXON pins. Also check if 100 pF
capacitors are connected between MXOP (MXON) and
IF2P (IF2N) pins.
SYCNB pin is low.
Capacitor value (between the MXOP/MXON and
IF2P/IF2N pins) is larger than 100 pF.
Table 20 lists the recommended sequence of the SPI commands for
tuning the ADC, and Table 21 lists all of the SPI registers associated
with band-pass Σ-∆ ADC. Note that the recommended sequence
includes additional steps for robustness. These steps are additional
measures to prevent the back-end ADC from generating an
internal unstable signal that locks the state machine, thus
preventing resonator tuning. It also allows five attempts to calibrate
the resonators. As a further safeguard, the user can save the settings
for Register 0x1D, Register 0x1E, and Register 0x1F determined
during factory test and reload these settings after five attempts.
Note that that the occurrence of this tuning issue is extremely
rare among devices shipped to date, and the occurrence among
any suspect devices being quite low (<0.1%).
Table 20. Tuning Sequence
Address
(Hex)
0x3E
Value
(Hex)
0x47
0x38
0x39
0x00
0x01
0x0F
0x45
0x1C
0x03
0x00
0x44
0x1C
0x00
0x38
0x3E
0x00
0x00
0x00
0x40
1
Comments
Disable internal ADC unstable signal.
Enable RC Q enhancement.
Bypass RC and SC resonators.
Enable manual control of feedback DAC
Set DAC to midscale
LO synthesizer, LNA/mixer, and ADC are
placed in standby.1
Set TUNE_LC and TUNE_RC. Wait for CLK to
stabilize if CLK synthesizer used.
Take the ADC out of standby. Wait for 0x1C
to clear (<6 ms).
If 0x1C does not clear, clear 0x1C and make
five attempts before exiting loop.
Disable manual control of feedback DAC
Re-enable ADC unstable signal.
Disable RC Q enhancement.
Disable bypass of RC and SC resonators.
LNA/mixer (and LO SYN if used) can now be
taken out of standby
If external CLK VCO or source is used, the CLK oscillator must also be
disabled. Large IF or LO signals can corrupt the calibration; these signals
must be disabled during the calibration sequence.
Rev. A | Page 30 of 47
Data Sheet
AD9864
12
0x1D
0x1E
0x1F
0x38
0x39
0x3E
Value
[7:0]
1
0
[2:0]
[5:0]
[7:0]
0
[7:0]
0
Width
8
1
1
3
6
8
1
8
1
Default Value
0xFF
0
0
0
0x00
0x00
0
0x00
0
Name
STBY
TUNE_LC
TUNE_RC
CAPL1 [2:0]
CAPL1 [5:0]
CAPR
DACCR
DACDATA
ADCCR
BW = 75kHz
11
NF (dB)
Address (Hex)
0x00
0x1C
When the AD9864 is tuned, the noise figure degradation
attributed solely to the temperature drift of the LC and RC
resonators is minimal. Because the drift of the RC resonator is
actually negligible compared to that of the LC resonator, the
external L and C components temperature drift characteristics
tend to dominate. Figure 51 shows the degradation in noise
figure as the product of the LC value is allowed to vary from
−12.5% to +12.5%. Note that the noise figure remains relatively
constant over a ±3.5% range (±35,000 ppm), suggesting that
most applications do not need to be retuned over the operating
temperature range.
Rev. A | Page 31 of 47
10
BW = 30kHz
9
BW = 10kHz
8
–15
–10
–5
0
5
10
LC ERROR (%)
Figure 51. Typical Noise Figure Degradation from L and C
Component Drift (fCLK = 18 MSPS, fIF = 73.3501 MHz)
15
04319-0-051
Table 21. SPI Registers Associated with Band-Pass Σ-∆ ADC
AD9864
Data Sheet
DECIMATION FILTER
DEC1
DATA
FROM Σ-Δ
MODULATOR
SIN
SINC4
FILTER
12
M
K
DEC2
DEC3
I
COMPLEX
4 DATA TO
FIR
OR SSI PORT
FILTER
5 Q
SINC4 M + 1
FILTER
04319-0-052
COS
Figure 52. Decimation Filter Architecture
Figure 53 shows the response of the decimation filter at a
decimation factor of 900 (K = 0, M = 14) and a sampling clock
frequency of 18 MHz. In this example, the output data rate (fOUT)
is 20 kSPS, with a usable complex signal bandwidth of 10 kHz
centered around dc. As this figure shows, the first and second
alias bands (occurring at even integer multiples of fOUT/2) have
the least attenuation but provide at least 88 dB of attenuation.
Note that signals falling around frequency offsets that are odd
integer multiples of fOUT/2 (that is, 10 kHz, 30 kHz, and 50 kHz)
fall back into the transition band of the digital filter.
0
±135.466kHz PASS BAND
–40
–60
–98dB
–80
–120
–94dB
–115dB
–100
0
0.5
1.0
1.5
FREQUENCY
EQUENCY (MHz)
2.0
2.5
Figure 54. Decimation Filter Frequency Response for
fOUT = 541.666 kSPS (fCLK = 26 MHz, OSR = 48)
Figure 55 and Figure 56 show expanded views of the pass band
for the two possible configurations of the third decimation
filter. When decimating by 60n (K = 0), the pass-band gain
variation is 1.2 dB; when decimating by 48n (K = 1), the passband gain variation is 0.9 dB. Normalization of full scale at
band center is accurate to within 0.14 dB across all decimation
modes. Figure 57 and Figure 58 show the folded frequency
response of the decimator for K = 0 and K = 1, respectively.
±5.0kHz PASS BAND
FOLDING
POINT
2
–88dB
–60
–88dB
–101dB
–103dB
–80
0
10
20
30
50
60
40
FREQUENCY (kHz)
70
80
90
100
0
–1
–2
04319-0-053
–100
–3
Figure 53. Decimation Filter Frequency Response for
fOUT = 20 kSPS (fCLK = 18 MHz, OSR = 900)
Figure 54 shows the response of the decimation filter with a
decimation factor of 48 and a sampling clock rate of 26 MHz.
PASS-BAND GAIN FREQUENCY = 1.2dB
1
0
0.125
NORMALIZED FREQUENCY (RELATIVE TO fOUT)
0.250
04319-0-055
–40
FILTER RESPONSE (dB)
FILTER RESPONSE (dB)
–20
3
–20
–120
120
0
04319-0-054
The output data rate (fOUT) is equal to the modulator clock
frequency (fCLK) divided by the decimation factor of the digital
filter. Due to the transition region associated with the frequency
response of the decimation filter, the decimation factor must be
selected so that fOUT is equal to or greater than twice the signal
bandwidth, which ensures low amplitude ripple in the pass
band along with the ability to provide further applicationspecific digital filtering prior to demodulation.
The alias attenuation is at least 94 dB and occurs for frequencies
at the edges of the fourth alias band. The difference between
the alias attenuation characteristics of Figure 53 and those of
Figure 54 is due to the fact that the third decimation stage
decimates by a factor of 5 for Figure 53 compared with a factor
of 4 for Figure 54.
FILTER RESPONSE (dB)
The decimation filter shown in Figure 52 consists of an fCLK/8
complex mixer and a cascade of three linear phase FIR filters:
DEC1, DEC2, and DEC3. DEC1 downsamples by a factor of 12
using a fourth-order comb filter. DEC2 also uses a fourth-order
comb filter, but its decimation factor is set by the M field of
Register 0x07. DEC3 is either a decimate-by-5 FIR filter or a
decimate-by-4 FIR filter, depending on the value of the K bit
within Register 0x07. Thus, the composite decimation factor
can be set to either 60 × M or 48 × M for K equal to 0 or 1,
respectively.
Figure 55. Pass-Band Frequency Response of the Decimator for K = 0
Rev. A | Page 32 of 47
Data Sheet
AD9864
3
VARIABLE GAIN AMPLIFIER OPERATION WITH
AUTOMATIC GAIN CONTROL
FILTER RESPONSE (dB)
2
PASS-BAND GAIN VARIATION = 0.9dB
1
0
–1
–3
0
0.125
NORMALIZED FREQUENCY (RELATIVE TO fOUT)
0.250
04319-0-056
–2
Figure 56. Pass-Band Frequency Response of the Decimator for K = 1
0
The purpose of the VGA is to extend the usable dynamic range
of the AD9864 by allowing the ADC to digitize a desired signal
over a large input power range as well as recover a low level
signal in the presence of larger unfiltered interferers without
saturating or clipping the ADC. The DVGA is most useful in
extending the dynamic range in narrow-band applications
requiring a 16-bit I and Q data format. In these applications,
quantization noise resulting from internal truncation to 16 bits
as well as external 16-bit fixed point post processing can degrade
the effective noise figure of the AD9864 by 1 dB or more.
FILTER RESPONSE (dB)
–20
–40
–60
MIN ALIAS ATTN = 87.7dB
–80
0
0.25
NORMALIZED FREQUENCY (RELATIVE TO fOUT)
0.50
04319-0-057
–100
–120
Figure 57. Folded Decimator Frequency Response for K = 0
0
–20
–40
–60
–80
The DVGA is enabled by writing a 1 to the AGCV field. The
VGA (and the DVGA) can operate in either a user controlled
variable gain mode or automatic gain control (AGC) mode. It is
worth noting that the VGA imparts negligible phase error upon
the desired signal as its gain is varied over a 12 dB range. This is
due to the bandwidth of the VGA being far greater than the
down converted desired signal (centered about fCLK/8) and
remaining relatively independent of gain setting. As a result,
phase modulated signals experience minimal phase error as the
AGC varies the VGA gain while tracking an interferer or the
desired signal under fading conditions. Note that the envelope
of the signal is still affected by the AGC settings.
MIN ALIAS ATTN = 97.2dB
–100
–120
0
0.25
NORMALIZED FREQUENCY (RELATIVE TO fOUT)
0.50
04319-0-058
FILTER RESPONSE (dB)
The AD9864 contains both a variable gain amplifier (VGA) and
a digital VGA (DVGA) along with all of the necessary signal
estimation and control circuitry required to implement automatic
gain control (AGC), as shown in Figure 59. The AGC control
circuitry provides a high degree of programmability, allowing
users to optimize the AGC response as well as the dynamic
range of the AD9864 for a given application. The VGA is
programmable over a 12 dB range and implemented within the
ADC by adjusting its full-scale reference level. Increasing the
full scale of the ADC is equivalent to attenuating the signal. An
additional 12 dB of digital gain range is achieved by scaling the
output of the decimation filter in the DVGA. Note that a slight
increase in the supply current (0.67 mA) is drawn from VDDI
and VDDF as the VGA changes from 0 dB to 12 dB attenuation.
Figure 58. Folded Decimator Frequency Response for K = 1
Rev. A | Page 33 of 47
AD9864
Data Sheet
DEC2
AND
DEC3
C1
DEC1
÷12
DVGA
I/Q DATA
TO SSI
AGCR
REF LEVEL
I + Q
SELECT
LARGER
+
K
I + Q
AGCA/AGCD
SCALING
VGA
DAC
1
(1 – Z–1)
AGCV
CV
SETTING
RSSI DATA
GCP
CDAC
TO SSI
04319-0-059
Σ-Δ ADC
FS
Figure 59. Functional Block Diagram of VGA and AGC
Variable Gain Control
The variable gain control is enabled by setting the AGCR field
of Register 0x06 to 0. In this mode, the gain of the VGA (and
the DVGA) can be adjusted by writing to the 16-bit AGCG
register. The maximum update rate of the AGCG register via
the SPI port is fCLK/240. The MSB of this register is the bit that
enables 16 dB of attenuation in the mixer. This feature allows
the AD9864 to cope with large level signals beyond the VGA
range (that is, > −18 dBm at LNA input) to prevent overloading
of the ADC.
The lower 15 bits specify the attenuation in the remainder of the
signal path. If the DVGA is enabled, the attenuation range is
from −12 dB to +12 dB because the DVGA provides 12 dB of
digital gain. In this case, all 15 bits are significant. However,
with the DVGA disabled, the attenuation range extends from
0 dB to 12 dB and only the lower 14 bits are useful. Figure 60
shows the relationship between the amount of attenuation and
the AGC register setting for both cases.
12
VGA
RANGE
6
DVGA AND
VGA ENABLED
0
DVGA
RANGE
–6
–12
0000
1FFF
3FFF
5FFF
7FFF
AGCG SETTING (HEX)
Figure 60. AGC Gain Range Characteristics vs. AGCG Register
Setting With and Without DVGA Enabled
04319-0-060
AGC ATTENUATION (dB)
ONLY
VGA ENABLED
Referring to Figure 59, the gain of the VGA is set by an 8-bit
control DAC that provides a control signal to the VGA
appearing at the gain control pin (GCP). For applications
implementing automatic gain control, the output resistance of
the DAC can be reduced by a factor of 9 to decrease the attack
time of the AGC response for faster signal acquisition. An
external capacitor, CDAC, from GCP to analog ground is
required to smooth the output of the DAC each time it updates
as well as to filter wideband noise. Note that CDAC, in
combination with the programmable output resistance of the
DAC, sets the −3 dB bandwidth and time constant associated
with this RC network.
A linear estimate of the received signal strength is performed at
the output of the first decimation stage (DEC1) and output of
the DVGA (if enabled), as discussed in the AGC section. This
data is available as a 6-bit RSSI field within an SSI frame with 60
corresponding to a full-scale signal for a given AGC attenuation
setting. The RSSI field is updated at fCLK/60 and can be used
with the 8-bit attenuation field (or AGCG attenuation setting)
to determine the absolute signal strength. Note that the RSSI data
must be post filtered to remove the ac ripple component that is
dependent on the frequency offset relative to the IF frequency.
The accuracy of the mean RSSI reading (relative to the IF input
power) depends on the input signal’s frequency offset relative to
the IF frequency because both the response of the DEC1 filter
as well as the signal transfer function of the ADC attenuate the
downconverted signal level of the mixer, centered at fCLK/8. As a
result, the estimated signal strength of input signals falling
within proximity to the IF is reported accurately, while those
signals at increasingly higher frequency offsets incur larger
measurement errors. Figure 61 shows the normalized error of
the RSSI reading as a function of the frequency offset from the
IF frequency. Note that the significance of this error becomes
apparent when determining the maximum input interferer (or
blocker) levels with the AGC enabled.
Rev. A | Page 34 of 47
Data Sheet
AD9864
0
the DVGA allows the AGC to minimize the effects of the 16-bit
truncation noise.
When the estimated signal level falls within the range of the
AGC, the AGC loop adjusts the VGA (or DVGA) attenuation
setting so that the estimated signal level is equal to the
programmed level specified in the AGCR field. The absolute
signal strength can be determined from the contents of the
ATTN and RSSI field that is available in the SSI data frame
when properly configured. Within this AGC tracking range, the
6-bit value in the RSSI field remains constant while the 8-bit
ATTN field varies according to the VGA/DVGA setting. Note
that the ATTN value is based on the 8 MSB contained in the
AGCG field of Register 0x03 and Register 0x04.
–6
–9
–12
–15
–18
0
0.01
0.02
0.03
0.04
NORMALIZED FREQUENCY OFFSET ((fIN–fIF) fCLK)
0.05
04319-0-061
MEASURED RSSI ERROR (dB)
–3
Figure 61. Normalized RSSI Error vs. Normalized IF Frequency Offset
Automatic Gain Control (AGC)
The gain of the VGA (and DVGA) is automatically adjusted
when the AGC is enabled via the AGCR field of Register 0x06.
In this mode, the gain of the VGA is continuously updated at
fCLK/60 in an attempt to ensure that the maximum analog signal
level into the ADC does not exceed the ADC clip level and that
the rms output level of the ADC is equal to a programmable
reference level. With the DVGA enabled, the AGC control loop
also attempts to minimize the effects of 16-bit truncation noise
prior to the SSI output by continuously adjusting the gain of the
DVGA to ensure maximum digital gain while not exceeding the
programmable reference level.
This programmable level can be set at 3 dB, 6 dB, 9 dB, 12 dB,
and 15 dB below the ADC saturation (clip) level by writing
values from 1 to 5 to the 3-bit AGCR field. Note that the ADC
clip level is defined to be 2 dB below its full scale (−18 dBm at
the LNA input for a matched input and maximum attenuation).
If AGCR is 0, automatic gain control is disabled. Because
clipping of the ADC input degrades the SNR performance, the
reference level must also take into consideration the peak-torms characteristics of the target (or interferer) signals.
Referring again to Figure 59, the majority of the AGC loop
operates in the discrete time domain. The sample rate of the
loop is fCLK/60; therefore, registers associated with the AGC
algorithm are updated at this rate. The number of overload and
ADC reset occurrences within the final I/Q update rate of the
AD9864, as well as the AGC value (8 MSB), can be read from
the SSI data upon proper configuration.
The AGC performs digital signal estimation at the output of the
first decimation stage (DEC1) as well as the DVGA output that
follows the last decimation stage (DEC3). The rms power of the
I and Q signal is estimated by the equation
Xest[n] = Abs(I[n] + Abs(Q[n])
(7)
Signal estimation after the first decimation stage allows the
AGC to cope with out-of-band interferers and in-band signals
that could otherwise overload the ADC. Signal estimation after
A description of the AGC control algorithm and the user
adjustable parameters follows. First, consider the case where the
in-band target signal is bigger than all out-of-band interferers
and the DVGA is disabled. With the DVGA disabled, a control
loop based only on the target signal power measured after
DEC1 is used to control the VGA gain, and the target signal is
tracked to the programmed reference level. If the signal is too
large, the attenuation is increased with a proportionality
constant determined by the AGCA setting. Large AGCA values
result in large gain changes, thus rapid tracking of changes in
signal strength. If the target signal is too small relative to the
reference level, the attenuation is reduced; however, now the
proportionality constant is determined by both the AGCA and
AGCD settings. The AGCD value is effectively subtracted from
AGCA, so a large AGCD results in smaller gain changes and
thus slower tracking of fading signals.
The 4-bit code in the AGCA field sets the raw bandwidth of the
AGC loop. With AGCA = 0, the AGC loop bandwidth is at its
minimum of 50 Hz, assuming fCLK = 18 MHz. Each increment of
AGCA increases the loop bandwidth by a factor of √2; thus the
maximum bandwidth is 9 kHz. A general expression for the
attack bandwidth is
BWA = 50 × (fCLK/18 MHz) × 2(AGCA/2) Hz
(8)
Assuming that the loop dynamics are essentially those of a
single-pole system, the corresponding attack time is
tATTACK = 2.2/(100 × π × 2(AGCA/2)) = 35/BWA
(9)
The 4-bit code in the AGCD field sets the ratio of the attack
time to the decay time in the amplitude estimation circuitry.
When AGCD is zero, this ratio is one. Incrementing AGCD
multiplies the decay time constant by 21/2, allowing a 180:1
range in the decay time relative to the attack time. The decay
time may be computed from
tDECAY = tATTACK × 2(AGCD/2)
(10)
Figure 62 shows the AGC response to a 30 Hz pulse-modulated
IF burst for different AGCA and AGCD settings. The 3-bit
value in the AGCO field determines the amount of attenuation
added in response to a reset event in the ADC. Each increment
in AGCO doubles the weighting factor. At the highest AGCO
setting, the attenuation changes from 0 dB to 12 dB in
Rev. A | Page 35 of 47
AD9864
Data Sheet
approximately 10 µs, while at the lowest setting the attenuation
changes from 0 dB to 12 dB in approximately 1.2 ms. In both
cases, assume that fCLK = 18 MHz. Figure 63 shows the AGC
attack time response for different AGCO settings.
AGCA = 0
AGCD = 8
64
AGCD = 0
Now consider the case described previously but with the DVGA
enabled to minimize the effects of 16-bit truncation. With the
DVGA enabled, a control loop based on the larger of the two
estimated signal levels (that is, the output of DEC1 and DVGA)
controls the DVGA gain. The DVGA multiplies the output of
the decimation filter by a factor of 1 to 4 (that is, 0 dB to 12 dB).
When signals are small, the DVGA gain is 4 and the 16-bit output
is extracted from the 24-bit data produced by the decimation
filter by dropping 2 MSB and taking the next 16 bits. As signals
become larger, the DVGA gain decreases to the point where the
DVGA gain is 1 and the 16-bit output data is simply the 16 MSB
of the internal 24-bit data. As signals become even larger,
attenuation is accomplished by the normal method of
increasing the full scale of the ADC.
16
VGA ATTENUATION SETTING
0
AGCA = 4
96
80
AGCD = 8
64
48
AGCD = 0
32
16
0
AGCA = 8
96
80
AGCD = 8
64
48
32
0
10
30
20
40
50
TIME (ms)
04319-0-062
AGCD = 0
16
0
Figure 62. AGC Response for Different AGCA and AGCD Settings with
fCLK = 18 MSPS, fCLKOUT = 20 kSPS, Decimate by 900, and AGCO = 0
128
VGA ATTENUATION SETTING
112
96
AGCO = 7
80
64
AGCO = 4
48
32
AGCD = 0
0
0.1
0.2
0.3
0.4
0.5
0.6
TIME (ms)
0.7
0.8
0.9
1.0
The extra 12 dB of gain range provided by the DVGA reduces
the input-referred truncation noise by 12 dB and makes the
data more tolerant of LSB corruption within the DSP. The price
paid for this extension to the gain range is that the start of AGC
action is 12 dB lower and that the AGC loop becomes unstable
if its bandwidth is set too wide. The latter difficulty results from
the large delay of the decimation filters, DEC2 and DEC3, when
the user implements a large decimation factor. As a result, given
the option, the use of 24-bit data is preferable to using the
DVGA. Figure 64 indicates which AGCA values are reasonable
for various decimation factors (DEC FAC). The white cells
indicate that the (decimation factor/AGCA) combination works
well; the light gray cells indicate ringing and an increase in the
AGC settling time; and the dark gray cells indicate that the
combination results in instability or near instability in the AGC
loop. Setting AGCF = 1 improves the time-domain behavior at
the expense of increased spectral spreading.
04319-0-063
16
0
where:
R is the resistance between the GCP pin and ground
(72.5 kΩ ±30% if AGCF = 0, < 8 kΩ if AGCF = 1).
BW is the raw loop bandwidth.
Note that with C chosen at this upper limit, the loop bandwidth
increases by approximately 30%.
48
32
(11)
Figure 63. AGC Response for Different AGCO Settings with fCLK = 18 MSPS,
fCLKOUT = 300 kSPS, Decimate by 60 and AGCA = AGCD = 0
Lastly, the AGCF bit reduces the DAC source resistance by at
least a factor of 10, which facilitates fast acquisition by lowering
the RC time constant that is formed with the external capacitors
connected from the GCP pin to ground (GCN pin). For an
overshoot-free step response in the AGC loop, choose the
capacitor connected from the GCP pin to the GCN ground
pin so that the RC time constant is less than one quarter of the
raw loop.
AGCA
M
Rev. A | Page 36 of 47
60
0
120
1
300
4
540
8
900
E
4
5
6
7
8
9
10
11
12
Figure 64. AGCA Limits when DVGA is Enabled
13
14
15
04319-0-164
80
RC < 1/(8πBW)
DECIMATION FACTOR
96
Specifically,
Data Sheet
AD9864
Finally, consider the case of a strong out-of-band interferer (that
is, −18 dBm to −32 dBm for matched IF input) that is larger
than the target signal and large enough to be tracked by the
control loop based on the output of the DEC1. The ability of the
control loop to track this interferer and set the VGA attenuation
to prevent clipping of the ADC is limited by the accuracy of the
digital signal estimation occurring at the output of DEC1. The
accuracy of the digital signal estimation is a function of the
frequency offset of the out-of-band interferer relative to the IF
frequency as shown in Figure 61. Interferers at increasingly
higher frequency offsets incur larger measurement errors,
potentially causing the control loop to inadvertently reduce the
amount of VGA attenuation that may result in clipping of the
ADC. Figure 65 shows the maximum measured interferer signal
level versus the normalized IF offset frequency (relative to fCLK)
tolerated by the AD9864 relative to its maximum target input
signal level (0 dBFS = −18 dBm). Note that the increase in
allowable interferer level occurring beyond 0.04 × fCLK results
from the inherent signal attenuation provided by the signal
transfer function of the ADC.
–3
The system noise figure of the AD9864 is a function of the ACG
attenuation and output signal bandwidth. Figure 66 plots the
nominal system NF as a function of the AGC attenuation for
both narrow-band (20 kHz) and wideband (150 kHz) modes
with fCLK = 18 MHz. Also shown on the plot is the SNR that is
observed at the output for a −2 dBFS input. The high dynamic
range of the ADC within the AD9864 ensures that the system
NF increases gradually as the AGC attenuation is increased. In
narrow-band (BW = 20 kHz) mode, the system noise figure
increases by less than 3 dB over a 12 dB AGC range, while in
wideband (BW = 150 kHz) mode, the degradation is
approximately 5 dB. Therefore, the highest instantaneous
dynamic range for the AD9864 occurs with 12 dB of AGC
attenuation, since the AD9864 can accommodate an additional
12 dB peak signal level with only a moderate increase in its
noise floor.
As Figure 66 shows, the AD9864 can achieve an SNR in excess
of 100 dB in narrow-band applications. To realize the full
performance of the AD9864 in such applications, it is
recommended that the I/Q data be represented with 24 bits. If
16-bit data is used, the effective system NF increases because of
the quantization noise present in the 16-bit data after truncation.
15
–12
0.01
0.02
0.03
0.04
NORMALIZED FREQUENCY OFFSET = (fIN–fIF)/fCLK
0.05
0x06
Width
1
7
8
4
4
1
3
1
3
Default Value
0
0x00
0x00
0
0x00
0
0x00
0
0x00
8
Name
ATTEN
AGCG [14:8]
AGCG [7:0]
AGCA
AGCD
AGCV
AGCO
AGCF
AGCR
BW = 150kHz
11
SNR = 103.2dB
SNR = 82.9dBFS
BW = 10kHz
9
Table 22. SPI Registers Associated with AGC
Bit(s)
7
[6:0]
[7:0]
[7:4]
[3:0]
7
[6:4]
3
[2:0]
BW = 50kHz
12
10
Figure 65. Maximum Interferer (or Blocker) Input Level vs. Normalized IF
Frequency Offset
Address (Hex)
0x03
13
SNR = 95.1dBFS
0
3
6
VGA ATTENUATION (dB)
9
12
04319-0-065
–9
NOISE FIGURE (dB)
14
–15
0
0x04
0x05
SNR = 90.1dBFS
–6
04319-0-064
RELATIVE TO CLIP POINT (dBFS)
0
System Noise Figure (NF) vs. VGA (or AGC) Control
Figure 66. Nominal System Noise Figure and Peak SNR vs. AGCG Setting
(fIF = 73.35 MHz, fCLK = 18 MSPS, and 24-bit I/Q Data)
Figure 67 plots the nominal system NF with 16-bit output data
as a function of AGC in both narrow-band and wideband
mode. In wideband mode, the NF curve is virtually unchanged
relative to the 24-bit output data because the output SNR before
truncation is always less than the 96 dB SNR that 16-bit data
can support. However, in narrow-band mode, where the output
SNR approaches or exceeds the SNR that can be supported with
16-bit data, the degradation in system NF is more severe.
Furthermore, if the signal processing within the DSP adds noise
at the level of an LSB, the system noise figure can be degraded
even more than Figure 67 shows. For example, this might occur
in a fixed 16-bit DSP whose code is not optimized to process
the AD9864 16-bit data with minimal quantization effects. To
limit the quantization effects within the AD9864, the 24-bit data
Rev. A | Page 37 of 47
AD9864
Data Sheet
undergoes noise shaping just prior to 16-bit truncation, thus
reducing the in-band quantization noise by 5 dB (with 2×
oversampling). Therefore, 98.8 dBFS SNR performance is still
achievable with 16-bit data in a 10 kHz BW.
17
SNR = 98.8dBFS
m × fLO = (n ± 1/8) × fCLK
BW = 10kHz
13
BW = 150kHz
12
A second source of spurs is a large block of digital circuitry that
is clocked at fCLK/3. Problematic LO frequencies associated with
this spur source are given by
SNR = 89.9dBFS
11
SNR = 94.1dBFS
10
SNR = 83dBFS
0
3
6
VGA ATTENUATION (dB)
9
12
(13)
where n = 1, 2, 3...
04319-0-066
8
fLO = fCLK/3 + n × fCLK ± fCLK/8
BW = 50kHz
9
(12)
where:
m = 1, 3, 5...
n = 1, 2, 3...
14
Figure 67. Nominal System Noise Figure and Peak SNR vs. AGCG Setting
(fIF = 73.35 MHz, fCLK = 18 MSPS, and 16-bit I/Q Data)
APPLICATIONS CONSIDERATIONS
Frequency Planning
The LO frequency (and/or ADC clock frequency) must be
chosen carefully to prevent known internally generated spurs
from mixing down along with the desired signal and thus
degrading the SNR performance. The major sources of spurs in
the AD9864 are the ADC clock and digital circuitry operating
at 1/3 of fCLK. Therefore, the clock frequency (fCLK) is the most
important variable in determining which LO (and therefore IF)
frequencies are viable.
Many applications have frequency plans that take advantage of
industry-standard IF frequencies due to the large selection of
low cost crystal or SAW filters. If the selected IF frequency and
ADC clock rate result in a problematic spurious component,
select an alternative ADC clock rate by slightly modifying the
decimation factor and CLK synthesizer settings (if used) so that
the output sample rate remains the same. Also, applications
requiring a certain degree of tuning range must take into
consideration the location and magnitude of these spurs when
determining the tuning range as well as optimum IF and ADC
clock frequency.
Figure 70 shows that omitting the LO frequencies given by
Equation 12 for m = 1, 3, and 5, and by Equation 13 accounts
for most of the spurs. Some of the remaining low level spurs can
be attributed to coupling from the SSI digital output. Therefore,
users are also advised to optimize the output bit rate (fCLKOUT via
the SSIORD register) and the digital output driver strength to
achieve the lowest spurious and noise figure performance for a
particular LO frequency and fCLK setting. This is especially the
case for particularly narrow-band channels where low level
spurs can degrade the sensitivity performance of the AD9864.
Despite the many spurs, sweet spots in the LO frequency are
generally wide enough to accommodate the maximum signal
bandwidth of the AD9864. As evidence of this property,
Figure 68 shows that the in-band noise is quite constant for LO
frequencies ranging from 70 MHz to 71 MHz.
–50
IN-BAND POWER (dBFS)
NOISE FIGURE (dB)
15
Figure 69 plots the measured in-band noise power as a function
of the LO frequency for fCLK = 18 MHz and an output signal
bandwidth of 150 kHz when no signal is present. Any LO
frequency resulting in large spurs must be avoided. As this
figure shows, large spurs result when the LO is fCLK/8 = 2.25 MHz
away from a harmonic of 18 MHz , that is, n fCLK ± fCLK/8. Also
Rev. A | Page 38 of 47
–60
–70
–80
–90
70.0
70.5
LO FREQUENCY (MHz)
Figure 68. Expanded View from 70 MHz to 71 MHz
71.0
04319-0-069
16
problematic are LO frequencies whose odd order harmonics (that
is, m × fLO) mix with harmonics of fCLK to fCLK/8. This spur
mechanism is a result of the mixer being internally driven by a
squared-up version of the LO input consisting of the LO frequency
and its odd order harmonics. These spur frequencies can be
calculated from the relation
Data Sheet
AD9864
–60
–70
–80
–90
50
0
100
150
200
300
250
LO FREQUENCY (MHz)
04319-0-067
IN-BAND POWER (dBFS)
–50
Figure 69. Total In-Band Noise + Spur Power with No Signal Applied as a Function of the LO Frequency (fCLK = 18 MHz and Output Signal Bandwidth = 150 kHz)
–60
–70
–80
–90
0
50
100
150
200
300
250
LO FREQUENCY (MHz)
04319-0-068
IN-BAND POWER (dBFS)
–50
Figure 70. Same as Figure 69 Excluding LO Frequencies Known to Produce Large In-Band Spurs
The spectral purity of the LO (including its phase noise) is an
important consideration because LO spurs can mix with undesired
signals present at the AD9864 IFIN input to produce an in-band
response. To demonstrate the low LO spur level introduced
within the AD9864, Figure 71 plots the demodulated output
power as a function of the input IF frequency for an LO
frequency of 71.1 MHz and a clock frequency of 18 MHz.
The two large −10 dBFS spikes near the center of the plot are the
desired responses at fLO, ± fIF2_ADC, where fIF2_ADC = fCLK/8, that is,
at 68.85 MHz and 73.35 MHz. LO spurs at fLO ± fSPUR result in
spurious responses at offsets of ± fSPUR around the desired
responses. Close-in spurs of this kind are not visible on the plot;
however, small spurious responses at fLO ± fIF2_ADC ± fCLK (at
50.85 MHz, 55.35 MHz, 86.85 MHz, and 91.35 MHz) are visible
at the −90 dBFS level. This data indicates that the AD9864 does
an excellent job of preserving the purity of the LO signal.
Figure 71 can also be used to gauge how well the AD9864
rejects undesired signals. For example, the half-IF response
(at 69.975 MHz and 72.225 MHz) is approximately −100 dBFS,
giving a selectivity of 90 dB for this spurious response. The
largest spurious response at approximately −70 dBFS occurs
with input frequencies of 70.35 MHz and 71.85 MHz. These
spurs result from third-order nonlinearity in the signal path
(that is, abs [3 × fLO − 3 × fIF_INPUT] = fCLK/8).
0
D = fCLK/4 = 4.5MHz
–20
–40
dBFS
Spurious Responses
DESIRED
RESPONSES
–60
–80
–120
50
60
70
80
IF FREQUENCY (MHz)
90
100
04319-0-070
–100
Figure 71. Response of AD9864 to a −20 dBm IF Input when fLO = 71.1 MHz
Rev. A | Page 39 of 47
AD9864
Data Sheet
EXTERNAL PASSIVE COMPONENT REQUIREMENTS
2.2nF
10nF
10nF
1nF
100nF
4 IF2N
5 IF2P
GNDH 32
AD9864
FS 31
DOUTB 30
DOUTA 29
9 VDDA
CLKOUT 28
VDDH 27
10 GNDA
PE 25
PC
GNDD
GNDS
CLKN
CLKP
GNDC
VDDC
GNDQ
100pF
IOUTC
VDDD 26
VDDQ
11 VREFP
12 VREFN
RREF
10nF
GNDL 36
FREF 35
GNDS 34
SYNCB 33
6 VDDF
7 GCP
8 GCN
100pF
GNDP
IOUTL
VDDL
VDDP
CXVM
LOP
LON
GNDI
CXVL
IFIN
CXIF
3 GNDF
13 14 15 16 17 18 19 20 21 22 23 24
100kΩ
10nF
04319-0-071
100pF
1 MXOP
2 MXON
PD
100
pF
Although power supply decoupling capacitors are not shown, it
is recommended that a 0.1 µF surface-mount capacitor be
placed as close as possible to each power supply pin for
maximum effectiveness. Also not shown is the input impedance
matching network used to match the IF input of the AD9864 to
the external IF filter. Lastly, the loop filter components
associated with the LO and CLK synthesizers are not shown.
48 47 46 45 44 43 42 41 40 39 38 37
VDDI
180pF
10mH
10mH
LC TANK
50Ω
100nF
10nF
Figure 72 shows an example circuit using the AD9864 and
Table 23 shows the nominal dc bias voltages seen at the different
pins. The purpose is to show the various external passive
components required by the AD9864, along with nominal dc
voltages for troubleshooting purposes.
10nF
Figure 72. Example Circuit Showing Recommended Component Values
Mnemonic
MXOP
MXON
IF2N
IF2P
VREFP
VREFN
RREF
CLKP
CLKN
FREF
CXVM
LON
LOP
CXVL
CXIF
IFIN
LC component values for fCLK = 18 MHz are given in Figure 72.
For other clock frequencies, the two inductors and the capacitor
of the LC tank must be scaled in inverse proportion to the
clock. For example, if fCLK = 26 MHz, the two inductors must be
6.9 µH and the capacitor must be approximately 120 pF. A
tolerance of 10% is sufficient for these components because
tuning of the LC tank is performed upon system startup.
APPLICATIONS
Superheterodyne Receiver Example
Table 23. Nominal DC Bias Voltages
Pin Number
1
2
4
5
11
12
13
19
20
35
41
42
43
44
46
47
The LO, CLK, and IFIN signals are coupled to their respective
inputs using 10 nF capacitors. The output of the mixer is
coupled to the input of the ADC using 100 pF. An external
100 kΩ resistor from the RREF pin to GND sets up the internal
bias currents of the AD9864. VREFP and VREFN provide a
differential reference voltage to the Σ-∆ ADC of the AD9864,
and must be decoupled by a 0.01 µF differential capacitor along
with two 100 pF capacitors to GND. The remaining capacitors
are used to decouple other sensitive internal nodes to GND.
Note that SYNCB is tied to VDDH because it is unused.
Nominal DC Bias (V)
VDDI − 0.2
VDDI − 0.2
1.3 − 1.7
1.3 − 1.7
VDDA/2 + 0.250
VDDA/2 − 0.250
1.2
VDDC − 1.3
VDDC − 1.3
VDDC/2
1.6 − 2.0
1.65 − 1.9
1.65 − 1.9
VDDI − 0.05
1.6 − 2.0
0.9 − 1.1
The AD9864 is well suited for analog and/or digital narrowband radio systems based on a superheterodyne receiver
architecture. The superheterodyne architecture is noted for
achieving exceptional dynamic range and selectivity by using
two or more downconversion stages to provide amplification of
the target signal while filtering the undesired signals. The
AD9864 greatly simplifies the design of these radio systems by
integrating the complete IF strip (excluding the LO VCO) while
providing an I/Q digital output (along with other system
parameters) for the demodulation of both analog and digital
modulated signals. The exceptional dynamic range of the
AD9864 often simplifies the IF filtering requirements and
eliminates the need for an external AGC.
Rev. A | Page 40 of 47
Data Sheet
AD9864
VDDA
TUNER
AD9864
DAC AGC
–16dB
DOUTA
IFIN
LNA
GCN
GCP
II-2P
IF CRYSTAL OR
SAW FILTER
II-2N
PRESELECT
FILTER
VXOP
RF
INPUT
VXON
IF2 = fCLK/8
Σ-∆ ADC
LNA
DECIMATION
FILTER
FORMATTING/SSI
DOUTB
FS
TO
DSP
CLKOUT
CONTROL LOGIC
VCO
SAMPLE CLOCK
SYNTHESIZER
LO
SYNTH.
LOOP
FILTER
VCO
SYNCB
PE
PD
PC
RREF
VREFN
VREFP
CLKN
CLKP
IOUTC
LON
REFIN
SPI
LOP
IOUTC
ADF42xx
PLL SYN
VOLTAGE
REFERENCE
LOOP
FILTER
FROM DSP
04319-0-072
VDDC
CRYSTAL
OSCILLATOR
Figure 73. Typical Dual Conversion Superheterodyne Application Using the AD9864
Figure 73 shows a typical dual conversion superheterodyne
receiver using the AD9864. An RF tuner is used to select and
downconvert the target signal to a suitable first IF for the
AD9864. A preselect filter may precede the tuner to limit the RF
input to the band of interest. The output of the tuner drives an
IF filter that provides partial suppression of adjacent channels
and interferers that could otherwise limit the dynamic range of
the receiver. Set the conversion gain of the tuner such that the
peak IF input signal level into the AD9864 is no greater than
−18 dBm to prevent clipping. The AD9864 downconverts the
first IF signal to a second IF that is exactly 1/8 of the clock rate
of the Σ-Δ ADC (fCLK/8) to simplify the digital quadrature
demodulation process.
This second IF signal is then digitized by the Σ-Δ ADC,
demodulated into its quadrature I and Q components, filtered
via matching decimation filters, and reformatted to enable a
synchronous serial interface to a DSP. In this example, the LO
and CLK synthesizers of the AD9864 are both enabled,
requiring some additional passive components (for the loop
filters and CLK oscillator of the synthesizer) and a VCO for the
LO synthesizer. Note that not all of the required decoupling
capacitors are shown. See the External Passive Component
Requirements section and Figure 72 for more information on
required external passive components.
The selection of the first IF frequency is often based on the
availability of low cost standard crystal or SAW filters as well as
system frequency planning considerations. In general, crystal
filters are often used for narrow-band radios having channel
bandwidths below 50 kHz with IFs below 120 MHz, while SAW
filters are more suited for channel bandwidths greater than
50 kHz with IFs greater than 70 MHz. The ultimate stop-band
rejection required by the IF filter depends on how much
suppression is required at the AD9864 image band resulting
from downconversion to the second IF. This image band is
offset from the first IF by twice the second IF frequency
(± fCLK/4, depending on high-side or low-side injection).
The selectivity and bandwidth of the IF filter depends on both
the magnitude and frequency offset(s) of the adjacent channel
blocker(s) that could overdrive the input of the AD9864 or
generate in-band intermodulation components. Further
suppression is performed within the AD9864 by its inherent
band-pass response and digital decimation filters. Note that
some applications require additional application-specific
filtering performed in the DSP that follows the AD9864 to
remove the adjacent channel and/or implement a matched filter
for optimum signal detection.
Choose the output data rate of the AD9864, fOUT, to be at least
twice the bandwidth or symbol rate of the desired signal to ensure
that the decimation filters provide a flat pass-band response as
well as to allow for postprocessing by a DSP. After fOUT is
determined, the decimation factor of the digital filters must be
set such that the input clock rate, fCLK, falls between the AD9864
Rev. A | Page 41 of 47
AD9864
Data Sheet
VDDC
0.1µF
RBIAS
COSC
LOOP
FILTER
RD
LOSC
CVAR
CP
CZ
15
IOUTC
47 IFIN
FS 31
DOUTA 29
TO DSP
CLKOUT 28
43 LOP
Some applications, such as receiver diversity and beam steering,
may require two or more AD9864 devices operating in parallel
while maintaining synchronization. Figure 73 shows an example
of how multiple AD9864 devices can be cascaded, with one device
serving as the master and the other devices serving as the slaves.
In this example, all of the devices have the same SPI register
configuration since they share the same SPI interface to the
DSP. Because the state of each of the internal counters of the
AD9864 devices is unknown upon initialization, synchronization
of the devices is required via a SYNCB pulse (see Figure 37) to
synchronize their digital filters and to ensure precise time
alignment of the data streams.
Rev. A | Page 42 of 47
FROM
CRYSTAL
OSCILLATION
20 CLKN
42 LON
Note that although the VCO output of the LO synthesizer is accoupled to the LO input(s) of the slave, all of the CLK inputs of
the devices must be dc-coupled if the CLK oscillators of the
AD9864 are enabled. This is because of the dc current required
by the CLK oscillators in each device. In essence, these negative
impedance cores are operating in parallel, increasing the
effective Q of the LC resonator circuit. RBIAS must be sized such
that the sum of the dc bias currents of the oscillators maintains
a common-mode voltage of approximately 1.6 V.
FREF 35
19 CLKP
Synchronization of Multiple AD9864 Devices
Although the synthesizers of all of the device are enabled, the
LO and CLK signals for the slave(s) are derived from the
synthesizers of the master and are referenced to an external
crystal oscillator. All of the necessary external components (the
loop filters, varactor, LC, and VCO) required to ensure proper
closed-loop operation of the master synthesizers are included.
Note that the FREF input of the slave devices must be tied to
ground.
RF
PE 25
PD 24
AD9864
IOUTL
FROM
DSP
PC 23
MASTER
SYNCB 33
38
VCO
LOOP
FILTER
15
IOUTC
PE 25
47 IFIN
PD 24
PC 23
43 LOP
SYNCB 33
42 LON
AD9864
SLAVE
TO OTHER
AD9864s
19 CLKP
20 CLKN
FS 31
DOUTA 29
TO OTHER
AD9864s
CLKOUT 28
TO
DSP
FREF 35
Figure 74. Example of Synchronizing Multiple AD9864 Devices
04319-0-073
rated operating range of 13 MHz to 26 MHz and no significant
spurious products related to fCLK fall within the desired pass band,
resulting in a reduction in sensitivity performance. If a spurious
component is found to limit the sensitivity performance, the
decimation factor can often be modified slightly to find a
spurious free pass band. Selecting a higher fCLK is typically more
desirable given a choice, because the filtering requirements of
the first IF often depend on the transition region between the IF
frequency and the image band (±fCLK/4). Lastly, the output SSI
clock rate, fCLKOUT, and digital driver strength must be set to their
lowest possible settings to minimize the potential harmful
effects of digital induced noise while preserving a reliable data
link to the DSP. Note that the SSICRA, SSICRB, and SSIORD
registers (0x18, 0x19, and 0x1A) provide a large degree of
flexibility for optimization of the SSI interface.
Data Sheet
AD9864
Split Path Rx Architecture
A split path Rx architecture may be attractive for those
applications whose instantaneous dynamic range requirements
exceed the capability of a single AD9864 device. To cope with
these higher dynamic range requirements, two AD9864 devices
can be operated in parallel with their respective clip points
offset by a fixed amount. Adding a fixed amount of attenuation
in front of the AD9864 and/or programming the attenuation
setting of its internal VGA can adjust the input-referred clip
point. To save power and simplify hardware, the LO and CLK
circuits of the device can also be shared. Connecting the
SYNCB pins of the two devices and pulsing this line low
synchronizes the two devices.
An example of this concept for possible use in a GSM base
station is shown in Figure 75. The signal chain consists of a high
linearity RF front end and IF stage followed by two AD9864
devices operating in parallel. The RF front end consists of a
duplexer and preselect filter to pass the GSM RF band of
interest. A high performance LNA isolates the duplexer from
the preselect filter while providing sufficient gain to minimize
system NF. An RF mixer is used to downconvert the entire GSM
band to a suitable IF, where much of the channel selectivity is
accomplished. The 170.6 MHz IF is chosen to avoid any self
induced spurs from the AD9864. The IF stage consists of two
SAW filters isolated by a 15 dB gain stage.
The cascaded SAW filter response must provide sufficient blocker
rejection for the receiver to meet its sensitivity requirements
under worst-case blocker conditions. A composite response
having 27 dB, 60 dB, and 100 dB rejection at frequency offsets
of ±0.8 MHz, ±1.6 MHz, and ±6.5 MHz, respectively, provides
enough blocker suppression to ensure that the AD9864 with the
lower clip point is not overdriven by any blocker. This
configuration results in the best possible receiver sensitivity
under all blocking conditions.
The output of the last SAW filters drives the two AD9864
devices via a direct signal path and an attenuated signal path.
The direct path corresponds to the AD9864 having the lowest
clip point and provides the highest receiver sensitivity with a
system noise figure of 4.7 dB. The VGA of this device is set for
maximum attenuation, so its clip point is approximately
−17 dBm. Since conversion gain from the antenna to the
AD9864 is 19 dB, the digital output of this path is nominally
selected unless the target signal’s power exceeds −36 dBm at the
antenna. The attenuated path corresponds to the AD9864
having the highest input-referred clip point, and its digital
output point of this path is set to 7 dBm by inserting a 30 dB
attenuator and setting the VGA of the AD9864 to the middle of
its 12 dB range. This setting results in a ±6 dB adjustment of the
clip point, allowing the clip point difference to be calibrated to
exactly 24 dB, so that a simple 5-bit shift would make up the
gain difference. The attenuated path can handle signal levels up
to −12 dB at the antenna before being overdriven. Because the
SAW filters provide sufficient blocker suppression, the digital
data from this path need only be selected when the target signal
exceeds −36 dBm. Although the sensitivity of the receiver with
the attenuated path is 20 dB lower than the direct path, the
strong target signal ensures a sufficiently high carrier-to-noise
ratio.
Because GSM is based on a TDMA scheme, digital data (or
path) selection can occur on a slot-by-slot basis. Configure the
AD9864 to provide serial I and Q data at a frame rate of
541.67 kSPS, as well as additional information including a 2-bit
reset field and a 6-bit RSSI field. These two fields contain the
information needed to decide whether the direct or attenuated
path is to be used for the current time slot.
Rev. A | Page 43 of 47
AD9864
Data Sheet
VDDC
LOOP
FILTER
RBIAS
COSC
0.1µF
RD
LOSC
CP
CVAR
RF
CZ
ATTENUATED PATH WITH
CLIP POINT = 7.0dBm
15
19 CLKP
13MHz
IO
FREF 35
20 CLKN
FS 31
47 IFIN
DOUTA 29
CLKOUT 28
36dB
PAD
43 LOP
42 LON
AD9864
PE 25
MASTER
PD 24
PC 23
IOUTL
SYNCB 33
38
VCO
LOOP
FILTER
IF SAW 1I
LNA
MIXER
GAIN = –2dB
NF = 2dB
GAIN = 22dB
NF = 1dB
GAIN = –3dB
NF = 3dB
F SAW 2
15
IO
IF
AMP
PE 25
47 IFIN
PD 24
PC 23
43 LOP
GAIN = 5dB GAIN = 15dB GAIN = –9dB
NF = 12dB NF = 2dB
NF = –9dB
SYNCB 33
42 LON
AD9864
SLAVE
DIRECT PATH WITH
CLIP POINT = –17dBm
FS 31
19 CLKP
DOUTA 29
CLKOUT 28
20 CLKN
04319-0-074
DUPLEXER PRESELECT
DSP
OR
ASIC
FREF 35
Figure 75. Example of Split Path Rx Architecture to Increase Receiver Dynamic Range Capabilities
105
fCLK = 18MSPS
100
Because the mixer does not have any losses associated with the
mixing operation, the conversion gain through the LNA and
mixer is higher resulting in a nominal input clip point of
−24 dBm. The SNR performance is dependent on the VGA
attenuation setting, I/Q data resolution, and output bandwidth
as shown in Figure 76. Applications requiring the highest
instantaneous dynamic range must set the VGA for maximum
Rev. A | Page 44 of 47
MAX ATTEN WITH
24-BIT I/Q DATA
95
MAX ATTEN WITH
16-BIT I/Q DATA
90
MIN ATTEN WITH
16-BIT I/Q DATA
85
80
MIN ATTEN WITH
24-BIT I/Q DATA
0
20
40
60
80
100
120
140
BW (kHz)
Figure 76. Hung Mixer SNR vs. BW and VGA
160
04319-0-075
The AD9864 can operate in hung mixer mode by tying one of
the self biasing inputs of the LO to ground (that is, GNDI), or
the positive supply (VDDI). In this mode, the AD9864 acts as a
narrow-band, band-pass Σ-∆ ADC, because its mixer passes the
IFIN signal without any frequency translation. The IFIN signal
must be centered around the resonant frequency of the Σ-∆
ADC, fCLK/8, and the clock rate, fCLK, and decimation factors
must be selected to accommodate the bandwidth of the desired
input signal. Note that the LO synthesizer can be disabled
because it is no longer required.
attenuation. Several extra decibels in SNR performance can be
gained at lower signal bandwidths by using 24-bit I/Q data.
SNR (dB)
Hung Mixer Mode
Data Sheet
AD9864
LAYOUT EXAMPLE, EVALUATION BOARD, AND
SOFTWARE
The evaluation platform and its accompanying software provide
a simple way to evaluate the AD9864. Figure 77 shows the
AD9864 evaluation board connected to the HSC-ADCEVALCZ data capture card in its simplest configuration with
only an external 5 V lab supply and RF signal generator
required for the AD6676 evaluation board. The evaluation
board is designed to be flexible, supporting different IFs as well
as LO and CLK generation schemes. An alternative right angle
18-pin header is available for monitoring digital input/output
signals or interfacing to other FPGA, DSP, or microcontroller
development platforms. The evaluation board also serves as a
layout example.
The power supply distribution block provides filtered,
adjustable voltages to the main core of the AD9864 as well as
the digital input/output supply. In the IF input signal path,
component pads are available to implement different IF
impedance matching networks. The LO and CLK signals can be
externally applied or internally derived from on-board VCOs
supporting the on-chip LO and CLK synthesizers. The reference
for the on-chip LO and CLK synthesizers can be applied via the
external fREF input or an via the on-board crystal oscillator.
A simple software SPI control graphical user interface (GUI)
provides a ways to configure the AD9864. Analog Devices
VisualAnalog software is used for real-time data analysis. These
programs have a convenient GUI that allows easy access to the
various SPI port configuration registers and frequency/time
domain analysis of the output data.
SPI INITIALIZATION EXAMPLE
Table 24 shows an example SPI initialization sequence that is
used to configure the device when using both the on-chip LO
and CLK synthesizer. Note the following:


04319-0-177

Figure 77. Evaluation Board Platform
Rev. A | Page 45 of 47
Step 4 through Step 10 can be avoided if an external clock
is supplied. Note that clock synthesizer and oscillator must
also be disabled in Step 3 (0x00 = 0x7F) and remain disabled
in Step 15 (0x70 or 0x30 depending on the LO synthesizer
being enabled).
Step 20 to Step 24 can be avoided if an external LO source
is supplied.
Wait states are included for both tuning procedure and
clock synthesizer (if used).
AD9864
Data Sheet
DEVICE SPI INITIALIZATION
Table 24 shows an example SPI initialization script when both the CLK SYN/OSC and LO SYN are used.
Table 24. SPI Initialization Example, fCLKIN = 16.8 MHz, fADC = 18 MHz, fIF = 73.35 MHz, LO = 71.10 MHz, fDATA_IQ = 60 kSPS with
Decimate by 300
Step
1
2
Address (Hex)
0x3F
0x19
Write Value
0x99
0x87
3
0x00
0x45
Configure Clock Synthesizer
4
0x01
0x0C
5
0x10
0x00
6
0x11
0x38
7
0x12
0x00
8
0x13
0x3C
9
0x14
0x03
10
Not applicable Not applicable
Begin LC and RC Resonator Calibration
11
0x3E
0x47
12
13
14
15
16
0x38
0x39
0x1C
0x00
0x1C
0x01
0x0F
0x03
0x44
Readback value
17
18
0x38
0x3E
0x00
0x00
End of LC and RC Resonator Calibration
19
0x00
0x00
Configure LO Synthesizer
20
0x08
0x00
21
0x09
0x38
22
0x0A
0xA0
23
0x0B
0x1D
24
0x0C
0x03
Configure Remaining SPI Registers
25
0x03 to 0x07
26
0x07
0x04
27
0x1A
0x01
28
0x18
0x40
Description
Software reset.
Enable 4-wire SPI readback while keeping default 16-bit I/Q word width and maximum CMOS
output drive strength.
Take REF, GC, and CLK SYN/LO out of standby.
Set the CK oscillator bias to 0.65 mA for maximum swing.
Set the CLK synthesizer MSB and LSB reference frequency divider. (Note that default setting is
shown resulting in divide by 56.)
Set the CLK synthesizer MSB and LSB reference frequency divider. (Note that default setting is
shown resulting in divide by 300.)
Set the CLK synthesizer charge pump to 0.625 mA. (Note that setting depends on PLL loop
configuration.)
Wait until the CLK SYN output frequency settles to within 0.01% of final frequency.
Disable the internal ADC unstable signal. Enable RC Q enhancement, and bypass the RC and SC
resonators.
Enable manual control of feedback DAC.
Set DAC to midscale.
Set LC and RC tuning bits.
Bring ADC out of standby to initiate LC and RC calibration.
Wait 6 ms and read back Register 0x1C. If Register 0x1C clears, proceed to Step 17. If Register 0x1C
does not clear, reset Register 0x1C and return to Step 14. Make five attempts before exiting loop.
Disable manual control of feedback DAC.
Re-enable ADC unstable signal, disable RC_Q enhancement, and disable bypass of RC and SC
resonators.
Bring LNA/Mixer and LO synthesizer out of standby.
Set the LO synthesizer MSB and LSB reference frequency divider. (Note that the default setting
is shown resulting in divide by 56.)
Set the LO synthesizer A and B counters. (Note that the default setting is shown resulting in
divide by 237.)
Set the LO synthesizer charge pump to 0.625 mA. (Note that setting depends on PLL loop
configuration.)
Set AGC registers.
Set decimation factor to 300.
Set SSIORD register, which determines the CLKOUT frequency.
Take FS and CLKOUT out of tristate and configure SSI frame format.
Rev. A | Page 46 of 47
Data Sheet
AD9864
OUTLINE DIMENSIONS
0.30
0.23
0.18
0.60 MAX
0.60 MAX
37
48
1
36
PIN 1
INDICATOR
6.85
6.75 SQ
6.65
0.50
REF
5.25
5.10 SQ
4.95
EXPOSED
PAD
12
25
TOP VIEW
1.00
0.85
0.80
12° MAX
0.80 MAX
0.65 TYP
0.50
0.40
0.30
13
24
0.25 MIN
5.50 REF
0.05 MAX
0.02 NOM
COPLANARITY
0.08
0.20 REF
SEATING
PLANE
PIN 1
INDICATOR
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-VKKD-2
06-05-2012-A
7.10
7.00 SQ
6.90
Figure 78. 48-Lead Lead Frame Chip Scale Package [LFCSP]
7 mm × 7 mm Body and 0.85 mm Package Height
(CP-48-1)
Dimensions shown in millimeters
ORDERING GUIDE
Model1
AD9864BCPZ
AD9864BCPZRL
AD9864-EBZ
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
Package Description
48-Lead Lead Frame Chip Scale Package [LFCSP]
48-Lead Lead Frame Chip Scale Package [LFCSP]
Evaluation Board
Z = RoHS Compliant Part.
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registered trademarks are the property of their respective owners.
D04319-0-2/16(A)
Rev. A | Page 47 of 47
Package Option
CP-48-1
CP-48-1