AD ADL5382 700 mhz to 2.7 ghz quadrature demodulator Datasheet

700 MHz to 2.7 GHz
Quadrature Demodulator
ADL5382
FEATURES
FUNCTIONAL BLOCK DIAGRAM
CMRF CMRF RFIP RFIN CMRF VPX
23
24
22
21
20
19
VPA
1
ADL5382
18 VPB
COM
2
BIAS
GEN
17 VPB
BIAS
3
16 QHI
0°
90°
15 QLO
VPL
4
VPL
5
14 IHI
VPL
6
13 ILO
7
CML
8
9
10
LOIP LOIN CML
11
12
CML
COM
07208-001
Operating RF and LO frequency: 700 MHz to 2.7 GHz
Input IP3
33.5 dBm @ 900 MHz
30.5 dBm @1900 MHz
Input IP2: >70 dBm @ 900 MHz
Input P1dB: 14.7 dBm @ 900 MHz
Noise figure (NF)
14.0 dB @ 900 MHz
15.6 dB @ 1900 MHz
Voltage conversion gain: ~4 dB
Quadrature demodulation accuracy
Phase accuracy: ~0.2°
Amplitude balance: ~0.05 dB
Demodulation bandwidth: ~370 MHz
Baseband I/Q drive: 2 V p-p into 200 Ω
Single 5 V supply
Figure 1.
APPLICATIONS
Cellular W-CDMA/CDMA/CDMA2000/GSM
Microwave point-to-(multi)point radios
Broadband wireless and WiMAX
GENERAL DESCRIPTION
The ADL5382 is a broadband quadrature I-Q demodulator that
covers an RF input frequency range from 700 MHz to 2.7 GHz.
With a NF = 14 dB, IP1dB = 14.7 dBm, and IIP3 = 33.5 dBm at
900 MHz, the ADL5382 demodulator offers outstanding dynamic
range suitable for the demanding infrastructure direct-conversion
requirements. The differential RF inputs provide a well-behaved
broadband input impedance of 50 Ω and are best driven from a
1:1 balun for optimum performance.
The fully balanced design minimizes effects from second-order
distortion. The leakage from the LO port to the RF port is
<−65 dBc. Differential dc offsets at the I and Q outputs are typically
<10 mV. Both of these factors contribute to the excellent IIP2
specifications which is >60 dBm.
Excellent demodulation accuracy is achieved with amplitude and
phase balances ~0.05 dB and ~0.2°, respectively. The demodulated
in-phase (I) and quadrature (Q) differential outputs are fully
buffered and provide a voltage conversion gain of ~4 dB. The
buffered baseband outputs are capable of driving a 2 V p-p
differential signal into 200 Ω.
The ADL5382 is fabricated using the Analog Devices, Inc.,
advanced Silicon-Germanium bipolar process and is available
in a 24-lead exposed paddle LFCSP.
The ADL5382 operates off a single 4.75 V to 5.25 V supply. The
supply current is adjustable with an external resistor from the
BIAS pin to ground.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2008 Analog Devices, Inc. All rights reserved.
ADL5382
TABLE OF CONTENTS
Features .............................................................................................. 1
Emitter Follower Buffers ........................................................... 13
Applications....................................................................................... 1
Bias Circuit.................................................................................. 13
Functional Block Diagram .............................................................. 1
Applications Information .............................................................. 14
General Description ......................................................................... 1
Basic Connections...................................................................... 14
Revision History ............................................................................... 2
Power Supply............................................................................... 14
Specifications..................................................................................... 3
Local Oscillator (LO) Input ...................................................... 14
Absolute Maximum Ratings............................................................ 5
RF Input....................................................................................... 15
ESD Caution.................................................................................. 5
Baseband Outputs ...................................................................... 15
Pin Configuration and Function Descriptions............................. 6
Error Vector Magnitude (EVM) Performance........................... 16
Typical Performance Characteristics ............................................. 7
Low IF Image Rejection............................................................. 17
Distributions for fRF = 900 MHz ............................................... 10
Example Baseband Interface..................................................... 17
Distributions for fRF = 1900 MHz............................................. 11
Characterization Setups................................................................. 21
Distributions for fRF = 2700 MHz............................................. 12
Evaluation Board ............................................................................ 23
Circuit Description......................................................................... 13
Outline Dimensions ....................................................................... 27
LO Interface................................................................................. 13
Ordering Guide .......................................................................... 27
V-to-I Converter......................................................................... 13
Mixers .......................................................................................... 13
REVISION HISTORY
3/08—Revision 0: Initial Version
Rev. 0 | Page 2 of 28
ADL5382
SPECIFICATIONS
VS = 5 V, TA = 25°C, fLO = 900 MHz, fIF = 4.5 MHz, PLO = 0 dBm, BIAS pin open, ZO = 50 Ω, unless otherwise noted. Baseband outputs
differentially loaded with 450 Ω. Loss of the balun used to drive the RF port was de-embedded from these measurements.
Table 1.
Parameter
OPERATING CONDITIONS
LO and RF Frequency Range
LO INPUT
Input Return Loss
LO Input Level
I/Q BASEBAND OUTPUTS
Voltage Conversion Gain
Demodulation Bandwidth
Quadrature Phase Error
I/Q Amplitude Imbalance
Output DC Offset (Differential)
Output Common Mode
0.1 dB Gain Flatness
Output Swing
Peak Output Current
POWER SUPPLIES
Voltage
Current
DYNAMIC PERFORMANCE at RF = 900 MHz
Conversion Gain
Input P1dB
Second-Order Input Intercept (IIP2)
Third-Order Input Intercept (IIP3)
LO to RF
RF to LO
IQ Magnitude Imbalance
IQ Phase Imbalance
LO to IQ
Noise Figure
Noise Figure under Blocking Conditions
DYNAMIC PERFORMANCE at RF = 1900 MHz
Conversion Gain
Input P1dB
Second-Order Input Intercept (IIP2)
Third-Order Input Intercept (IIP3)
LO to RF
RF to LO
IQ Magnitude Imbalance
IQ Phase Imbalance
LO to IQ
Noise Figure
Noise Figure under Blocking Conditions
Condition
Min
Typ
0.7
LOIP, LOIN
LO driven differentially through a balun at 900 MHz
−6
QHI, QLO, IHI, ILO
450 Ω differential load on I and Q outputs at 900 MHz
200 Ω differential load on I and Q outputs at 900 MHz
1 V p-p signal, 3 dB bandwidth
At 900 MHz
−11
0
Differential 200 Ω load
Each pin
VPA, VPL, VPB, VPX
4.75
−5 dBm each input tone
−5 dBm each input tone
RFIN, RFIP terminated in 50 Ω
LOIN, LOIP terminated in 50 Ω
RFIN, RFIP terminated in 50 Ω
With a −5 dBm interferer 5 MHz away
−5 dBm each input tone
−5 dBm each input tone
RFIN, RFIP terminated in 50 Ω
LOIN, LOIP terminated in 50 Ω
RFIN, RFIP terminated in 50 Ω
With a −5 dBm interferer 5 MHz away
Rev. 0 | Page 3 of 28
Unit
2.7
GHz
+6
dB
dBm
3.9
3.0
370
0.2
0.05
±5
VPOS − 2.8
50
2
12
0 dBm LO input at 900 MHz
BIAS pin open
RBIAS = 4 kΩ
Max
dB
dB
MHz
Degrees
dB
mV
V
MHz
V p-p
mA
220
196
5.25
V
mA
mA
3.9
14.7
73
33.5
−92
−89
0.05
0.2
−43
14.0
19.9
dB
dBm
dBm
dBm
dBm
dBc
dB
Degrees
dBm
dB
dB
3.9
14.4
65
30.5
−71
−78
0.05
0.2
−41
15.6
20.5
dB
dBm
dBm
dBm
dBm
dBc
dB
Degrees
dBm
dB
dB
ADL5382
Parameter
DYNAMIC PERFORMANCE at RF = 2700 MHz
Conversion Gain
Input P1dB
Second-Order Input Intercept (IIP2)
Third-Order Input Intercept (IIP3)
LO to RF
RF to LO
IQ Magnitude Imbalance
IQ Phase Imbalance
LO to IQ
Noise Figure
Condition
RFIP, RFIN
Min
−5 dBm each input tone
−5 dBm each input tone
RFIN, RFIP terminated in 50 Ω, 1xLO appearing at RF port
LOIN, LOIP terminated in 50 Ω
RFIN, RFIP terminated in 50 Ω, 1xLO appearing at BB port
Rev. 0 | Page 4 of 28
Typ
3.3
14.5
52
28.3
−70
−55
0.16
0.1
−42
17.6
Max
Unit
dB
dBm
dBm
dBm
dBm
dBc
dB
Degrees
dBm
dB
ADL5382
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Supply Voltage (VPA, VPL, VPB, VPX)
LO Input Power
RF Input Power
Internal Maximum Power Dissipation
θJA
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Rating
5.5 V
13 dBm (re: 50 Ω)
15 dBm (re: 50 Ω)
1230 mW
54°C/W
150°C
−40°C to +85°C
−65°C to +125°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
Rev. 0 | Page 5 of 28
ADL5382
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
23
22
21
20
19
1
2
COM
3
BIAS
4
VPL
5
VPL
6
VPL
CML
7
VPB 17
QHI 16
ADL5382
TOP VIEW
(Not to Scale)
QLO 15
IHI 14
ILO 13
LOIP LOIN CML
8
9
10
CML
COM
11
12
07208-002
24
CMRF CMRF RFIP RFIN CMRF VPX
VPA
VPB 18
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1, 4 to 6,
17 to 19
2, 7, 10 to 12,
20, 23, 24
3
Mnemonic
VPA, VPL, VPB, VPX
BIAS
8, 9
LOIP, LOIN
13 to 16
ILO, IHI, QLO, QHI
21, 22
RFIN, RFIP
COM, CML, CMRF
EP
Description
Supply. Positive supply for LO, IF, biasing, and baseband sections. These pins should be decoupled
to the board ground using appropriate-sized capacitors.
Ground. Connect to a low impedance ground plane.
Bias Control. A resistor (RBIAS) can be connected between BIAS and COM to reduce the mixer core
current. The default setting for this pin is open.
Local Oscillator Input. Pins must be ac-coupled. A differential drive through a balun (recommended
balun is the M/A-COM ETC1-1-13) is necessary to achieve optimal performance.
I Channel and Q Channel Mixer Baseband Outputs. These outputs have a 50 Ω differential output
impedance (25 Ω per pin). The bias level on these pins is equal to VPOS − 2.8 V. Each output pair can
swing 2 V p-p (differential) into a load of 200 Ω. Output 3 dB bandwidth is 370 MHz.
RF Input. A single-ended 50 Ω signal can be applied to the RF inputs through a 1:1 balun (recommended
balun is the M/A-COM ETC1-1-13). Ground-referenced inductors must also be connected to RFIP and
RFIN (recommended values = 33 nH).
Exposed Paddle. Connect to a low impedance thermal and electrical ground plane.
Rev. 0 | Page 6 of 28
ADL5382
TYPICAL PERFORMANCE CHARACTERISTICS
VS = 5 V, TA = 25°C, LO drive level = 0 dBm, RBIAS = open, RF input balun loss is de-embedded, unless otherwise noted.
20
2
TA = –40°C
TA = +25°C
TA = +85°C
1
0
BASEBAND RESPONSE (dB)
GAIN (dB), IP1dB (dBm)
INPUT P1dB
15
10
GAIN
5
–1
–2
–3
–4
–5
–6
1100 1300 1500 1700 1900 2100 2300 2500 2700
RF FREQUENCY (MHz)
–8
07208-003
10
Figure 3. Conversion Gain and Input IP1 dB Compression Point (IP1dB) vs.
RF Frequency
80
19
70
INPUT IP3 (I AND Q CHANNELS)
900
16
15
14
13
TA = –40°C
TA = +25°C
TA = +85°C
1100 1300 1500 1700 1900 2100 2300 2500 2700
RF FREQUENCY (MHz)
12
700
07208-004
10
700
17
900
RF FREQUENCY (MHz)
Figure 4. Input Third-Order Intercept (IIP3) and
Input Second-Order Intercept Point (IIP2) vs. RF Frequency
Figure 7. Noise Figure vs. RF Frequency
4
1.5
3
QUADRATURE PHASE ERROR (Degrees)
2.0
0.5
0
–0.5
–1.0
–2.0
700
TA = –40°C
TA = +25°C
TA = +85°C
900
1100 1300 1500 1700 1900 2100 2300 2500 2700
RF FREQUENCY (MHz)
TA = –40°C
TA = +25°C
TA = +85°C
2
1
0
–1
–2
–3
–4
700
07208-005
GAIN MISMATCH (dB)
1.0
–1.5
1100 1300 1500 1700 1900 2100 2300 2500 2700
07208-007
IIP3, IIP2 (dBm)
NOISE FIGURE (dB)
INPUT IP2
30
20
TA = –40°C
TA = +25°C
TA = +85°C
18
50
40
1000
Figure 6. Normalized IQ Baseband Frequency Response
I CHANNEL
Q CHANNEL
60
100
BASEBAND FREQUENCY (MHz)
900
1100 1300 1500 1700 1900 2100 2300 2500 2700
RF FREQUENCY (MHz)
Figure 8. IQ Quadrature Phase Error vs. RF Frequency
Figure 5. IQ Gain Mismatch vs. RF Frequency
Rev. 0 | Page 7 of 28
07208-008
900
07208-006
–7
0
700
IIP3
5
35
GAIN
0
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
20
LO LEVEL (dBm)
8
4
25
GAIN
2
–5
32
IIP3 (dBm) AND NOISE FIGURE (dB)
220
INPUT IP3
210
200
190
18
NOISE FIGURE
180
14
170
160
100
10
RBIAS (kΩ)
–1
0
1
2
3
4
5
6
15
NOISE FIGURE
12
GAIN (dB), IP1dB (dBm),
IIP2 I AND Q CHANNEL (dBm)
19
900MHz
17
10
100
100
Figure 13. IIP3 and Noise Figure vs. RBIAS, fRF = 1900 MHz
70
21
1
RBIAS (kΩ)
27
23
INPUT IP3
16
80
25
NOISE FIGURE (dB)
–2
20
29
15
60
50
900MHz: GAIN
900MHz: IP1dB
900MHz: IIP2, I CHANNEL
900MHz: IIP2, Q CHANNEL
1900MHz: GAIN
1900MHz: IP1dB
1900MHz: IIP2, I CHANNEL
1900MHz: IIP2, Q CHANNEL
40
30
20
10
–25
–20
–15
–10
–5
RF BLOCKER INPUT POWER (dBm)
0
5
07208-011
13
–30
–3
24
Figure 10. IIP3, Noise Figure, and Supply Current vs. RBIAS, fRF = 900 MHz
1900MHz
–4
TA = –40°C
TA = +25°C
TA = +85°C
28
8
07208-010
1
35
IIP3
6
Figure 12. Conversion Gain, IP1dB, Noise Figure, IIP3, and IIP2 vs.
LO Level, fRF = 1900 MHz
230
22
10
45
LO LEVEL (dBm)
SUPPLY CURRENT (mA)
IIP3 (dBm) AND NOISE FIGURE (dB)
26
IP1dB
10
240
SUPPLY CURRENT
55
NOISE FIGURE
12
250
TA = –40°C
TA = +25°C
TA = +85°C
30
14
0
–6
Figure 9. Conversion Gain, IP1dB, Noise Figure, IIP3, and IIP2 vs.
LO Level, fRF = 900 MHz
34
65
16
IIP3, IIP2 (dBm)
50
10
IIP2, I CHANNEL
07208-013
NOISE FIGURE
IIP3, IIP2 (dBm)
65
15
18
07208-012
IP1dB
IIP2, Q CHANNEL
07208-014
IIP2, I CHANNEL
75
20
80
IIP2, Q CHANNEL
GAIN (dB), IP1dB (dBm), NOISE FIGURE (dB)
20
07208-009
GAIN (dB), IP1dB (dBm), NOISE FIGURE (dB)
ADL5382
Figure 11. Noise Figure vs. Input Blocker Level, fRF = 900 MHz, 1900 MHz
(RF Blocker 5 MHz Offset)
Rev. 0 | Page 8 of 28
0
1
10
RBIAS (kΩ)
Figure 14. Conversion Gain, IP1dB, IIP2_I, and IIP2_Q vs.
RBIAS, fRF = 900 MHz, 1900MHz
ADL5382
85
I CHANNEL
Q CHANNEL
35
–30
80
25
75
IIP2
70
15
IP1dB
65
TA = –40°C
TA = +25°C
TA = +85°C
0
0
10
–60
–70
–80
–90
20
30
40
60
50
BASEBAND FREQUENCY (MHz)
–100
700
–10
–30
–20
–40
LEAKAGE (dBc)
–20
–30
–40
–50
–50
–60
–70
–60
–80
–70
–90
1100 1300 1500 1700 1900 2100 2300 2500 2700
LO FREQUENCY (MHz)
–100
700
07208-016
LEAKAGE (dBm)
Figure 18. LO-to-RF Leakage vs. LO Frequency
0
900
1100 1300 1500 1700 1900 2100 2300 2500 2700
LO FREQUENCY (MHz)
Figure 15. IP1dB, IIP3, and IIP2 vs. Baseband Frequency
–80
700
900
07208-018
5
–50
900
1100 1300 1500 1700 1900 2100 2300 2500 2700
RF FREQUENCY (MHz)
07208-019
10
–40
07208-015
IP1dB, IIP3 (dBm)
30
20
–20
LEAKAGE (dBm)
IIP3
IIP2, I AND Q CHANNELS (dBm)
40
Figure 19. RF-to-LO Leakage vs. RF Frequency
Figure 16. LO-to-BB Leakage vs. LO Frequency
0
0
–5
–5
–15
RETURN LOSS (dB)
RETURN LOSS (dB)
–10
–20
–25
–30
–35
–40
–10
–15
–20
–25
900
1100 1300 1500 1700 1900 2100 2300 2500 2700
RF FREQUENCY (MHz)
Figure 17. RF Port Return Loss vs. RF Frequency Measured on a Characterization
Board through an ETC1-1-13 Balun with 33 nH Bias Inductors
Rev. 0 | Page 9 of 28
–30
700
900
1100 1300 1500 1700 1900 2100 2300 2500 2700
LO FREQUENCY (MHz)
Figure 20. LO Port Return Loss vs. LO Frequency Measured on
Characterization Board through an ETC1-1-13 Balun
07208-020
–50
700
07208-017
–45
ADL5382
DISTRIBUTIONS FOR fRF = 900 MHz
100
100
TA = –40°C
TA = +25°C
TA = +85°C
80
PERCENTAGE (%)
60
40
20
32
33
34
35
36
37
Figure 21. IIP3 Distributions, fRF = 900 MHz
100
0
45
55
60
65
70
75
80
85
Figure 24. IIP2 Distributions for I Channel and Q Channel, fRF = 900 MHz
100
TA = –40°C
TA = +25°C
TA = +85°C
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
80
60
40
20
60
40
13
14
15
16
17
0
12.5
07208-022
12
14.0
14.5
15.0
15.5
1.00
Figure 25. Noise Figure Distributions, fRF = 900 MHz
100
TA = –40°C
TA = +25°C
TA = +85°C
80
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
80
60
40
20
60
40
20
–0.1
0
0.1
GAIN MISMATCH (dB)
0.2
07208-023
0
–0.2
13.5
NOISE FIGURE (dB)
Figure 22. IP1dB Distributions, fRF = 900 MHz
100
13.0
07208-025
20
INPUT P1dB (dBm)
PERCENTAGE (%)
50
INPUT IP2 (dBm)
80
0
I CHANNEL
Q CHANNEL
07208-024
31
INPUT IP3 (dBm)
PERCENTAGE (%)
40
20
07208-021
0
60
07208-026
PERCENTAGE (%)
80
TA = –40°C
TA = +25°C
TA = +85°C
Figure 23. IQ Gain Mismatch Distributions, fRF = 900 MHz
0
–1.00
–0.75
–0.50
–0.25
0
0.25
0.50
0.75
QUADRATURE PHASE ERROR (Degrees)
Figure 26. IQ Quadrature Phase Error Distributions, fRF = 900 MHz
Rev. 0 | Page 10 of 28
ADL5382
DISTRIBUTIONS FOR fRF = 1900 MHz
100
TA = –40°C
TA = +25°C
TA = +85°C
80
PERCENTAGE (%)
60
40
20
30
31
32
33
0
45
Figure 27. IIP3 Distributions, fRF = 1900 MHz
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
PERCENTAGE (%)
65
70
75
80
85
TA = –40°C
TA = +25°C
TA = +85°C
60
40
20
20
14
15
16
17
0
14.0
07208-028
13
INPUT P1dB (dBm)
14.5
15.0
15.5
16.0
16.5
17.0
1.00
NOISE FIGURE (dB)
Figure 28. IP1dB Distributions, fRF = 1900 MHz
Figure 31. Noise Figure Distributions, fRF = 1900 MHz
100
100
TA = –40°C
TA = +25°C
TA = +85°C
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
80
60
40
60
40
20
20
–0.1
0
0.1
GAIN MISMATCH (dB)
0.2
07208-029
PERCENTAGE (%)
60
80
40
0
–0.2
55
Figure 30. IIP2 Distributions for I Channel and Q Channel, fRF = 1900 MHz
60
80
50
INPUT IP2 (dBm)
80
0
12
I CHANNEL
Q CHANNEL
07208-030
29
INPUT IP3 (dBm)
100
40
20
07208-027
0
28
60
07208-031
PERCENTAGE (%)
80
TA = –40°C
TA = +25°C
TA = +85°C
07208-032
100
0
–1.00
–0.75
–0.50
–0.25
0
0.25
0.50
0.75
QUADRATURE PHASE ERROR (Degrees)
Figure 32. IQ Quadrature Phase Error Distributions, fRF = 1900 MHz
Figure 29. IQ Gain Mismatch Distributions, fRF = 1900 MHz
Rev. 0 | Page 11 of 28
ADL5382
DISTRIBUTIONS FOR fRF = 2700 MHz
100
TA = –40°C
TA = +25°C
TA = +85°C
80
PERCENTAGE (%)
60
40
20
28
29
30
31
0
45
Figure 33. IIP3 Distributions, fRF = 2700 MHz
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
PERCENTAGE (%)
20
65
70
75
80
85
TA = –40°C
TA = +25°C
TA = +85°C
60
40
20
14
15
16
17
0
16.0
07208-034
13
INPUT P1dB (dBm)
16.5
17.0
17.5
18.0
18.5
19.0
1.00
NOISE FIGURE (dB)
Figure 34. IP1dB Distributions, fRF = 2700 MHz
Figure 37. Noise Figure Distributions, fRF = 2700 MHz
100
TA = –40°C
TA = +25°C
TA = +85°C
80
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
80
60
40
20
60
40
20
0
0.1
0.2
0.3
GAIN MISMATCH (dB)
07208-035
PERCENTAGE (%)
60
80
40
0
–0.1
55
Figure 36. IIP2 Distributions for I Channel and Q Channel, fRF = 2700 MHz
60
100
50
INPUT IP2 (dBm)
80
0
12
I CHANNEL
Q CHANNEL
07208-036
27
INPUT IP3 (dBm)
100
40
20
07208-033
0
26
60
07208-037
PERCENTAGE (%)
80
TA = –40°C
TA = +25°C
TA = +85°C
07208-038
100
Figure 35. IQ Gain Mismatch Distributions, fRF = 2700 MHz
0
–1.00
–0.75
–0.50
–0.25
0
0.25
0.50
0.75
QUADRATURE PHASE ERROR (Degrees)
Figure 38. IQ Quadrature Phase Error Distributions, fRF = 2700 MHz
Rev. 0 | Page 12 of 28
ADL5382
CIRCUIT DESCRIPTION
The ADL5382 can be divided into five sections: the local
oscillator (LO) interface, the RF voltage-to-current (V-to-I)
converter, the mixers, the differential emitter follower outputs,
and the bias circuit. A detailed block diagram of the device is
shown in Figure 39.
BIAS
The ADL5382 has two double-balanced mixers: one for the
in-phase channel (I channel) and one for the quadrature channel
(Q channel). These mixers are based on the Gilbert cell design
of four cross-connected transistors. The output currents from
the two mixers are summed together in the resistive loads that
then feed into the subsequent emitter follower buffers.
ILO
LOIP
RFIN
The differential RF input signal is applied to a resistively
degenerated common base stage, which converts the differential
input voltage to output currents. The output currents then
modulate the two half frequency LO carriers in the mixer stage.
MIXERS
IHI
RFIP
V-TO-I CONVERTER
POLYPHASE
QUADRATURE
PHASE SPLITTER
EMITTER FOLLOWER BUFFERS
The output emitter followers drive the differential I and Q
signals off-chip. The output impedance is set by on-chip 25 Ω
series resistors that yield a 50 Ω differential output impedance
for each baseband port. The fixed output impedance forms a
voltage divider with the load impedance that reduces the effective
gain. For example, a 500 Ω differential load has 1 dB lower
effective gain than a high (10 kΩ) differential load impedance.
LOIN
QLO
07208-039
QHI
Figure 39. Block Diagram
The LO interface generates two LO signals at 90° of phase
difference to drive two mixers in quadrature. RF signals are
converted into currents by the V-to-I converters that feed into
the two mixers. The differential I and Q outputs of the mixers
are buffered via emitter followers. Reference currents to each
section are generated by the bias circuit. A detailed description
of each section follows.
LO INTERFACE
The LO interface consists of a polyphase quadrature splitter
followed by a limiting amplifier. The LO input impedance is set
by the polyphase, which splits the LO signal into two differential
signals in quadrature. Each quadrature LO signal then passes
through a limiting amplifier that provides the mixer with a
limited drive signal. For optimal performance, the LO inputs
must be driven differentially.
BIAS CIRCUIT
A band gap reference circuit generates the proportional-toabsolute temperature (PTAT) as well as temperature-independent
reference currents used by different sections. The mixer current
can be reduced via an external resistor between the BIAS pin
and ground. When the BIAS pin is open, the mixer runs at
maximum current and therefore the greatest dynamic range.
The mixer current can be reduced by placing a resistance to
ground; therefore, reducing overall power consumption, noise
figure, and IIP3. The effect on each of these parameters is
shown in Figure 10, Figure 13, and Figure 14.
Rev. 0 | Page 13 of 28
ADL5382
APPLICATIONS INFORMATION
BASIC CONNECTIONS
LOCAL OSCILLATOR (LO) INPUT
Figure 41 shows the basic connections schematic for the ADL5382.
For optimum performance, the LO port should be driven
differentially through a balun. The recommended balun is
the M/A-COM ETC1-1-13. The LO inputs to the device should
be ac-coupled with 1000 pF capacitors. The LO port is designed
for a broadband 50 Ω match from 700 MHz to 2.7 GHz. The
LO return loss can be seen in Figure 20. Figure 40 shows the LO
input configuration.
POWER SUPPLY
The nominal voltage supply for the ADL5382 is 5 V and is
applied to the VPA, VPB, VPL, and VPX pins. Ground should
be connected to the COM, CML, and CMRF pins. The exposed
paddle on the underside of the package should also be soldered
to a low thermal and electrical impedance ground plane. If the
ground plane spans multiple layers on the circuit board, these
layers should be stitched together with nine vias under the
exposed paddle. The Application Note AN-772 discusses the
thermal and electrical grounding of the LFCSP in detail. Each
of the supply pins should be decoupled using two capacitors;
recommended capacitor values are 100 pF and 0.1 μF.
LO INPUT
8
LOIP
9
LOIN
1000pF
07208-040
1000pF
ETC1-1-13
Figure 40. Differential LO Drive
The recommended LO drive level is between −6 dBm and +6 dBm.
The applied LO frequency range is between 700 MHz and 2.7 GHz.
ETC1-1-13
RFC
1000pF
1000pF
24
23
22
21
20
19
CMRF
RFIP
RFIN
CMRF
VPX
VPOS
1 VPA
0.1µF
33nH
CMRF
33nH
100pF
2 COM
QHI 16
ADL5382
4 VPL
5 VPL
IHI 14
6 VPL
ILO 13
LOIP
LOIN
CML
CML
COM
100pF
7
8
9
10
11
12
1000pF
QHI
QLO 15
CML
QLO
IHI
ILO
1000pF
ETC1-1-13
07208-041
0.1µF
0.1µF
VPB 17
3 BIAS
VPOS
VPOS
VPB 18
100pF
LO
Figure 41. Basic Connections Schematic
Rev. 0 | Page 14 of 28
ADL5382
33nH
21 RFIN
1000pF
ETC1-1-13
1000pF
22 RFIP
RF INPUT
07208-042
33nH
–10
–12
–14
–16
–18
–20
–22
–24
0.7
0.9
1.1
1.3
1.5
1.7
1.9
2.1
2.3
2.5
FREQUENCY (GHz)
2.7
2.9
07208-043
The RF inputs have a differential input impedance of
approximately 50 Ω. For optimum performance, the RF port
should be driven differentially through a balun. The recommended
balun is the M/A-COM ETC1-1-13. The RF inputs to the device
should be ac-coupled with 1000 pF capacitors. Ground-referenced
choke inductors must also be connected to RFIP and RFIN (the
recommended value is 33 nH, Coilcraft 0603CS-33NX) for
appropriate biasing. Several important aspects must be taken
into account when selecting an appropriate choke inductor for
this application. First, the inductor must be able to handle the
approximately 40 mA of standing dc current being delivered
from each of the RF input pins (RFIP, RFIN). The suggested
0603 inductor has a 600 mA current rating. The purpose of the
choke inductors is to provide a very low resistance dc path to
ground and high ac impedance at the RF frequency so as not to
affect the RF input impedance. A choke inductor that has a selfresonant frequency greater than the RF input frequency ensures
that the choke is still looking inductive and therefore has a more
predictable ac impedance (jωL) at the RF frequency. Figure 42
shows the RF input configuration.
The differential RF port return loss is characterized as shown in
Figure 43.
S11 (dB)
RF INPUT
Figure 43. Differential RF Port Return Loss
BASEBAND OUTPUTS
The baseband outputs QHI, QLO, IHI, and ILO are fixed
impedance ports. Each baseband pair has a 50 Ω differential
output impedance. The outputs can be presented with differential
loads as low as 200 Ω (with some degradation in gain) or high
impedance differential loads (500 Ω or greater impedance yields
the same excellent linearity) that is typical of an ADC. The
TCM9-1 9:1 balun converts the differential IF output to singleended. When loaded with 50 Ω, this balun presents a 450 Ω
load to the device. The typical maximum linear voltage swing
for these outputs is 2 V p-p differential. The bias level on these
pins is equal to VPOS − 2.8 V. The output 3 dB bandwidth is
370 MHz. Figure 44 shows the baseband output configuration.
QHI 16
QHI
QLO 15
QLO
IHI 14
IHI
ILO 13
ILO
07208-044
Figure 42. RF Input
Figure 44. Baseband Output Configuration
Rev. 0 | Page 15 of 28
ADL5382
0
ERROR VECTOR MAGNITUDE (EVM) PERFORMANCE
–10
–15
EVM (dB)
EVM is a measure used to quantify the performance of a digital
radio transmitter or receiver. A signal received by a receiver
would have all constellation points at the ideal locations; however,
various imperfections in the implementation (such as magnitude
imbalance, noise floor, and phase imbalance) cause the actual
constellation points to deviate from the ideal locations.
–5
The ADL5382 shows excellent EVM performance for various
modulation schemes. Figure 45 shows the EVM performance of
the ADL5382 with a 16 QAM, 200 kHz low IF.
–20
–25
–30
–35
–40
–50
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
–5
10
–20
Figure 47 exhibits multiple W-CDMA low-IF EVM
performance curves over a wide RF input power range into the
ADL5382. In the case of zero-IF, the noise contribution by the
vector signal analyzer becomes predominant at lower power
levels, making it difficult to measure SNR accurately.
–25
–30
–35
–40
0
–45
–65
–55
–45
–35
–25
–15
RF INPUT POWER (dBm)
–5
–10
–15
Figure 45. EVM, RF = 900 MHz, IF = 200 kHz vs.
RF Input Power for a 16 QAM 160 ksym/s Signal
Figure 46 shows the zero-IF EVM performance of a 10 MHz
IEEE 802.16e WiMAX signal through the ADL5382. The
differential dc offsets on the ADL5382 are in the order of a few
millivolts. However, ac coupling the baseband outputs with
10 μF capacitors eliminates dc offsets and enhances EVM
performance. With a 10 MHz BW signal, 10 μF ac coupling
capacitors with the 500 Ω differential load results in a high-pass
corner frequency of ~64 Hz, which absorbs an insignificant
amount of modulated signal energy from the baseband signal.
By using ac-coupling capacitors at the baseband outputs, the dc
offset effects, which can limit dynamic range at low input power
levels, can be eliminated.
EVM (dB)
–75
07208-045
–5
–20
–25
–30
–35
0Hz
5MHz
–40
–45
2.5MHz
7.5MHz
–50
–75 –70 –65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
RF INPUT POWER (dBm)
07208-047
EVM (dB)
5
Figure 46. EVM, RF = 2.6 GHz, IF = 0 Hz vs. RF Input Power for a 16 QAM
10 MHz Bandwidth Mobile WiMAX Signal (AC-Coupled Baseband Outputs)
–15
–50
–85
0
RF INPUT POWER (dBm)
–10
07208-046
–45
0
Figure 47. EVM, RF = 1900 MHz, IF = 0 Hz, 2.5 MHz, 5 MHz, and 7.5 MHz vs. RF
Input Power for a W-CDMA Signal (AC-Coupled Baseband Outputs)
Rev. 0 | Page 16 of 28
ADL5382
COSωLOt
0°
ωIF
ωIF
–ωIF
0
+ωIF
–90°
0
+ωIF
0
+ωIF
+90°
ωLSB
ωLO
ωUSB
0°
0
+ωIF
07208-048
–ωIF
SINωLOt
Figure 48. Illustration of the Image Problem
LOW IF IMAGE REJECTION
EXAMPLE BASEBAND INTERFACE
The image rejection ratio is the ratio of the intermediate frequency
(IF) signal level produced by the desired input frequency to that
produced by the image frequency. The image rejection ratio is
expressed in decibels. Appropriate image rejection is critical
because the image power can be much higher than that of the
desired signal, thereby plaguing the down conversion process.
Figure 48 illustrates the image problem. If the upper sideband
(lower sideband) is the desired band, a 90° shift to the Q channel
(I channel) cancels the image at the lower sideband (upper
sideband). Phase and gain balance between I and Q channels
are critical for high levels of image rejection.
In most direct conversion receiver designs, it is desirable to
select a wanted carrier within a specified band. The desired
channel can be demodulated by tuning the LO to the appropriate
carrier frequency. If the desired RF band contains multiple
carriers of interest, the adjacent carriers would also be down
converted to a lower IF frequency. These adjacent carriers can
be problematic if they are large relative to the wanted carrier as
they can overdrive the baseband signal detection circuitry. As a
result, it is often necessary to insert a filter to provide sufficient
rejection of the adjacent carriers.
Figure 49 shows the excellent image rejection capabilities of
the ADL5382 for low IF applications, such as W-CDMA. The
ADL5382 exhibits image rejection greater than 45 dB over a
broad frequency range.
70
2.5MHz LOW IF
5MHz LOW IF
IMAGE REJECTION (dB)
60
50
7.5MHz LOW IF
40
30
20
0
700
900
1100 1300 1500 1700 1900 2100 2300 2500 2700
RF FREQUENCY (MHz)
07208-049
10
It is necessary to consider the overall source and load impedance
presented by the ADL5382 and ADC input to design the filter
network. The differential baseband output impedance of the
ADL5382 is 50 Ω. The ADL5382 is designed to drive a high
impedance ADC input. It may be desirable to terminate the
ADC input down to lower impedance by using a terminating
resistor, such as 500 Ω. The terminating resistor helps to better
define the input impedance at the ADC input at the cost of a
slightly reduced gain (see the Circuit Description section for
details on the emitter-follower output loading effects). The order
and type of filter network depends on the desired high frequency
rejection required, pass-band ripple, and group delay. Filter
design tables provide outlines for various filter types and orders,
illustrating the normalized inductor and capacitor values for a
1 Hz cutoff frequency and 1 Ω load. After scaling the normalized
prototype element values by the actual desired cut-off frequency
and load impedance, the series reactance elements are halved to
realize the final balanced filter network component values.
Figure 49. Image Rejection vs. RF Frequency for a W-CDMA Signal,
IF = 2.5 MHz, 5 MHz, and 7.5 MHz
Rev. 0 | Page 17 of 28
ADL5382
The balanced configuration is realized as the 0.54 μH inductor
is split in half to realize the network shown in Figure 50.
LN = 0.074H
NORMALIZED
SINGLE-ENDED
CONFIGURATION
VS
CN
14.814F
RS
= 0.1
RL
RS = 50Ω
5
0
–5
–10
–15
RL= 500Ω
–20
fC = 1Hz
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
FREQUENCY (MHz)
0.54µH
DENORMALIZED
SINGLE-ENDED
EQUIVALENT
VS
10
07208-051
RS = 50Ω
Figure 51 and Figure 52 show the measured frequency response
and group delay of the filter.
MAGNITUDE RESPONSE (dB)
As an example, a second-order Butterworth, low-pass filter
design is shown in Figure 50 where the differential load impedance
is 500 Ω and the source impedance of the ADL5382 is 50 Ω. The
normalized series inductor value for the 10-to-1, load-to-source
impedance ratio is 0.074 H, and the normalized shunt capacitor
is 14.814 F. For a 10.9 MHz cutoff frequency, the single-ended
equivalent circuit consists of a 0.54 μH series inductor followed
by a 433 pF shunt capacitor.
Figure 51. Sixth-Order Baseband Filter Response
433pF
900
RL= 500Ω
800
RL
2 = 250Ω
RL
= 250Ω
2
0.27µH
Figure 50. Second-Order Butterworth, Low-Pass Filter Design Example
A complete design example is shown in Figure 53. A sixth-order
Butterworth differential filter having a 1.9 MHz corner frequency
interfaces the output of the ADL5382 to that of an ADC input.
The 500 Ω load resistor defines the input impedance of the
ADC. The filter adheres to typical direct conversion W-CDMA
applications, where 1.92 MHz away from the carrier IF frequency,
1 dB of rejection is desired and 2.7 MHz away 10 dB of rejection
is desired.
Rev. 0 | Page 18 of 28
600
500
400
300
200
100
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
FREQUENCY (MHz)
Figure 52. Sixth-Order Baseband Filter Group Delay
1.8
07208-052
RS
= 25Ω
2
433pF
DELAY (ns)
BALANCED
CONFIGURATION
VS
700
0.27µH
07208-050
RS
= 25Ω
2
fC = 10.9MHz
ADL5382
ETC1-1-13
RFC
1000pF
1000pF
33nH
RFIP
RFIN
CMRF
VPX
1 VPA
0.1µF
VPOS
VPB 18
100pF
100pF
CAC
10µF
0.1µF
VPB 17
2 COM
27µH
27µH
10µH
27µH
27µH
10µH
ADC INPUT
19
ADC INPUT
20
500Ω
21
10µH
500Ω
22
27µH
68pF
23
27µH
100pF
24
CMRF
VPOS
CMRF
CAC
10µF
270pF
33nH
QHI 16
3 BIAS
ADL5382
VPOS
QLO 15
5 VPL
IHI 14
6 VPL
ILO 13
COM
10
11
12
1000pF
CAC
10µF
1000pF
27µH
ETC1-1-13
LO
Figure 53. Sixth-Order Low-Pass Butterworth, Baseband Filter Schematic
Rev. 0 | Page 19 of 28
27µH
10µH
07208-053
CML
9
68pF
CML
8
100pF
LOIN
7
CAC
10µF
270pF
LOIP
100pF
CML
0.1µF
4 VPL
ADL5382
0
–5
–10
–15
200Ω
–20
0.2
0.6
1.0
1.4
1.8
2.2
2.6
3.0
3.4
3.8
8µH
Figure 55. Fourth-Order Low-Pass W-CDMA Filter Magnitude Response
900
Figure 54. Fourth-Order Low-Pass W-CDMA Filter Schematic
800
Figure 55 and Figure 56 illustrate the magnitude response and
group delay response of the fourth-order filter, respectively.
DELAY (ns)
700
600
500
400
300
200
100
0.2
0.4
0.6
0.8
1.0
1.2
1.4
FREQUENCY (MHz)
1.6
1.8
2.0
07208-056
10µH
07208-054
FREQUENCY (MHz)
07208-055
8µH
100pF
680pF
50Ω
10µH
5
MAGNITUDE RESPONSE (dB)
As the load impedance of the filter increases, the filter design
becomes more challenging in terms of meeting the required
rejection and pass band specifications. In the previous WCDMA example, the 500 Ω load impedance resulted in the
design of a sixth-order filter that has relatively large inductor
values and small capacitor values. If the load impedance is 200 Ω,
the filter design becomes much more manageable. As shown in
Figure 54, the resultant inductor and capacitor values become
much more practical.
Figure 56. Fourth-Order Low-Pass W-CDMA Filter Group Delay Response
Rev. 0 | Page 20 of 28
ADL5382
CHARACTERIZATION SETUPS
Figure 57 to Figure 59 show the general characterization bench
setups used extensively for the ADL5382. The setup shown in
Figure 59 was used to do the bulk of the testing and used sinusoidal
signals on both the LO and RF inputs. An automated Agilent
VEE program was used to control the equipment over the IEEE
bus. This setup was used to measure gain, IP1dB, IIP2, IIP3, I/Q
gain match, and quadrature error. The ADL5382 characterization
board had a 9-to-1 impedance transformer on each of the
differential baseband ports to do the differential-to-singleended conversion, which presented a 450 Ω differential load to
each baseband port, when interfaced with 50 Ω test equipment.
For all measurements of the ADL5382, the loss of the RF input
balun (the M/A-COM ETC1-1-13 was used on RF input during
characterization) was de-embedded.
The two setups shown in Figure 57 and Figure 58 were used for
making NF measurements. Figure 57 shows the setup for
measuring NF with no blocker signal applied while Figure 58
was used to measure NF in the presence of a blocker. For both
setups, the noise was measured at a baseband frequency of
10 MHz. For the case where a blocker was applied, the output
blocker was at a 15 MHz baseband frequency. Note that great
care must be taken when measuring NF in the presence of a
blocker. The RF blocker generator must be filtered to prevent its
noise (which increases with increasing generator output power)
from swamping the noise contribution of the ADL5382. At least
30 dB of attention at the RF and image frequencies is desired.
For example, assume a 915 MHz signal applied to the LO inputs of
the ADL5382. To obtain a 15 MHz output blocker signal, the RF
blocker generator is set to 930 MHz and the filters tuned such
that there is at least 30 dB of attenuation from the generator at
both the desired RF frequency (925 MHz) and the image RF
frequency (905 MHz). Finally, the blocker must be removed
from the output (by the 10 MHz low-pass filter) to prevent the
blocker from swamping the analyzer.
SNS
OUTPUT
RF
ADL5382
VPOS CHAR BOARD
I
LO
INPUT
6dB PAD
HP 6235A
POWER SUPPLY
R1
50Ω
AGILENT N8974A
NOISE FIGURE ANALYZER
LOW-PASS
FILTER
IEEE
GND
Q
FROM SNS PORT
CONTROL
AGILENT 8665B
SIGNAL GENERATOR
PC CONTROLLER
Figure 57. General Noise Figure Measurement Setup
Rev. 0 | Page 21 of 28
07208-057
IEEE
ADL5382
BAND-PASS
TUNABLE FILTER
BAND-REJECT
TUNABLE FILTER
6dB PAD
R&S SMT03
SIGNAL GENERATOR
RF
GND
ADL5382
6dB PAD
VPOS CHAR BOARD
LOW-PASS
FILTER
I
LO
6dB PAD
HP 6235A
POWER SUPPLY
R&S FSEA30
SPECTRUM ANALYZER
R1
50Ω
Q
HP87405
LOW NOISE
PREAMP
07208-058
BAND-PASS
CAVITY FILTER
AGILENT 8665B
SIGNAL GENERATOR
Figure 58. Measurement Setup for Noise Figure in the Presence of a Blocker
3dB PAD
RF
AMPLIFIER
3dB PAD IN
RF
OUT 3dB PAD
IEEE
VP GND
3dB PAD
AGILENT
11636A
R&S SMT06
6dB PAD
IEEE
RF
SWITCH
MATRIX
VPOS CHAR BOARD
LO
I 6dB PAD
IEEE
6dB PAD
AGILENT E3631
PWER SUPPLY
RF
INPUT
AGILENT E8257D
SIGNAL GENERATOR
IEEE
PC CONTROLLER
IEEE
R&S FSEA30
SPECTRUM ANALYZER
Figure 59. General Characterization Setup
Rev. 0 | Page 22 of 28
HP 8508A
VECTOR VOLTMETER
07208-059
IEEE
Q 6dB PAD
ADL5382
IEEE
RF
GND
INPUT CHANNELS
A AND B
R&S SMT06
ADL5382
EVALUATION BOARD
The ADL5382 evaluation board is available. The board can be
used for single-ended or differential baseband analysis. The default
configuration of the board is for single-ended baseband analysis.
T1
RFC
C1
23
22
21
20
19
CMRF
RFIP
RFIN
CMRF
VPX
R7
R6
VPB 18
VPOS
C8
C2
2 COM
VPB 17
3 BIAS
QHI 16
ADL5382
4 VPL
C9
R9
Q OUTPUT OR QHI
R14
R15
QLO 15
T2
C12
R3
5 VPL
IHI 14
6 VPL
ILO 13
R16
LOIP
LOIN
CML
CML
COM
C4
CML
7
8
9
10
11
12
QLO
R10
R11
I OUTPUT OR IHI
R5
R4
T3
C13
R13
C6
C7
ILO
R12
T4
07208-060
C3
L1
24
1 VPA
R2
VPOS
C10
CMRF
VPOS
R1
C11
L2
R8
LO
Figure 60. Evaluation Board Schematic
Rev. 0 | Page 23 of 28
ADL5382
Table 4. Evaluation Board Configuration Options
Component
VPOS, GND
R1, R3, R6
C1, C2, C3,
C4, C8, C9
C6, C7,
C10, C11
R4, R5,
R9 to R16
L1, L2,
R7, R8
T2, T3
C12, C13
T4
T1
R2
Function
Power Supply and Ground Vector Pins.
Power Supply Decoupling. Shorts or power supply decoupling resistors.
These capacitors provide the required decoupling up to 2.7 GHz.
AC Coupling Capacitors. These capacitors provide the required ac coupling from
700 MHz to 2.7 GHz.
Single-Ended Baseband Output Path. This is the default configuration of the evaluation
board. R14 to R16 and R4, R5, and R13 are populated for appropriate balun interface.
R9, R10 and R11, R12 are not populated. Baseband outputs are taken from QHI and IHI.
The user can reconfigure the board to use full differential baseband outputs. R9 to R12
provide a means to bypass the 9:1 TCM9-1 transformer to allow for differential baseband
outputs. Access the differential baseband signals by populating R9 to R12 with 0 Ω and
not populating R4, R5, R13 to R16. This way the transformer does not need to be
removed. The baseband outputs are taken from the SMAs of Q_HI, Q_LO, I_HI, and I_LO.
Input Biasing. Inductance and resistance sets the input biasing of the common
base input stage. The default value is 33 nH.
IF Output Interface. TCM9-1 converts a differential high impedance IF output to a singleended output. When loaded with 50 Ω, this balun presents a 450 Ω load to the device.
The center tap can be decoupled through a capacitor to ground.
Decoupling Capacitors. C12 and C13 are the decoupling capacitors used to reject noise
on the center tap of the TCM9-1.
LO Input Interface. The LO is driven differentially. ETC1-1-13 is a 1:1 RF balun that
converts the single-ended RF input to differential signal.
RF Input Interface. ETC1-1-13 is a 1:1 RF balun that converts the single-ended RF input
to differential signal.
RBIAS. Optional bias setting resistor. See the Bias Circuit section to see how to use this feature.
Rev. 0 | Page 24 of 28
Default Condition
Not applicable
R1, R3, R6 = 0 Ω (0603)
C2, C4, C8 = 100 pF (0402)
C1, C3, C9 = 0.1 μF (0603)
C6, C10, C11 = 1000 pF (0402)
C7 = open
R4, R5, R13 to R16 = 0 Ω (0402)
R9 to R12 = open
L1, L2 = 33 nH (0603CS-33NX,
Coilcraft)
R7, R8 = 0 Ω (0402)
T2, T3 = TCM9-1, 9:1
(Mini-Circuits)
C12, C13 = 0.1 μF (0402)
T4 = ETC1-1-13, 1:1 (M/A-COM)
T1 = ETC1-1-13, 1:1 (M/A-COM)
R2 = open
07208-062
07208-061
ADL5382
Figure 62. Evaluation Board Top Layer Silkscreen
Figure 61. Evaluation Board Top Layer
Rev. 0 | Page 25 of 28
07208-063
07208-064
ADL5382
Figure 63. Evaluation Board Bottom Layer
Figure 64. Evaluation Board Bottom Layer Silkscreen
Rev. 0 | Page 26 of 28
ADL5382
OUTLINE DIMENSIONS
0.60 MAX
4.00
BSC SQ
PIN 1
INDICATOR
0.60 MAX
TOP
VIEW
0.50
BSC
3.75
BSC SQ
0.50
0.40
0.30
1.00
0.85
0.80
12° MAX
SEATING
PLANE
0.80 MAX
0.65 TYP
0.30
0.23
0.18
PIN 1
INDICATOR
19
18
24 1
*2.45
EXPOSED
PAD
2.30 SQ
2.15
(BOTTOMVIEW)
13
12
7
6
0.23 MIN
2.50 REF
0.05 MAX
0.02 NOM
0.20 REF
COPLANARITY
0.08
*COMPLIANT TO JEDEC STANDARDS MO-220-VGGD-2
EXCEPT FOR EXPOSED PAD DIMENSION
Figure 65. 24-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
4 mm × 4 mm Body, Very Thin Quad
(CP-24-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADL5382ACPZ-R7 1
ADL5382ACPZ-WP1
ADL5382-EVALZ1
1
Temperature Range
–40°C to +85°C
–40°C to +85°C
Package Description
24-Lead LFCSP_VQ, 7” Tape and Reel
24-Lead LFCSP_VQ, Waffle Pack
Evaluation Board
Z = RoHS Compliant Part.
Rev. 0 | Page 27 of 28
Package Option
CP-24-2
CP-24-2
Ordering Quantity
1,500
64
ADL5382
NOTES
©2008 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D07208-0-3/08(0)
T
T
Rev. 0 | Page 28 of 28