LINER LTC3873ETS8-5-TRPBF No rsensetm constant frequency current mode boost/flyback/sepic dc/dc controller Datasheet

LTC3873-5
No RSENSETM Constant
Frequency Current Mode
Boost/Flyback/SEPIC
DC/DC Controller
FEATURES
DESCRIPTION
n
The LTC®3873-5 is a constant frequency current mode
controller for boost, flyback or SEPIC DC/DC converters
designed to drive an N-channel MOSFET for high input
and output voltage converter applications.
n
n
n
n
n
n
n
n
n
VIN and VOUT Limited Only by External Components
Internal or Programmable External Soft-Start
Constant Frequency 200kHz Operation
Adjustable Current Limit
Current Sense Resistor Optional
±1.5% Voltage Reference Accuracy
Current Mode Operation for Excellent Line and Load
Transient Response
4.1V Undervoltage Threshold for Logic Level
MOSFET Applications
Low Quiescent Current: 300μA
Low Profile (1mm) ThinSOTTM and (0.75mm)
2mm × 3mm DFN Package
The LTC3873-5 is available in 8-lead ThinSOT and 2mm
× 3mm DFN packages.
APPLICATIONS
n
n
n
The LTC3873-5 provides ±1.5% output voltage accuracy
and consumes only 300μA quiescent current during normal operation and only 50μA during micropower start-up.
Using a 9.3V internal shunt regulator, the LTC3873-5 can
be powered from a high input voltage through a resistor
or it can be powered directly from a low impedance DC
voltage of 9V or less. Soft-start can be programmed using
an external capacitor.
PARAMETER
VCC UV+
VCC UV–
Telecom Power Supplies
Automotive Power Supplies
PoE Applications
LTC3873
8.4V
4V
LTC3873-5
4.1V
2.9V
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
TYPICAL APPLICATION
Efficiency and Power Loss vs Load Current
High Efficiency 5V Input, 12V Output Boost Converter
100
1nF
80
RUN/SS
11.8k
47pF
VIN
5V
VCC
LTC3873-5
10μH
10μF
IPRG
12k
GND
VFB
VOUT
12V
2A
SW
NGATE
22μF
70
POWER LOSS (mW)
ITH
EFFICIENCY (%)
4.7nF
10000
90
1000
60
50
40
100
30
20
108k
38735 TA01a
10
0
10
10
100
IOUT (mA)
1000
3873 TA01b
38735fb
1
LTC3873-5
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VCC to GND
Low Impedance Source ........................... –0.3V to 9V
Current Fed ..........................................25mA Into VCC
RUN/SS........................................................ –0.3V to 9V
IPRG Voltage.................................–0.3V to (VCC + 0.3V)
VFB, ITH Voltages ....................................... –0.3V to 2.4V
SW Voltage ................................................ –0.3V to 60V
Operating Temperature Range (Note 2)
LTC3873E-5 ......................................... –40°C to 85°C
LTC3873I-5 ........................................ –40°C to 125°C
Junction Temperature (Note 3) ............................. 125°C
Storage Temperature Range................... –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
TS8 Package ..................................................... 300°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
IPRG 1
ITH 2
VFB 3
GND 4
8 SW
7 RUN/SS
6 VCC
5 NGATE
8
VFB 2
7
VCC
6
RUN/SS
5
SW
ITH 3
9
IPRG 4
TS8 PACKAGE
8-LEAD PLASTIC TSOT-23
NGATE
GND 1
DDB PACKAGE
8-LEAD (3mm s 2mm) PLASTIC DFN
TJMAX = 125°C, θJA = 230°C/W
TJMAX = 125°C, θJA = 76°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3873ETS8-5#PBF
LTC3873ETS8-5#TRPBF
LTCSP
8-Lead Plastic TSOT-23
–40°C to 85°C
LTC3873ITS8-5#PBF
LTC3873ITS8-5#TRPBF
LTCSP
8-Lead Plastic TSOT-23
–40°C to 125°C
LTC3873EDDB-5#PBF
LTC3873EDDB-5#TRPBF
LCSM
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LTC3873IDDB-5#PBF
LTC3873IDDB-5#TRPBF
LCSM
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 125°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3873ETS8-5
LTC3873ETS8-5#TR
LTCSP
8-Lead Plastic TSOT-23
–40°C to 85°C
LTC3873ITS8-5
LTC3873ITS8-5#TR
LTCSP
8-Lead Plastic TSOT-23
–40°C to 125°C
LTC3873EDDB-5
LTC3873EDDB-5#TR
LCSM
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LTC3873IDDB-5
LTC3873IDDB-5#TR
LCSM
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
38735fb
2
LTC3873-5
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = 5V unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
Input DC Supply Current
Normal Operation
Shutdown
UVLO
Typicals at VCC = 5V (Note 4)
VITH = 1.9V
VRUN/SS = 0V
VCC = UVLO Threshold – 100mV, VRUN/SS = VCC
Undervoltage Lockout Threshold
VCC Rising
VCC Falling
VCC Hysteresis
l
l
l
Shutdown Threshold (at RUN/SS)
VRUN/SS Falling
VRUN/SS Rising
Regulated Feedback Voltage
(Note 5)
Feedback Voltage Line Regulation
3.5V < VCC < 9V (Note 5)
Feedback Voltage Load Regulation
VITH = 1.6V (Note 5)
VITH = 1V (Note 5)
VFB Input Current
(Note 5)
RUN/SS Pull Up Current
VRUN/SS = 0V
VRUN/SS = 1.3V
MIN
TYP
MAX
UNITS
300
50
35
400
80
50
μA
μA
μA
3.8
2.5
0.9
4.1
2.9
1.25
4.4
3.3
1.7
V
V
V
l
0.5
0.6
0.7
0.8
0.9
1.0
V
V
l
1.182
1.2
1.218
V
0.1
mV/V
0.05
–0.05
Maximum Duty Cycle
%
%
25
50
nA
1.5
5
3
15
4.5
25
μA
μA
70
78
84
%
ISLMAX, Peak Slope Compensation Current
20
Oscillator Frequency
160
200
μA
240
Gate Drive Rise Time
CLOAD = 3000pF (Note 6)
Gate Drive Fall Time
CLOAD = 3000pF (Note 6)
Peak Current Sense Voltage
IPRG = GND
IPRG = Float
IPRG = VIN
l
l
l
95
165
265
110
185
295
125
210
325
mV
mV
mV
VIN Shunt Regulator Voltage
IIN = 1mA, IIN = 25mA, VRUN/SS = 0V
l
9
9.3
9.6
V
Default Internal Soft-Start
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3873E-5 is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LTC3873I-5 is guaranteed to meet
performance specifications over the full –40°C to 125°C operating
temperature range.
40
kHz
ns
40
3.3
ns
ms
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • θJA)
Note 4: The dynamic input supply current is higher due to power MOSFET
gate charging (QG • fOSC). See Applications Information.
Note 5: The LTC3873-5 is tested in a feedback loop which servos VFB to
the reference voltage with the ITH pin forced to the midpoint of its voltage
range (0.7V ≤ VITH ≤ 1.9V, midpoint = 1.3V).
Note 6: Rise and fall times are measured at 10% and 90% levels.
VCC = 5.6V.
38735fb
3
LTC3873-5
TYPICAL PERFORMANCE CHARACTERISTICS
Regulated Feedback Voltage vs
Temperature
Regulated Feedback Voltage Line
Regulation
1.25
1.2025
1.24
1.2020
1.23
1.2015
ITH Voltage vs RUN/SS Voltage
2.5
VIN = 5V
1.22
1.21
ITH VOLTAGE (V)
VFB VOLTAGE (V)
VFB VOLTAGE (V)
2.0
1.2010
1.2005
1.20
1.2000
1.19
1.1995
1.5
1.0
0.5
1.18
–60 –40 –20
0
20
40 60 80 100 120
TEMPERATURE (°C)
1.1990
3
4
5
7
6
VIN (V)
38735 G01
70
Gate Drive Rise and Fall Time
vs CLOAD
100
80
65
90
70
50
45
40
35
30
80
60
70
50
TIME (ns)
SHUTDOWN MODE IQ (μA)
55
40
30
20
3
4
7
6
VIN (V)
5
8
9
10
RISE TIME
60
FALL TIME
50
40
30
20
20
10
25
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
RUN/SS VOLTAGE (V)
38735 G03
Shutdown IQ vs Temperature
60
0
38735 G02
Shutdown Mode IQ vs VIN
SHUTDOWN MODE IQ (μA)
0
10
9
8
10
0
0
–60 –40 –20 0 20 40 60 80 100 120
TEMPERATURE (°C)
3873-5 G04
0
2000
6000
4000
CLOAD (pF)
8000
10000
38735 G06
3873-5 G05
Shunt Regulator Voltage
vs ISHUNT
RUN Threshold vs Temperature
1.0
10.2
10.1
0.8
0.7
REGULATION VOLTAGE (V)
RUN THRESHOLDS (V)
0.9
RISING
FALLING
0.6
10.0
9.9
9.8
9.7
9.6
9.5
9.4
9.3
0.5
–60 –40 –20 0 20 40 60 80 100 120
TEMPERATURE (°C)
38735 G07
9.2
0
5
10
15
20 25 30
ISHUNT (mA)
35
40
45
3873-5 G08
38735fb
4
LTC3873-5
TYPICAL PERFORMANCE CHARACTERISTICS
Maximum Sense Threshold
vs Temperature
Frequency vs Temperature
300
VIN = 5V
MAXIMUM SENSE THRESHOLD (mV)
250
FREQUECY (kHz)
230
210
190
170
150
–60 –40 –20 0 20 40 60 80 100 120
TEMPERATURE (°C)
3873-5 G09
PIN FUNCTIONS
IPRG = VIN
250
200
IPRG = FLOAT
150
IPRG = GND
100
50
0
–60 –40 –20 0 20 40 60 80 100 120
TEMPERATURE (°C)
38735 G10
(TS8/DD8)
IPRG (Pin 1/Pin 4): Current Sense Limit Select Pin.
ITH (Pin 2/Pin 3): This pin serves as the error amplifier
compensation point. Nominal voltage range for this pin
is 0.7V to 1.9V.
VFB (Pin 3/Pin 2): This pin receives the feedback voltage
from an external resistor divider across the output.
GND (Pin 4/Pin 1): Ground Pin.
NGATE (Pin 5/Pin 8): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to VIN.
RUN/SS (Pin 7/Pin 6): Shutdown and External Soft-Start
Pin. In shutdown, all functions are disabled and the NGATE
pin is held low.
SW (Pin 8/Pin 5): Switch node connection to inductor and
current sense input pin through external slope compensation resistor. Normally, the external N-channel MOSFET’s
drain is connected to this pin.
Exposed Pad (NA/Pin 9): Ground. Must be soldered to PCB
for electrical contact and rated thermal performance.
VCC (Pin 6/Pin 7): Supply Pin. This pin must be closely
decoupled to GND (Pin 4).
38735fb
5
LTC3873-5
FUNCTIONAL DIAGRAM
GND
VCC
UV
UNDERVOLTAGE
LOCKOUT
VOLTAGE
REFERENCE
1.2V
SW
VCC
SHUNT
REGULATOR
SLOPE
COMPENSATION
SHUTDOWN
COMPARATOR
–
+
IPRG
CURRENT
COMPARATOR
3μA
ILIM
+
SHDN
ITH
BUFFER
RS
LATCH
–
RUN/SS
R
S
Q
CURRENT LIMIT
CLAMP
VIN
SWITCHING
LOGIC CIRCUIT
INTERNAL
SOFT-START
RAMP
NGATE
VFB
–
ERROR
AMPLIFIER
200kHz
OSCILLATOR AND
MAX DUTY CYCLE
1.2V
+
ITH
38735 FD
38735fb
6
LTC3873-5
OPERATION
Main Control Loop
The LTC3873-5 is a general purpose N-channel switching
DC/DC converter for boost, flyback and SEPIC applications.
Its No RSENSE sensing technique improves efficiency,
increases power density and reduces the cost of the
overall solution.
For circuit operation, please refer to the Functional Diagram
of the IC and the Typical Application on the front page.
During normal operation, the power MOSFET is turned
on when the oscillator sets the PWM latch and is turned
off when the current comparator resets the latch. The
divided-down output voltage is compared to an internal
1.2V reference by the error amplifier, which outputs an
error signal at the ITH pin. The voltage on the ITH pin
sets the current comparator input threshold. When the
load current increases, a fall in the VFB voltage relative to
the reference voltage causes the ITH pin to rise, causing
the current comparator to trip at a higher peak inductor
current value. The average inductor current will therefore
rise until it equals the load current, thereby maintaining
output regulation.
D
L
VIN
VOUT
VCC
+
SW
COUT
VSW
LTC3873-5
The LTC3873-5 can be used either by sensing the voltage
drop across the power MOSFET or by connecting the SW
pin to a conventional sensing resistor in the source of the
power MOSFET. Sensing the voltage across the power
MOSFET maximizes converter efficiency and minimizes the
component count; the maximum rating for this pin, 60V,
allows MOSFET sensing in a wide output voltage range.
Shunt Regulator
A built-in shunt regulator from the VCC pin to GND limits
the voltage on the VCC pin to approximately 9.3V as long as
the shunt regulator is not forced to sink more than 25mA.
The shunt regulator permits the use of a wide variety of
powering schemes that exceed the LTC3873-5’s absolute
maximum ratings. Further details on powering schemes
are described in the Application Information section.
Start-Up/Shutdown
The LTC3873-5 has two shutdown mechanisms to disable
and enable operation: an undervoltage lockout on the
VCC supply pin voltage and a threshold RUN/SS pin. The
LTC3873-5 transitions into and out of shutdown according
to the state diagram shown in Figure 3.
The undervoltage lockout (UVLO) mechanism prevents
the LTC3873-5 from trying to drive a MOSFET with insufficient VGS. The voltage at the VCC pin must exceed
NGATE
GND
GND
38735 F01
LTC3873-5
SHUT DOWN
Figure 1. SW Pin (Internal Sense Pin)
Connection for Maximum Efficiency
D
L
VIN
VOUT
VSW
VCC
VRUN/SS < VSHDN
(NOMINALLY 0.7V)
VRUN/SS > VSHDN
AND VIN > VTURNON
(NOMINALLY 4.1V)
+
NGATE
LTC3873-5
VIN < VTURNOFF
(NOMINALLY 2.9V)
COUT
RSL
SW
GND
RSENSE
LTC3873-5
ENABLED
GND
38735 F02
Figure 2. SW Pin (Internal Sense Pin)
Connection for Sensing Resistor
38735 F03
Figure 3. Start-Up/Shut Down State Diagram
38735fb
7
LTC3873-5
OPERATION
Soft-Start
Leave the RUN/SS pin open to use the internal 3.3ms
soft-start. During the internal soft-start, a voltage ramp
limits the VITH. 3.3ms is required for ITH to ramp from
zero current level to full current level. The soft-start can
be lengthened by placing an external capacitor from the
RUN/SS pin to the GND. A 3μA current will charge the
capacitor, pulling the RUN/SS pin above the shutdown
threshold and a 15μA pull-up current will continue to ramp
RUN/SS to limit VITH during the start-up. When RUN/SS
is driven by an external logic, a minimum of 2.75V logic
is recommended to allow the maximum ITH range.
Light Load Operation
Under very light load current conditions, the ITH pin voltage
will be very close to 0.85V. As the load current decreases
further, an internal offset at the current comparator input
will assure that the current comparator remains tripped
(even at zero load current) and the regulator will start to
skip cycles in order to maintain regulation. This behavior
allows the regulator to maintain constant frequency down
to very light loads, resulting in low output ripple as well
as low audible noise and reduced RF interference while
providing high light load efficiency.
Current Sense
During the switch on-time, the control circuit limits the
maximum voltage drop across the current sense component to about 295mV, 110mV and 185mV at low duty
cycle with IPRG tied to VIN, GND or left floating respectively. It is reduced with increasing duty cycle as shown
in Figure 4.
MAXIMUM CURRENT SENSE VOLTAGE (mV)
VTURNON (nominally 4.1V) at least momentarily to enable
LTC3873-5 operation. The VCC voltage is then allowed to fall
to VTURNOFF (nominally 2.9V) before undervoltage lockout
disables the LTC3873-5. The RUN/SS pin can be driven
below VSHDN (nominally 0.7V) to force the LTC3873-5 into
shutdown. When the chip is off, the input supply current is
typically only 50μA. Keep in mind that VCC should exceed
the gate threshold voltage of the switching MOSFET for
safe operation.
300
250
IPRG = HIGH
200
IPRG = FLOAT
150
IPRG = LOW
100
50
0
1
20
40
60
DUTY CYCLE (%)
80
100
3873-5 F04
Figure 4. Maximum SENSE Threshold Voltage vs Duty Cycle
38735fb
8
LTC3873-5
APPLICATIONS INFORMATION
VCC Bias Power
The VCC pin must be bypassed to the GND pin with a
minimum 10μF ceramic or tantalum capacitor located
immediately adjacent to the two pins. Proper supply bypassing is necessary to supply the high transient currents
required by the MOSFET gate driver.
For maximum flexibility, the LTC3873-5 is designed so
that it can be operated from voltages well beyond the
LTC3873-5’s absolute maximum ratings. In the simplest
case, the LTC3873-5 can be powered with a resistor connected between the input voltage and VCC. The built-in shunt
regulator limits the voltage on the VCC pin to around 9.3V
as long as the shunt regulator is not forced to sink more
than 25mA. This powering scheme has the drawback that
the power loss in the resistor reduces converter efficiency
and the 25mA shunt regulator maximum may limit the
maximum-minimum range of input voltage.
The circuit in Figure 5 shows a second way to power the
LTC3873-5. An external series pre-regulator consisting of
series pass transistor Q1, zener diode D1 and bias resistor RB brings VCC to at least 7.6V nominal, well above the
undervoltage lockout threshold.
Duty Cycle – 6%
20μA • RSLOPE
80%
Note the external programmable slope compensation is
only needed when the internal slope compensation is not
sufficient. In some applications RSL can be shorted. For
the LTC3873-5, when the RDS(ON) sensing technique is
used, the ringing on the SW pin disrupts the tiny slope
compensation current out of the pin. It is not recommended
to add external slope compensation in this case.
Output Voltage Programming
The output voltage is set by a resistor divider according
to the following formula:
LTC3873-5
Q1
VCC
D1
8.2V
ΔVSENSE =
⎛ R2⎞
VO = 1.2V • ⎜1+ ⎟
⎝ R1⎠
VIN
RB
resistor (RSL) connecting the SW pin to the current sense
resistor (RSENSE) thus develops a ramping voltage drop.
From the perspective of the SW pin, this ramping voltage
adds to the voltage across the sense resistor, effectively
reducing the current comparator threshold in proportion
to duty cycle. The amount of reduction in the current
comparator threshold (ΔVSENSE) can be calculated using
the following equation:
CVCC
0.1μF
GND
38735 F05
Figure 5. External Pre-Regulator for VCC Bias Power
Slope Compensation
The LTC3873-5 has built-in internal slope compensation
to stabilize the control loop against sub-harmonic oscillation. It also provides the ability to externally increase slope
compensation by injecting a ramping current out of its
SW pin into an external slope compensation resistor (RSL
in Figure 2). This current ramp starts at zero right after
the NGATE pin has been high. The current rises linearly
towards a peak of 20μA at the maximum duty cycle of
80%, shutting off once the NGATE pin goes low. A series
The external resistor divider is connected to the output
as shown in Figure 4, allowing remote voltage sensing.
Choose resistance values for R1 and R2 to be as large as
possible in order to minimize any efficiency loss due to
the static current drawn from VOUT, but just small enough
so that when VOUT is in regulation, the error caused by
the nonzero input current to the VFB pin is less than 1%.
A good rule of thumb is to choose R1 to be 24k or less.
Transformer Design Considerations
Transformer specification and design is perhaps the most
critical part of applying the LTC3873-5 successfully. In
addition to the usual list of caveats dealing with high frequency power transformer design, the following should
prove useful.
38735fb
9
LTC3873-5
APPLICATIONS INFORMATION
Turns Ratios
Due to the use of the external feedback resistor divider
ratio to set output voltage, the user has relative freedom
in selecting a transformer turns ratio to suit a given application. Simple ratios of small integers, e.g., 1:1, 2:1, 3:2,
etc. can be employed which yield more freedom in setting
total turns and mutual inductance. Simple integer turns
ratios also facilitate the use of “off-the-shelf” configurable
transformers such as the Coiltronics VERSA-PAC series
in applications with high input to output voltage ratios.
For example, if a 6-winding VERSA-PAC is used with three
windings in series on the primary and three windings in
parallel on the secondary, a 3:1 turns ratio will be achieved.
Turns ratio can be chosen on the basis of desired duty
cycle. However, remember that the input supply voltage
plus the secondary-to-primary referred version of the
flyback pulse (including leakage spike) must not exceed
the allowed external MOSFET breakdown rating.
Leakage Inductance
Transformer leakage inductance (on either the primary
or secondary) causes a voltage spike to occur after the
output switch (Q1) turn-off. This is increasingly prominent
at higher load currents where more stored energy must
be dissipated. In some cases a “snubber” circuit will be
required to avoid overvoltage breakdown at the MOSFET’s
drain node. Application Note 19 is a good reference on
snubber design. A bifilar or similar winding technique is a
good way to minimize troublesome leakage inductances.
However, remember that this will limit the primary-tosecondary breakdown voltage, so bifilar winding is not
always practical.
Power MOSFET Selection
The power MOSFET serves two purposes in the LTC3873-5:
it represents the main switching element in the power path
and its RDS(ON) represents the current sensing element
for the control loop. Important parameters for the power
MOSFET include the drain-to-source breakdown voltage
(BVDSS), the threshold voltage (VGS(TH)), the on-resistance
(RDS(ON)) versus gate-to-source voltage, the gate-to-source
and gate-to-drain charges (QGS and QGD, respectively),
the maximum drain current (ID(MAX)) and the MOSFET’s
thermal resistances (RTH(JC) and RTH(JA)).
Current Limit Programming
During the switch on-time, the control circuit limits the
maximum voltage drop across the current sense component to about 270mV, 100mV and 170mV at low duty
cycle with IPRG tied to VIN, GND or left floating respectively. For boost applications with RDS(ON) sensing, refer
to the LTC3872 data sheet for the selection of MOSFET
RDS(ON).
MOSFETs have conduction losses (I2R) and switching
losses. For VDS < 20V, high current efficiency generally
improves with large MOSFETs with low RDS(ON), while
for VDS > 20V the transition losses rapidly increase to the
point that the use of a higher RDS(ON) device with lower
reverse transfer capacitance, CRSS, actually provides
higher efficiency.
Output Capacitors
The output capacitor is normally chosen by its effective
series resistance (ESR), which determines output ripple
voltage and affects efficiency. Low ESR ceramic capacitors are often used to minimize the output ripple. Boost
regulators have large RMS ripple current in the output
capacitor that must be rated to handle the current. The
output ripple current (RMS) is:
IRMS(COUT ) ≈ IOUT(MAX ) •
VOUT – VIN(MIN)
VIN(MIN)
Output ripple is then simply:
VOUT = RESR(ΔIL(RMS))
The output capacitor for flyback converter should have a
ripple current rating greater than:
IRMS = IOUT •
DMAX
1 – DMAX
38735fb
10
LTC3873-5
APPLICATIONS INFORMATION
Input Capacitors
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular, and
does not contain large square wave currents as found in
the output capacitor. The input voltage source impedance
determines the size of the capacitor that is typically 10μF to
100μF. A low ESR is recommended although not as critical
as the output capacitor can be on the order of 0.3Ω.
The RMS input ripple current for a boost converter is:
IRMS(CIN) = 0.3 •
VIN(MIN)
L•f
• DMAX
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to the
input of the converter and solid tantalum capacitors can
fail catastrophically under these conditions.
In a flyback converter, the input flows in pulses placing
severe demands on the input capacitors. Select an input
capacitor with a ripple current rating greater than:
IRMS =
PIN
VIN(MIN)
1 – DMAX
DMAX
Duty Cycle Considerations
The LTC3873-5 imposes a maximum duty cycle limit of
80% typical. For a flyback converter, the maximum duty
cycle prevents the transformer core from saturation. In
a boost converter application, however, it sets a limit on
the maximum step-up ratio or maximum output voltage
with the given input voltage of:
VOUT(MAX ) =
VIN(MIN)
1 – 0.8%
– VD
Current and voltage stress on the power switch and
synchronous rectifiers, input and output capacitor RMS
currents and transformer utilization (size vs power) are
impacted by duty factor. Unfortunately duty factor cannot be adjusted to simultaneously optimize all of these
requirements. In general, avoid extreme duty factors since
this severely impacts the current stress on most of the
components. A reasonable target for duty factor is 50% at
nominal input voltage. Using this rule of thumb, the ideal
transformer turns ratio is:
NIDEAL =
VOUT 1 – D VOUT
•
=
VIN
D
VIN
Output Diode Selection
To maximize efficiency, a fast switching diode with low
forward drop and low reverse leakage is desired. The output
diode in a boost converter conducts current during the
switch off-time. The peak reverse voltage that the diode
must withstand is equal to the regulator output voltage.
The average forward current in normal operation is equal
to the output current, and the peak current is equal to the
peak inductor current.
38735fb
11
LTC3873-5
TYPICAL APPLICATIONS
1W Isolated Housekeeping Telecom Converter
BAS516
PRIMARY SIDE
10V, 100mA
OUTPUT
T1
•
2.2μF
VIN
36V TO 75V
1μF
BAS516
•
221k
2.2μF
100Ω
BAS516
8.8k
•
8.2V
1nF
SECONDARY
SIDE GROUND
LTC3873-5
NGATE
ITH
22k
FDC2512
VCC
GND
10nF
SECONDARY SIDE
10V, 100mA
OUTPUT
T1: PULSE ENGINEERING PA0648
OR TYCO TTI8698
RUN/SS
1.2k
VFB
SW
IPRG
10μF
0.1Ω
PRIMARY GROUND
38735 TA04
9V to 15V VIN, 12V VOUT SEPIC Converter
T1
4.56μH
BH510-1009
BH ELECTRONICS
1
4
VIN
9V TO 15V
10μF
s3
+
100μF
20V 2
•
•
3
10μF
25V
UPS840
301Ω
+
LTC3873-5
100k
1
2
3
10nF
4
33.2k
11k
IPRG
ITH
SW
RUN/SS
VFB = 1.2V VCC
GND
NGATE
Si4840
8
47μF
16V
s3
10μF
16V
VOUT
12V
2A
7
6
5
4.7μF
0.1μF
3873-5 TA04
38735fb
12
LTC3873-5
TYPICAL APPLICATIONS
–10V to –15V VIN, –5V VOUT Negative-to-Negative (Negative Buck) Converter
J1
GND
C4
10μF
25V
R1
301Ω
1
SW
IPRG
D1
L1
6.4μH
8
J3
GND
COUT
100μF
6.3V
s3
R4
4.99k J4
–VOUT
–5V
3A
LTC3873-5
2
3
R3
15k
J2
–VIN
–10V TO –15V
C3
4.7nF
R2
1.2k
4
ITH
VFB = 1.2V
GND
RUN/SS
VCC
NGATE
7
6
6
5
5
4
Q2
BC857
Q1
C1
1μF
C2
0.1μF
R5
10k
1
2
3
3873-3 TA05
38735fb
13
LTC3873-5
PACKAGE DESCRIPTION
TS8 Package
8-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1637)
0.52
MAX
2.90 BSC
(NOTE 4)
0.65
REF
1.22 REF
1.4 MIN
3.85 MAX 2.62 REF
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.22 – 0.36
8 PLCS (NOTE 3)
0.65 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
1.95 BSC
TS8 TSOT-23 0802
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
38735fb
14
LTC3873-5
PACKAGE DESCRIPTION
DDB Package
8-Lead Plastic DFN (3mm × 2mm)
(Reference LTC DWG # 05-08-1702 Rev B)
0.61 p0.05
(2 SIDES)
0.70 p0.05
2.55 p0.05
1.15 p0.05
PACKAGE
OUTLINE
0.25 p 0.05
0.50 BSC
2.20 p0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
3.00 p0.10
(2 SIDES)
R = 0.115
TYP
5
R = 0.05
TYP
0.40 p 0.10
8
2.00 p0.10
(2 SIDES)
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
0.56 p 0.05
(2 SIDES)
0.200 REF
0.75 p0.05
0 – 0.05
4
0.25 p 0.05
1
PIN 1
R = 0.20 OR
0.25 s 45o
CHAMFER
(DDB8) DFN 0905 REV B
0.50 BSC
2.15 p0.05
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING CONFORMS TO VERSION (WECD-1) IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
38735fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3873-5
TYPICAL APPLICATION
5V Output Nonisolated Telecom Housekeeping Power Supply
VIN
36V TO 72V
4.7μF
100V
X5R
UPS840
221k
•
T1
D1
5.1V
•
300μF
6.3V
X5R
s3
VOUT
5V
2A MAX
10μF
56k
0.1μF
2.2nF
VCC
ITH
NGATE
LTC3873-5
RUN/SS
GND
SW
IPRG
VFB
12k
FDC2512
68mΩ
36k
38735 TA03
D1: MBR0540T1
T1: COOPER CTX02-15242
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT 1619
Current Mode PWM Controller
300kHz Fixed Frequency, Boost, SEPIC, Flyback Topology
LTC1624
Current Mode DC/DC Controller
SO-8; 300kHz Operating Frequency; Buck, Boost, SEPIC Design;
VIN Up to 36V
LTC1700
No RSENSE Synchronous Step-Up Controller
Up to 95% Efficiency, Operating as Low as 0.9V Input
®
LTC1871-7
Wide Input Range Controller
No RSENSE, 7V Gate Drive, Current Mode Control
LTC1872/LTC1872B
SOT-23 Boost Controller
Delievers Up to 5A, 550kHz Fixed Frequency, Current Mode
LT1930
1.2MHz, SOT-23 Boost Converter
Up to 34V Output, 2.6V VIN 16V, Miniature Design
LT1931
Inverting 1.2MHz, SOT-23 Converter
Positive-to Negative DC/DC Conversion, Miniature Design
LTC3401/LTC3402
1A/2A 3MHz Synchronous Boost Converters
Up to 97% Efficiency, Very Small Solution, 0.5V ≤ VIN ≤ 5V
LTC3704
Positive-to Negative DC/DC Controller
No RSENSE, Current Mode Control, 50kHz to 1MHz
LTC1871/LTC1871-7
No RSENSE, Wide Input Range DC/DC Boost Controller
No RSENSE, Current Mode Control, 2.5V ≤ VIN ≤ 36V
LTC3703/LTC3703-5
100V Synchronous Controller
Step-Up or Step Down, 600kHz, SSOP-16, SSOP-28
LTC3803/LTC3803-5
200kHz Flyback DC/DC Controller
VIN and VOUT Limited Only by External Components
LTC3805
Adjustable Frequency Flyback Controller
VIN and VOUT Limited Only by External Components
LT3825
Isolated No-Opto Synchronous Flyback Controller
VIN: 24V to 75V, Up to 80W, Current Mode Control
LT3837
Isolated No-Opto Synchronous Flyback Controller
VIN: 4.5V to 20V, Up to 60W, Current Mode Control
LTC3873
No RSENSE Constant Frequency Boost/Flyback/SEPIC
Controller
VIN and VOUT Limited Only by External Components, 200kHz Frequency,
ThinSOT or DFN Package
38735fb
16 Linear Technology Corporation
LT 0908 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2007
Similar pages