Intersil ADC0803LD 8-bit, microprocessor- compatible, a/d converter Datasheet

ADC0802, ADC0803
ADC0804
8-Bit, MicroprocessorCompatible, A/D Converters
August 1997
Features
Description
• 80C48 and 80C80/85 Bus Compatible - No Interfacing
Logic Required
The ADC0802 family are CMOS 8-Bit, successive-approximation A/D converters which use a modified potentiometric
ladder and are designed to operate with the 8080A control
bus via three-state outputs. These converters appear to the
processor as memory locations or I/O ports, and hence no
interfacing logic is required.
• Conversion Time < 100µs
• Easy Interface to Most Microprocessors
• Will Operate in a “Stand Alone” Mode
• Differential Analog Voltage Inputs
The differential analog voltage input has good commonmode-rejection and permits offsetting the analog zero-inputvoltage value. In addition, the voltage reference input can be
adjusted to allow encoding any smaller analog voltage span
to the full 8 bits of resolution.
• Works with Bandgap Voltage References
• TTL Compatible Inputs and Outputs
• On-Chip Clock Generator
• 0V to 5V Analog Voltage Input Range (Single + 5V Supply)
• No Zero-Adjust Required
Ordering Information
PART NUMBER
ERROR
ADC0802LCN
±1/2 LSB
ADC0802LCD
±3/4 LSB
±1 LSB
ADC0802LD
ADC0803LCN
±1/2 LSB
ADC0803LCD
±3/4 LSB
TEMP. RANGE (oC)
EXTERNAL CONDITIONS
VREF/2 = 2.500VDC (No Adjustments)
0 to 70
PACKAGE
PKG. NO
20 Ld PDIP
E20.3
-40 to 85
20 Ld CERDIP
F20.3
-55 to 125
20 Ld CERDIP
F20.3
20 Ld PDIP
E20.3
-40 to 85
20 Ld CERDIP
F20.3
VREF/2 Adjusted for Correct Full Scale
Reading
0 to 70
ADC0803LCWM
±1 LSB
-40 to 85
20 Ld SOIC
M20.3
ADC0803LD
±1 LSB
-55 to 125
20 Ld CERDIP
F20.3
ADC0804LCN
±1 LSB
20 Ld PDIP
E20.3
ADC0804LCD
±1 LSB
-40 to 85
20 Ld CERDIP
F20.3
ADC0804LCWM
±1 LSB
-40 to 85
20 Ld SOIC
M20.3
VREF/2 = 2.500VDC (No Adjustments)
Pinout
0 to 70
Typical Application Schematic
ADC0802, ADC0803, ADC0804
(PDIP, CERDIP)
TOP VIEW
1
CS
2
RD
V+ 20
CLK R 19
3
WR
CLK IN
4
5
INTR
11
DB7
DB4
VIN (+)
6
DB3
VIN (-)
1
20 V+ OR VREF
RD
2
19 CLK R
WR
3
18 DB0 (LSB)
CLK IN
4
17 DB1
INTR
5
16 DB2
15
VIN (+)
6
15 DB3
16
VIN (-)
7
14 DB4
17
AGND
8
13 DB5
18
VREF/2
9
12 DB6
DGND 10
ANY
µPROCESSOR
µP BUS
CS
12
13
14
+5V
150pF
10K
DB6
DB5
DB2
DB1
DB0
7
AGND 8
VREF/2 9
DGND 10
DIFF
INPUTS
8-BIT RESOLUTION
OVER ANY
DESIRED
ANALOG INPUT
VOLTAGE RANGE
VREF/2
11 DB7 (MSB)
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 | Copyright © Intersil Corporation 1999
6-5
File Number
3094.1
ADC0802, ADC0803, ADC0804
Functional Diagram
RD
CS
WR
2
READ
1
3
SET
“1” = RESET SHIFT REGISTER
“0” = BUSY AND RESET STATE
Q
RESET
INPUT PROTECTION
FOR ALL LOGIC INPUTS
CLK R
19
CLK
INPUT
CLK A
CLK IN
TO INTERNAL
CIRCUITS
G1
RESET
4
CLK OSC
D
BV = 30V
CLK
GEN CLKS
DFF1
Q
START F/F
10
DGND
START
CONVERSION
CLK B
MSB
V+
(VREF)
VREF/2
D
20
LADDER
AND
DECODER
SUCCESSIVE
APPROX.
REGISTER
AND LATCH
9
8-BIT
SHIFT
REGISTER
IF RESET = “0”
R
RESET
AGND
DAC
VOUT
8
LSB
INTR F/F
Q
CLK A
V+
D
VIN (+)
VIN (-)
6
+
∑
-
DFF2
COMP
Q
+
-
Q
XFER
THREE-STATE
OUTPUT LATCHES
7
G2
SET
5
LSB
MSB
CONV. COMPL.
11 12 13 14 15 16 17 18
8 X 1/f
DIGITAL OUTPUTS
THREE-STATE CONTROL
“1” = OUTPUT ENABLE
6-6
INTR
ADC0802, ADC0803, ADC0804
Absolute Maximum Ratings
Thermal Information
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.5V
Voltage at Any Input . . . . . . . . . . . . . . . . . . . . . . -0.3V to (V+ +0.3V)
Thermal Resistance (Typical, Note 1)
θJA (oC/W) θJC (oC/W)
PDIP Package . . . . . . . . . . . . . . . . . . . . .
125
N/A
CERDIP Package . . . . . . . . . . . . . . . . . .
80
20
SOIC Package . . . . . . . . . . . . . . . . . . . . .
120
N/A
Maximum Junction Temperature
Hermetic Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175oC
Plastic Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150oC
Maximum Storage Temperature Range . . . . . . . . . .-65oC to 150oC
Maximum Lead Temperature (Soldering, 10s) . . . . . . . . . . . . 300oC
(SOIC - Lead Tips Only)
Operating Conditions
Temperature Range
ADC0802/03LD. . . . . . . . . . . . . . . . . . . . . . . . . . . -55oC to 125oC
ADC0802/03/04LCD . . . . . . . . . . . . . . . . . . . . . . . . -40oC to 85oC
ADC0802/03/04LCN . . . . . . . . . . . . . . . . . . . . . . . . . .0oC to 70oC
ADC0803/04LCWM . . . . . . . . . . . . . . . . . . . . . . . . -40oC to 85oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation
of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on an evaluation PC board in free air.
Electrical Specifications
PARAMETER
(Notes 1, 7)
TEST CONDITIONS
MIN
TYP
MAX
UNITS
CONVERTER SPECIFICATIONS V+ = 5V, TA = 25oC and fCLK = 640kHz, Unless Otherwise Specified
Total Unadjusted Error
ADC0802
VREF/2 = 2.500V
-
-
±1/2
LSB
ADC0803
VREF/2 Adjusted for Correct Full
Scale Reading
-
-
±1/2
LSB
ADC0804
VREF/2 = 2.500V
-
-
±1
LSB
1.0
1.3
-
kΩ
VREF/2 Input Resistance
Input Resistance at Pin 9
Analog Input Voltage Range
(Note 2)
GND-0.05
-
(V+) + 0.05
V
DC Common-Mode Rejection
Over Analog Input Voltage Range
-
±1/16
±1/8
LSB
Power Supply Sensitivity
V+ = 5V ±10% Over Allowed Input
Voltage Range
-
±1/16
±1/8
LSB
-
±1/2
LSB
LSB
CONVERTER SPECIFICATIONS V+ = 5V, 0oC to 70oC and fCLK = 640kHz, Unless Otherwise Specified
Total Unadjusted Error
ADC0802
VREF/2 = 2.500V
-
ADC0803
VREF/2 Adjusted for Correct Full
Scale Reading
-
-
±1/
ADC0804
VREF/2 = 2.500V
-
-
±1
LSB
VREF/2 Input Resistance
Input Resistance at Pin 9
Analog Input Voltage Range
(Note 2)
2
1.0
1.3
-
kΩ
GND-0.05
-
(V+) + 0.05
V
±1/8
±1/16
±1/4
±1/8
LSB
-
±3/4
LSB
-
±3/
LSB
-
-
±1
LSB
1.0
1.3
-
kΩ
DC Common-Mode Rejection
Over Analog Input Voltage Range
-
Power Supply Sensitivity
V+ = 5V ±10% Over Allowed Input
Voltage Range
-
LSB
CONVERTER SPECIFICATIONS V+ = 5V, -25oC to 85oC and fCLK = 640kHz, Unless Otherwise Specified
Total Unadjusted Error
ADC0802
VREF/2 = 2.500V
-
ADC0803
VREF/2 Adjusted for Correct Full
Scale Reading
ADC0804
VREF/2 = 2.500V
-
4
VREF/2 Input Resistance
Input Resistance at Pin 9
Analog Input Voltage Range
(Note 2)
GND-0.05
-
(V+) + 0.05
V
DC Common-Mode Rejection
Over Analog Input Voltage Range
-
±1/8
±1/4
LSB
Power Supply Sensitivity
V+ = 5V ±10% Over Allowed Input
Voltage Range
-
±1/16
±1/8
LSB
6-7
ADC0802, ADC0803, ADC0804
Electrical Specifications
PARAMETER
(Notes 1, 7) (Continued)
TEST CONDITIONS
MIN
TYP
MAX
UNITS
CONVERTER SPECIFICATIONS V+ = 5V, -55oC to 125oC and fCLK = 640kHz, Unless Otherwise Specified
Total Unadjusted Error
ADC0802
VREF/2 = 2.500V
-
-
±1
LSB
ADC0803
VREF/2 Adjusted for Correct Full
Scale Reading
-
-
±1
LSB
VREF/2 Input Resistance
Input Resistance at Pin 9
Analog Input Voltage Range
(Note 2)
1.0
1.3
-
kΩ
GND-0.05
-
(V+) + 0.05
V
±1/8
±1/8
±1/4
±1/4
LSB
kHz
DC Common-Mode Rejection
Over Analog Input Voltage Range
-
Power Supply Sensitivity
V+ = 5V ±10% Over Allowed Input
Voltage Range
-
LSB
AC TIMING SPECIFICATIONS V+ = 5V, and TA = 25oC, Unless Otherwise Specified
V+ = 6V (Note 3)
100
640
1280
V+ = 5V
100
640
800
kHz
62
-
73
Clocks/Conv
-
-
8888
Conv/s
100
-
-
ns
Access Time (Delay from Falling CL = 100pF (Use Bus Driver IC for
Edge of RD to Output Data Valid), Larger CL)
tACC
-
135
200
ns
Three-State Control (Delay from
Rising Edge of RD to Hl-Z State),
t1H, t0H
-
125
250
ns
Delay from Falling Edge of WR to
Reset of INTR, tWI, tRI
-
300
450
ns
Input Capacitance of Logic
Control Inputs, CIN
-
5
-
pF
Three-State Output Capacitance
(Data Buffers), COUT
-
5
-
pF
Clock Frequency, fCLK
Clock Periods per Conversion
(Note 4), tCONV
Conversion Rate In Free-Running INTR tied to WR with CS = 0V,
Mode, CR
fCLK = 640kHz
Width of WR Input (Start Pulse
Width), tW(WR)I
CS = 0V (Note 5)
CL = 10pF, RL= 10K
(See Three-State Test Circuits)
DC DIGITAL LEVELS AND DC SPECIFICATIONS V+ = 5V, and TMIN to TMAX , Unless Otherwise Specified
CONTROL INPUTS (Note 6)
Logic “1“ Input Voltage (Except
Pin 4 CLK IN), VINH
V+ = 5.25V
2.0
-
V+
V
Logic “0“ Input Voltage (Except
Pin 4 CLK IN), VINL
V+ = 4.75V
-
-
0.8
V
CLK IN (Pin 4) Positive Going
Threshold Voltage, V+CLK
2.7
3.1
3.5
V
CLK IN (Pin 4) Negative Going
Threshold Voltage, V-CLK
1.5
1.8
2.1
V
CLK IN (Pin 4) Hysteresis, VH
0.6
1.3
2.0
V
Logic “1” Input Current
(All Inputs), IINHI
VlN = 5V
-
0.005
1
µΑ
Logic “0” Input Current
(All Inputs), IINLO
VlN = 0V
-1
-0.005
-
µA
Supply Current (Includes Ladder
Current), I+
fCLK = 640kHz,TA = 25oC
and CS = Hl
-
1.3
2.5
mA
lO = 1.6mA, V+ = 4.75V
-
-
0.4
V
DATA OUTPUTS AND INTR
Logic “0” Output Voltage, VOL
6-8
ADC0802, ADC0803, ADC0804
Electrical Specifications
(Notes 1, 7) (Continued)
MIN
TYP
MAX
UNITS
Logic “1” Output Voltage, VOH
PARAMETER
lO = -360µA, V+ = 4.75V
TEST CONDITIONS
2.4
-
-
V
Three-State Disabled Output
Leakage (All Data Buffers), ILO
VOUT = 0V
-3
-
-
µA
-
-
3
µA
Output Short Circuit Current,
ISOURCE
VOUT Short to Gnd TA = 25oC
4.5
6
-
mA
Output Short Circuit Current,
ISINK
VOUT Short to V+ TA = 25oC
9.0
16
-
mA
VOUT = 5V
NOTES:
1. All voltages are measured with respect to GND, unless otherwise specified. The separate AGND point should always be wired to the
DGND, being careful to avoid ground loops.
2. For VIN(-) ≥ VIN(+) the digital output code will be 0000 0000. Two on-chip diodes are tied to each analog input (see Block Diagram) which
will forward conduct for analog input voltages one diode drop below ground or one diode drop greater than the V+ supply. Be careful,
during testing at low V+ levels (4.5V), as high level analog inputs (5V) can cause this input diode to conduct - especially at elevated temperatures, and cause errors for analog inputs near full scale. As long as the analog VIN does not exceed the supply voltage by more than
50mV, the output code will be correct. To achieve an absolute 0V to 5V input voltage range will therefore require a minimum supply voltage of 4.950V over temperature variations, initial tolerance and loading.
3. With V+ = 6V, the digital logic interfaces are no longer TTL compatible.
4. With an asynchronous start pulse, up to 8 clock periods may be required before the internal clock phases are proper to start the conversion
process.
5. The CS input is assumed to bracket the WR strobe input so that timing is dependent on the WR pulse width. An arbitrarily wide pulse
width will hold the converter in a reset mode and the start of conversion is initiated by the low to high transition of the WR pulse (see
Timing Diagrams).
6. CLK IN (pin 4) is the input of a Schmitt trigger circuit and is therefore specified separately.
7. None of these A/Ds requires a zero-adjust. However, if an all zero code is desired for an analog input other than 0V, or if a narrow full scale span
exists (for example: 0.5V to 4V full scale) the VIN(-) input can be adjusted to achieve this. See the Zero Error description in this data sheet.
Timing Waveforms
2.4V
V+
tr = 20ns
tr
90%
50%
RD
RD
0.8V
DATA
OUTPUT
CS
CL
10%
t1H
VOH
10K
90%
DATA
OUTPUTS
GND
FIGURE 1A. t1H
FIGURE 1B. t1H , CL = 10pF
tr = 20ns
V+
V+
tr
2.4V
RD
10K
0.8V
RD
DATA
OUTPUT
CS
V+
CL
DATA
OUTPUTS
VOI
FIGURE 1C. t0H
90%
50%
10%
t0H
10%
FIGURE 1D. t0H , CL = 10pF
FIGURE 1. THREE-STATE CIRCUITS AND WAVEFORMS
6-9
ADC0802, ADC0803, ADC0804
Typical Performance Curves
500
-55oC TO 125oC
1.7
400
DELAY (ns)
LOGIC INPUT THRESHOLD VOLTAGE (V)
1.8
1.6
1.5
300
200
1.4
100
1.3
4.50
4.75
5.00
5.25
5.50
0
200
V+ SUPPLY VOLTAGE (V)
FIGURE 2. LOGIC INPUT THRESHOLD VOLTAGE vs SUPPLY
VOLTAGE
1000
R = 10K
3.1
VT(+)
R = 50K
fCLK (kHz)
2.7
-55oC TO 125oC
2.3
1.9
VT(-)
1.5
4.50
4.75
5.00
R = 20K
5.25
100
5.50
10
V+ SUPPLY VOLTAGE (V)
FIGURE 4. CLK IN SCHMITT TRIP LEVELS vs SUPPLY VOLTAGE
100
CLOCK CAPACITOR (pF)
16
6
VIN(+) = VIN(-) = 0V
14
ASSUMES VOS = 2mV
12
THIS SHOWS THE NEED
FOR A ZERO ADJUSTMENT
IF THE SPAN IS REDUCED
OFFSET ERROR (LSBs)
V+ = 4.5V
5
4
3
V+ = 5V
2
1000
FIGURE 5. fCLK vs CLOCK CAPACITOR
7
FULL SCALE ERROR (LSBs)
1000
FIGURE 3. DELAY FROM FALLING EDGE OF RD TO OUTPUT
DATA VALID vs LOAD CAPACITANCE
3.5
CLK IN THRESHOLD VOLTAGE (V)
400
600
800
LOAD CAPACITANCE (pF)
10
8
6
4
1
2
V+ = 6V
0
0
400
800
1200
fCLK (kHz)
1600
0
0.01
2000
0.1
1.0
5
VREF/2 (V)
FIGURE 6. FULL SCALE ERROR vs fCLK
FIGURE 7. EFFECT OF UNADJUSTED OFFSET ERROR
6-10
ADC0802, ADC0803, ADC0804
Typical Performance Curves
(Continued)
8
1.6
V+ = 5V
POWER SUPPLY CURRENT (mA)
fCLK = 640kHz
OUTPUT CURRENT (mA)
7
DATA OUTPUT
BUFFERS
6
ISOURCE
VOUT = 2.4V
5
4
3
-ISINK
VOUT = 0.4V
2
-50
-25
0
25
50
75
100
1.5
V+ = 5.5V
1.4
1.3
V+ = 5.0V
1.2
V+ = 4.5V
1.1
1.0
125
-50
-25
TA AMBIENT TEMPERATURE (oC)
FIGURE 8. OUTPUT CURRENT vs TEMPERATURE
0
25
50
75
100
TA AMBIENT TEMPERATURE (oC)
FIGURE 9. POWER SUPPLY CURRENT vs TEMPERATURE
Timing Diagrams
CS
WR
tWI
ACTUAL INTERNAL
STATUS OF THE
CONVERTER
“BUSY”
tW(WR)I
DATA IS VALID IN
OUTPUT LATCHES
“NOT BUSY”
1 TO 8 x 1/fCLK
INTERNAL TC
(LAST DATA READ)
INTR
INTR
ASSERTED
(LAST DATA NOT READ)
tVI
FIGURE 10A. START CONVERSION
INTR
CS
INTR RESET
tRI
RD
VALID
DATA
DATA
OUTPUTS
tACC
125
THREE-STATE
(HI-Z)
VALID
DATA
t1H , t0H
FIGURE 10B. OUTPUT ENABLE AND RESET INTR
6-11
1/ f
2 CLK
+1 LSB
D+1
5 6
D
ERROR
DIGITAL OUTPUT CODE
ADC0802, ADC0803, ADC0804
3 4
D-1
1
+1/2 LSB
3
* QUANTIZATION ERROR
0
-1/2 LSB
1 2
5
2
4
6
-1 LSB
A-1
A-1
A+1
A
A
A+1
ANALOG INPUT (VIN)
ANALOG INPUT (VIN)
TRANSFER FUNCTION
ERROR PLOT
+1 LSB
1
5
D+1
6
ERROR
DIGITAL OUTPUT CODE
FIGURE 11A. ACCURACY = ±0 LSB; PERFECT A/D
3
D
4
3
6
0
*
QUANTIZATION
ERROR
1
D-1
2
4
2
-1 LSB
A-1
A
A+1
A-1
A
A+1
ANALOG INPUT (VIN)
ANALOG INPUT (VIN)
TRANSFER FUNCTION
ERROR PLOT
FIGURE 11B. ACCURACY = ±1/2 LSB
FIGURE 11. CLARIFYING THE ERROR SPECS OF AN A/D CONVERTER
Understanding A/D Error Specs
A perfect A/D transfer characteristic (staircase wave-form) is
shown in Figure 11A. The horizontal scale is analog input voltage and the particular points labeled are in steps of 1 LSB
(19.53mV with 2.5V tied to the VREF/2 pin). The digital output
codes which correspond to these inputs are shown as D-1, D,
and D+1. For the perfect A/D, not only will center-value (A - 1,
A, A + 1, . . .) analog inputs produce the correct output digital
codes, but also each riser (the transitions between adjacent
output codes) will be located ±1/2 LSB away from each centervalue. As shown, the risers are ideal and have no width. Correct
digital output codes will be provided for a range of analog input
voltages which extend ±1/2 LSB from the ideal center-values.
Each tread (the range of analog input voltage which provides
the same digital output code) is therefore 1 LSB wide.
The error curve of Figure 11B shows the worst case transfer
function for the ADC0802. Here the specification guarantees
that if we apply an analog input equal to the LSB analog voltage center-value, the A/D will produce the correct digital code.
Next to each transfer function is shown the corresponding error
plot. Notice that the error includes the quantization uncertainty of
the A/D. For example, the error at point 1 of Figure 11A is
+1/2 LSB because the digital code appeared 1/2 LSB in advance
of the center-value of the tread. The error plots always have a
constant negative slope and the abrupt upside steps are always
1 LSB in magnitude, unless the device has missing codes.
Detailed Description
The functional diagram of the ADC0802 series of A/D
converters operates on the successive approximation principle (see Application Notes AN016 and AN020 for a more
detailed description of this principle). Analog switches are
closed sequentially by successive-approximation logic until
the analog differential input voltage [VlN(+) - VlN(-)] matches
a voltage derived from a tapped resistor string across the
reference voltage. The most significant bit is tested first and
after 8 comparisons (64 clock cycles), an 8-bit binary code
(1111 1111 = full scale) is transferred to an output latch.
The normal operation proceeds as follows. On the high-to-low
transition of the WR input, the internal SAR latches and the
shift-register stages are reset, and the INTR output will be set
high. As long as the CS input and WR input remain low, the
A/D will remain in a reset state. Conversion will start from 1 to
8 clock periods after at least one of these inputs makes a lowto-high transition. After the requisite number of clock pulses to
complete the conversion, the INTR pin will make a high-to-low
transition. This can be used to interrupt a processor, or
otherwise signal the availability of a new conversion. A RD
operation (with CS low) will clear the INTR line high again.
6-12
ADC0802, ADC0803, ADC0804
The device may be operated in the free-running mode by connecting INTR to the WR input with CS = 0. To ensure start-up
under all possible conditions, an external WR pulse is
required during the first power-up cycle. A conversion-in-process can be interrupted by issuing a second start command.
Digital Operation
The converter is started by having CS and WR simultaneously
low. This sets the start flip-flop (F/F) and the resulting “1” level
resets the 8-bit shift register, resets the Interrupt (INTR) F/F
and inputs a “1” to the D flip-flop, DFF1, which is at the input
end of the 8-bit shift register. Internal clock signals then transfer this “1” to the Q output of DFF1. The AND gate, G1, combines this “1” output with a clock signal to provide a reset
signal to the start F/F. If the set signal is no longer present
(either WR or CS is a “1”), the start F/F is reset and the 8-bit
shift register then can have the “1” clocked in, which starts the
conversion process. If the set signal were to still be present,
this reset pulse would have no effect (both outputs of the start
F/F would be at a “1” level) and the 8-bit shift register would
continue to be held in the reset mode. This allows for asynchronous or wide CS and WR signals.
After the “1” is clocked through the 8-bit shift register (which
completes the SAR operation) it appears as the input to
DFF2. As soon as this “1” is output from the shift register, the
AND gate, G2, causes the new digital word to transfer to the
Three-State output latches. When DFF2 is subsequently
clocked, the Q output makes a high-to-low transition which
causes the INTR F/F to set. An inverting buffer then supplies
the INTR output signal.
When data is to be read, the combination of both CS and RD
being low will cause the INTR F/F to be reset and the threestate output latches will be enabled to provide the 8-bit digital
outputs.
Digital Control Inputs
The digital control inputs (CS, RD, and WR) meet standard
TTL logic voltage levels. These signals are essentially equivalent to the standard A/D Start and Output Enable control signals, and are active low to allow an easy interface to
microprocessor control busses. For non-microprocessor
based applications, the CS input (pin 1) can be grounded and
the standard A/D Start function obtained by an active low
pulse at the WR input (pin 3). The Output Enable function is
achieved by an active low pulse at the RD input (pin 2).
Analog Operation
The analog comparisons are performed by a capacitive
charge summing circuit. Three capacitors (with precise
ratioed values) share a common node with the input to an
auto-zeroed comparator. The input capacitor is switched
between VlN(+) and VlN(-) , while two ratioed reference capacitors are switched between taps on the reference voltage
divider string. The net charge corresponds to the weighted difference between the input and the current total value set by
the successive approximation register. A correction is made to
offset the comparison by 1/2 LSB (see Figure 11A).
Analog Differential Voltage Inputs and Common-Mode
Rejection
This A/D gains considerable applications flexibility from the analog differential voltage input. The VlN(-) input (pin 7) can be used
to automatically subtract a fixed voltage value from the input
reading (tare correction). This is also useful in 4mA - 20mA current loop conversion. In addition, common-mode noise can be
reduced by use of the differential input.
The time interval between sampling VIN(+) and VlN(-) is 41/2
clock periods. The maximum error voltage due to this slight
time difference between the input voltage samples is given by:
4.5
∆V E ( MAX ) = (V PEAK ) ( 2πf CM ) -----------f CLK
where:
∆VE is the error voltage due to sampling delay,
VPEAK is the peak value of the common-mode voltage,
fCM is the common-mode frequency.
For example, with a 60Hz common-mode frequency, fCM ,
and a 640kHz A/D clock, fCLK , keeping this error to 1/4 LSB
(~5mV) would allow a common-mode voltage, VPEAK , given
by:
∆V E ( MAX ) ( f
CLK )
V PEAK = -------------------------------------------------- ,
( 2πf CM ) ( 4.5 )
or
–3
3
( 5 × 10 ) ( 640 × 10 )
V PEAK = ---------------------------------------------------------- ≅ 1.9V .
( 6.28 ) ( 60 ) ( 4.5 )
The allowed range of analog input voltage usually places
more severe restrictions on input common-mode voltage
levels than this.
An analog input voltage with a reduced span and a relatively
large zero offset can be easily handled by making use of the
differential input (see Reference Voltage Span Adjust).
Analog Input Current
The internal switching action causes displacement currents to
flow at the analog inputs. The voltage on the on-chip capacitance to ground is switched through the analog differential
input voltage, resulting in proportional currents entering the
VIN(+) input and leaving the VIN(-) input. These current transients occur at the leading edge of the internal clocks. They
rapidly decay and do not inherently cause errors as the onchip comparator is strobed at the end of the clock perIod.
Input Bypass Capacitors
Bypass capacitors at the inputs will average these charges
and cause a DC current to flow through the output resistances
of the analog signal sources. This charge pumping action is
worse for continuous conversions with the VIN(+) input voltage
at full scale. For a 640kHz clock frequency with the VIN(+)
input at 5V, this DC current is at a maximum of approximately
5µA. Therefore, bypass capacitors should not be used at
the analog inputs or the VREF/2 pin for high resistance
sources (>1kΩ). If input bypass capacitors are necessary for
noise filtering and high source resistance is desirable to minimize capacitor size, the effects of the voltage drop across this
input resistance, due to the average value of the input current,
can be compensated by a full scale adjustment while the
given source resistor and input bypass capacitor are both in
place. This is possible because the average value of the input
current is a precise linear function of the differential input
voltage at a constant conversion rate.
6-13
ADC0802, ADC0803, ADC0804
Input Source Resistance
V+
(VREF)
Large values of source resistance where an input bypass
capacitor is not used will not cause errors since the input
currents settle out prior to the comparison time. If a lowpass filter is required in the system, use a low-value series
resistor (≤1kΩ) for a passive RC section or add an op amp
RC active low-pass filter. For low-source-resistance
applications (≤1kΩ), a 0.1µF bypass capacitor at the inputs
will minimize EMI due to the series lead inductance of a long
wire. A 100Ω series resistor can be used to isolate this
capacitor (both the R and C are placed outside the feedback
loop) from the output of an op amp, if used.
20
R
VREF/2
9
DIGITAL
CIRCUITS
Stray Pickup
The leads to the analog inputs (pins 6 and 7) should be kept
as short as possible to minimize stray signal pickup (EMI).
Both EMI and undesired digital-clock coupling to these inputs
can cause system errors. The source resistance for these
inputs should, in general, be kept below 5kΩ. Larger values of
source resistance can cause undesired signal pickup. Input
bypass capacitors, placed from the analog inputs to ground,
will eliminate this pickup but can create analog scale errors as
these capacitors will average the transient input switching currents of the A/D (see Analog Input Current). This scale error
depends on both a large source resistance and the use of an
input bypass capacitor. This error can be compensated by a
full scale adjustment of the A/D (see Full Scale Adjustment)
with the source resistance and input bypass capacitor in
place, and the desired conversion rate.
8
AGND
10
DGND
FIGURE 12. THE VREFERENCE DESIGN ON THE IC
Reference Voltage Span Adjust
VREF
(5V)
For maximum application flexibility, these A/Ds have been
designed to accommodate a 5V, 2.5V or an adjusted voltage
reference. This has been achieved in the design of the IC as
shown in Figure 12.
ICL7611
Notice that the reference voltage for the IC is either 1/2 of the
FS
ADJ.
voltage which is applied to the V+ supply pin, or is equal to
the voltage which is externally forced at the VREF /2 pin. This
allows for a pseudo-ratiometric voltage reference using, for
the V+ supply, a 5V reference voltage. Alternatively, a voltage less than 2.5V can be applied to the VREF/2 input. The
internal gain to the VREF/2 input is 2 to allow this factor of 2
reduction in the reference voltage.
Such an adjusted reference voltage can accommodate a
reduced span or dynamic voltage range of the analog input
voltage. If the analog input voltage were to range from 0.5V to
3.5V, instead of 0V to 5V, the span would be 3V. With 0.5V
applied to the VlN(-) pin to absorb the offset, the reference
voltage can be made equal to 1/2 of the 3V span or 1.5V. The
A/D now will encode the VlN(+) signal from 0.5V to 3.5V with
the 0.5V input corresponding to zero and the 3.5V input corresponding to full scale. The full 8 bits of resolution are therefore
applied over this reduced analog input voltage range. The requisite connections are shown in Figure 13. For expanded
scale inputs, the circuits of Figures 14 and 15 can be used.
ANALOG
CIRCUITS
DECODE
R
“SPAN”/2
5V
300
-
TO VREF/2
+
0.1µF
TO VIN(-)
ZERO SHIFT VOLTAGE
FIGURE 13. OFFSETTING THE ZERO OF THE ADC0802 AND
PERFORMING AN INPUT RANGE (SPAN)
ADJUSTMENT
5V
(VREF)
R
VIN ± 10V
2R
6
VIN(+)
V+
ADC0802ADC0804
2R
7
20
+
10µF
VIN(-)
FIGURE 14. HANDLING ±10V ANALOG INPUT RANGE
6-14
ADC0802, ADC0803, ADC0804
Full Scale Adjust
5V
(VREF)
The full scale adjustment can be made by applying a
differential input voltage which is 11/2 LSB down from the
desired analog full scale voltage range and then adjusting
the magnitude of the VREF/2 input (pin 9) for a digital output
code which is just changing from 1111 1110 to 1111 1111.
When offsetting the zero and using a span-adjusted VREF/2
voltage, the full scale adjustment is made by inputting VMlN
to the VIN(-) input of the A/D and applying a voltage to the
VIN(+) input which is given by:
R
VIN ±5V
R
6
VIN(+)
V+
ADC0802ADC0804
7
20
+
10µF
VIN(-)
( V MAX – V MIN )
V IN ( + ) f SADJ = V MAX – 1.5 ----------------------------------------- ,
256
where:
FIGURE 15. HANDLING ±5V ANALOG INPUT RANGE
VMAX = the high end of the analog input range,
Reference Accuracy Requirements
and
The converter can be operated in a pseudo-ratiometric
mode or an absolute mode. In ratiometric converter applications, the magnitude of the reference voltage is a factor in
both the output of the source transducer and the output of
the A/D converter and therefore cancels out in the final digital output code. In absolute conversion applicatIons, both the
initial value and the temperature stability of the reference
voltage are important accuracy factors in the operation of the
A/D converter. For VREF/2 voltages of 2.5V nominal value,
initial errors of ±10mV will cause conversion errors of ±1
LSB due to the gain of 2 of the VREF/2 input. In reduced
span applications, the initial value and the stability of the
VREF/2 input voltage become even more important. For
example, if the span is reduced to 2.5V, the analog input LSB
voltage value is correspondingly reduced from 20mV (5V
span) to 10mV and 1 LSB at the VREF/2 input becomes
5mV. As can be seen, this reduces the allowed initial tolerance of the reference voltage and requires correspondingly
less absolute change with temperature variations. Note that
spans smaller than 2.5V place even tighter requirements on
the initial accuracy and stability of the reference source.
VMIN = the low end (the offset zero) of the analog range.
(Both are ground referenced.)
In general, the reference voltage will require an initial
adjustment. Errors due to an improper value of reference
voltage appear as full scale errors in the A/D transfer function. IC voltage regulators may be used for references if the
ambient temperature changes are not excessive.
FIGURE 16. SELF-CLOCKING THE A/D
Zero Error
The zero of the A/D does not require adjustment. If the
minimum analog input voltage value, VlN(MlN) , is not ground, a
zero offset can be done. The converter can be made to output
0000 0000 digital code for this minimum input voltage by biasing the A/D VIN(-) input at this VlN(MlN) value (see Applications
section). This utilizes the differential mode operation of the A/D.
The zero error of the A/D converter relates to the location of
the first riser of the transfer function and can be measured by
grounding the VIN(-) input and applying a small magnitude
positive voltage to the VIN(+) input. Zero error is the difference
between the actual DC input voltage which is necessary to
just cause an output digital code transition from 0000 0000 to
0000 0001 and the ideal 1/2 LSB value (1/2 LSB = 9.8mV for
VREF/2 = 2.500V).
Clocking Option
The clock for the A/D can be derived from an external source
such as the CPU clock or an external RC network can be
added to provIde self-clocking. The CLK IN (pin 4) makes
use of a Schmitt trigger as shown in Figure 16.
CLK R
19
ADC0802ADC0804
R
CLK IN
C
4
fCLK ≅
1
1.1 RC
R ≅ 10kΩ
CLK
Heavy capacitive or DC loading of the CLK R pin should be
avoided as this will disturb normal converter operation.
Loads less than 50pF, such as driving up to 7 A/D converter
clock inputs from a single CLK R pin of 1 converter, are
allowed. For larger clock line loading, a CMOS or low power
TTL buffer or PNP input logic should be used to minimize the
loading on the CLK R pin (do not use a standard TTL buffer).
Restart During a Conversion
If the A/D is restarted (CS and WR go low and return high)
during a conversion, the converter is reset and a new conversion is started. The output data latch is not updated if the
conversion in progress is not completed. The data from the
previous conversion remain in this latch.
Continuous Conversions
In this application, the CS input is grounded and the WR
input is tied to the INTR output. This WR and INTR node
should be momentarily forced to logic low following a powerup cycle to insure circuit operation. See Figure 17 for details.
6-15
ADC0802, ADC0803, ADC0804
10K
signal leads. Exposed leads to the analog inputs can cause
undesired digital noise and hum pickup; therefore, shielded
leads may be necessary in many applications.
5V (VREF)
ADC0802 - ADC0804
150pF
1 CS
N.O.
START
ANALOG
INPUTS
V+ 20
2 RD
CLK R 19
3 WR
DB0 18
4 CLK IN
DB1 17
5 INTR
DB2 16
6 VIN (+)
DB3 15
7 VIN (-)
DB4 14
8 AGND
DB5 13
9 VREF/2
DB6 12
10 DGND
DB7 11
+
10µF
LSB
DATA
OUTPUTS
MSB
A single-point analog ground should be used which is separate
from the logic ground points. The power supply bypass capacitor and the self-clockIng capacitor (if used) should both be
returned to digital ground. Any VREF/2 bypass capacitors, analog input filter capacitors, or input signal shielding should be
returned to the analog ground point. A test for proper grounding
is to measure the zero error of the A/D converter. Zero errors in
excess of 1/4 LSB can usually be traced to improper board
layout and wiring (see Zero Error for measurement). Further
information can be found in Application Note AN018.
Testing the A/D Converter
There are many degrees of complexity associated with testing
an A/D converter. One of the simplest tests is to apply a
known analog input voltage to the converter and use LEDs to
display the resulting digital output code as shown in Figure 18.
FIGURE 17. FREE-RUNNING CONNECTION
Driving the Data Bus
This CMOS A/D, like MOS microprocessors and memories,
will require a bus driver when the total capacitance of the
data bus gets large. Other circuItry, which is tied to the data
bus, will add to the total capacitive loading, even in threestate (high-impedance mode). Back plane busing also
greatly adds to the stray capacitance of the data bus.
There are some alternatives available to the designer to handle this problem. Basically, the capacitive loading of the data
bus slows down the response time, even though DC specifications are still met. For systems operating with a relatively
slow CPU clock frequency, more time is available in which to
establish proper logic levels on the bus and therefore higher
capacitive loads can be driven (see Typical Performance
Curves).
At higher CPU clock frequencies time can be extended for
I/O reads (and/or writes) by inserting wait states (8080) or
using clock-extending circuits (6800).
Finally, if time is short and capacitive loading is high,
external bus drivers must be used. These can be three-state
buffers (low power Schottky is recommended, such as the
74LS240 series) or special higher-drive-current products
which are designed as bus drivers. High-current bipolar bus
drivers with PNP inputs are recommended.
For ease of testing, the VREF/2 (pin 9) should be supplied
with 2.560V and a V+ supply voltage of 5.12V should be
used. This provides an LSB value of 20mV.
If a full scale adjustment is to be made, an analog input voltage of 5.090V (5.120 - 11/2 LSB) should be applied to the
VIN(+) pin with the VIN(-) pin grounded. The value of the
VREF/2 input voltage should be adjusted until the digital output code is just changing from 1111 1110 to 1111 1111. This
value of VREF/2 should then be used for all the tests.
The digital-output LED display can be decoded by dividing the 8
bits into 2 hex characters, one with the 4 most-significant bits
(MS) and one with the 4 least-significant bits (LS). The output is
then interpreted as a sum of fractions times the full scale voltage:
MS LS
VO UT =  --------- + ---------- ( 5.12 )V .
 16 256
10kΩ
150pF
1
N.O.
START
VIN (+)
Noise spikes on the V+ supply line can cause conversion
errors as the comparator will respond to this noise. A
low-inductance tantalum filter capacitor should be used
close to the converter V+ pin, and values of 1µF or greater
are recommended. If an unregulated voltage is available in
the system, a separate 5V voltage regulator for the converter
(and other analog circuitry) will greatly reduce digital noise
on the V+ supply. An lCL7663 can be used to regulate such
a supply from an input as low as 5.2V.
Wiring and Hook-Up Precautions
Standard digital wire-wrap sockets are not satisfactory for
breadboarding with this A/D converter. Sockets on PC
boards can be used. All logic signal wires and leads should
be grouped and kept as far away as possible from the analog
2
19
3
18
4
17
5
Power Supplies
0.1µF
AGND
2.560V
VREF/2
0.1µF
20
6
ADC0802ADC0804
5.120V
10µF
TANTALUM
LSB
16
15
7
14
8
13
9
12
10
11
DGND
+
5V
MSB
1.3kΩ LEDs
(8)
(8)
FIGURE 18. BASIC TESTER FOR THE A/D
For example, for an output LED display of 1011 0110, the
MS character is hex B (decimal 11) and the LS character is
hex (and decimal) 6, so:
11
6
VO UT =  ------ + ---------- ( 5.12 ) = 3.64V.
 16 256
6-16
ADC0802, ADC0803, ADC0804
Figures 19 and 20 show more sophisticated test circuits.
8-BIT
A/D UNDER
TEST
Interfacing the Z-80 and 8085
VANALOG OUTPUT
10-BIT
DAC
R
R
“B”
ANALOG
INPUTS
-
+
A1
“C”
R
100R
R
-
+
“A”
A2
100X ANALOG
ERROR VOLTAGE
Additional I/O advantages exist as software DMA routines are
available and use can be made of the output data transfer
which exists on the upper 8 address lines (A8 to A15) during
I/O input instructions. For example, MUX channel selection for
the A/D can be accomplished with this operating mode.
FIGURE 19. A/D TESTER WITH ANALOG ERROR OUTPUT. THIS
CIRCUIT CAN BE USED TO GENERATE “ERROR
PLOTS” OF FIGURE 11.
DIGITAL
INPUTS
10-BIT
DAC
The 8085 also provides a generalized RD and WR strobe, with
an IO/M line to distinguish I/O and memory requests. The circuit of Figure 22 can again be used, with IO/M in place of IORQ
for a memory-mapped interface, and an extra inverter (or the
logic equivalent) to provide IO/M for an I/O-mapped connection.
DIGITAL
OUTPUTS
VANALOG
The Z-80 and 8085 control buses are slightly different from
that of the 8080. General RD and WR strobes are provided
and separate memory request, MREQ, and I/O request,
IORQ, signals have to be combined with the generalized
strobes to provide the appropriate signals. An advantage of
operating the A/D in I/O space with the Z-80 is that the CPU
will automatically insert one wait state (the RD and WR
strobes are extended one clock period) to allow more time
for the I/O devices to respond. Logic to map the A/D in I/O
space is shown in Figure 22. By using MREQ in place of
IORQ, a memory-mapped configuration results.
A/D UNDER
TEST
Interfacing 6800 Microprocessor Derivatives (6502, etc.)
FIGURE 20. BASIC “DIGITAL” A/D TESTER
Typical Applications
Interfacing 8080/85 or Z-80 Microprocessors
This converter has been designed to directly interface with
8080/85 or Z-80 Microprocessors. The three-state output
capability of the A/D eliminates the need for a peripheral
interface device, although address decoding is still required
to generate the appropriate CS for the converter. The A/D
can be mapped into memory space (using standard memory-address decoding for CS and the MEMR and MEMW
strobes) or it can be controlled as an I/O device by using the
I/OR and I/OW strobes and decoding the address bits A0 →
A7 (or address bits A8 → A15, since they will contain the
same 8-bit address information) to obtain the CS input.
Using the I/O space provides 256 additional addresses and
may allow a simpler 8-bit address decoder, but the data can
only be input to the accumulator. To make use of the additional memory reference instructions, the A/D should be
mapped into memory space. See AN020 for more discussion of memory-mapped vs I/O-mapped interfaces. An
example of an A/D in I/O space is shown in Figure 21.
The control bus for the 6800 microprocessor derivatives does
not use the RD and WR strobe signals. Instead it employs a
single R/W line and additional timing, if needed, can be derived
from the φ2 clock. All I/O devices are memory-mapped in the
6800 system, and a special signal, VMA, indicates that the current address is valid. Figure 23 shows an interface schematic
where the A/D is memory-mapped in the 6800 system. For simplicity, the CS decoding is shown using 1/2 DM8092. Note that
in many 6800 systems, an already decoded 4/5 line is brought
out to the common bus at pin 21. This can be tied directly to the
CS pin of the A/D, provided that no other devices are
addressed at HEX ADDR: 4XXX or 5XXX.
In Figure 24 the ADC0802 series is interfaced to the MC6800
microprocessor through (the arbitrarily chosen) Port B of the
MC6820 or MC6821 Peripheral Interface Adapter (PlA). Here
the CS pin of the A/D is grounded since the PlA is already
memory-mapped in the MC6800 system and no CS decoding
is necessary. Also notice that the A/D output data lines are connected to the microprocessor bus under program control
through the PlA and therefore the A/D RD pin can be grounded.
Application Notes
The standard control-bus signals of the 8080 (CS, RD and
WR) can be directly wired to the digital control inputs of the
A/D, since the bus timing requirements, to allow both starting
the converter, and outputting the data onto the data bus, are
met. A bus driver should be used for larger microprocessor
systems where the data bus leaves the PC board and/or
must drive capacitive loads larger than 100pF.
It is useful to note that in systems where the A/D converter is
1 of 8 or fewer I/O-mapped devices, no address-decoding
circuitry is necessary. Each of the 8 address bits (A0 to A7)
can be directly used as CS inputs, one for each I/O device.
6-17
NOTE #
DESCRIPTION
AnswerFAX
DOC. #
AN016
“Selecting A/D Converters”
9016
AN018
“Do’s and Don’ts of Applying A/D
Converters”
9018
AN020
“A Cookbook Approach to High Speed
Data Acquisition and Microprocessor
Interfacing”
9020
AN030
“The ICL7104 - A Binary Output A/D
Converter for Microprocessors”
9030
ADC0802, ADC0803, ADC0804
INT (14)
I/O WR (27) (NOTE)
I/O RD (25) (NOTE)
10K
ADC0802 - ADC0804
ANALOG
INPUTS
150pF
1 CS
V+ 20
2 RD
CLK R 19
5V
+
10µF
3 WR
DB0 18 LSB
DB0 (13) (NOTE)
4 CLK IN
DB1 17
DB1 (16) (NOTE)
5 INTR
DB2 16
DB2 (11) (NOTE)
6 VIN (+)
DB3 15
DB3 (9) (NOTE)
7 VIN (-)
DB4 14
DB4 (5) (NOTE)
8 AGND
DB5 13
DB5 (18) (NOTE)
9 VREF/2
DB6 12
10 DGND
DB7 11
DB6 (20) (NOTE)
MSB
DB7 (7) (NOTE)
5V
OUT
V+
B5
AD15 (36)
B4
AD14 (39)
B3
AD13 (38)
B2
AD12 (37)
T1
B1
AD11 (40)
T0
B0
AD10 (1)
T5
T4
T3
T2
8131
BUS
COMPARATOR
NOTE: Pin numbers for 8228 System Controller: Others are 8080A.
FIGURE 21. ADC0802 TO 8080A CPU INTERFACE
6-18
ADC0802, ADC0803, ADC0804
IRQ (4) † [D] ††
R/W (34) [6]
10K
+
ADC0802 - ADC0804
RD
RD
ANALOG
INPUTS
2
IORQ
ADC0802ADC0804
150pF
V+ 20
2 RD
CLK R 19
10µF
ABC
5V (8) 1 2 3
3 WR
DB0 18 LSB
D0 (33) [31]
4 CLK IN
DB1 17
D1 (32) [29]
5 INTR
DB2 16
D2 (31) [K]
6 VIN (+)
DB3 15
D3 (30) [H]
7 VIN (-)
DB4 14
D4 (29) [32]
8 AGND
DB5 13
D5 (28) [30]
9 VREF/2
DB6 12
10 DGND
WR
WR
1 CS
DB7 11
D6 (27) [L]
MSB
D7 (26) [J]
3
1
74C32
A12 (22) [34]
2
6
3
1/ DM8092
2
†
††
FIGURE 22. MAPPING THE A/D AS AN
I/O DEVICE FOR USE
WITH THE Z-80 CPU
A13 (23) [N]
A14 (24) [M]
4
A15 (25) [33]
5
VMA (5) [F]
Numbers in parentheses refer to MC6800 CPU Pinout.
Numbers or letters in brackets refer to standard MC6800 System Common Bus Code.
FIGURE 23. ADC0802 TO MC6800 CPU INTERFACE
18
19
CB1
CB2
10K
ADC0802 - ADC0804
ANALOG
INPUTS
150pF
1 CS
V+ 20
2 RD
CLK R 19
MC6820
(MCS6520)
5V
PIA
3 WR
DB0 18 LSB
10
PB0
4 CLK IN
DB1 17
11
PB1
5 INTR
DB2 16
12
PB2
6 VIN (+)
DB3 15
13
PB3
7 VIN (-)
DB4 14
14
PB4
8 AGND
DB5 13
15
PB5
9 VREF/2
DB6 12
16
PB6
17
PB7
10 DGND
DB7 11
MSB
FIGURE 24. ADC0802 TO MC6820 PIA INTERFACE
6-19
ADC0802, ADC0803, ADC0804
Die Characteristics
DIE DIMENSIONS:
PASSIVATION:
(101 mils x 93 mils) x 525µm x 25µm
Type: Nitride over Silox
Nitride Thickness: 8kÅ
Silox Thickness: 7kÅ
METALLIZATION:
Type: Al
Thickness: 10kÅ ±1kÅ
Metallization Mask Layout
ADC0802, ADC0803, ADC0804
AGND
VIN (-)
VIN (+)
INTR
CLK IN
WR
VREF/2
RD
DGND
CS
DB7 (MSB)
DB6
V+ OR VREF
V+ OR VREF
DB5
CLK R
DB4
DB3
DB2
DB1
DB0
All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification.
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate
and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which
may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
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6-20
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