AD ADDC02808PB 28 v, 200 w pulsed dc/dc converter with integral emi filter Datasheet

28 V, 200 W Pulsed DC/DC Converter
with Integral EMI Filter
ADDC02808PB
a
FEATURES
28 V dc Input, 8 V dc @ 25 A, 200 W Pulse Output
Integral EMI Filter
Ultrafast Transient Response
Minimal Output Voltage Deviation
Low Weight: 80 Grams
NAVMAT Derated
Many Protection and System Features
OBS
APPLICATIONS
Distributed Power Architecture for Driving T/R Modules
Motor and Actuator Drivers
GENERAL DESCRIPTION
FUNCTIONAL BLOCK DIAGRAM
– SENSE
+ SENSE
ADJUST
OUTPUT SIDE
CONTROL
CIRCUIT
STATUS
RETURN
VAUX
INHIBIT
SYNC
ISHARE
FIXED
FREQUENCY
DUAL
INTERLEAVED
POWER TRAIN
INPUT SIDE
CONTROL
CIRCUIT
–VIN
RETURN
+VOUT
+VOUT
+VOUT
TEMP
+VIN
RETURN
OUTPUT
FILTER
EMI FILTER
OLE
TE
PRODUCT HIGHLIGHTS
The ADDC02808PB hybrid military dc/dc converter is compensated specifically for pulse applications where fast transient
response and minimum output voltage deviation are required.
It is also designed to deliver very high, pulsed output power.
The unit is designed for high reliability and high performance
applications where saving space and/or weight are critical.
1. 120 W/cubic inch pulsed power density with an integral EMI
filter
The ADDC02808PB has been characterized over a wide variety
of load conditions. Its transient response has been set to insure
output stability over a broad range of load capacitance. For
applications that require factory modified compensation
optimized for a specific load, or for applications that require a
different output voltage than 8 V dc, contact the factory.
4. Operational and survivable over a wide range of input
conditions: 16 V–50 V dc; survives low line, high line
The ADDC02808PB is available in a hermetically sealed,
molybdenum based hybrid package and is easily heatsink
mountable. For MIL-STD-883 devices, contact the factory for
availability.
2. Ultrafast transient response time with minimum output
voltage deviation
3. Light weight: 80 grams
5. High reliability; NAVMAT derated
6. Protection features include:
Output Overvoltage Protection
Output Short Circuit Current Protection
Thermal Monitor/Shutdown
Input Overvoltage Shutdown
Input Transient Protection
7. System level features include:
Current Sharing for Parallel Operation
Logic Level Disable
Output Status Signal
Synchronization for Multiple Units
Input Referenced Auxiliary Voltage Supply
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
World Wide Web Site: http://www.analog.com
Fax: 617/326-8703
© Analog Devices, Inc., 1996
ADDC02808PB–SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
(TC = +258C, VIN = 28 V dc 60.5 V dc, unless otherwise noted; full temperature range is
–558C to +908C; all temperatures are case and TC is the temperature measured at the center of the package bottom.)
Parameter
INPUT CHARACTERISTICS
Steady State Operating Input Voltage Range1
Abnormal Operating Input Voltage Range
(Per MIL-STD-704D)1
Input Overvoltage Shutdown
No Load Input Current
Disabled Input Current
OUTPUT CHARACTERISTICS2, 3
Output Voltage (VO)
Case
Temp
Test
Level Conditions
Min
Full
VI
IO = 1.25 A to 25 A Pulsed
18
Full
+25°C
+25°C
+25°C
VI
I
VI
VI
IO = 1.25 A to 25 A Pulsed
16
50
+25°C
Full
Full
+25°C
+25°C
+25°C
Full
+25°C
+25°C
+25°C
I
VI
VI
VI
VI
I
VI
V
V
I
IO = 1.25 A to 25 A, VIN = 18 V to 40 V dc
IO = 1.25 A to 25 A, VIN = 18 V to 40 V dc
IO = 1.25 A to 20 A, VIN = 16 V to 50 V dc
IO = 25 A Pulsed, VIN = 18 V to 40 V dc
VIN = 28 V dc, IO = 1.25 A to 25 A Pulsed
IO = 25 A, 5 kHz – 2 MHz BW
VIN = 18 V to 40 V dc, Pulsed
IO = 25 A, Open Remote Sense Connection
VO = 90% VOUT Nom
45 mΩ ≤ RSHORT Circuit ≤ 60 mΩ
7.92
7.84
7.84
I
Input to Output or Any Pin to Case at 500 V dc
100
I
I
(Reference Section Entitled “Transient Response”)
(Reference Section Entitled “Response at End
of Pulse”)
IO = 25 A, From Inhibit High to Status High
I
VI
I
VI
V
IO = 12.5 A
IO = 12.5 A
IO = 25 A
IO = 25 A
IO = 25 A
Full
+25°C
VI
I
IO = 2 A
+25°C
+25°C
I
I
IOH = 400 µA
IOL = 1 mA
2.4
+25°C
I
IAUX = 5 mA, Load Current = 12.5 A
14.9
+25°C
+25°C
+25°C
I
I
I
VIL = 0.5 V
+25°C
+25°C
+25°C
+25°C
I
I
I
V
VIH = 7.0 V
IO = 20 A
OBS
Line Regulation
Load Regulation
Output Ripple/Noise4
Output Current (IO)5
Output Overvoltage Protection
Output Current Limit
Output Short Circuit Current
ISOLATION CHARACTERISTICS
Isolation Resistance
+25°C
4
DYNAMIC CHARACTERISTICS
Step Changes In Load (min to max)
Step Changes In Load (max to min)
Soft Start Turn-On Time
THERMAL CHARACTERISTICS
Efficiency
Hottest Junction Temperature5
CONTROL CHARACTERISTICS
Clock Frequency
ADJUST (Pin 3) V ADJ
STATUS (Pin 4)
VOH
VOL
VAUX (Pin 5)
VO (nom)
INHIBIT (Pin 6)
VIL
IIL
VI (Open Circuit)
SYNC (Pin 7)6
VIH
IIH
ISHARE (Pin 8)
TEMP (Pin 9)
+25°C
+25°C
+25°C
Full
+25°C
Full
+90°C
ADDC02808PB
Typ
Max
28
40
V
52.5
45
1
50
55
100
5
V
V
mA
mA
8.08
8.16
8.16
5
10
60
25
V
V
V
mV
mV
mV p-p
A
% VO nom
% IO max
A
8.00
1
2.5
40
1.25
OLE
Units
120
130
40
MΩ
TE
6
79
78
75
72.5
10
80
%
%
%
%
°C
76
110
0.85
3.1
3.2
0.99
3.3
MHz
V
4.0
0.15
0.7
V
V
15.15
15.4
V
0.5
1.2
15
V
mA
V
150
2.65
V
µA
V
V
4.0
2.45
ms
2.55
3.90
NOTES
1
50 V dc upper limit rated for transient condition of up to 50 ms. 16 V dc lower limit rated for continuous operation during emergency condition. Steady state and
abnormal input voltage range require source impedance sufficient to insure input stability at low line. See sections entitled System Instability Considerations and Input
Voltage Range.
2
Measured at the remote sense points.
3
Unit regulates output voltage to zero load; tests performed at low continuous load and 200 W pulsed load.
4
CLOAD = 1,000 µF. Output ripple/noise measured at converter output; may be smaller at external load capacitance. Unit is stable for C LOAD ranging from 500 µF
to 4,000 µF.
5
Refer to section entitled “Pulse Output Power vs. Pulse Length” for more information.
6
Unit has internal pull-down; refer to section entitled Pin 7 (SYNC).
Specifications subject to change without notice.
–2–
REV. 0
ADDC02808PB
PIN DESCRIPTIONS
ABSOLUTE MAXIMUM RATINGS*
INHIBIT . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V dc, –0.5 V dc
SYNC . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.0 V dc, –0.5 V dc
ISHARE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V dc, –0.5 V dc
TEMP . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 V dc, –0.3 V dc
Common-Mode Voltage, Input to Output . . . . . . . . . 500 V dc
Lead Soldering Temp (10 sec) . . . . . . . . . . . . . . . . . . . +300°C
Storage Temperature . . . . . . . . . . . . . . . . . . –65°C to +150°C
Maximum Junction Temperature . . . . . . . . . . . . . . . . . +150°C
Maximum Case Operating Temperature . . . . . . . . . . . +125°C
OBS
ADDC02808PBKV
–40°C to +85°C
ADDC02808PBTV
–55°C to +90°C
ADDC02808PBTV/883B* –55°C to +125°C
*Contact factory.
EXPLANATION OF TEST LEVELS
Test Level
I
–
100% production tested.
6
Function
Feedback loop connection for remote sensing
output voltage. Must always be connected to
output return for proper operation.
+SENSE Feedback loop connection for remote sensing
output voltage. Must always be connected to
+VOUT for proper operation.
ADJUST Adjusts output voltage setpoint.
STATUS Indicates output voltage is within ± 5% of
nominal. Active high referenced to –SENSE
(Pin 1).
VAUX
Low level dc auxiliary voltage supply referenced to input return (Pin 10).
INHIBIT Power Supply Inhibit. Active low and referenced to input return (Pin 10).
SYNC
Clock synchronization input for multiple
units; referenced to input return (Pin 10).
ISHARE
Current share pin which allows paralleled
units to share current typically within ± 5% at
full load; referenced to input return (Pin 10).
TEMP
Case temperature indicator and temperature
shutdown override; referenced to input return
(Pin 10).
–VIN
Input Return.
+VIN
+28 V Nominal Input Bus.
+8 V dc Output.
+VOUT
+VOUT
+8 V dc Output.
+VOUT
+8 V dc Output.
RETURN Output Return.
RETURN Output Return.
RETURN Output Return.
OLE
Description
7
Hermetic Package
Hermetic Package
Hermetic Package
8
100% production tested at +25°C, and sample tested at
specified temperatures.
III –
Sample tested only.
IV –
Parameter is guaranteed by design and characterization
testing.
V
Parameter is a typical value only.
VI –
–SENSE
5
II –
–
1
3
4
ORDERING GUIDE
Device
Name
2
*Absolute maximum ratings are limiting values, to be applied individually, and
beyond which the serviceability of the circuit may be impaired. Functional
operability under any of these conditions is not necessarily implied. Exposure of
absolute maximum rating conditions for extended periods of time may affect device
reliability.
Operating
Temperature
Range (Case)
Pin
No.
9
10
11
12
13
14
15
16
17
TE
All devices are 100% production tested at +25°C. 100%
production tested at temperature extremes for military
temperature devices; guaranteed by design and
characterization testing for industrial devices.
PIN CONFIGURATION
1
17
TOP
VIEW
11
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Therefore, proper ESD precautions are recommended to avoid performance degradation or loss
of functionality.
REV. 0
–3–
12
WARNING!
ESD SENSITIVE DEVICE
ADDC02808PB–Typical Performance Curves
1.0
82
28VIN, 150W PEAK
80
0.5
VOUT DEVIATION – %
40V
PERCENTAGE
78
28V
76
74
18V
72
0.0
–0.5
70
68
0
20
40
60
80
100 120 140
OUTPUT POWER – Watts
160
OBS
180
–1.0
–55 –45 –35 –25 –15 –5
200
Figure 1. Efficiency vs. Line and Load at +25 °C
(Load Pulse Width of 50 ms)
15 25 35 45 55 65 75 85 95
TCASE
5
Figure 4. Output Voltage vs. Case Temperature (°C)
OLE
79
28VIN, 150W PEAK
EFFICIENCY – Percentage
79
78
78
100
2V
90
10
77
0%
10V
VINHIBIT
77
76
–55 –45 –35 –25 –15 –5
5
TE
VO
1ms
15 25 35 45 55 65 75 85 95
TCASE
Figure 2. Efficiency vs. Case Temperature ( °C)
(at Nominal VIN, 75% Max Load, Load Pulse Width
of 50 ms)
Figure 5. Output Voltage Transient During Turn-On with
0.1 A Load Displaying Soft Start When Supply Is Enabled
16.0
15.5
INPUT VOLTAGE
100
15.0
90
100mV
VO
14.5
14.0
10
0%
13.5
13.0
140
100µs
150
160
170
180
PEAK OUTPUT POWER – Watts
190
200
Figure 3. Low Line Dropout vs. Load at 90 °C Case
Temperature (Load Pulse Width of 50 ms)
Figure 6. Output Voltage Transient Response to
a 1 A to 25 A Step Change in Load, di/dt = 12 A/µ s,
with 1,000 µ F Load Capacitance (RESR = 10 mΩ)
–4–
REV. 0
ADDC02808PB
0
1
–10
–20
0.1
–40
|ZOUT| – Ω
|AS| – dB
–30
–50
–60
0.01
–70
–80
–90
–100
10
100
1k
FREQUENCY – Hz
10k
OBS
0.001
10
50k
Figure 7. Audio Susceptibility (Magnitude of VOUT /VIN)
10k
100k
Figure 9. Incremental Output Impedance (Magnitude)
10
1
5mV
1mV
VOLTS
|ZIN| – Ω
100
.5mV
100µV
100
1k
FREQUENCY – Hz
10k
50µV
100k
0
Figure 8. Incremental Input Impedance (Magnitude)
REV. 0
1k
FREQUENCY – Hz
OLE
1k
0.1
10
100
TE
10
FREQUENCY – MHz
Figure 10. Output Frequency Spectrum
–5–
20
ADDC02808PB–Typical EMI Curves & Test Setup for 28 VIN, 5 V dc Out, 100 W Converter
130
166
CONDUCTED EMISSIONS CE–101
146
EMISSION LEVEL – dB/pT
EMISSION LEVEL – dB µV
110
RE101 MIL-STD-461D
90
CE101–1 4.5 AMPS
70
50
126
RE101–1
106
86
30
0.0001
0.001
FREQUENCY – MHz
66
0.01
OBS
0.0001
Figure 11. Conducted Emissions, MIL-STD-461D, CE101,
+28 V Hot Line 100 W Load
130
0.001
0.01
FREQUENCY – MHz
Figure 13. Radiated Emissions, MIL-STD-461D, RE101,
100 W Load
OLE
90
RADIATED EMISSIONS RE–102
EMISSION LEVEL – dB µV
90
70
LIMIT 28VDC
EMISSION LEVEL – dB µV/m
CONDUCTED EMISSIONS CE–102
110
50
30
0.01
0.1
1
FREQUENCY – MHz
70
30
10
LISN
100µF
0.1
RE102–2
10
1
FREQUENCY – MHz
100
1000
Figure 14. Radiated Emissions, MIL-STD-461D, RE102,
Vertical Polarity, 100 W Load
+VIN
1Ω
TE
50
–30
0.01
10
Figure 12. Conducted Emissions, MIL-STD-461D, CE102,
+28 V Hot Line 100 W Load
LISN
0.1
+VOUT
82nF
0.1µF
2µF
1/4Ω
82nF
–VIN
TWO METERS OF
TWISTED CABLE
RETURN
CASE
GROUND PLANE
NOTE: 100µF CAPACITOR AND 1Ω RESISTOR PROVIDE STABILIZATION FOR 100µH DIFFERENTIAL SOURCE INDUCTANCE
INTRODUCED BY THE LISNs. REFER TO SECTION ON EMI CONSIDERATIONS FOR MORE INFORMATION.
Figure 15. Schematic of Test Setup for EMI Measurements
–6–
REV. 0
ADDC02808PB
multiple converters and to reduce input power draw during
extended time in a no load condition.
BASIC OPERATION
The ADDC02808PB converter uses a flyback topology with
dual interleaved power trains operating 180° out of phase. Each
power train switches at a fixed frequency of 500 kHz, resulting
in a 1 MHz fixed switching frequency as seen at the input and
output of the converter. In a flyback topology, energy is stored
in the inductor during one half portion of the switching cycle
and is then transferred to the output filter during the next half
portion. With two interleaved power trains, energy is transferred
to the output filter during both halves of the switching cycle,
resulting in smaller filters to meet the required ripple.
A SYNC pin, referenced to the input return line (Pin 10), is
available to synchronize multiple units to one switching
frequency. This feature is particularly useful in eliminating
beat frequencies which may cause increased output ripple on
paralleled units. A current share pin (ISHARE) is available which
permits paralleled units to share current typically within 5% at
full load.
A low level dc auxiliary voltage supply referenced to the input
return line is provided for miscellaneous system use.
A five pole differential input EMI filter, along with a commonmode EMI capacitor and careful attention to layout parasitics, is
designed to meet all applicable requirements in MIL-STD-461D
when installed in a typical system setup. Due to the higher output
level in this product compared to the 100 W continuous output
products, input stability is more of a concern. As a result, the two
inductors in the internal input EMI filter have smaller values of
inductance to mitigate input stability concerns. The effectiveness of
the internal input EMI filter is, therefore, slightly diminished
compared to these other products. A more detailed discussion
of CE102 and other EMI issues is included in the section
entitled “EMI Considerations.”
PULSED OUTPUT POWER VS. PULSE LENGTH
OBS
The maximum specified pulsed output power in the standard
configuration of the ADDC02808PB is 200 W. This limit is
based on issues of working down to the minimum input
voltage, of providing a reasonable short circuit current limit,
and so on. However, this power level assumes that the
junction temperatures of the converter’s power semiconductor
devices have not exceeded 110°C. For short pulse lengths and
low duty cycles, this condition will be met. Otherwise, the
pulsed output power will have to be reduced to keep the
junction temperatures below 110°C if NAVMAT guidelines are
to be followed.
OLE
TE
The maximum available peak power out is 200 W and is based
on a combination of maximum junction temperatures, maximum
pulse width, and maximum duty cycle. Refer to section entitled,
“Pulsed Output Power vs. Pulse Length,” for peak power
derating curves for varying conditions.
Figures 16 and 17 show the tradeoff that must be made
between the highest allowable pulsed output power and the
pulse length. Notice that for each curve, as the pulse length is
made longer, the pulsed power that causes a 110°C junction
temperature to be reached is lower. The curves are provided
for two baseplate temperatures (25°C and 90°C) and three
average output powers. The duty cycle that corresponds to any
point on a curve can be calculated by dividing the average
power by the pulsed power for that point. The curves
represent typical upper limits; operation anywhere below the
curves is acceptable and will result in cooler junctions.
The unit is compensated for ultrafast transient response with
minimum output voltage deviation. The compensation has been
optimized and output stability insured for an external load
capacitance in the range of 500 µF, 20 mΩ ESR to 4,000 µF,
2.5 mΩ ESR. Peak performance and output stability are dependent on minimizing parasitic inductance and resistance in the connection from the converter to the load.
The converter uses current mode control and employs a high
performance opto-isolator in its feedback path to maintain
isolation between input and output. The control circuit is
designed to give a nearly constant output current as the output
voltage drops from VO nom to VSC during a short circuit
condition. It does not let the current fold back below the
maximum rated output current. The output overvoltage
protection circuitry, which is independent from the normal
feedback loop, protects the load against a break in the remote
sense leads. Remote sense connections, which can be made at
the load, can adjust for voltage drops of as much as 0.25 V dc
between the converter and the load, thereby maintaining an
accurate voltage level at the load.
The ADDC02808PB is designed to deliver a continuous 100
watts to its output while keeping its hottest junction temperature below 110°C with a baseplate temperature of 90°C.
250
MAXIMUM PEAK
POWER LIMIT
200
PEAK POWER – W
10W ave
An input overvoltage protection feature shuts down the converter when the input voltage exceeds (nominally) 52.5 V dc.
150
50W ave
100
MAXIMUM CONTINUOUS
POWER LIMIT
50
An internal temperature sensor shuts down the unit and
prevents it from becoming too hot if the heat removal system
fails. The temperature sensed is the case temperature and is
factory set to trip at a nominal case temperature of 110°C to
115°C. The shut down temperature setting can be raised
externally or disabled by the user.
0
0
100
200
300
400
500
PULSE WIDTH – ms
600
700
Figure 16. Largest On-State Power vs. Pulse Width that
Maintains TJMAX ≤ 110°C at 25°C Baseplate
Each unit has an INHIBIT pin that can be used to turn off the
converter. This feature can be used to sequence the turn-on of
REV. 0
100W ave
–7–
ADDC02808PB
250
8.1
MAXIMUM PEAK
POWER LIMIT
8
200
7.9
150
VO – V
PEAK POWER – W
10W ave
100W ave
50W ave
12A STEP CHANGE
7.7
100
0
24A STEP CHANGE
7.6
MAXIMUM CONTINUOUS
POWER LIMIT
7.5
50
7.4
–200 –100
0
25
75
50
100
PULSE WIDTH – ms
125
OBS
TRANSIENT RESPONSE
100
200
300
400
500
600
700
800
TIME – µs
Figure 19. Predicted Response to 12 A and 24 A Step
Change in Load Current, di/dt = 12 A/µ s, for CLOAD =
1000 µ F and RESR = 10 mΩ
Step Change
If the step change is less than 24 A, the pulse response will improve. For instance, with a 12 A step change, Figure 19 shows
a comparison of the response for a 24 A step change and a 12 A
step change in load.
OLE
The standard ADDC02808PB is designed to deliver large
changes, or pulses, in load current with minimum output
voltage deviation and an ultrafast return to the nominal output
voltage. The compensation of the feedback loop is optimized,
and output stability is insured, for a broad range of external load
capacitance extending from 500 µF (RESR = 20 mΩ) to 4,000 µF
(RESR = 2.5 mΩ). The variables that impact pulse performance
(the maximum output voltage deviation and the settling time)
are:
1. Size of step change in the output current.
2. Amount of external load capacitance.
3. Internal compensation of the feedback loop (factory set).
4. Connection from converter output to load.
Extensive modeling of the converter with ADI proprietary
software permits analysis and prediction of the impact each of
these parameters has on the pulse response. The analyses in this
data sheet are based on the load capacitance being comprised of
100 µF, 100 mΩ tantalum load capacitors such as the CSR21
style. Figure 18 is the prediction of the standard converter’s
response to a 24 A step change in load current (from 1 A to
25 A) with a load capacitance of 1,000 µF (RESR = 10 mΩ).
This is very close to the measured pulse response under the
same conditions shown in Figure 6.
TE
Load Capacitance
Varying the external load capacitance and associated RESR between the range of CLOAD = 500 µF (RESR = 20 mΩ) and CLOAD
= 4,000 µF (RESR = 2.5 mΩ) results in the predicted waveforms
shown in Figures 20, 21, and 22. As can be seen, the larger the
capacitor, the smaller the deviation, but the longer the settling
time. Table I lists the maximum output voltage deviations and
settling times for the four combinations of CLOAD and RESR mentioned above. Note that these are based on the standard compensation for the feedback loop.
Table I. Output Response to a 24 A (1 A–25 A) Step in Load
Current (Standard Compensation)
Typical
Settling Time
See
CLOAD
RESR
Deviation
(Within 1%)
Figure
500 µF
1,000 µF
2,000 µF
4,000 µF
20 mΩ
10 mΩ
5 mΩ
2.5 mΩ
–7%
–6%
–5%
–4%
150 µs
175 µs
200 µs
250 µs
20
18
21
22
8.1
8.1
8
7.9
7.9
7.8
7.8
VO – V
8
7.7
7.7
7.6
7.6
7.5
7.5
7.4
–200 –100
0
150
Figure 17. Largest On-State Power vs. Pulse Width that
Maintains TJMAX ≤ 110°C at 90°C Baseplate
VO – V
7.8
0
100
200
300
400
500
600
700
7.4
–200 –100
800
0
100
200
300
400
500
600
700
800
TIME – µs
TIME – µs
Figure 20. Predicted Response for 24 A Step Load
Change in Load Current, di/dt = 12 A/µ s, for CLOAD
= 500 µ F and RESR = 20 mΩ
Figure 18. Predicted Response to 24 A Step Change in
Load Current, di/dt = 12 A/µ s, for CLOAD = 1,000 µ F and
RESR = 10 mΩ
–8–
REV. 0
ADDC02808PB
8.1
Table II. Output Response to a 24 A (1 A-25 A) Step in
Load Current (Compensation Optimized)
8
Typical
Settling Time
See
CLOAD
RESR
Deviation
(Within 1%)
Figure
1,000 µF
2,000 µF
4,000 µF
10 mΩ
5 mΩ
2.5 mΩ
–4%
–2.5%
–1%
125 µs
100 µs
0 µs
23
24
25
VO – V
7.9
7.8
7.7
7.6
Connection to Load
Pulse performance is dependent on minimizing the parasitic impedances in the connection between the converter output and
the load and external capacitors. Low inductance and low resistance connections should be used. Multilayer connections
should be avoided to minimize stray capacitance. The converter
should be placed as close to the load and external capacitors as
possible.
7.5
7.4
–200 –100
0
100
200
300
400
500
OBS
600
700
800
TIME – µs
Figure 21. Predicted Response for 24 A Step Load
Change in Load Current, di/dt = 12 A/µ s, for CLOAD =
2,000 µ F and RESR = 5 mΩ
OLE
8
VO – V
7.9
7.8
7.7
8.1
8
7.9
VO – V
8.1
7.8
7.7
7.6
7.6
7.5
7.5
7.4
–200 –100
7.4
–200 –100
0
100
200
300
400
500
600
700
0
100
TE
200
300
400
500
600
700
800
TIME – µs
800
TIME – µs
Figure 23. Predicted Response for 24 A Step Load Change,
di/dt = 12 A/µs, with Factory Set Internal Compensation
Optimized for CLOAD = 1,000 µF and RESR = 10 mΩ
Figure 22. Predicted Response for 24 A Step Load
Change in Load Current, di/dt = 12 A/µ s, for CLOAD =
4,000 µ F and RESR = 2.5 mΩ
8.1
Factory Set Internal Compensation
If the user knows the external load capacitance and RESR to be
used in the application and if the application requires better
pulse response than is summarized in Table I, then the internal
feedback compensation can be modified at the factory to improve the transient response. In these instances, the compensation is optimized for a particular CLOAD at the expense of
performing well over a broader range of CLOAD. The predicted
maximum output voltage deviation and settling times for factory-modified feedback compensation are shown in Figures 23,
24, and 25, and summarized in Table II, for three combinations
of CLOAD and RESR. As can be seen, optimizing the compensation for a given load capacitance gives the best transient response in terms of both voltage deviation and settling time.
8
VO – V
7.9
7.8
7.7
7.6
7.5
7.4
–200 –100
0
100
200
300 400
TIME – µs
500
600
700
800
Figure 24. Predicted Response for 24 A Step Load Change,
di/dt = 12 A/µ s, with Factory Set Internal Compensation
Optimized for CLOAD = 2,000 µ F and RESR = 5 mΩ
REV. 0
–9–
ADDC02808PB
RESR = 10 mΩ. The di/dt is 12 A/µs. As can be seen, the peak
deviations for these curves are close to each other and
comparable to the negative deviation shown in Figure 6 for
a similarly sized positive step change in load current.
8.1
8
VO – V
7.9
7.8
100
7.7
90
VO
100mV
7.6
7.5
10
7.4
–200 –100
0
100
200
300
400
500
600
OBS
700
0%
800
100µs
TIME – µs
Figure 25. Predicted Response for 24 A Step Load Change,
di/dt = 12 A/µ s, with Factory Set Internal Compensation
Optimized for CLOAD = 4,000 µ F and RESR = 2.5 mΩ
Figure 28. Output Voltage Transient Response to a 25 A
to 1 A Step Change in Load, di/dt/ = 12 A/µ s, with
1,000 µ F Load Capacitance (RESR = 10 mΩ)
OLE
RESPONSE AT END OF PULSE
The previous section describes how the ADDC02808PB
converter responds to the positive step change in load current
that occurs at the beginning of a power pulse. This section will
discuss the converter’s response at the end of the power pulse
when the load current is abruptly returned to a small value.
Figures 26-29 show the converter’s measured output voltage as
the load current is stepped from 25 A down to 4 A, 2 A, 1 A, and
0.1 A, respectively. The load capacitance is 1,000 µF with
100
90
100mV
10
0%
100
100mV
90
VO
1ms
Figure 29. Output Voltage Transient Response to a 25 A
to 0.1 A Step Change in Load, di/dt/ = 12 A/µ s, with
1,000 µ F Load Capacitance (RESR = 10 mΩ)
10
0%
100µs
Figure 26. Output Voltage Transient Response to a 25 A
to 4 A Step Change in Load, di/dt/ = 12 A/µ s, with 1,000 µ F
Load Capacitance (RESR = 10 mΩ)
100
100mV
TE
VO
90
VO
10
0%
100µs
Figure 27. Output Voltage Transient Response to a 25 A
to 2 A Step Change in Load, di/dt/ = 12 A/µ s, with 1,000 µ F
Load Capacitance (RESR = 10 mΩ)
What is different about these curves is the settling time. Once
the converter’s output voltage rises above nominal, the
converter cannot help to discharge the load capacitor. It can
only reduce its output current to zero; it cannot draw a negative
current. As such, the time it takes to bring the output voltage
back down to its nominal value depends on the load current
during the low load portion of the cycle. The rate at which the
output voltage falls to its nominal value is the load current
divided by the load capacitance (including the 150 µF
capacitance that is inside the converter). The smaller the load
current, the longer it takes to get the output voltage back to its
nominal value.
During the time that the output voltage is too high, the
integrator in the converter’s feedback circuitry is continuing to
ramp out of range. As the output voltage then falls below its
nominal value, it must have an undershoot error to bring the
integrator back into range. As can be seen from these figures,
the lower the load current, the longer the output voltage
remains too high, and the longer and the greater the output
voltage undershoot is.
Even when the load current steps down to 0.1 A, the maximum
deviation of the output voltage is only about 400 mV, or 5%.
However, it is important to realize that if the next power pulse
occurs before this transient is over, then the output voltage will
–10–
REV. 0
ADDC02808PB
not have the response depicted in the last section. This is
because the feedback integrator will not have had time to return
to its normal state, and so the converter’s ability to respond to a
positive step change in load current will be reduced. The
maximum negative going deviation in the output voltage under
this circumstance will then be greater than is shown in the
figures of that section.
8
7
RESISTANCE – MΩ
6
Should this situation arise, one approach would be to step the
load current down to an intermediate value (e.g., 4 A) at the
end of the power pulse, and then let this current decay to a
smaller value (e.g., 0.1 A) with a time constant in the 100 µs to
200 µs range. This should permit a rapid return to a steady state
condition at the end of the power pulse without requiring a large
average load current during the low power portion of the cycle.
5
4
3
2
1
99
OBS
PIN CONNECTIONS
Pins 1 and 2 (6SENSE)
98
97
OUTPUT VOLTAGE – %
96
95
Figure 30. External Resistor Value for Reducing Output
Voltage
Pins 1 and 2 must always be connected for proper operation,
although failure to make these connections will not be catastrophic to the converter under normal operating conditions.
Pin 1 must always be connected to the output return and Pin 2
must always be connected to +VOUT. These connections can
be made at any one of the output pins of the converter, or
remotely at the load. A remote connection at the load can
adjust for voltage drops of as much as 0.25 V dc between the
converter and the load.
Long remote sense leads can affect converter stability, although
this condition is rare. The impedance of the long power leads
between the converter and the remote sense point could affect
the converter’s unity gain crossover frequency and phase margin.
Consult factory if long remote sense leads are to be used.
OLE
5
RESISTANCE – MΩ
4
3
2
1
0
101
Pin 3 (ADJUST)
An adjustment pin is provided so that the user can change the
nominal output voltage during the prototype stage. Since very
low temperature coefficient resistors are used to set the output
voltage and maintain tight regulation over temperature, using
standard external resistors to adjust the output voltage will
loosen output regulation over temperature. Furthermore, since
the status trip point is not changed when the output voltage is
adjusted using external resistors, the status line will no longer
trip at the standard levels of the newly adjusted output voltage.
If necessary, modified standard units can be ordered with the
necessary changes made inside the package at the factory. The
ADJUST function is sensitive to noise, and care should be
taken in the routing of connections.
To make the output voltage higher, place a resistor from
ADJUST (Pin 3) to –SENSE (Pin 1). To make the output
voltage lower, place a resistor from ADJUST (Pin 3) to
+SENSE (Pin 2). Figures 30 and 31 show resistor values for a
± 5% change in output voltage.
102
TE
103
104
OUTPUT VOLTAGE – %
105
Figure 31. External Resistor for Increasing Output Voltage
Pin 4 (STATUS)
Pin 4 is active high referenced to –SENSE (Pin 1), indicating
that the output voltage is typically within ± 5%. The pin is both
pulled up and down by internal circuitry. Figures 32 and 33
show the typical source and sink capabilities of the status
output. Refer to the paragraphs describing Pin 3 (ADJUST) for
effect on status trip point.
5
VOH – V
4
3
2
With regard to the range that the output voltage can be adjusted
by the user, there are two concerns. As the output voltage is
raised it may become difficult to maintain regulation at full
power and low input voltage. As the output voltage is lowered,
it may become difficult to maintain regulation at minimum
power and high input line.
1
0
0.2
0.4
0.6
0.8
1.0
IOH – mA
1.2
1.4
1.6
Figure 32. Source Capability of Status Output
REV. 0
–11–
ADDC02808PB
When Pin 6 is disconnected from input return, the converter
will restart in the soft-start mode. Pin 6 must be kept low for at
least 2 milli- seconds to initiate a full soft start. Shorter off times
will result in a partial soft start. Figure 35 shows the input
characteristics of Pin 6.
1.0
0.8
0.6
VOL – V
Pin 7 (SYNC)
0.4
0.2
0.0
1.0
4.0
7.0
10.0
IOL – mA
13.0
16.0
19.0
OBS
Figure 33. Sink Capability of Status Output
Pin 5 (VAUX)
Pin 5 is referenced to the input return and provides a semiregulated 14 V to 16 V dc voltage supply for miscellaneous
system use. The maximum permissible current draw is 5 mA
and the voltage varies with the output load of the converter
shown in Figure 34.
OLE
16
15
VAUX – V
13
11
2.5
5.0
7.5 10.0 12.5 15.0 17.5 20.0
CONVERTER OUTPUT CURRENT – A
22.5
25.0
Figure 34.
Pin 6 (INHIBIT)
Pin 6 is active low and is referenced to the input return of the
converter. Connecting it to the input return will turn the
converter off. For normal operation, the inhibit pin is internally
pulled up to 12 V. Use of an open collector circuit is
recommended.
1.2
Frequency: 1.00 MHz min
Duty cycle: 7% min, 14% max
High state voltage high level: 4 V min to 7 V max
Low state voltage low level: 0 V min to 3.0 V max
TE
Pin 8 (ISHARE)
Pin 8 allows paralleled converters to share the total load current, typically within ± 5% at full load. To use the current share
feature, connect all current share pins to each other and connect the SENSE pins on each of the converters. The current
sharing function is sensitive to the differential voltage between the
input return pins of paralleled converters. The current sharing
function is also sensitive to noise, and care should be taken in
the routing of connections. Refer to Figure 45 for typical application circuits using paralleled converters.
Pin 9 (TEMP)
1.1
1.0
I IL – mA
a.
b.
c.
d.
Users should be careful about the frequency selected for the
external master clock. Higher switching frequencies will reduce
efficiency and may reduce the amount of output power available
at minimum input line. Consult factory for modified standard
switching frequency to accommodate system clock
characteristics.
12
0.9
0.8
0.7
0.5
For user-supplied master clocks with no external circuitry, the
following specifications must be met:
Users should note that the SYNC pin is referenced to the input
return of the converter. If the user-supplied master clock is
generated on the output side of the converter, the signal should
be isolated.
14
10
0.0
Pin 7 can be used for connecting multiple converters to a master
clock. This master clock can be either an externally usersupplied clock or it can be a converter that has been modified
and designated as a master unit. Consult factory for availability
of these devices. Capacitive coupling of the clock signal will
insure that if the master clock stops working the individual units
will continue to operate at their own internal clock frequency,
thereby eliminating a potential single point failure. Capacitive
coupling will also permit a wider duty cycle to be used. Consult
factory for more information. The SYNC pin has an internal
pull-down so it is not necessary to sink any current when driving
the pin low.
1.0
1.5
2.0
Pin 9 can be used to indicate case temperature or to raise or
disable the temperature at which thermal shutdown occurs.
Typically, 3.90 V corresponds to +25°C, with a +13.1 mV/°C
change for every 1°C rise. The sensor IC (connected from Pin
9 to the input return (Pin 10)) has a 13.1 kΩ impedance.
The thermal shutdown feature has been set to shut down the
converter when the case temperature is nominally 110°C to
115°C. To raise the temperature at which shutdown occurs,
connect a resistor with the value shown in Figure 36 from Pin 9
to the input return (Pin 10). To completely disable the
temperature shutdown feature, connect a 50 kΩ resistor from
Pin 9 to the input return (Pin 10).
VI L – V
Figure 35. Input Characteristics of Pin 6 When Pulled Low
–12–
REV. 0
ADDC02808PB
Once the pulse is over, the converter initiates a soft-start, which
is completed before the next pulse. No degradation of converter
performance occurs.
1400
RESISTANCE – kΩ
1200
1000
THERMAL CHARACTERISTICS
800
600
400
200
0
120
125
130
135
140
145
SHUTDOWN CASE TEMPERATURE – °C
150
OBS
Figure 36. External Resistor Value for Raising
Temperature Shutdown Point
INPUT VOLTAGE RANGE
The steady state operating input voltage range for the converter
is defined as 18 V to 40 V. The abnormal operating input
voltage range is defined as 16 V to 50 V. In accordance with
MIL-STD-704D, the converter can operate up to 50 V dc input
for transient conditions as long as 50 milliseconds, and it can
operate down to 16 V dc input for continuous operation during
emergency conditions. Figure 3 (typical low line dropout vs.
load) shows that the converter can work continuously down to
and below 16 V dc under reduced load conditions.
The ADDC02808PB can be modified to survive, but not work
through, the upper limit input voltages defined in MIL-STD704A (aircraft) and MIL-STD-1275A (military vehicles). MILSTD-704A defines an 80 V surge that lasts for 1 second before
it falls below 50 V, while MIL-STD-1275A defines a 100 V
surge that lasts for 200 milliseconds before it falls below 50 V.
In both cases, the ADDC02808PB can be modified to operate
to specification up to the 50 V input voltage limit and to shut
down and protect itself during the time the input voltage
exceeds 50 V. When the input voltage falls below 50 V as the
surge ends, the converter will automatically initiate a soft start.
In order to survive these higher input voltage surges, the modified
converter will no longer have input transient protection, however, as described below.
Contact the factory for information on units surviving high
input voltage surges.
Junction and Case Temperatures: It is important for the
user to know how hot the hottest semiconductor junctions
within the converter get and to understand the relationship
between junction, case, and ambient temperatures. The hottest
semiconductors in the 100 W product line of Analog Devices’
high density power supplies are the switching MOSFETs and
the output rectifiers. There is an area inside the main power
transformers that is hotter than these semiconductors, but it is
within NAVMAT guidelines and well below the Curie temperature of the ferrite. (The Curie temperature is the point at which
the ferrite begins to lose its magnetic properties.)
Since NAVMAT guidelines require that the maximum junction
temperature be 110°C, the power supply manufacturer must
specify the temperature rise above the case for the hottest semiconductors so the user can determine what case temperature
is required to meet NAVMAT guidelines. The thermal characteristics section of the specification table states the hottest junction
temperature for maximum output power at a specified case
temperature. The unit can operate to higher case temperatures
than 90°C, but 90°C is the maximum temperature that permits
NAVMAT guidelines to be met.
OLE
Input Voltage Transient Protection: The converter has a
transient voltage suppressor connected across its input leads to
protect the unit against high voltage pulses (both positive and
negative) of short duration. With the power supply connected
in the typical system setup shown in Figure 15, a transient
voltage pulse is created across the converter in the following
manner. A 20 µF capacitor is first charged to 400 V. It is then
connected directly across the converter’s end of the two meter
power lead cable through a 2 Ω on-state resistance MOSFET.
The duration of this connection is 10 µs. The pulse is repeated
every second for 30 minutes. This test is repeated with the
connection of the 20 µF capacitor reversed to create a negative
pulse on the supply leads. (If continuous reverse voltage
protection is required, a diode can be added externally in series
at the expense of lower efficiency for the power system.)
The converter responds to this input transient voltage test by
shutting down due to its input overvoltage protection feature.
REV. 0
TE
Case and Ambient Temperatures: It is the user’s
responsibility to properly heat sink the power supply in order to
maintain the appropriate case temperature and, in turn, the
maximum junction temperature. Maintaining the appropriate
case temperature is a function of the ambient temperature and
the mechanical heat removal system. The static relationship of
these variables is established by the following formula:
TC = T A + ( P D × Rθ
CA
)
where
TC
TA
PD
RθCA
= case temperature measured at the center of the package
bottom,
= ambient temperature of the air available for cooling,
= the power, in watts, dissipated in the power supply,
= the thermal resistance from the center of the package
to free air, or case to ambient.
The power dissipated in the power supply, PD, can be calculated
from the efficiency, h, given in the data sheets and the actual
output power, PO , in the user’s application by the following
formula:


P D = PO  1
– 1
η


For example, at 80 W of output power and 80% efficiency, the
power dissipated in the power supply is 20 W. If under these
conditions, the user wants to maintain NAVMAT deratings
(i.e., a case temperature of approximately 90°C) with an
ambient temperature of 75°C, the required thermal resistance,
case to ambient, can be calculated as
90 = 75 + (20 × RθCA)
or RθCA = 0.75°C/W
This thermal resistance, case to ambient, will determine what
kind of heat sink and whether convection cooling or forced air
cooling is required to meet the constraints of the system.
–13–
ADDC02808PB
SYSTEM INSTABILITY CONSIDERATIONS
In a distributed power supply architecture, a power source
provides power to many “point-of-load” (POL) converters. At
low frequencies, the POL converters appear incrementally as
negative resistance loads. This negative resistance could cause
system instability problems.
Incremental Negative Resistance: A POL converter is
designed to hold its output voltage constant no matter how its
input voltage varies. Given a constant load current, the power
drawn from the input bus is therefore also a constant. If the
input voltage increases by some factor, the input current must
decrease by the same factor to keep the power level constant. In
incremental terms, a positive incremental change in the input
voltage results in a negative incremental change in the input
current. The POL converter therefore looks, incrementally, as a
negative resistor.
OBS
The value of this negative resistor at a particular operating
point, VIN , IIN , is:
Potential System Instability: The preceding analysis assumes
dc voltages and currents. For ac waveforms the incremental
input model for the POL converter must also include the effects
of its input filter and control loop dynamics. When the POL
converter is connected to a power source, modeled as a voltage
source, VS, in series with an inductor, LS, and some positive
resistor, RS, the network of Figure 37 results.
VS
LS
INPUT
TERMINALS
The analysis so far has assumed the source of power was a
voltage source (e.g., a battery) with some source impedance. In
some cases, this source may be the output of a front-end (FE)
converter. Although each FE converter is different, a model for
a typical one would have an LC output filter driven by a voltage
source whose value was determined by the feedback loop. The
LC filter usually has a high Q, so the compensation of the
feedback loop is chosen to help dampen any oscillations that
result from load transients. In effect, the feedback loop adds
“positive resistance” to the LC network.
When the POL converter is connected to the output of this FE
converter, the POL’s “negative resistance” counteracts the
effects of the FE’s “positive resistance” offered by the feedback
loop. Depending on the specific details, this might simply mean
that the FE converter’s transient response is slightly more
oscillatory, or it may cause the entire system to be unstable.
OLE
–VIN
I IN
Note that this resistance is a function of the operating point. At
full load and low input line, the resistance is its smallest, while
at light load and high input line, it is its largest.
RN =
RS
LISN requirement, one possible solution is to place a capacitor
across the input of the POL converter. Another possibility is to
place a small resistor in series with this extra capacitor.
LP
CP
–|RN|
ADI DC/DC CONVERTER
Figure 37. Model of Power Source and POL Converter
Connection
For the ADDC02808PB, LP is approximately 0.5 µH and CP is
approximately 4 µF. Figure 8 shows a more accurate depiction
of the input impedance of the converter as a function of
frequency. The negative resistance is, itself, a very good
incremental model for the power state of the converter for
frequencies into the several kHz range (see Figure 8).
NAVMAT DERATING
Resistors
The network shown in Figure 37 is second order and has the
following characteristic equation:
80% voltage derating
50% power derating
 (L + LP )

s (LS + LP )C + s  S
+ RS CP  + 1 = 0
–
R
|
|


N
For the power delivery to be efficient, it is required that RS <<
RN. For the system to be stable, however, the following
relationship must hold:
Capacitors
2
(L + L )
(L + L )
S
P
P
or
R > S
S
R
C |R |
S
P N
Notice from this result that if (L S + L P) is too large, or if RS is
too small, the system might be unstable. This condition would
first be observed at low input line and full load since the
absolute value of RN is smallest at this operating condition.
C |R |>
P N
If an instability results and it cannot be corrected by changing
LS or RS, such as during the MIL-STD-461D tests due to the
TE
NAVMAT is a Navy power supply reliability manual that is
frequently cited by specifiers of power supplies. A key section
of NAVMAT P4855-1A discusses guidelines for derating designs
and their components. The two key derating criteria are voltage
derating and power derating. Voltage derating is done to reduce
the possibility of electrical breakdown, whereas power derating
is done to maintain the component material below a specified
maximum temperature. While power deratings are typically
stated in terms of current limits (e.g., derate to x% of maximum
rating), NAVMAT also specifies a maximum junction temperature of the semiconductor devices in a power supply. The
NAVMAT component deratings applicable to the ADDC02808PB
are as follows:
50% voltage and ripple voltage derating
70% ripple current derating
Transformers and Inductors
60% continuous voltage and current derating
90% surge voltage and current derating
20°C less than rated core temperature
30°C below insulation rating for hot spot temperature
25% insulation breakdown voltage derating
40°C maximum temperature rise
Transistors
50% power derating
60% forward current (continuous) derating
75% voltage and transient peak voltage derating
110°C maximum junction temperature
–14–
REV. 0
ADDC02808PB
Diodes (Switching, General Purpose, Rectifiers)
ADDC02805SA dc/dc converter (28 VIN, 5 VOUT, 100 W)
using the test setup shown in Figure 15. The EMI performance
of the ADDC02808PB dc/dc converter will be different for
several reasons. The purpose of this section is to describe the
various MIL-STD-461D baseline tests and the
ADDC02805SA converter’s corresponding performance and
then explain how the EMI performance of the ADDC02808PB
will differ from this baseline.
70% current (surge and continuous) derating
65% peak inverse voltage derating
110°C maximum junction temperature
Diodes (Zeners)
70% surge current derating
60% continuous current derating
50% power derating
110°C maximum junction temperature
28 VIN, 100 W Out, Baseline Performance: The
ADDC02805SA has an integral differential- and commonmode EMI filter that is designed to meet all applicable
requirements in MIL-STD-461D when the power converter is
installed in a typical system setup (described below). The
converter also contains transient protection circuitry that
permits the unit to survive short, high voltage transients across
its input power leads.
Microcircuits (Linears)
70% continuous current derating
75% signal voltage derating
110°C maximum junction temperature
OBS
The ADDC02808PB can meet all the derating criteria listed
above. However, there are a few areas of the NAVMAT
deratings where meeting the guidelines unduly sacrifices
performance of the circuit. Therefore, the standard unit makes
the following exceptions.
Electromagnetic interference (EMI) is governed by MIL-STD461D, which establishes design requirements, and MIL-STD462D, which defines test methods. EMI requirements are
categorized as follows (xxx designates a three digit number):
OLE
Common-Mode EMI Filter Capacitors: The standard
supply uses 500 V capacitors to filter common-mode EMI.
NAVMAT guidelines would require 1000 V capacitors to meet
the 50% voltage derating (500 V dc input to output isolation),
resulting in less common-mode capacitance for the same space.
In typical electrical power supply systems, where the load
ground is eventually connected to the source ground, commonmode voltages never get near the 500 V dc rating of the standard
supply. Therefore, a lower voltage rating capacitor (500 V)
was chosen to fit more capacitance in the same space in order
to better meet the conducted emissions requirement of MILSTD-461D (CE102). For those applications which require 250
V or less of isolation from input to output, the present designs
would meet NAVMAT guidelines.
Switching Transistors: 100 V MOSFETs are used in the
standard unit to switch the primary side of the transformers.
Their nominal off-state voltage meets the NAVMAT derating
guidelines. When the MOSFETs are turned off, however,
momentary spikes occur that reach 100 V. The present
generation of MOSFETs are rated for repetitive avalanche, a
condition that was not considered by the NAVMAT deratings.
In the worst case condition, the energy dissipated during
avalanche is 1% of the device’s rated repetitive avalanche
energy. To meet the NAVMAT derating, 200 V MOSFETs
could be used. The 100 V MOSFETs are used instead for their
lower on-state resistance, resulting in higher efficiency for the
power supply.
NAVMAT Junction Temperatures: The two types of power
deratings (current and temperature) can be independent of one
another. For instance, a switching diode can meet its derating
of 70% of its maximum current, but its junction temperature
can be higher than 110°C if the case temperature of the
converter, which is not controlled by the manufacturer, is
allowed to go higher. Since some users may choose to operate
the power supply at a case temperature higher than 90°C, it then
becomes important to know the temperature rise of the hottest
semiconductors. This is covered in the specification table in the
section entitled “Thermal Characteristics”.
EMI CONSIDERATIONS
• CExxx: conducted emissions (EMI produced internal to the
power supply which is conducted externally through its input
power leads)
TE
• CSxxx: conducted susceptibility (EMI produced external to
the power supply which is conducted internally through the
input power leads and may interfere with the supply’s
operation)
• RExxx: radiated emissions (EMI produced internal to the
power supply which is radiated into the surrounding space)
• RSxxx: radiated susceptibility (EMI produced external to
the power supply which radiates into or through the power
supply and may interfere with its proper operation)
It should be noted that there are several areas of ambiguity with
respect to CE102 measurements that may concern the systems
engineer. One area of ambiguity in this measurement is the
nature of the load. If it is constant, then the ripple voltage on
the converter’s input leads is due only to the operation of the
converter. If, on the other hand, the load is changing over
time, this variation causes an additional input current and
voltage ripple to be drawn at the same frequency. If the
frequency is high enough, the converter’s filter will help
attenuate this second source of ripple, but if it is below
approximately 100 kHz, it will not. The system may then not
meet the CE102 requirement, even though the converter is not
the source of the EMI. If this is the case, additional
capacitance may be needed across the load or across the input
to the converter.
Another ambiguity in the CE102 measurement concerns
common-mode voltage. If the load is left unconnected from the
ground plane (even though the case is grounded), the commonmode ripple voltages will be smaller than if the load is
grounded. The test specifications do not state which procedure
should be used. However, in neither case (load grounded or
floating) will the typical EMI test setup described below be
exactly representative of the final system configuration EMI
test. For the following reasons, the same is true if separately
packaged EMI filters are used.
Figures 11 through 14 show the results of EMI measurements
conducted in accordance with MIL-STD-461D/462D for the
REV. 0
–15–
ADDC02808PB
In almost all systems the output ground of the converter is
ultimately connected to the input ground of the system. The
parasitic capacitances and inductances in this connection will
affect the common-mode voltage and the CE102
measurement. In addition, the inductive impedance of this
ground connection can cause resonances, thereby affecting the
performance of the common-mode filter in the power supply.
In response to these ambiguities, the Analog Devices’
converter has been tested for CE102 under a constant load
and with the output ground floating. While these
measurements are a good indication of how the converter will
operate in the final system configuration, the user should
confirm CE102 testing in the final system configuration.
OBS
CE101: This test measures emissions on the input leads in the
frequency range between 30 Hz and 10 kHz. The intent of
this requirement is to ensure that the dc/dc converter does not
corrupt the power quality (allowable voltage distortion) on the
power busses present on the platform. There are several
CE101 limit curves in MIL-STD-461D. The most stringent
one app-licable for the converter is the one for submarine
applications. Figure 11 shows that the converter easily meets
this requirement (the return line measurement is similar). The
components at 60 Hz and its harmonics are a result of ripple in
the output of the power source used to supply the converter.
CS114: This test measures the ability of the converter to
operate correctly during and after being subjected to currents
injected into bulk cables in the 10 kHz to 400 MHz range. Its
purpose is to simulate currents that would be developed in these
cables due to electromagnetic fields generated by antenna
transmissions. The converter is designed to meet the
requirements of this test when the current is injected on the
input power leads cable. Consult factory for more information.
CS115: This test measures the ability of the converter to
operate correctly during and after being subjected to 30 ns long
pulses of current injected into bulk cables. Its purpose is to
simulate transients caused by lightning or electromagnetic
pulses. The converter is designed to meet this requirement
when applied to its input power leads cable. Consult factory for
more information.
CS116: This test measures the ability of the converter to
operate correctly during and after being subjected to damped
sinusoid transients in the 10 kHz to 100 MHz range. Its
purpose is to simulate current and voltage waveforms that would
occur when natural resonances in the system are excited. The
converter is designed to meet this requirement when applied to
its input power leads cable. Consult factory for more information.
OLE
CE102: This test measures emissions in the frequency range
between 10 kHz and 10 MHz. The measurements are made
on both of the input leads of the converter which are
connected to the power source through LISNs. The intent of
this requirement in the lower frequency portion of the
requirement is to ensure that the dc/dc converter does not
corrupt the power quality (allowable voltage distortion) on the
power busses present on the platform. At higher frequencies,
the intent is to serve as a separate control from RE102 on
potential radiation from power leads which may couple into
sensitive electronic equipment.
Figure 12 shows the CE102 limit and the measurement taken
from the +VIN line. While the measurement taken from the
input return line is slightly different, both comfortably meet
the MIL-STD-461D, CE102 limit.
CS101: This test measures the ability of the converter to
reject low frequency differential signals, 30 Hz to 50 kHz,
injected on the dc inputs. The measurement is taken on the
output power leads. The intent is to ensure that equipment
performance is not degraded from ripple voltages associated
with allowable distortion of power source voltage waveforms.
Figure 7 shows a typical audio susceptibility graph. Note that
according to the MIL-STD-461D test requirements, the
injected signal between 30 Hz and 5 kHz has an amplitude of
2 V rms and from 5 kHz to 50 kHz the amplitude decreases
inversely with frequency to 0.2 V rms. The curve of the
injected signal should be multiplied by the audio susceptibility
curve to determine the output ripple at any frequency. When
this is done, the worst case output ripple at the frequency of
the input ripple occurs at 5 kHz, at which point there is
typically a 25 mV peak-to-peak output ripple.
It should be noted that MIL-STD-704 has a more relaxed
requirement for rejection of low frequency differential signals
injected on the dc inputs than MIL-STD-461D. MIL-STD704 calls for a lower amplitude ripple to be injected on the
input in a narrower frequency band, 10 Hz to 20 kHz.
RE101: This requirement limits the strength of the magnetic
field created by the converter in order to avoid interference with
sensitive equipment located nearby. The measurement is made
from 30 Hz to 100 kHz. The most stringent requirement is for
the Navy. Figure 13 shows the test results when the pickup coil
is held 7 cm above the converter. As can be seen, the converter
easily meets this requirement.
TE
RE102: This requirements limits the strength of the electric
field emissions from the power converter to protect sensitive
receivers from interference. The measurement is made from
10 kHz to 18 GHz with the antenna oriented in the vertical
plane. For the 30 MHz and above range the standard calls for
the measurement to be made with the antenna oriented in the
horizontal plane, as well.
In a typical power converter system setup, the radiated emissions can come from two sources: 1) the input power leads as
they extend over the two meter distance between the LISNs and
the converter, as required for this test, and 2) the converter
output leads and load. The latter is likely to create significant
emissions if left uncovered since minimal EMI filtering is
provided at the converter’s output. It is typical, however,
that the power supply and its load would be contained in a
conductive enclosure in applications where this test is applicable.
A metal screen enclosure was therefore used to cover the
converter and its load for this test.
Figure 14 shows test results for the vertical measurement and
compares them against the most stringent RE102 requirement;
the horizontal measurement (30 MHz and above) was similar.
As can be seen, the emissions just meet the standard in the
18 MHz–28 MHz range. This component of the emissions is
due to common-mode currents flowing through the input power
leads. As mentioned in the section on CE102 above, the level
of common-mode current that flows is dependent on how the
load is connected. This measurement is therefore a good
indication of how well the converter will perform in the final
configuration, but the user should confirm RE102 testing in the
final system.
–16–
REV. 0
ADDC02808PB
3. A repetitively pulsed load will cause large input currents at
the fundamental frequency (and harmonics) of the pulse
waveform.
RS101: This requirement is specialized and is intended to check
for sensitivity to low frequency magnetic fields in the 30 Hz to
50 kHz range. The converter is designed to meet this requirement. Consult factory for more information.
RS103: This test calls for correct operation during and after the
unit under test is subjected to radiated electric fields in the 10 kHz
to 40 GHz range. The intent is to simulate electromagnetic
fields generated by antenna transmissions. The converter is
designed to meet this requirement. Consult factory for more
information.
Circuit Setup for EMI Test
Figure 15 shows a schematic of the test setup used for the EMI
measurements discussed above. The output of the converter is
connected to a resistive load designed to draw full power. There
is a 0.1 µF capacitor placed across this resistor that typifies bypass capacitance normally used in this application. At the input
of the converter there are two differential capacitors (the larger
one having a series resistance) and two small common-mode
capacitors connected to case ground. The case itself was connected to the metal ground plane in the test chamber. For the
RE102 test, a metal screen box was used to cover both the
converter and its load (but not the two meters of input power
lead cables). This box was also electrically connected to the
metal ground plane.
OBS
The result of Items 1 and 2 is that the ADDC02808PB
converter will have higher conducted and radiated emissions in
the 1/2 MHz to 10 MHz range. The emissions in this range are
dominated by differential currents. These currents are
proportional to power, so we would expect a factor of two
increase in emissions due to the 200 W operating level. It does
not matter that the average power of this pulsed unit is 100 W
or lower. MIL-STD-462D calls for measurements to be made
with peak detectors that will determine the emissions during the
200 W pulse, and not average them in any way with the lower
power part of the cycle.
In addition, the differential EMI filter in the ADDC02808PB
converter is less effective at attenuating the ripple currents than
is the filter in the ADDC02805SA converter due to smaller
value inductors. Figure 38 shows the transfer functions of these
two filters in the frequency range of interest.
OLE
TE
For both conducted and radiated tests, this increase would
require some additional differential filtering to meet the most
stringent MIL-STD-461D levels shown in the figures. This
could be done, for example, by increasing the 2 µF ceramic (low
parasitic inductance) capacitor placed across the input of the
converter in Figure 15 to 30 µF or, a small 0.5 µH, 16 A
inductor could be placed in series between the top of the 2 µF
capacitor and the +VIN pin. Figures 40 and 41 show the ratios,
G(S) – TRANSFER FUNCTION (DIMENSIONLESS)
With regard to the components added to the input power lines,
the 100 µF capacitor with its 1 Ω series resistance is required to
achieve system stability when the unit is powered through the
LISNs, as the MIL-STD-461D standard requires. These
LISNs have a series inductance of 50 µH at low frequencies,
giving a total differential inductance of 100 µH. As explained
earlier in the System Instability section, such a large series
source inductance will cause an instability as it interacts with the
converter’s negative incremental input resistance unless some
corrective action is taken. The 100 µF capacitor and 1 Ω
resistor provide the stabilization required.
Combining the factor of two and the reduced filter attenuation,
Figure 39 shows the ratio, in dB, by which the emissions of
Figures 12 and 14 should be increased to estimate the emissions
of the ADDC02808PB converter in this frequency range. From
this curve the 1/2 MHz component should increase by 25 dB,
and the 1 MHz and higher components should increase by 22 dB.
It should be noted that the values of these stabilization components
are appropriate for a single converter load. If the system makes
use of several converters, the values of the components will need
to be changed slightly, but not such that they are repeated for
every converter. It should also be noted that most system
applications will not have a source inductance as large as the
100 µH built into the LISNs. For those systems, a much
smaller input capacitor could be used.
The 2 µF differential-mode capacitor and the two 82 nF
common-mode capacitors were added to achieve the results
shown in the EMI measurement figures described above.
ADDC02808PB EMI Performance
The EMI performance of the ADDC02808PB power converter
will be different from the ADDC02805SA baseline previously
discussed for several reasons:
1. Its maximum power is 200 W, or twice that of the
ADDC02805SA converter.
2. Its differential input filter inductors are smaller in value by a
factor of two compared to those in the ADDC02805SA
converter to accommodate input stability at the higher power
level.
REV. 0
100
10
1
0.1
0.01
0.001
ADDC02808PB
10–4
10–5
ADDC02805SA
10–6
10–7
10–8
1•104
1•105
1•106
FREQUENCY – Hz
1•107
Figure 38. Comparison of Transfer Functions for the Input
EMI Filters in ADDC02805SA and ADDC02808PB
–17–
ADDC02808PB
30
The peak in the radiated emissions in the 20 MHz–30 MHz
range of Figure 14 is dominated by common-mode noise. This
common-mode noise emission is changed only slightly between
the ADDC02808PB and the ADDC02805SA converters since it
does not depend on the power level or the differential input
filter. The turns ratio on the transformer has been changed, so
we expect the common-mode emissions might be 2–4 times
larger. This increase could be countered by increasing the 82 nF
common-mode capacitors of Figure 15 correspondingly. Again,
this solution is a suggestion; it has not been tested.
20
dB
10
0
–10
–20
1•105
1•106
FREQUENCY – Hz
OBS
1•107
Figure 39. Change in ADDC02808PB Differential Emissions vs. ADDC02805SA Emissions with the Same Test
Setup
OLE
in dB, by which the differential emissions would change if either
of these approaches were followed. Notice that the inductor
solution provides substantial attenuation in the 1 MHz and
higher frequency range, while the larger capacitor solution has a
more uniform effect. These proposed solutions are suggestions;
they have not been tested.
30
20
dB
0
RELIABILITY CONSIDERATIONS
MTBF (Mean Time Between Failure) is a commonly used
reliability concept that applies to repairable items in which
failed elements are replaced upon failure. The expression for
MTBF is
–10
1•106
FREQUENCY – Hz
MTBF = T/r
1•107
where
Figure 40. Change in ADDC02808PB Differential Emissions
vs. the ADDC02805SA Emissions with External 30 µ F
Capacitor
30
T = total operating time
r = number of failures
In lieu of actual field data, MTBF can be predicted per
MIL-HDBK-217.
MTBF, Failure Rate, and Probability of Failure: A proper
understanding of MTBF begins with its relationship to lambda
(λ), which is the failure rate. If a constant failure rate is assumed,
then MTBF = 1/λ, or λ = 1/MTBF. If a power supply has an
MTBF of 1,000,000 hours, this does not mean it will last
1,000,000 hours before it fails. Instead, the MTBF describes
the failure rate. For 1,000,000 hours MTBF, the failure rate
during any hour is 1/1,000,000, or 0.0001%. Thus, a power
supply with an MTBF of 500,000 hours would have twice the
failure rate (0.0002%) of one with 1,000,000 hours.
20
dB
10
0
–10
–20
1•105
TE
With the test setup in Figure 15, given the impedances of the
LISNs and the 100 µF capacitor with its 1 Ω series resistance, a
3.6 V rms waveform would result from this fundamental
component of the input current. The MIL-STD-461D limit
shown in Figure 11 calls for approximately 100 mV at the 1 kHz
frequency. If this limit is to be met, substantial filtering at the
lower frequencies will have to be added to the system.
10
–20
1•105
Finally, the pulsed nature of the load means there will be a
substantial ripple in the input current at the fundamental pulse
rate and its harmonics. This ripple can be calculated once the
power is known as a function of time by dividing by the input
voltage. For instance, if the load switches between zero and
200 W (260 W at the input) at 1 kHz with a duty ratio of 50%,
the current drawn by the converter will have a 9.3 A on, 0 A off,
50% duty ratio input current waveform (260 W/28 V = 9.3 A).
This waveform has an average of 4.65 A and a square wave of
plus and minus 4.65 A around this average. This square wave of
current has a fundamental component as well as odd harmonics
(3rd, 5th, 7th, . . .). The peak of the fundamental component is
(4/π) 4.65. The rms value of this component is .707 times the
peak, or 4.2 A.
1•106
FREQUENCY – Hz
1•107
What users should be interested in is the probability of a power
supply not failing prior to some time t. Given the assumption of
a constant failure rate, this probability is defined as
R(t ) = e –λt
Figure 41. Change in ADDC02808PB Differential Emissions
vs. the ADDC02805SA Emissions with External 0.5 µ H,
16 A Inductor
–18–
REV. 0
ADDC02808PB
where R(t) is the probability of a device not failing prior to some
time t.
If we substitute λ = 1/MTBF in the above formula, then the
expression becomes
–t
R(t ) = e MTBF
This formula is the correct way to interpret the meaning of
MTBF.
circuit board through holes. In order to maintain the hermetic
integrity of the seals around the pins, a fixture should be used
for bending the pins without stressing the pin-to-sidewall seals.
It is recommended that the minimum distance between the
package edge and the inside of the pin be 100 mils (2.54 mm)
for the 40 mil (1.02 mm) diameter pins; 120 mils (3.05 mm)
from the package edge to the center of the pin as shown in
Figure 43.
If we assume t = MTBF = 1,000,000 hours, then the probability
that a power supply will not fail prior to 1,000,000 hours of use
is e–1, or 36.8%. This is quite different from saying the power
supply will last 1,000,000 hours before it fails. The probability
that the power supply will not fail prior to 50,000 hours of use is
e–.05, or 95%. For t = 10,000 hours, the probability of no failure
is e–.01, or 99%.
OBS
Temperature and Environmental Factors: Although the
calculation of MTBF per MIL-HDBK-217 is a detailed process,
there are two key variables that give the manufacturer significant leeway in predicting an MTBF rating. These two
variables are temperature and environmental factor. Therefore,
for users to properly compare MTBF numbers from two
different manufacturers, the environmental factor and the
temperature must be identical. Contact the factory for MTBF
calculations for specific environmental factors and temperatures.
MECHANICAL CONSIDERATIONS
0.100"
(2.54mm)
0.120"
(3.05mm)
OLE
Figure 43. Minimum Bend Radius of 40 Mil (1.02 mm)
Pins
PS1
1
2
TE
ADDC02808PB
When mounting the converter into the next higher level
assembly, it is important to insure good thermal contact is made
between the converter and the external heat sink. Poor thermal
connection can result in the converter shutting off, due to the
temperature shutdown feature (Pin 9), or reduced reliability for
the converter due to higher than anticipated junction and case
temperatures. For these reasons the mounting tab locations
were selected to insure good thermal contact is made near the
hot spots of the converter which are shown in the shaded areas
of Figure 42.
10
11
+28VDC
C1
28RTN
17
16
15
14
13
12
RLOAD
NOTE: VALUE OF C1 IS DEPENDENT ON SOURCE IMPEDANCE.
REFER TO SECTION ON SYSTEM INSTABILITY CONSIDERATIONS.
Figure 44. Typical Power Connections and External Parts
for Converter
PS1
1
2
ADDC02808PB
8
10
11
+28VDC
17
16
15
14
13
12
VOUT+
+SENSE PS1
+SENSE PS2
–SENSE PS1
C1
RLOAD
–SENSE PS2
PS2
28RTN
1
2
I SHARE
8
10
11
ADDC02808PB
17
16
15
14
13
12
VOUT –
NOTE: VALUE OF C1 IS DEPENDENT ON SOURCE IMPEDANCE.
REFER TO SECTION ON SYSTEM INSTABILITY CONSIDERATIONS.
Figure 45. Typical Connections for Paralleling Two
Converters
Figure 42. Hot Spots (Shaded Areas) of DC/DC Converter
The pins of the converter are typically connected to the next
higher level assembly by bending them at right angles, either
down or up, and cutting them shorter for insertion in printed
REV. 0
–19–
ADDC02808PB
Screening Steps
Industrial (KV)
Ruggedized Industrial (TV)
MIL-STD-883B/SMD (TV/883B)
Pre-Cap Visual
Temp Cycle
Constant Acceleration
Fine Leak
100%
N/A
N/A
Guaranteed to Meet
MIL-STD-883, TM1014
Guaranteed to Meet
MIL-STD-883, TM1014
N/A
MIL-STD-883, TM2017
N/A
N/A
Guaranteed to Meet
MIL-STD-883, TM1014
Guaranteed to Meet
MIL-STD-883, TM1014
MIL-STD-883, TM1015,
96 Hrs at +115°C Case
At +25°C, Per Specification
Table
Compliant to MIL-PRF-38534
Gross Leak
Burn-In
OBS
Final Electrical Test
At +25°C, Per Specification
Table
C2183–7.5–10/96
Screening Levels for ADDC02808PB
NOMINAL CASE DIMENSIONS IN INCHES AND (mm)
OLE
[All tolerances ± .005" (± .13 mm) unless otherwise specified]
0.150 (3.81)
4 PLCS
0.100 (2.54)
8 PLCS
0.149 (3.78)
DIA TYP
0.300 (7.62) SQ
± 0.010
4 PLCS
TOP VIEW
0.200 (5.08)
0.200 (5.08) 5 PLCS
0.200 (5.08)
0.150 (3.81)
TE
1.500 ± 0.010
(38.10 ± 0.25)
0.150 (3.81)
0.250 (6.35)
2 PLCS
0.800 ± 0.010
(20.32 ± 0.25)
4 PLCS
1.145 (29.08)
2 PLCS
1.800
(45.72)
TYP
2.100 ± 0.010
(53.34 ± 0.25)
0.040 ± 0.003
(1.02 ± 0.08)
2.745 ± 0.010
(69.72 ± 0.25)
0.090 ± 0.010
(2.29 ± 0.25)
0.390 ± 0.010
(9.91 ± 0.25)
2. The package base material is made of molybdenum and is
nominally 40 mils (1.02 mm) thick. The “runout” is less
than 2 mils per inch (0.02 mm per cm).
3. The high current pins (10–17) are 40 mil (1.02 mm) diameter;
are 99.8% copper; and are plated with gold over nickel.
4. The signal carrying pins (1–9) are 18 mil (0.46 mm)
diameter; are Kovar; and are plated with gold over nickel.
5. All pins are a minimum length of 0.740 inches (18.80 mm)
when the product is shipped. The pins are typically bent up
or down and cut shorter for proper connection into the user’s
system.
6. All pin-to-sidewall spacings are guaranteed for a minimum of
500 V dc breakdown at standard air pressure.
7. The case outline was originally designed using the inch-pound
units of measurement. In the event of conflict between the
metric and inch-pound units, the inch-pound shall take
precedence.
–20–
REV. 0
PRINTED IN U.S.A.
NOTES
1. The final product weight is 85 grams maximum.