LT1961 1.5A, 1.25MHz Step-Up Switching Regulator U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO 1.5A Switch in a Small MSOP Package Constant 1.25MHz Switching Frequency Wide Operating Voltage Range: 3V to 25V High Efficiency 0.2Ω Switch 1.2V Feedback Reference Voltage ±2% Overall Output Voltage Tolerance Uses Low Profile Surface Mount External Components Low Shutdown Current: 6μA Synchronizable from 1.5MHz to 2MHz Current-Mode Loop Control Constant Maximum Switch Current Rating at All Duty Cycles* Thermally Enhanced Exposed Pad 8-Lead Plastic MSOP Package U APPLICATIO S ■ ■ ■ ■ DSL Modems Portable Computers Battery-Powered Systems Distributed Power The LT®1961 is a 1.25MHz monolithic boost switching regulator. A high efficiency 1.5A, 0.2Ω switch is included on the die together with all the control circuitry required to complete a high frequency, current-mode switching regulator. Current-mode control provides fast transient response and excellent loop stability. New design techniques achieve high efficiency at high switching frequencies over a wide operating voltage range. A low dropout internal regulator maintains consistent performance over a wide range of inputs from 24V systems to Li-Ion batteries. An operating supply current of 1mA maintains high efficiency, especially at lower output currents. Shutdown reduces quiescent current to 6μA. Maximum switch current remains constant at all duty cycles. Synchronization allows an external logic level signal to increase the internal oscillator from 1.5MHz to 2MHz. The LT1961 is available in an exposed pad, 8-pin MSOP package. Full cycle-by-cycle switch current limit protection and thermal shutdown are provided. High frequency operation allows the reduction of input and output filtering components and permits the use of chip inductors. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. *Patent Pending U TYPICAL APPLICATIO Efficiency vs Load Current 5V to 12V Boost Converter 90 6.8μH VIN 5V 2.2μF CERAMIC 1 VIN VSW VOUT 12V 0.5A* 2 LT1961 OPEN OR 5 6 SHDN FB HIGH VC SYNC GND = ON 8 3,4 7 90.9k 6800pF 100pF 10k 1% 10μF CERAMIC EFFICIENCY (%) 85 UPS120 80 75 70 65 VIN = 5V VOUT = 12V 6.8k 60 *MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING. 1961 TA01 0 100 200 300 400 LOAD CURRENT (mA) 500 1961 TA01a 1961fa 1 LT1961 U U U U W W W ABSOLUTE MAXIMUM RATINGS PI CO FIGURATIO (Note 1) Input Voltage .......................................................... 25V Switch Voltage ......................................................... 35V SHDN Pin ............................................................... 25V FB Pin Current ....................................................... 1mA SYNC Pin Current .................................................. 1mA Operating Junction Temperature Range (Note 2) LT1961E, LT1961I ........................... – 40°C to 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C TOP VIEW VIN SW GND GND 8 7 6 5 1 2 3 4 SYNC VC FB SHDN MS8E PACKAGE 8-LEAD PLASTIC MSOP GROUND PAD CONNECTED TO LARGE COPPER AREA TJMAX = 125°C, θJA = 50°C/W U W U ORDER I FOR ATIO LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT1961EMS8E#PBF LT1961EMS8E#TRPBF LTQY 8-Lead Plastic MSOP –40°C to 125°C LT1961IMS8E#PBF LT1961IMS8E#TRPBF LTQY 8-Lead Plastic MSOP –40°C to 125°C LEAD BASED FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT1961EMS8E LT1961EMS8E#TR LTQY 8-Lead Plastic MSOP –40°C to 125°C LT1961IMS8E LT1961IMS8E#TR LTQY 8-Lead Plastic MSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *Temperature grades are identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VC = 0.8V, SHDN, SYNC and switch open unless otherwise noted. PARAMETER CONDITION MIN Recommended Operating Voltage ● 3 Maximum Switch Current Limit ● 1.5 1 Oscillator Frequency 3.3V < VIN < 25V ● Switch On Voltage Drop ISW = 1.5A ● VIN Undervoltage Lockout (Note 3) ● VIN Supply Current ISW = 0A ● VIN Supply Current/ISW ISW = 1.5A Shutdown Supply Current VSHDN = 0V, VIN = 25V, VSW = 25V 2.47 TYP V 2 3 A 1.5 MHz 310 500 mV 2.6 2.73 V 0.9 1.3 27 3V < VIN < 25V, 0.4V < VC < 0.9V ● 1.182 1.176 UNITS 25 mA mA/A 6 20 45 μA μA 1.2 1.218 1.224 V V ● Feedback Voltage MAX 1961fa 2 LT1961 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VC = 0.8V, SHDN, SYNC and switch open unless otherwise noted. PARAMETER CONDITION FB Input Current MIN TYP MAX UNITS ● 0 – 0.2 – 0.4 μA FB to VC Voltage Gain 0.4V < VC < 0.9V 150 350 FB to VC Transconductance ΔIVC = ±10μA ● 500 850 1300 μMho VC Pin Source Current VFB = 1V ● – 85 – 120 – 165 μA VC Pin Sink Current VFB = 1.4V ● 70 110 165 VC Pin to Switch Current Transconductance VC Pin Minimum Switching Threshold Duty Cycle = 0% VC Pin 1.5A ISW Threshold Maximum Switch Duty Cycle VC = 1.2V, ISW = 100mA VC = 1.2V, ISW = 1A, 25°C ≤ TA ≤ 125°C VC = 1.2V, ISW = 1A, TA ≤ 25°C SHDN Threshold Voltage SHDN Input Current (Shutting Down) SHDN = 60mV Above Threshold SHDN Threshold Current Hysteresis SHDN = 100mV Below Threshold SYNC Pin Resistance A/V 0.3 V 0.9 V ● 80 75 70 90 80 75 % % % ● 1.28 1.35 1.42 V ● –7 –10 –13 μA 4 7 10 μA 1.5 2.2 SYNC Threshold Voltage SYNC Input Frequency 1.5 ISYNC = 1mA Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT1961E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the – 40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT1961I is guaranteed over the – 40ºC to 125ºC operating junction temperature range. μA 2.4 2 20 V MHz kΩ Note 3: Minimum input voltage is defined as the voltage where the internal regulator enters lockout. Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. See Applications Information. 1961fa 3 LT1961 U W TYPICAL PERFORMANCE CHARACTERISTICS FB vs Temperature Switch On Voltage Drop 1.22 Oscillator Frequency 400 1.5 TA = 25°C 125°C 350 1.20 1.19 300 1.4 25°C FREQUENCY (MHz) SWITCH VOLTAGE (mV) FB VOLTAGE (V) 1.21 250 –40°C 200 150 1.3 1.2 100 50 1.18 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 0 125 0 0.5 1 SWITCH CURRENT (A) 1.1 –50 1.5 1961 G01 SHDN Threshold vs Temperature SHDN IP Current vs Temperature TA = 25°C SHDN = 0V 6 –10 SHDN INPUT (μA) VIN CURRENT (μA) SHDN THRESHOLD (V) 5 4 3 2 1.32 0 0 25 50 75 TEMPERATURE (°C) 100 125 0 5 10 15 VIN (V) 20 25 1961 G04 TA = 25°C VIN = 15V 0 –50 30 100 TA = 25°C 0 0.2 0.4 0.6 0.8 1 1.2 SHUTDOWN VOLTAGE (V) 1.4 1961 G07 100 800 MINIMUM INPUT VOLTAGE 600 125 400 40 TA = 25°C 1.5 30 SWITCH CURRENT 1.0 20 0.5 10 200 50 0 25 50 75 TEMPERATURE (°C) Current Limit Foldback SWITCH PEAK CURRENT (A) VIN CURRENT (μA) 150 –25 1961G06 2.0 1000 200 STARTING UP FB CURRENT 0 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 1961 G08 0 0 0.2 0.4 0.6 0.8 FEEDBACK VOLTAGE (V) 1 0 1.2 1961 G09 1961fa 4 FB INPUT CURRENT (μA) VIN CURRENT (μA) –4 Input Supply Current 1200 250 0 –6 1961 G05 SHDN Supply Current 300 SHUTTING DOWN –8 –2 1 –25 125 –12 1.38 1.30 –50 100 1961 G03 SHDN Supply Current vs VIN 7 1.34 0 25 50 75 TEMPERATURE (°C) 1961 G02 1.40 1.36 –25 LT1961 U U U PIN FUNCTIONS FB: The feedback pin is used to set output voltage using an external voltage divider that generates 1.2V at the pin with the desired output voltage. If required, the current limit can be reduced during start up when the FB pin is below 0.5V (see the Current Limit Foldback graph in the Typical Performance Characteristics section). An impedance of less than 5kΩ at the FB pin is needed for this feature to operate. VIN: This pin powers the internal circuitry and internal regulator. Keep the external bypass capacitor close to this pin. GND: Short GND pins 3 and 4 and the exposed pad on the PCB. The GND is the reference for the regulated output, so load regulation will suffer if the “ground” end of the load is not at the same voltage as the GND of the IC. This condition occurs when the load current flows through the metal path between the GND pins and the load ground point. Keep the ground path short between the GND pins and the load and use a ground plane when possible. Keep the path between the input bypass and the GND pins short. The exposed pad should be attached to a large copper area to improve thermal resistance. VSW: The switch pin is the collector of the on-chip power NPN switch and has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. SYNC: The sync pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 20% and 80% duty cycle. The synchronizing range is equal to initial operating frequency, up to 2MHz. See Synchronization section in Applications Information for details. When not in use, this pin should be grounded. SHDN: The shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. The 1.35V threshold can function as an accurate undervoltage lockout (UVLO), preventing the regulator from operating until the input voltage has reached a predetermined level. Float or pull high to put the regulator in the operating mode. VC: The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation, but can do double duty as a current clamp or control loop override. This pin sits at about 0.3V for very light loads and 0.9V at maximum load. 1961fa 5 LT1961 W BLOCK DIAGRAM The LT1961 is a constant frequency, current-mode boost converter. This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. In addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS flip-flop to turn the switch on. When switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. This technique means that the error VIN amplifier commands current to be delivered to the output rather than voltage. A voltage fed system will have low phase shift up to the resonant frequency of the inductor and output capacitor, then an abrupt 180° shift will occur. The current fed system will have 90° phase shift at a much lower frequency, but will not have the additional 90° shift until well beyond the LC resonant frequency. This makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. A comparator connected to the shutdown pin disables the internal regulator, reducing supply current. 1 2.5V BIAS REGULATOR INTERNAL VCC SLOPE COMP Σ 0.3V SYNC 1.25MHz OSCILLATOR 8 S + – SHUTDOWN COMPARATOR 7μA + DRIVER CIRCUITRY RS FLIP-FLOP CURRENT COMPARATOR R 2 SW 6 FB 3 GND Q1 POWER SWITCH CURRENT SENSE AMPLIFIER VOLTAGE GAIN = 40 – + 1.35V – 0.01Ω 5 – 3μA 7 VC ERROR AMPLIFIER gm = 850μMho + SHDN 1.2V 4 GND 1767 F01 Figure 1. Block Diagram 1961fa 6 LT1961 U U W U APPLICATIONS INFORMATION FB RESISTOR NETWORK The suggested resistance (R2) from FB to ground is 10k 1%. This reduces the contribution of FB input bias current to output voltage to less than 0.2%. The formula for the resistor (R1) from VOUT to FB is: R1 = ( ) R2 VOUT − 1. 2 1.2 − R2(0.2μA) VSW LT1961 OUTPUT ERROR AMPLIFIER + 1.2V FB R1 + – Table 1. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E Case Size ESR (Max, Ω ) Ripple Current (A) AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1 AVX TAJ 0.7 to 0.9 0.4 0.1 to 0.3 0.7 to 1.1 0.2 (typ) 0.5 (typ) AVX TPS, Sprague 593D C Case Size 1961 F02 GND Figure 2. Feedback Network OUTPUT CAPACITOR Step-up regulators supply current to the output in pulses. The rise and fall times of these pulses are very fast. The output capacitor is required to reduce the voltage ripple this causes. The RMS ripple current can be calculated from: IRIPPLE(RMS) = IOUT Tantalum capacitors are usually chosen for their bulk capacitance properties, useful in high transient load applications. ESR rather than absolute value defines output ripple at 1.25MHz. Values in the 22μF to 100μF range are generally needed to minimize ESR and meet ripple current ratings. Care should be taken to ensure the ripple ratings are not exceeded. D Case Size R2 10k VC defines the pole frequency of the output stage, an X7R or X5R type ceramic, which have good temperature stability, is recommended. (VOUT − VIN) / VIN The LT1961 will operate with both ceramic and tantalum output capacitors. Ceramic capacitors are generally chosen for their small size, very low ESR (effective series resistance), and good high frequency operation, reducing output ripple voltage. Their low ESR removes a useful zero in the loop frequency response, common to tantalum capacitors. To compensate for this, the VC loop compensation pole frequency must typically be reduced by a factor of 10. Typical ceramic output capacitors are in the 1μF to 10μF range. Since the absolute value of capacitance AVX TPS INPUT CAPACITOR Unlike the output capacitor, RMS ripple current in the input capacitor is normally low enough that ripple current rating is not an issue. The current waveform is triangular, with an RMS value given by: IRIPPLE(RMS) = ( )( ) (L)(f)(VOUT ) 0.29 VIN VOUT − VIN At higher switching frequency, the energy storage requirement of the input capacitor is reduced so values in the range of 1μF to 4.7μF are suitable for most applications. Y5V or similar type ceramics can be used since the absolute value of capacitance is less important and has no significant effect on loop stability. If operation is required close to the minimum input voltage required by either the output or the LT1961, a larger value may be necessary. This is to prevent excessive ripple causing dips below the minimum operating voltage resulting in erratic operation. 1961fa 7 LT1961 U W U U APPLICATIONS INFORMATION INDUCTOR CHOICE AND MAXIMUM OUTPUT CURRENT When choosing an inductor, there are 2 conditions that limit the minimum inductance; required output current, and avoidance of subharmonic oscillation. The maximum output current for the LT1961 in a standard boost converter configuration with an infinitely large inductor is: IOUT (MAX) = 1.5A VIN • η VOUT Where η = converter efficiency (typically 0.87 at high current). As the value of inductance is reduced, ripple current increases and IOUT(MAX) is reduced. The minimum inductance for a required output current is given by: LMIN = VIN (VOUT – VIN ) ⎛ (V )(I )⎞ 2VOUT (f)⎜ 1.5 – OUT OUT ⎟ VIN • η ⎠ ⎝ The second condition, avoidance of subharmonic oscillation, must be met if the operating duty cycle is greater than 50%. The slope compensation circuit within the LT1961 prevents subharmonic oscillation for inductor ripple currents of up to 0.7AP-P, defining the minimum inductor value to be: LMIN = VIN (VOUT – VIN ) 0.7VOUT (f) These conditions define the absolute minimum inductance. However, it is generally recommended that to prevent excessive output noise, and difficulty in obtaining stability, the ripple current is no more than 40% of the average inductor current. Since inductor ripple is: V (V –V ) IP −P RIPPLE = IN OUT IN VOUT (L)(f) The recommended minimum inductance is: LMIN = (VIN )2 (VOUT – VIN ) 0.4(VOUT )2 (IOUT )(f) The inductor value may need further adjustment for other factors such as output voltage ripple and filtering requirements. Remember also, inductance can drop significantly with DC current and manufacturing tolerance. The inductor must have a rating greater than its peak operating current to prevent saturation resulting in efficiency loss. Peak inductor current is given by: ILPEAK = (VOUT )(IOUT ) VIN (VOUT − VIN ) + VIN • η 2VOUT (L)(f) Also, consideration should be given to the DC resistance of the inductor. Inductor resistance contributes directly to the efficiency losses in the overall converter. Suitable inductors are available from Coilcraft, Coiltronics, Dale, Sumida, Toko, Murata, Panasonic and other manufactures. Table 2 PART NUMBER VALUE (uH) ISAT(DC) (Amps) DCR (Ω) HEIGHT (mm) Coiltronics TP1-2R2 2.2 1.3 0.188 1.8 TP2-2R2 2.2 1.5 0.111 2.2 TP3-4R7 4.7 1.5 0.181 2.2 TP4- 100 10 1.5 0.146 3.0 LQH1C1R0M04 1.0 0.51 0.28 1.8 LQH3C1R0M24 1.0 1.0 0.06 2.0 LQH3C2R2M24 2.2 0.79 0.1 2.0 LQH4C1R5M04 1.5 1 0.09 2.6 CD73- 100 10 1.44 0.080 3.5 CDRH4D18-2R2 2.2 1.32 0.058 1.8 CDRH5D18-6R2 6.2 1.4 0.071 1.8 CDRH5D28-100 10 1.3 0.048 2.8 1008PS-272M 2.7 1.3 0.14 2.7 LPO1704-222M 2.2 1.6 0.12 1.0 LPO1704-332M 3.3 1.3 0.16 1.0 Murata Sumida Coilcraft 1961fa 8 LT1961 U U W U APPLICATIONS INFORMATION CATCH DIODE The suggested catch diode (D1) is a UPS120 or 1N5818 Schottky. It is rated at 1A average forward current and 20V/30V reverse voltage. Typical forward voltage is 0.5V at 1A. The diode conducts current only during switch off time. Peak reverse voltage is equal to regulator output voltage. Average forward current in normal operation is equal to output current. SHUTDOWN AND UNDERVOLTAGE LOCKOUT Figure 4 shows how to add undervoltage lockout (UVLO) to the LT1961. Typically, UVLO is used in situations where the input supply is current limited, or has a relatively high source resistance. A switching regulator draws constant power from the source, so source current increases as source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur. shutdown pin can be used. The threshold voltage of the shutdown pin comparator is 1.35V. A 3μA internal current source defaults the open pin condition to be operating (see Typical Performance Graphs). Current hysteresis is added above the SHDN threshold. This can be used to set voltage hysteresis of the UVLO using the following: R1 = R2 = VH − VL 7μA 1.35V (VH − 1.35V) + 3μA R1 VH – Turn-on threshold VL – Turn-off threshold Example: switching should not start until the input is above 4.75V and is to stop if the input falls below 3.75V. VH = 4.75V VL = 3.75V LT1961 R1 = 7μA IN INPUT 1.35V R1 3μA VCC SHDN C1 R2 GND 1961 F04 Figure 4. Undervoltage Lockout An internal comparator will force the part into shutdown below the minimum VIN of 2.6V. This feature can be used to prevent excessive discharge of battery-operated systems. If an adjustable UVLO threshold is required, the R2 = 4.75V − 3.75V = 143k 7μA 1.35V (4.75V − 1.35V) + 3μA = 50.4k 143k Keep the connections from the resistors to the SHDN pin short and make sure that the interplane or surface capacitance to the switching nodes are minimized. If high resistor values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from the switch node. 1961fa 9 LT1961 U W U U APPLICATIONS INFORMATION SYNCHRONIZATION The SYNC pin, is used to synchronize the internal oscillator to an external signal. The SYNC input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 20% and 80%. The input can be driven directly from a logic level output. The synchronizing range is equal to initial operating frequency up to 2MHz. This means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (1.5MHz), not the typical operating frequency of 1.25MHz. Caution should be used when synchronizing above 1.7MHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. Higher inductor values will tend to eliminate this problem. See Frequency Compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. Application Note 19 has more details on the theory of slope compensation. LAYOUT CONSIDERATIONS As with all high frequency switchers, when considering layout, care must be taken to achieve optimal electrical, thermal and noise performance. For maximum efficiency, switch rise and fall times are typically in the nanosecond range. To prevent noise both radiated and conducted, the high speed switching current path, shown in Figure 5, must be kept as short as possible. This is implemented in the suggested layout of Figure 6. Shortening this path will also reduce the parasitic trace inductance of approximately 25nH/inch. At switch off, this parasitic inductance produces a flyback spike across the LT1961 switch. When operating at higher currents and output voltages, with poor layout, this spike can generate voltages across the LT1961 that may exceed its absolute maximum rating. A ground plane should always be used under the switcher circuitry to prevent interplane coupling and overall noise. The VC and FB components should be kept as far away as possible from the switch node. The LT1961 pinout has been designed to aid in this. The ground for these components should be separated from the switch current path. Failure to do so will result in poor stability or subharmonic like oscillation. Board layout also has a significant effect on thermal resistance. The exposed pad is the copper plate that runs under the LT1961 die. This is the best thermal path for heat out of the package. Soldering the pad onto the board will reduce die temperature and increase the power capability of the LT1961. Provide as much copper area as possible around this pad. Adding multiple solder filled feedthroughs under and around the pad to the ground plane will also help. Similar treatment to the catch diode and inductor terminations will reduce any additional heating effects. L1 D1 C3 VOUT SW LT1961 VIN HIGH FREQUENCY SWITCHING PATH C1 LOAD GND 1961 F05 Figure 5. High Speed Switching Path 1961fa 10 LT1961 U W U U APPLICATIONS INFORMATION L1 6.8μH D1 UPS120 INPUT 5V C3 2.2μF CERAMIC OPEN OR HIGH = ON LT1961 SHDN SYNC OUTPUT 12V 0.5A* VSW VIN GND R1 90.9k VC FB C2 6800pF R3 6.8k C4 100pF C1 10μF CERAMIC R2 10k 1% *MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING. INPUT GND L1 R3 C4 C3 LT1961EMS8E C2 KEEP FB AND VC COMPONENTS AWAY FROM HIGH FREQUENCY, HIGH INPUT COMPONENTS D1 MINIMIZE LT1961, C1, D1 LOOP U1 C1 GND R2 R1 VOUT KELVIN SENSE VOUT PLACE FEEDTHROUGHS AROUND GROUND PIN FOR GOOD THERMAL CONDUCTIVITY SOLDER EXPOSED GROUND PAD TO BOARD Figure 6. Typical Application and Suggested Layout (Topside Only Shown) 1961fa 11 LT1961 U U W U APPLICATIONS INFORMATION THERMAL CALCULATIONS Power dissipation in the LT1961 chip comes from four sources: switch DC loss, switch AC loss, drive current, and input quiescent current. The following formulas show how to calculate each of these losses. These formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. (VOUT − VIN ) VOUT (V )(I ) = OUT OUT VIN DC, duty cycle = ISW Switch loss: ( )( )( ) PSW = (DC )(ISW )2 (RSW ) + 17n ISW VOUT f VIN loss: (VIN )(ISW )(DC ) + 1mA(VIN ) 50 RSW = Switch resistance (≈ 0.27Ω hot) with no device power, in an oven. The same measurement can then be used in operation to indicate the die temperature. FREQUENCY COMPENSATION Loop frequency compensation is performed on the output of the error amplifier (VC pin) with a series RC network. The main pole is formed by the series capacitor and the output impedance (≈500kΩ) of the error amplifier. The pole falls in the range of 2Hz to 20Hz. The series resistor creates a “zero” at 1kHz to 5kHz, which improves loop stability and transient response. A second capacitor, typically one-tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the VC pin. VC pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor, VC pin ripple is: PVIN = Example: VIN = 5V, VOUT = 12V and IOUT = 0.5A Total power dissipation = 0.23 + 0.31 + 0.07 + 0.005 = 0.62W Thermal resistance for LT1961 package is influenced by the presence of internal or backside planes. With a full plane under the package, thermal resistance will be about 50°C/W. To calculate die temperature, use the appropriate thermal resistance number and add in worst-case ambient temperature: TJ = TA + θJA (PTOT) If a true die temperature is required, a measurement of the SYNC to GND pin resistance can be used. The SYNC pin resistance across temperature must first be calibrated, VC Pin Ripple = 1.2(VRIPPLE)(gm)(RC) (VOUT) VRIPPLE = Output ripple (VP–P) gm = Error amplifier transconductance (≈850μmho) RC = Series resistor on VC pin VOUT = DC output voltage To prevent irregular switching, VC pin ripple should be kept below 50mVP–P. Worst-case VC pin ripple occurs at maximum output load current and will also be increased if poor quality (high ESR) output capacitors are used. The addition of a 47pF capacitor on the VC pin reduces switching frequency ripple to only a few millivolts. A low value for RC will also reduce VC pin ripple, but loop phase margin may be inadequate. 1961fa 12 LT1961 U TYPICAL APPLICATIO S Dual Output Flyback Converter R2 10k 1% R1 115k 1% UPS140 T1* VIN 5V TO 10V + 2, 3 P6KE-20A • C1 4.7μF OFF S/S 8, 9 • LT1961 VC VOUT 15V + •4 10 1N4148 VIN VSW FB ON 7 C4 47μF + C5 47μF 1 –VOUT –15V UPS140 GND C2 2.2nF R3 10k C3 100pF *DALE LPE-4841-100MB LT1961 • TA02 4V-9VIN to 5VOUT SEPIC Converter** VIN** 4V TO 9V L1A* 10μH VIN OFF ON VSW S/S LT1961 + C1 4.7μF 20V FB GND D1 UPS120 • R2 31.6k 1% C2 4.7μF • VC + L1B* 10μH R1 10k C4 2.2nF R3 10k 1% C5 100pF †MAX I OUT * BH ELECTRONICS 511-1012 ** INPUT VOLTAGE MAY BE GREATER OR LESS THAN OUTPUT VOLTAGE IOUT 0.59A 0.65A 0.70A 0.74A 0.80A VOUT† 5V C3 47μF 10V LT1961 • TA03 VIN 4V 5V 6V 7V 9V 1961fa 13 LT1961 U TYPICAL APPLICATIO S Single Li-Ion Cell to 5V L1 4.7μH D1 UPS120 VOUT 5V VSW R1 31.6k 1% FB + VIN OFF ON S/S LT1961 + SINGLE Li-Ion CELL + C1 10μF VC GND C2 2.2nF R3 10k C4 47μF 10V R2 10k 1% C3 100pF LT1961 • TA04 IOUT VIN 0.75A 2.7V 0.93A 3.3V 1.0A 3.6V 1961fa 14 LT1961 U PACKAGE DESCRIPTION MS8E Package 8-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1662 Rev D) BOTTOM VIEW OF EXPOSED PAD OPTION 2.06 ± 0.102 (.081 ± .004) 1 5.23 (.206) MIN 1.83 ± 0.102 (.072 ± .004) 0.889 ± 0.127 (.035 ± .005) 2.794 ± 0.102 (.110 ± .004) 2.083 ± 0.102 3.20 – 3.45 (.082 ± .004) (.126 – .136) 8 0.42 ± 0.038 (.0165 ± .0015) TYP 3.00 ± 0.102 (.118 ± .004) (NOTE 3) 0.65 (.0256) BSC 8 7 6 5 0.52 (.0205) REF RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 4.90 ± 0.152 (.193 ± .006) DETAIL “A” 0° – 6° TYP GAUGE PLANE 0.53 ± 0.152 (.021 ± .006) DETAIL “A” 1 2 3 4 1.10 (.043) MAX 0.86 (.034) REF 0.18 (.007) SEATING PLANE 0.22 – 0.38 (.009 – .015) TYP 0.65 (.0256) BSC 0.1016 ± 0.0508 (.004 ± .002) MSOP (MS8E) 0307 REV D NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 1961fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1961 U TYPICAL APPLICATIO High Voltage Laser Power Supply 0.01μF 5kV 1800pF 10kV 47k 5W 1800pF 10kV 8 11 L1 1 4 5 HV DIODES 3 2 LASER + 2.2μF Q1 0.47μF 150Ω L2 10μH MUR405 VIN 12V TO 25V Q2 VSW + 10k 10k VIN FB LT1961 2.2μF VC 0.1μF VIN 1N4002 (ALL) 190Ω 1% GND + 10μF LT1961 • TA05 L1 = COILTRONICS CTX02-11128 Q1, Q2 = ZETEX ZTX849 0.47μF = WIMA 3X 0.15μF TYPE MKP-20 HV DIODES = SEMTECH-FM-50 LASER = HUGHES 3121H-P COILTRONICS (407) 241-7876 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1308A 600kHz, 2A, Step-Up Regulator 30V Switch, VIN = 1V to 6V, Low Battery Comparator, S8 Package LT1310 4.5MHz, 1.5A Step-Up with Phase Lock Loop 34V Switch, VIN = 2.75V to 18V, VOUT up to 35V, MS10E Package LT1370 High Efficiency DC/DC Converter 42V Switch, 6A, 500kHz Switch, DD-Pak, TO-220 Package LT1371 High Efficiency DC/DC Converter 35V Switch, 3A, 500kHz Switch, DD-Pak, TO-220 Package LT1372/LT1377 500kHz and 1MHz High Efficiency 1.5A Switching Regulators Boost Topology, VIN(MIN) = 2.7V, S8 Package LT1946/LT1946A 1.2MHz/2.7MHz, 1.5A, Monolithic Step-Up Regulator VIN = 2.6V to 16V, VOUT up to 34V, Integrated SS, MS8 Package LTC3400/ LTC3400B 1.2MHz, 600mA, Synchronous Step-Up VIN = 0.85V to 5V, VOUT to 5.5V, Up to 95% Efficiency, ThinSOT Package LTC3401 Single Cell, High Current (1A), Micropower, Synchronous 3MHz Step-Up DC/DC Converter VIN = 0.85V to 5V, VOUT to 5.5V, Up to 97% Efficiency Synchronizable, Oscillator from 100kHz to 3MHz, MS10 Package LTC3402 Single Cell, High Current (2A), Micropower, Synchronous 3MHz Step-Up DC/DC Converter VIN = 0.85V to 5V, VOUT to 5.5V, Up to 95% Efficiency Synchronizable, Oscillator from 100kHz to 3MHz, MS10 Package LTC3405/ LTC3405A 1.5MHz High Efficiency, IOUT = 300mA, Monolithic Synchronous Step-Down Regulator VIN = 2.5V to 5.5V, VOUT to 0.8V, Up to 95% Efficiency, 100% Duty Cycle, IQ = 20μA, ThinSOT Package ThinSOT is a trademark of Linear Technology Corporation. 1961fa 16 Linear Technology Corporation LT 0707 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2001