AD AD745JRZ-16 Ultralow noise, high speed, bifet op amp Datasheet

a
Ultralow Noise,
High Speed, BiFET Op Amp
AD745
FEATURES
ULTRALOW NOISE PERFORMANCE
2.9 nV/冑Hz at 10 kHz
0.38 V p-p, 0.1 Hz to 10 Hz
6.9 fA/冑Hz Current Noise at 1 kHz
CONNECTION DIAGRAM
16-Lead SOIC (R) Package
EXCELLENT AC PERFORMANCE
12.5 V/s Slew Rate
20 MHz Gain Bandwidth Product
THD = 0.0002% @ 1 kHz
Internally Compensated for Gains of +5 (or –4) or
Greater
EXCELLENT DC PERFORMANCE
0.5 mV Max Offset Voltage
250 pA Max Input Bias Current
2000 V/mV Min Open Loop Gain
Available in Tape and Reel in Accordance with
EIA-481A Standard
APPLICATIONS
Sonar
Photodiode and IR Detector Amplifiers
Accelerometers
Low Noise Preamplifiers
High Performance Audio
amplifier for high-speed applications demanding low noise and
high dc precision. Furthermore, the AD745 does not exhibit an
output phase reversal.
The AD745 also has excellent dc performance with 250 pA
maximum input bias current and 0.5 mV maximum offset voltage.
PRODUCT DESCRIPTION
The AD745 is an ultralow noise, high-speed, FET input operational amplifier. It offers both the ultralow voltage noise and
high speed generally associated with bipolar input op amps and
the very low input currents of FET input devices. Its 20 MHz
bandwidth and 12.5 V/µs slew rate makes the AD745 an ideal
The internal compensation of the AD745 is optimized for higher
gains, providing a much higher bandwidth and a faster slew
rate. This makes the AD745 especially useful as a preamplifier
where low level signals require an amplifier that provides both
high amplification and wide bandwidth at these higher gains.
The AD745 is available in two performance grades. The AD745J
and AD745K are rated over the commercial temperature range
of 0°C to 70°C, and are available in the 16-lead SOIC package.
120
1000
120
100
OP37 AND
RESISTOR
RSOURCE
100
AD745 AND RESISTOR
OR
OP37 AND RESISTOR
100
PHASE
AD745 AND
RESISTOR
10
80
80
60
60
GAIN
40
40
20
20
0
PHASE MARGIN – Degrees
EO
OPEN-LOOP GAIN – dB
INPUT NOISE VOLTAGE – nV/ Hz
RSOURCE
0
RESISTOR NOISE ONLY
1
100
1k
10k
100k
SOURCE RESISTANCE – 1M
10M
–20
100
1k
10k
100k
1M
FREQUENCY – Hz
10M
–20
100M
Figure 1.
Figure 2.
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 2002
REV. D
AD745–SPECIFICATIONS
AD745 ELECTRICAL CHARACTERISTICS
Model
Conditions
INPUT OFFSET VOLTAGE 1
Initial Offset
Initial Offset
vs. Temp.
vs. Supply (PSRR)
vs. Supply (PSRR)
INPUT BIAS CURRENT 3
Either Input
Either Input
@ TMAX
Either Input
Either Input, VS = ± 5 V
INPUT OFFSET CURRENT
Offset Current
@ TMAX
FREQUENCY RESPONSE
Gain BW, Small Signal
Full Power Response
Slew Rate
Settling Time to 0.01%
Total Harmonic
Distortion4
Min
INPUT VOLTAGE NOISE
AD745J
Typ
0.25
TMIN to TMAX
TMIN to TMAX
12 V to 18 V2
TMIN to TMAX
90
88
Max
Unit
0.5
1.0
mV
mV
µV/°C
dB
dB
pA
2
106
105
150
400
150
250
VCM = 0 V
VCM = +10 V
VCM = 0 V
250
30
8.8
600
200
250
30
5.5
400
125
nA
pA
pA
VCM = 0 V
40
150
30
75
pA
1.1
nA
VCM = 0 V
2.2
G = –4
VO = 20 V p-p
G = –4
f = 1 kHz
G = –4
20
120
12.5
5
20
120
12.5
5
MHz
kHz
V/µs
µs
0.0002
0.0002
%
1 × 1010储20
3 × 1011储18
1 × 1010储20
3 × 1011储18
Ω储pF
Ω储pF
± 20
+13.3, –10.7
± 20
+13.3, –10.7
V
V
V
–10
VCM = ± 10 V
TMIN to TMAX
80
78
+12
95
f = 1 kHz
6.9
OPEN LOOP GAIN
VO = ± 10 V
RLOAD ≥ 2 kΩ
TMIN to TMAX
RLOAD = 600 Ω
RLOAD ≥ 600 Ω
RLOAD ≥ 600 Ω
TMIN to TMAX
RLOAD ≥ 2 kΩ
Short Circuit
POWER SUPPLY
Rated Performance
Operating Range
Quiescent Current
1000
800
5.0
4.0
4000
2000
1800
102
dB
dB
1.0
10.0
6.0
5.0
4.0
fA/√Hz
4000
V/mV
V/mV
V/mV
+13, –12
+13.6, –12.6
V
+13.6, –12.6
V
+13.8, –13.1
40
V
+12, –10
+13.8, –13.1
40
± 15
8
50
20
± 18
10.0
± 4.8
µV p-p
nV/√Hz
nV/√Hz
nV/√Hz
nV/√Hz
6.9
1200
+13, –12
+12, –10
± 12
20
+12
0.38
5.5
3.6
3.2
2.9
1200
± 4.8
# of Transistors
–10
90
88
INPUT CURRENT NOISE
TRANSISTOR COUNT
AD745K
Typ
0.1
100
98
0.38
5.5
3.6
3.2
2.9
Current
Min
1.0
1.5
2
96
0.1 to 10 Hz
f = 10 Hz
f = 100 Hz
f = 1 kHz
f = 10 kHz
OUTPUT CHARACTERISTICS
Voltage
Max
VCM = 0 V
INPUT IMPEDANCE
Differential
Common Mode
INPUT VOLTAGE RANGE
Differential5
Common-Mode Voltage
Over Max Operating Range 6
Common-Mode
Rejection Ratio
(@ +25C and 15 V dc, unless otherwise noted.)
V
± 15
8
mA
± 18
10.0
V
V
mA
50
NOTES
1
Input offset voltage specifications are guaranteed after five minutes of operations at T A = 25°C.
2
Test conditions: +VS = 15 V, –VS = 12 V to 18 V and +VS = 12 V to +18 V, –VS = 15 V.
3
Bias current specifications are guaranteed maximum at either input after five minutes of operation at T A = 25°C. For higher temperature, the current doubles every 10°C.
4
Gain = –4, RL = 2 kΩ, CL = 10 pF.
5
Defined as voltage between inputs, such that neither exceeds ± 10 V from common.
6
The AD745 does not exhibit an output phase reversal when the negative common-mode limit is exceeded.
All min and max specifications are guaranteed.
Specifications subject to change without notice.
–2–
REV. D
AD745
ABSOLUTE MAXIMUM RATINGS 1
ESD SUSCEPTIBILITY
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V
Internal Power Dissipation2
SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.2 W
Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± VS
Output Short-Circuit Duration . . . . . . . . . . . . . . . . Indefinite
Differential Input Voltage . . . . . . . . . . . . . . . . . . +VS and –VS
Storage Temperature Range (R) . . . . . . . . . –65°C to +125°C
Operating Temperature Range
AD745J/K . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . . 300°C
An ESD classification per method 3015.6 of MIL-STD-883C
has been performed on the AD745, which is a class 1 device.
Using an IMCS 5000 automated ESD tester, the two null pins
will pass at voltages up to 1,000 volts, while all other pins will
pass at voltages exceeding 2,500 volts.
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to Absolute Maximum Rating
conditions for extended periods may affect device reliability.
2
16-Pin Plastic SOIC Package: θJA = 100°C/W, θJC = 30°C/W
ORDERING GUIDE
Model
Temperature Range
Package
Option*
AD745JR-16
AD745KR-16
0°C to 70°C
0°C to 70°C
R-16
R-16
*
R = Small Outline IC.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD745 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
REV. D
–3–
WARNING!
ESD SENSITIVE DEVICE
AD745 –Typical Performance Characteristics
20
35
20
15
+VIN
10
–VIN
5
0
15
POSITIVE
SUPPLY
10
NEGATIVE
SUPPLY
5
0
20
0
TPC 1. Input Voltage Swing vs.
Supply Voltage
INPUT BIAS CURRENT – Amps
12
9
6
3
0
5
10
15
SUPPLY VOLTAGE VOLTS
0
–9
–6
–3
0
3
6
9
COMMON-MODE VOLTAGE – V
TPC 7. Input Bias Current vs.
Common-Mode Voltage
12
15
10
5
100
1k
LOAD RESISTANCE – 10k
10–6
200
100
10–7
10–8
10–9
10–10
10
1
CLOSED LOOP GAIN = –5
0.1
10–11
0 20 40 60 80 100 120 140
TEMPERATURE – C
0.01
10k
1M
10M
FREQUENCY – Hz
100M
28
10–7
10–8
10–9
10–10
10–11
10–12
–60 –40 –20
100k
TPC 6. Output Impedance vs.
Frequency
GAIN BANDWIDTH PRODUCT – MHz
INPUT BIAS CURRENT – Amps
100
20
TPC 3. Output Voltage Swing vs.
Load Resistance
10–6
200
25
0
10
20
TPC 5. Input Bias Current vs.
Temperature
300
30
TPC 2. Output Voltage Swing vs.
Supply Voltage
10–12
–60 –40 –20
20
TPC 4. Quiescent Current vs.
Supply Voltage
0
–12
5
10
15
SUPPLY VOLTAGE VOLTS
OUTPUT IMPEDANCE – 5
10
15
SUPPLY VOLTAGE VOLTS
0
QUIESCENT CURRENT – mA
OUTPUT VOLTAGE SWING – V p-p
RLOAD = 10k
INPUT VOLTAGE SWING – V
INPUT VOLTAGE SWING – V
RLOAD = 10k
INPUT BIAS CURRENT – pA
(@ + 25C, VS = 15 V, unless otherwise noted.)
0 20 40 60 80 100 120 140
TEMPERATURE – C
TPC 8. Short Circuit Current Limit vs.
Temperature
–4–
26
24
22
20
18
16
14
–60 –40 –20
0 20 40 60 80 100 120 140
TEMPERATURE – C
TPC 9. Gain Bandwidth Product vs.
Temperature
REV. D
AD745
14
120
150
RL = 2k
100
SLEW RATE – V/s
60
40
GAIN
20
OPEN-LOOP GAIN – dB
OPEN-LOOP GAIN – dB
PHASE
80
12
CLOSED-LOOP GAIN = 5
10
140
130
120
100
0
–20
100
1k
10k 100k
1M
FREQUENCY – Hz
10M
8
–60 –40 –20 0 20 40 60 80 100 110 120
TEMPERATURE – C
100M
TPC 10. Open-Loop Gain and Phase
vs. Frequency
TPC 11. Slew Rate vs. Temperature
0
10
15
5
SUPPLY VOLTAGE VOLTS
20
TPC 12. Open-Loop Gain vs.
Supply Voltage
120
35
100
90
80
Vcm = 10V
70
60
1k
10k
100k
FREQUENCY – Hz
1M
1.0
–60
0.1
–80
0.01
GAIN = +10
0.001
–100
GAIN = –4
–120
0.0001
100
1k
10k
FREQUENCY – Hz
0.00001
100k
TPC 16. Total Harmonic Distortion
vs. Frequency
REV. D
80
60
–SUPPLY
40
20
0
100
1k
10k 100k
1M
FREQUENCY – Hz
10M
100
10
CLOSED-LOOP GAIN = 5
1.0
0.1
10
100
1k
10k 100k
FREQUENCY – Hz
1M
TPC 17. Input Noise Voltage
Spectral Density
–5–
10M
30
25
20
15
10
5
0
10k
100M
TPC 14. Power Supply Rejection
vs. Frequency
TOTAL HARMONIC DISTORTION (THD) – %
–40
–140
10
+SUPPLY
10M
TPC 13. Common-Mode Rejection vs.
Frequency
GAIN = +100
100
100k
1M
FREQUENCY – Hz
10M
TPC 15. Large Signal Frequency
Response
CURRENT NOISE SPECTRAL DENSITY – fA/ Hz
50
100
OUTPUT VOLTAGE SWING – V p-p
POWER SUPPLY REJECTION – dB
RL = 2k
110
NOISE VOLTAGE (referred to input) – nV/ Hz
COMMON-MODE REJECTION – dB
120
TOTAL HARMONIC DISTORTION (THD) – dB
80
1k
100
10
1.0
1
10
100
1k
FREQUENCY – Hz
10k
TPC 18. Input Noise Current
Spectral Density
100k
AD745
648
72
TOTAL UNITS = 760
540
54
486
48
42
36
30
24
432
378
324
270
216
18
162
12
108
6
54
0
–15
0
2.6 2.7 2.8 2.9 3.0 3.1 3.2 3.3 3.4
INPUT VOLTAGE NOISE @ 10kHz – nV Hz
–5
0
5
10
–10
15
INPUT OFFSET VOLTAGE DRIFT – V/C
TPC 19. Distribution of Offset
Voltage Drift. TA = 25°C to 125°C
TOTAL UNITS = 4100
594
60
NUMBER OF UNITS
NUMBER OF UNITS
66
TPC 20. Typical Input Noise Voltage
Distribution @ 10 kHz
TPC 21. Offset Null Configuration,
16-Lead Package Pinout
500ns
2µs
100
100
90
90
10
10
0%
0%
50mV
5V
TPC 22b. Gain of 5 Follower
Large Signal Pulse Response
TPC 22a. Gain of 5 Follower,
16-Lead Package Pinout
TPC 22c. Gain of 5 Follower Small
Signal Pulse Response
2µs
100
90
90
10
10
0%
0%
5V
TPC 23a. Gain of 4 Inverter,
16-Lead Package Pinout
500ns
100
50mV
TPC 23b. Gain of 4 Inverter Large
Signal Pulse Response
–6–
TPC 23c. Gain of 4 Inverter Small
Signal Pulse Response
REV. D
AD745
OP AMP PERFORMANCE JFET VERSUS BIPOLAR
The AD745 offers the low input voltage noise of an industry
standard bipolar opamp without its inherent input current
errors. This is demonstrated in Figure 3, which compares input
voltage noise vs. input source resistance of the OP37 and the
AD745 opamps. From this figure, it is clear that at high source
impedance the low current noise of the AD745 also provides
lower total noise. It is also important to note that with the AD745
this noise reduction extends all the way down to low source
impedances. The lower dc current errors of the AD745 also
reduce errors due to offset and drift at high source impedances
(Figure 4).
The internal compensation of the AD745 is optimized for higher
gains, providing a much higher bandwidth and a faster slew
rate. This makes the AD745 especially useful as a preamplifier,
where low-level signals require an amplifier that provides both
high amplification and wide bandwidth at these higher gains.
1000
INPUT NOISE VOLTAGE – nV/ Hz
RSOURCE
EO
OP37 AND
RESISTOR
AD745 AND RESISTOR
OR
OP37 AND RESISTOR
= 4kT/R∆ f
to compute the Johnson noise of a resistor, expressed as a
current, one can see that the current noise of the AD745 is
equivalent to that of a 3.45 × 108 Ω source resistance.
AD745 AND
RESISTOR
10
RESISTOR NOISE ONLY
1
100
1k
10k
100k
SOURCE RESISTANCE – 1M
10M
Figure 3. Total Input Noise Spectral Density @ 1 kHz
vs. Source Resistance
100
INPUT OFFSET VOLTAGE – mV
Low frequency current noise can be computed from the
magnitude of the dc bias current
 ~ = 2qI ∆f 
B
 In

and increases below approximately 100 Hz with a 1/f power
spectral density. For the AD745 the typical value of current
noise is 6.9 fA/√Hz at 1 kHz. Using the formula:
~I
RSOURCE
100
The 0.1 Hz to 10 Hz noise is typically 0.38 µV p-p. The user
should pay careful attention to several design details to optimize
low frequency noise performance. Random air currents can
generate varying thermocouple voltages that appear as low
frequency noise. Therefore, sensitive circuitry should be well
shielded from air flow. Keeping absolute chip temperature low
also reduces low frequency noise in two ways: first, the low
frequency noise is strongly dependent on the ambient temperature and increases above 25°C. Second, since the gradient of
temperature from the IC package to ambient is greater, the
noise generated by random air currents, as previously mentioned,
will be larger in magnitude. Chip temperature can be reduced
both by operation at reduced supply voltages and by the use of a
suitable clip-on heat sink, if possible.
OP37G
n
At high frequencies, the current noise of a FET increases proportionately to frequency. This noise is due to the “real” part of
the gate input impedance, which decreases with frequency. This
noise component usually is not important, since the voltage
noise of the amplifier impressed upon its input capacitance is an
apparent current noise of approximately the same magnitude.
In any FET input amplifier, the current noise of the internal
bias circuitry can be coupled externally via the gate-to-source
capacitances and appears as input current noise. This noise is
totally correlated at the inputs, so source impedance matching
will tend to cancel out its effect. Both input resistance and input
capacitance should be balanced whenever dealing with source
capacitances of less than 300 pF in value.
10
LOW NOISE CHARGE AMPLIFIERS
As stated, the AD745 provides both low voltage and low current
noise. This combination makes this device particularly suitable
in applications requiring very high charge sensitivity, such as
capacitive accelerometers and hydrophones. When dealing with
a high source capacitance, it is useful to consider the total input
charge uncertainty as a measure of system noise.
1.0
AD745 KN
0.1
100
1k
10k
100k
SOURCE RESISTANCE – 1M
10M
Charge (Q) is related to voltage and current by the simply stated
fundamental relationships:
Q = CV and I =
Figure 4. Input Offset Voltage vs. Source Resistance
DESIGNING CIRCUITS FOR LOW NOISE
An opamp’s input voltage noise performance is typically divided
into two regions: flatband and low frequency noise. The AD745
offers excellent performance with respect to both. The figure of
2.9 nV/冑Hz @ 10 kHz is excellent for a JFET input amplifier.
REV. D
dQ
dt
As shown, voltage, current and charge noise can all be directly
related. The change in open circuit voltage (∆V) on a capacitor
will equal the combination of the change in charge (∆Q/C) and
the change in capacitance with a built-in charge (Q/∆C).
–7–
AD745
Figures 5 and 6 show two ways to buffer and amplify the output
of a charge output transducer. Both require the use of an amplifier that has a very high input impedance, such as the AD745.
Figure 5 shows a model of a charge amplifier circuit. Here,
amplification depends on the principle of conservation of charge
at the input of amplifier A1, which requires that the charge on
capacitor CS be transferred to capacitor CF, thus yielding an
output voltage of ∆Q/CF. The amplifiers input voltage noise will
appear at the output amplified by the noise gain (1 + (CS/CF))
of the circuit.
–100
DECIBELS REFERENCED TO 1V/ Hz
–110
CF
R1
TOTAL
OUTPUT
NOISE
–150
–160
–170
–180
–190
NOISE DUE TO
RB ALONE
–200
NOISE DUE TO
IB ALONE
0.1
1
10
100
1k
10k
100k
FREQUENCY – Hz
Figure 7. Noise at the Outputs of the Circuits of Figures 5
and 6. Gain = 10, CS = 3000 pF, RB = 22 MΩ
A1
RB*
–140
–220
0.01
R2
CB*
–130
–210
RS
CS
–120
However, this does not change the noise contribution of RB
which, in this example, dominates at low frequencies. The graph
of Figure 8 shows how to select an RB large enough to minimize
this resistor’s contribution to overall circuit noise. When the
equivalent current noise of RB ((冑4 kT)/R) equals the noise of
I B 2qI B , there is diminishing return in making RB larger.
R1 CS
=
R2 CF
Figure 5. A Charge Amplifier Circuit
(
R1
CB*
)
5.2 1010
CS
A2
RB
5.2 109
RESISTANCE IN R2
RB*
*OPTIONAL, SEE TEXT.
Figure 6. Model for A High Z Follower with Gain
The second circuit, Figure 6, is simply a high impedance follower with gain. Here the noise gain (1 + (R1/R2)) is the same
as the gain from the transducer to the output. Resistor RB, in
both circuits, is required as a dc bias current return.
5.2 107
There are three important sources of noise in these circuits.
Amplifiers A1 and A2 contribute both voltage and current noise,
while resistor RB contributes a current noise of:
~
N =
4k
5.2 108
5.2 106
1pA
10pA
100pA
1nA
INPUT BIAS CURRENT
10nA
Figure 8. Graph of Resistance vs. Input Bias Current
Where the Equivalent Noise 兹4 kT/R, Equals the Noise
of the Bias Current I B 2qI B
T
∆f
RB
(
where:
)
To maximize dc performance over temperature, the source
resistances should be balanced on each input of the amplifier.
This is represented by the optional resistor RB in Figures 5 and 6.
As previously mentioned, for best noise performance care should
be taken to also balance the source capacitance designated by
CB The value for CB in Figure 5 would be equal to CS in
Figure 6. At values of CB over 300 pF, there is a diminishing
impact on noise; capacitor CB can then be simply a large mylar
bypass capacitor of 0.01 µF or greater.
k = Boltzman’s Constant = 1.381 × 10–23 Joules/Kelvin
T = Absolute Temperature, Kelvin (0°C = 273.2 Kelvin)
∆f = Bandwidth – in Hz (Assuming an Ideal “Brick Wall”
Filter)
This must be root-sum-squared with the amplifier’s own current
noise.
Figure 5 shows that these two circuits have an identical frequency
response and the same noise performance (provided that
CS/CF = R1/ R2). One feature of the first circuit is that a “T”
network is used to increase the effective resistance of RB and
improve the low frequency cutoff point by the same factor.
–8–
REV. D
AD745
HOW CHIP PACKAGE TYPE AND POWER DISSIPATION
AFFECT INPUT BIAS CURRENT
300
TA = 25C
INPUT BIAS CURRENT – Amps
As with all JFET input amplifiers, the input bias current of the
AD745 is a direct function of device junction temperature, IB
approximately doubling every 10°C. Figure 9 shows the relationship between bias current and junction temperature for the
AD745. This graph shows that lowering the junction temperature will dramatically improve IB.
10–6
INPUT BIAS CURRENT – Amps
VS = 15V
TA = 25C
200
JA = 165C/W
100
JA = 115C/W
10–7
JA = 0C/W
0
10–8
10–9
5
10
SUPPLY VOLTAGE – Volts
15
Figure 11. Input Bias Current vs. Supply Voltage for
Various Values of θJA
10–10
TJ
10–11
10–12
–60
A
(J TO DIE
MOUNT)
–40
–20
0
20
40
60
80
100
JUNCTION TEMPERATURE – C
120
140
Figure 9. Input Bias Current vs. Junction Temperature
The dc thermal properties of an IC can be closely approximated
by using the simple model of Figure 10 where current represents
power dissipation, voltage represents temperature, and resistors
represent thermal resistance (θ in °C/watt).
TJ
PIN
JC
CA
JA
TA
WHERE:
PIN = DEVICE DISSIPATION
TA = AMBIENT TEMPERATURE
TJ = JUNCTION TEMPERATURE
JC = THERMAL RESISTANCE – JUNCTION TO CASE
CA = THERMAL RESISTANCE – CASE TO AMBIENT
Figure 10. Device Thermal Model
B
(DIE MOUNT
TO CASE)
TA
A + B = JC
CASE
Figure 12. Breakdown of Various Package Thermal
Resistance
REDUCED POWER SUPPLY OPERATION FOR
LOWER IB
Reduced power supply operation lowers IB in two ways: first, by
lowering both the total power dissipation and, second, by reducing the basic gate-to-junction leakage (Figure 11). Figure 13
shows a 40 dB gain piezoelectric transducer amplifier, which
operates without an ac coupling capacitor, over the –40°C to
+85°C temperature range. If the optional coupling capacitor,
C1, is used, this circuit will operate over the entire –55°C to
+125°C temperature range.
100
From this model TJ = TA+θJA PIN. Therefore, IB can be determined in a particular application by using Figure 9 together with
the published data for θJA and power dissipation. The user can
modify θJA by use of an appropriate clip-on heat sink such as the
Aavid #5801. Figure 11 shows bias current versus supply voltage
with θJA as the third variable. This graph can be used to predict
bias current after θJA has been computed. Again bias current will
double for every 10°C.
10k
C1*
108**
TRANSDUCER
CT
108
CT**
+5V
AD745
–5V
*OPTIONAL DC BLOCKING CAPACITOR
**OPTIONAL, SEE TEXT
Figure 13. A Piezoelectric Transducer
REV. D
–9–
AD745
TWO HIGH PERFORMANCE ACCELEROMETER
AMPLIFIERS
Two of the most popular charge-out transducers are hydrophones
and accelerometers. Precision accelerometers are typically calibrated for a charge output (pC/g).* Figures 14 and 15 show two
ways in which to configure the AD745 as a low noise charge
amplifier for use with a wide variety of piezoelectric accelerometers. The input sensitivity of these circuits will be determined
by the value of capacitor C1 and is equal to:
∆V OUT =
∆QOUT
C1
The ratio of capacitor C1 to the internal capacitance (CT) of the
transducer determines the noise gain of this circuit (1 + CT/C1).
The amplifiers voltage noise will appear at its output amplified
by this amount. The low frequency bandwidth of these circuits
will be dependent on the value of resistor R1. If a “T” network
is used, the effective value is: R1 (1 + R2/R3).
*pC
= Picocoulombs
g = Earth’s Gravitational Constant
low frequency performance, the time constant of the servo loop
(R4C2 = R5C3) should be:

R2 
C1
Time Constant ≥10 R1 1+
R3 

A LOW NOISE HYDROPHONE AMPLIFIER
Hydrophones are usually calibrated in the voltage-out mode.
The circuit of Figures 16 can be used to amplify the output of a
typical hydrophone. If the optional ac coupling capacitor CC is
used, the circuit will have a low frequency cutoff determined by
an RC time constant equal to:
1
Time Constant ≥ 10 R1
2π × CC × 100 Ω
where the dc gain is 1 and the gain above the low frequency
cutoff (1/(2π CC(100 Ω))) is equal to (1 + R2/R3). The circuit
of Figure 17 uses a dc servo loop to keep the dc output at 0 V
and to maintain full dynamic range for IB’s up to 100 nA. The
time constant of R7 and C1 should be larger than that of R1
and CT for a smooth low frequency response.
R2
1900
C1
1250pF
R3
100
R1
110M
(5 22M)
R3
1k
B AND K TYPE 8100 HYDROPHONE
R1
108
CT
AD745
OUTPUT
INPUT SENSITIVITY = –179dB RE. 1V/mPa**
OUTPUT
0.8mV/pC
AD745
B AND K
4370 OR
EQUIVALENT
C1*
R4*
CC
R2
9k
*OPTIONAL DC BLOCKING CAPACITOR
**OPTIONAL, SEE TEXT
Figure 16. A Low Noise Hydrophone Amplifier
Figure 14. A Basic Accelerometer Circuit
C1
1250pF
R1
110M
(5 22M)
R3
1k
The transducer shown has a source capacitance of 7500 pF. For
smaller transducer capacitances (≤300 pF), lowest noise can be
achieved by adding a parallel RC network (R4 = R1, C1 = CT)
in series with the inverting input of the AD745.
R2
1900
R2
9k
R3
100
C2
2.2F
R4*
108
C1*
OUTPUT
R4
18M
AD711
R4
16M
AD745
R5
18M
C2
0.27F
C3
2.2F
B AND K
4370 OR
EQUIVALENT
AD745
R1
108
OUTPUT
0.8mV/pC
R5
100k
AD711K
CT
R6
1M
16M
Figure 15. An Accelerometer Circuit Employing a DC
Servo Amplifier
A dc servo loop (Figure 15) can be used to assure a dc output
<10 mV, without the need for a large compensating resistor
when dealing with bias currents as large as 100 nA. For optimal
DC OUTPUT 1mV FOR IB (AD745)
*OPTIONAL, SEE TEXT
100nA
Figure 17. A Hydrophone Amplifier Incorporating a DC
Servo Loop
–10–
REV. D
AD745
DESIGN CONSIDERATIONS FOR I-TO-V CONVERTERS
1F
+
There are some simple rules of thumb when designing an I-V
converter where there is significant source capacitance (as with
a photodiode) and bandwidth needs to be optimized. Consider
the circuit of Figure 18. The high frequency noise gain
(1 + CS/CL) is usually greater than five, so the AD745, with its
higher slew rate and bandwidth is ideally suited to this application.
+12V
0.01F
–12V
RB
CS
3
+12V
DIGITAL
INPUTS
15
AD1862
+12V
20-BIT D/A
CONVERTER
14
4
13
5
12
6
11
10F
+
ANALOG
COMMON
–12V
CL
0.1F
10
8
TOP VIEW
0.01F
0.1F
AD745
3k
7
RF
IS
2
0.01F
Here both the low current and low voltage noise of the AD745 can
be taken advantage of, since it is desirable in some instances to
have a large RF (which increases sensitivity to input current noise)
and, at the same time, operate the amplifier at high noise gain.
INPUT SOURCE: PHOTO DIODE,
ACCELEROMETER, ECT.
16
1
0.01F
OUTPUT
3 POLE
LOW
PASS
FILTER
9
–12V
DIGITAL
COMMON
2000pF
100pF
AD745
Figure 19. A High Performance Audio DAC Circuit
An important feature of this circuit is that high frequency energy, such as clock feedthrough, is shunted to common via a
high quality capacitor and not the output stage of the amplifier,
greatly reducing the error signal at the input of the amplifier and
subsequent opportunities for intermodulation distortions.
Figure 18. A Model for an l-to-V Converter
In this circuit, the RF CS time constant limits the practical bandwidth over which flat response can be obtained, in fact:
40
fC
2π RF CS
RTI NOISE VOLTAGE – nV/ Hz
fB ≈
where:
fB = signal bandwidth
fC = gain bandwidth product of the amplifier
With CL ≈ 1/(2 π RF CS) the net response can be adjusted to a
provide a two pole system with optimal flatness that has a corner
frequency of fB. Capacitor CL adjusts the damping of the circuit’s
response. Note that bandwidth and sensitivity are directly traded
off against each other via the selection of RF. For example, a
photodiode with CS = 300 pF and RF = 100 kΩ will have a maximum bandwidth of 360 kHz when capacitor CL ≈ 4.5 pF.
Conversely, if only a 100 kHz bandwidth were required, then
the maximum value of RF would be 360 kΩ and that of capacitor CL still ≈ 4.5 pF.
In either case, the AD745 provides impedance transformation,
the effective transresistance, i.e., the I/V conversion gain, may
be augmented with further gain. A wideband low noise amplifier
such as the AD829 is recommended in this application.
This principle can also be used to apply the AD745 in a high
performance audio application. Figure 19 shows that an I-V
converter of a high performance DAC, here the AD1862, can
be designed to take advantage of the low voltage noise of the
AD745 (2.9 nV/冑Hz) as well as the high slew rate and bandwidth provided by decompensation. This circuit, with component
values shown, has a 12 dB/octave rolloff at 728 kHz, with a
passband ripple of less than 0.001 dB and a phase deviation of
less than 2 degrees @ 20 kHz.
REV. D
30
20
UNBALANCED
10
BALANCED
2.9nV/ Hz
0
10
100
INPUT CAPACITANCE – pF
1k
Figure 20. RTI Noise Voltage vs. Input Capacitance
BALANCING SOURCE IMPEDANCES
As mentioned previously, it is good practice to balance the
source impedances (both resistive and reactive) as seen by the
inputs of the AD745. Balancing the resistive components will
optimize dc performance over temperature because balancing
will mitigate the effects of any bias current errors. Balancing
input capacitance will minimize ac response errors due to the
amplifier’s input capacitance and, as shown in Figure 20, noise
performance will be optimized. Figure 21 shows the required
external components for noninverting (A) and inverting (B)
configurations.
–11–
AD745
CF
R1
CB = CS
RB = RS
FOR
RS >> R1 OR R2
CB = CF || CS
RB = R1 || RS
CB
R1
RB
RS
AD745
AD745
CS
CB
CS
RS
OUTPUT
OUTPUT
NONINVERTING
CONNECTION
RB
INVERTING
CONNECTION
C00831–0–3/02(D)
R2
Figure 40. Optional External Components for Balancing Source Impedances
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
16-Lead SOIC (R) Package
0.4133 (10.50)
0.3977 (10.00)
9
16
0.2992 (7.60)
0.2914 (7.40)
PIN 1
0.4193 (10.65)
0.3937 (10.00)
8
1
0.050 (1.27)
BSC
0.0118 (0.30)
0.0040 (0.10)
0.1043 (2.65)
0.0926 (2.35)
8
0.0192 (0.49) SEATING
0
0.0125 (0.32)
0.0138 (0.35) PLANE
0.0091 (0.23)
0.0291 (0.74)
45
0.0098 (0.25)
0.0500 (1.27)
0.0157 (0.40)
Revision History
Location
Page
Deleted 8-Lead Plastic Mini-DIP (N) and 8-Lead Cerdip (Q) Packages from CONNECTION DIAGRAM . . . . . . . . . . . . . . . . . . 1
Edits to PRODUCT DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Edits to ELECTRICAL CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
Edits to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Edits to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Deleted to METALIZATION PHOTOGRAPH . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Deleted text from HOW CHIP PACKAGE TYPE AND POWER DISSIPATION AFFECT INPUT BIAS CURRENT . . . . . . . . 9
Deleted 8-Lead Plastic Mini-DIP (N) and 8-Lead Cerdip (Q) Packages from OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . 12
–12–
REV. D
PRINTED IN U.S.A.
Data Sheet changed from REV. C to REV. D.