LINER LT8303 100vin micropower isolated flyback converter with 150v/450ma switch Datasheet

LT8303
100VIN Micropower Isolated
Flyback Converter with
150V/450mA Switch
Description
Features
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5.5V to 100V Input Voltage Range
450mA, 150V Internal DMOS Power Switch
Up to 5W of Output Power
Low Quiescent Current:
70µA in Sleep Mode
280µA in Active Mode
Boundary Mode Operation at Heavy Load
Low-Ripple Burst Mode® Operation at Light Load
Minimum Load <0.5% (Typ) of Full Output
VOUT Set with a Single External Resistor
No Transformer Third Winding or Opto-Isolator
Required for Regulation
Accurate EN/UVLO Threshold and Hysteresis
Internal Compensation and Soft-Start
5-Lead TSOT-23 Package
The LT®8303 is a micropower high voltage isolated flyback
converter. By sampling the isolated output voltage directly
from the primary-side flyback waveform, the part requires
no third winding or opto-isolator for regulation. The output
voltage is programmed with a single external resistor.
Internal compensation and soft-start further reduce external
component count. Boundary mode operation provides a
small magnetic solution with excellent load regulation.
Low ripple Burst Mode operation maintains high efficiency
at light load while minimizing the output voltage ripple.
A 450mA, 150V DMOS power switch is integrated along
with all high voltage circuitry and control logic into a 5-lead
ThinSOT™ package.
The LT8303 operates from an input voltages range of
5.5V to 100V and can deliver up to 5W of isolated output
power. The high level of integration and the use of boundary
and low ripple burst modes result in a simple to use, low
component count, and high efficiency application solution
for isolated power delivery.
Applications
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Isolated Telecom, Datacom, Automotive, Industrial,
and Medical Power Supplies
Isolated Auxiliary/Housekeeping Power Supplies
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497,
and 7471522.
Typical Application
6V to 80VIN, 5VOUT Isolated Flyback Converter
4.7µF
150µH
VIN
•
LT8303
EN/UVLO
316k
GND
8303 TA01a
4.2µH
100µF
VOUT–
SW
RFB
•
100
VOUT+
5V
2.5mA TO 0.33A (VIN = 12V)
2.5mA TO 0.52A (VIN = 24V)
2.5mA TO 0.73A (VIN = 48V)
2.5mA TO 0.84A (VIN = 72V)
90
EFFICIENCY (%)
T1
6:1
VIN
6V TO 80V
Efficiency vs Load Current
80
70
60
VIN = 12V
VIN = 24V
VIN = 48V
VIN = 72V
50
40
0
100 200 300 400 500 600 700 800 900
LOAD CURRENT (mA)
8303 TA01b
8303f
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1
LT8303
Absolute Maximum Ratings
Pin Configuration
(Note 1)
TOP VIEW
SW (Note 2)............................................................ 150V
VIN.......................................................................... 100V
EN/UVLO.................................................................... VIN
RFB....................................................... VIN – 0.5V to VIN
Current into RFB.................................................... 200µA
Operating Junction Temperature Range (Notes 3, 4)
LT8303E, LT8303I.............................. –40°C to 125°C
Storage Temperature Range............... –65°C to 150°C
Order Information
EN/UVLO 1
5 VIN
GND 2
RFB 3
4 SW
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
θJA = 215°C/W
http://www.linear.com/product/LT8303#orderinfo
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT8303ES5#PBF
LT8303ES5#TRPBF
LTGXH
5-Lead Plastic TSOT-23
–40°C to 125°C
LT8303IS5#PBF
LT8303IS5#TRPBF
LTGXH
5-Lead Plastic TSOT-23
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
8303f
2
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LT8303
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted.
SYMBOL
PARAMETER
VIN
Input Voltage Range
CONDITIONS
MIN
TYP
5.5
MAX
UNIT
100
V
VIN UVLO Threshold
Rising
Falling
5.3
3.2
5.5
V
V
VIN Quiescent Current
VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
Sleep Mode (Switch Off)
Active Mode (Switch On)
1.5
200
70
280
2.5
µA
µA
µA
µA
EN/UVLO Shutdown Threshold
For Lowest Off IQ
l
0.3
0.75
EN/UVLO Enable Threshold
Falling
Hysteresis
l
1.186
1.223
0.016
1.284
V
V
IHYS
EN/UVLO Hysteresis Current
VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
VEN/UVLO = 1.3V
–0.1
2.1
–0.1
0
2.5
0
0.1
2.9
0.1
µA
µA
µA
fMAX
Maximum Switching Frequency
320
350
380
kHz
fMIN
Minimum Switching Frequency
5
7
9
kHz
IQ
V
tON(MIN)
Minimum Switch-On Time
160
ns
tOFF(MIN)
Minimum Switch-Off Time
VIN/VEN/UVLO = 12V
350
ns
tOFF(MAX)
Maximum Switch-Off Time
Backup Timer
200
µs
ISW(MAX)
Maximum SW Current Limit
450
535
620
mA
ISW(MIN)
Minimum SW Current Limit
70
105
140
mA
SW Over Current Limit
To Initiate Soft-Start
1
A
RDS(ON)
Switch On-Resistance
ISW = 100mA
3.2
Ω
ILKG
Switch Leakage Current
VIN = 100V, VSW = 150V
0.1
0.5
IRFB
RFB Regulation Current
100
102.5
µA
0.001
0.01
%/V
l
RFB Regulation Current Line Regulation
5.5V ≤ VIN ≤ 100V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The SW pin is rated to 150V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 150V as shown
in Figure 5.
Note 3: The LT8303E is guaranteed to meet performance specifications
from 0°C to 125°C operating junction temperature. Specifications over
97.5
µA
the –40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
The LT8303I is guaranteed over the full –40°C to 125°C operating junction
temperature range.
Note 4: The LT8303 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 150°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
8303f
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3
LT8303
Typical Performance Characteristics
Output Load and Line Regulation
5.3
5.3
VIN = 12V
VIN = 24V
VIN = 48V
VIN = 72V
4.8
4.7
0
300
5.1
5.0
4.9
4.7
–50 –25
0
200
150
0
Discontinuous Mode Waveforms
Burst Mode Waveforms
VSW
50V/DIV
VSW
50V/DIV
VOUT
50mV/DIV
VOUT
50mV/DIV
VOUT
50mV/DIV
8303 G04
8303 G05
2µs/DIV
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 200mA
VIN = 48V, IOUT = 3mA
VIN Shutdown Current
VIN Quiescent Current,
Sleep Mode
VIN Quiescent Current,
Active Mode
340
100
TJ = –50°C
TJ = 25°C
TJ = 150°C
90
320
TJ = 150°C
4
TJ = 25°C
70
TJ = –50°C
60
2
IQ (µA)
IQ (µA)
IQ (µA)
80
6
20
40
60
VIN (V)
80
100
8303 G07
40
TJ = 150°C
300
TJ = 25°C
280
TJ = –50°C
260
50
0
8303 G06
20µs/DIV
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 700mA
8
0
100 200 300 400 500 600 700 800 900
LOAD CURRENT (mA)
8303 G03
VSW
50V/DIV
10
0
8303 G02
Boundary Mode Waveforms
2µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
VIN = 24V
VIN = 48V
VIN = 72V
50
25 50 75 100 125 150
TEMPERATURE (°C)
8303 G01
250
100
IOUT = 3mA
IOUT = 200mA
IOUT = 700mA
4.8
100 200 300 400 500 600 700 800 900
LOAD CURRENT (mA)
FRONT PAGE APPLICATION
350
FREQUENCY (kHz)
4.9
400
FRONT PAGE APPLICATION
VIN = 48V
5.2
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
5.2
5.0
Switching Frequency
vs Load Current
Output Temperature Variation
FRONT PAGE APPLICATION
5.1
TA = 25°C, unless otherwise noted.
0
20
40
60
VIN (V)
80
100
8303 G08
240
0
20
40
60
VIN (V)
80
100
8303 G09
8303f
4
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LT8303
Typical Performance Characteristics
EN/UVLO Enable Threshold
TA = 25°C, unless otherwise noted.
RFB Regulation Current
EN/UVLO Hysteresis Current
1.28
5
105
104
1.27
4
103
102
RISING
1.24
1.23
FALLING
3
IRFB (µA)
1.25
IHYST (µA)
VEN/UVLO (V)
1.26
2
1
96
1.20
–50 –25
0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
8303 G10
RDS(ON)
Switch Current Limit
Maximum Switching Frequency
MAXIMUM CURRENT LIMIT
400
FREQUENCY (kHz)
500
400
ISW (mA)
4
300
200
2
MINIMUM CURRENT LIMIT
100
0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
300
300
200
8303 G16
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
Minimum Switch-Off Time
400
TIME (ns)
TIME (ns)
0
0
8303 G15
400
100
4
0
–50 –25
Minimum Switch-On Time
16
8
200
8303 G14
Minimum Switching Frequency
12
300
100
25 50 75 100 125 150
TEMPERATURE (°C)
8303 G13
20
25 50 75 100 125 150
TEMPERATURE (°C)
500
600
6
0
8303 G12
700
1SW = 100mA
0
–50 –25
95
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
8303 G11
8
RESISTANCE (Ω)
99
97
1.21
FREQUENCY (kHz)
100
98
1.22
10
101
200
100
0
25 50 75 100 125 150
TEMPERATURE (°C)
8303 G17
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
8303 G18
8303f
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5
LT8303
Pin Functions
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8303. Pull the pin
below 0.3V to shut down the LT8303. This pin has an accurate 1.223V threshold and can be used to program a VIN
undervoltage lockout (UVLO) threshold using a resistor
divider from VIN to ground. A 2.5µA current hysteresis
allows the programming of VIN UVLO hysteresis. If neither
function is used, tie this pin directly to VIN.
GND (Pin 2): Ground. Tie this pin directly to local ground
plane.
RFB (Pin 3): Input Pin for External Feedback Resistor.
Connect a resistor from this pin to the transformer primary SW pin. The ratio of the RFB resistor to the internal
trimmed 12.23k resistor, times the internal bandgap
reference, determines the output voltage (plus the effect
of any non-unity transformer turns ratio). Minimize trace
area at this pin.
SW (Pin 4): Drain of the 150V Internal DMOS Power
Switch. Minimize trace area at this pin to reduce EMI and
voltage spikes.
VIN (Pin 5): Input Supply. The VIN pin supplies current
to internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the RFB pin. Locally
bypass this pin to ground with a capacitor.
8303f
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LT8303
Block Diagram
T1
NPS:1
VIN
CIN
LPRI
•
•
DOUT
VOUT+
LSEC
COUT
RFB
5
3
VIN
VOUT–
4
RFB
SW
BOUNDARY
DETECTOR
1:4
M3
M2
OSCILLATOR
–
25µA
RREF
12.23kΩ
1.223V
+
–
gm
+
S
A3
R
Q
DRIVER
M1
R1
1
–
EN/UVLO
2.5µA
R2
1.223V
M4
+
+
RSENSE
A2
A1
VIN
REFERENCE
REGULATORS
–
GND
2
8303 BD
8303f
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7
LT8303
Operation
The LT8303 is a current mode switching regulator IC
designed specially for the isolated flyback topology. The
key problem in isolated topologies is how to communicate
the output voltage information from the isolated secondary
side of the transformer to the primary side for regulation.
Historically, opto-isolators or extra transformer windings
communicate this information across the isolation boundary. Opto-isolator circuits waste output power, and the
extra components increase the cost and physical size of
the power supply. Opto-isolators can also cause system
issues due to limited dynamic response, nonlinearity, unitto-unit variation and aging over lifetime. Circuits employing
extra transformer windings also exhibit deficiencies, as
using an extra winding adds to the transformer’s physical
size and cost, and dynamic response is often mediocre.
The LT8303 samples the isolated output voltage through
the primary-side flyback pulse waveform. In this manner,
neither opto-isolator nor extra transformer winding is required for regulation. Since the LT8303 operates in either
boundary conduction mode or discontinuous conduction
mode, the output voltage is always sampled on the SW
pin when the secondary current is zero. This method improves load regulation without the need of external load
compensation components.
The LT8303 is a simple to use micropower isolated flyback
converter housed in a 5-lead TSOT-23 package. The output
voltage is programmed with a single external resistor. By
integrating the loop compensation and soft-start inside, the
part further reduces the number of external components.
As shown in the Block Diagram, many of the blocks are
similar to those found in traditional switching regulators
including reference, regulators, oscillator, logic, current
amplifier, current comparator, driver, and power switch.
The novel sections include a flyback pulse sense circuit,
a sample-and-hold error amplifier, and a boundary mode
detector, as well as the additional logic for boundary
conduction mode, discontinuous conduction mode, and
low ripple Burst Mode operation.
Boundary Conduction Mode Operation
The LT8303 features boundary conduction mode operation
at heavy load, where the chip turns on the primary power
switch when the secondary current is zero. Boundary
conduction mode is a variable frequency, variable peakcurrent switching scheme. The power switch turns on
and the transformer primary current increases until an
internally controlled peak current limit. After the power
switch turns off, the voltage on the SW pin rises to the
output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
the SW pin voltage collapses and rings around VIN. A
boundary mode detector senses this event and turns the
power switch back on.
Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduction mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit sub-harmonic oscillation.
Discontinuous Conduction Mode Operation
As the load gets lighter, boundary conduction mode increases the switching frequency and decreases the switch
peak current at the same ratio. Running at a higher switching
frequency up to several MHz increases switching and gate
charge losses. To avoid this scenario, the LT8303 has an
additional internal oscillator, which clamps the maximum
switching frequency to be less than 350kHz (typical). Once
the switching frequency hits the internal frequency clamp,
the part starts to delay the switch turn-on and operates in
discontinuous conduction mode.
Low Ripple Burst Mode Operation
Unlike traditional flyback converters, the LT8303 has to
turn on and off at least for a minimum amount of time
and with a minimum frequency to allow accurate sampling
of the output voltage. The inherent minimum switch current limit and minimum switch-off time are necessary to
guarantee the correct operation of specific applications.
As the load gets very light, the LT8303 starts to fold back
the switching frequency while keeping the minimum switch
current limit. So the load current is able to decrease while
still allowing minimum switch-off time for the sample-andhold error amplifier. Meanwhile, the part switches between
sleep mode and active mode, thereby reducing the effec8303f
8
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LT8303
Operation
tive quiescent current to improve light load efficiency. In
this condition, the LT8303 operates in low ripple Burst
Mode. The typical 7kHz minimum switching frequency
determines how often the output voltage is sampled and
also the minimum load requirement.
Applications Information
Output Voltage
The RFB resistor as depicted in the Block Diagram is the
only external resistor used to program the output voltage.
The LT8303 operates similar to traditional current mode
switchers, except in the use of a unique flyback pulse
sense circuit and a sample-and-hold error amplifier, which
sample and therefore regulate the isolated output voltage
from the flyback pulse.
Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the VIN supply. The
amplitude of the flyback pulse, i.e., the difference between
the SW pin voltage and VIN supply, is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = Output diode forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current IRFB by
the flyback pulse sense circuit (M2 and M3). This current IRFB also flows through the internal trimmed 12.23k
RREF resistor to generate a ground-referred voltage. The
resulting voltage feeds to the inverting input of the sampleand-hold error amplifier. Since the sample-and-hold error
amplifier samples the voltage when the secondary current
is zero, the (ISEC • ESR) term in the VFLBK equation can be
assumed to be zero.
The bandgap reference voltage VBG, 1.223V, feeds to the
non-inverting input of the sample-and-hold error amplifier. The relatively high gain in the overall loop causes
the voltage across RREF resistor to be nearly equal to the
bandgap reference voltage VBG. The resulting relationship
between VFLBK and VBG can be expressed as:
 VFLBK 
 R  • RREF = VBG
FB
or
 V 
VFLBK =  BG  • RFB = IRFB • RFB
 RREF 
VBG = Bandgap reference voltage
IRFB = RFB regulation current = 100µA
Combination with the previous VFLBK equation yields an
equation for VOUT, in terms of the RFB resistor, transformer
turns ratio, and diode forward voltage:
R 
VOUT = 100µA •  FB  − VF
 NPS 
Output Temperature Coefficient
The first term in the VOUT equation does not have temperature dependence, but the output diode forward voltage VF
has a significant negative temperature coefficient (–1mV/°C
to –2mV/°C). Such a negative temperature coefficient produces approximately 200mV to 300mV voltage variation
on the output voltage across temperature.
For higher voltage outputs, such as 12V and 24V, the output
diode temperature coefficient has a negligible effect on the
output voltage regulation. For lower voltage outputs, such
as 3.3V and 5V, however, the output diode temperature
coefficient does count for an extra 2% to 5% output voltage
regulation. For customers requiring tight output voltage
regulation across temperature, please refer to other LTC
parts with integrated temperature compensation features.
8303f
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9
LT8303
Applications Information
Selecting Actual RFB Resistor Value
Output Power
The LT8303 uses a unique sampling scheme to regulate
the isolated output voltage. Due to the sampling nature,
the scheme contains repeatable delays and error sources,
which will affect the output voltage and force a re-evaluation
of the RFB resistor value. Therefore, a simple two-step
process is required to choose feedback resistor RFB.
A flyback converter has a complicated relationship between
the input and output currents compared to a buck or a
boost converter. A boost converter has a relatively constant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum output
current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flyback converter has both discontinuous input and output
currents which make it similar to a non-isolated buck-boost
converter. The duty cycle will affect the input and output
currents, making it hard to predict output power. In addition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.
Rearrangement of the expression for VOUT in the Output
Voltage section yields the starting value for RFB:
RFB =
(
NPS • VOUT + VF
100µA
)
VOUT = Output voltage
VF = Output diode forward voltage = ~0.3V
NPS = Transformer effective primary-to-secondary
turns ratio
Power up the application with the starting RFB value and
other components connected, and measure the regulated
output voltage, VOUT(MEAS). The final RFB value can be
adjusted to:
RFB(FINAL) =
VOUT
VOUT(MEAS)
• RFB
Once the final RFB value is selected, the regulation accuracy
from board to board for a given application will be very
consistent, typically under ±5% when including device
variation of all the components in the system (assuming
resistor tolerances and transformer windings matching
within ±1%). However, if the transformer or the output
diode is changed, or the layout is dramatically altered,
there may be some change in VOUT.
The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,
12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 120V during the switch-off time. 30V of margin is left for leakage
inductance voltage spike. To achieve this power level at
a given input, a winding ratio value must be calculated
to stress the switch to 120V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
One design example would be a 5V output converter with
a minimum input voltage of 30V and a maximum input
voltage of 80V. A six-to-one winding ratio fits this design
example perfectly and outputs equal to 4.35W at 80V but
lowers to 2.95W at 30V.
The following equations calculate output power:
POUT = η • VIN • D •ISW(MAX) • 0.5
η = Efficiency = 85%
( VOUT + VF ) • NPS
D = DutyCycle =
( VOUT + VF ) • NPS + VIN

ISW(MAX) = Maximum switch current limit = 450mA
8303f
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LT8303
Applications Information
6
6
OUTPUT POWER (W)
N = 8:1
4
N = 6:1
3
N = 4:1
2
1
0
MAXIMUM
OUTPUT
POWER
5
N = 12:1
OUTPUT POWER (W)
MAXIMUM
OUTPUT
POWER
5
N = 8:1
N = 6:1
4
N = 4:1
3
N = 2:1
2
1
ASSUME 80% EFFICIENCY
0
20
40
60
INPUT VOLTAGE (V)
80
0
100
ASSUME 85% EFFICIENCY
0
20
40
60
INPUT VOLTAGE (V)
80
8303 F01
8303 F02
Figure 1. Output Power for 3.3V Output
OUTPUT POWER (W)
6
MAXIMUM
OUTPUT
POWER
5
Figure 2. Output Power for 5V Output
5
N = 3:1
4
N = 2:1
3
N = 1:1
2
N = 2:1
N = 3:2
4
N = 1:1
3
N = 1:2
2
1
1
0
MAXIMUM
OUTPUT
POWER
N = 4:1
OUTPUT POWER (W)
6
ASSUME 85% EFFICIENCY
0
20
40
60
INPUT VOLTAGE (V)
80
0
100
ASSUME 85% EFFICIENCY
0
20
40
60
INPUT VOLTAGE (V)
Figure 3. Output Power for 12V Output
Primary Inductance Requirement
The LT8303 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction
of secondary current reflects the output voltage on the
primary SW pin. The sample-and-hold error amplifier needs
a minimum 350ns to settle and sample the reflected output
voltage. In order to ensure proper sampling, the secondary winding needs to conduct current for a minimum of
350ns. The following equation gives the minimum value
for primary-side magnetizing inductance:
(
tOFF(MIN) • NPS • VOUT + VF
ISW(MIN)
)
ISW(MIN) = Minimum switch current limit = 105mA
100
Figure 4. Output Power for 24V Output
In addition to the primary inductance requirement for
the minimum switch-off time, the LT8303 has minimum
switch-on time that prevents the chip from turning on
the power switch shorter than approximately 160ns. This
minimum switch-on time is mainly for leading-edge blanking the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loop will lose its ability to regulate. Therefore, the following
equation relating to maximum input voltage must also be
followed in selecting primary-side magnetizing inductance:
LPRI ≥
tOFF(MIN) = Minimum switch-off time = 350ns
80
8303 F04
8303 F03
LPRI ≥
100
tON(MIN) • VIN(MAX)
ISW(MIN)
tON(MIN) = Minimum Switch-On Time = 160ns
For more information www.linear.com/LT8303
8303f
11
LT8303
Applications Information
In general, choose a transformer with its primary magnetizing inductance about 40% to 60% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed
flyback transformers for use with the LT8303. Table 1
shows the details of these transformers.
Selecting a Transformer
Note that when choosing the RFB resistor to set output
voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast,
the use of simple ratios of small integers, e.g., 4:1, 2:1,
1:1, provides more freedom in settling total turns and
mutual inductance.
Transformer specification and design is perhaps the most
critical part of successfully applying the LT8303. In addition
to the usual list of guidelines dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
Turns Ratio
Table 1. Predesigned Transformers – Typical Specifications
TRANSFORMER
PART NUMBER
TARGET APPLICATION
DIMENSION
(W × L × H) (mm)
LPRI, µH
TYP
LLKG, µH
TYP (MAX)
NP: NS
VENDOR
VIN (V)
VOUT (V)
750315825
13.36 × 10.16 × 8.64
150
3 (6)
8:1
Wurth Elektronik
36 to 75
3.3
0.9
750315826
13.36 × 10.16 × 8.64
150
2 (4)
6:1
Wurth Elektronik
36 to 75
5
0.65
750315827
13.36 × 10.16 × 8.65
150
1.8 (3.6)
4:1
Wurth Elektronik
36 to 75
5
0.5
750315828
13.36 × 10.16 × 8.66
150
1.6 (3.2)
2:1
Wurth Elektronik
36 to 75
12
0.25
750315829
13.36 × 10.16 × 8.67
150
1.5 (3)
1:1
Wurth Elektronik
36 to 75
24
0.12
750315830
13.36 × 10.16 × 8.68
150
1.9 (3.8)
1:2
Wurth Elektronik
36 to 75
48
0.06
750315833
13.36 × 10.16 × 8.71
150
1.5 (3)
2:1:1
Wurth Elektronik
36 to 75
12/12
0.12/0.12
750315834
13.36 × 10.16 × 8.72
150
2.6 (5.2)
6:1:1
Wurth Elektronik
36 to 75
5/5
0.32/0.32
PS15-108
14 × 10 × 9.2
150
(5)
8:1
Sumida
36 to 75
3.3
0.9
PS15-109
14 × 10 × 9.2
150
(5)
6:1
Sumida
36 to 75
5
0.65
PS15-110
14 × 10 × 9.2
150
(5)
4:1
Sumida
36 to 75
5
0.5
PS15-111
14 × 10 × 9.2
150
(5)
2:1
Sumida
36 to 75
12
0.25
PS15-112
14 × 10 × 9.2
150
(5)
1:1
Sumida
36 to 75
24
0.12
PS15-113
14 × 10 × 9.2
150
(5)
1:2
Sumida
36 to 75
48
0.06
IOUT (A)
8303f
12
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LT8303
Applications Information
Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V or
5V), a larger N:1 turns ratio can be used with multiple
primary windings relative to the secondary to maximize
the transformer’s current gain (and output power).
However, remember that the SW pin sees a voltage that
is equal to the maximum input supply voltage plus the
output voltage multiplied by the turns ratio. In addition,
leakage inductance will cause a voltage spike (VLEAKAGE)
on top of this reflected voltage. This total quantity needs
to remain below the 150V absolute maximum rating of
the SW pin to prevent breakdown of the internal power
switch. Together these conditions place an upper limit
on the turns ratio, NPS, for a given application. Choose a
turns ratio low enough to ensure:
NPS <
150V − VIN(MAX) − VLEAKAGE
Saturation Current
The current in the transformer windings should not exceed
its rated saturation current. Energy injected once the core is
saturated will not be transferred to the secondary and will
instead be dissipated in the core. When designing custom
transformers to be used with the LT8303, the saturation
current should always be specified by the transformer
manufacturers.
Winding Resistance
Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output voltage
regulation will be maintained independent of winding resistance due to the boundary/discontinuous conduction
mode operation of the LT8303.
Leakage Inductance and Snubbers
VOUT + VF
For lower output power levels, choose a smaller N:1 turns
ratio to alleviate the SW pin voltage stress. Although a
1:N turns ratio makes it possible to have very high output
voltages without exceeding the breakdown voltage of the
internal power switch, the multiplied parasitic capacitance
through turns ratio coupled with the relatively resistive
150V internal power switch may cause the switch turn-on
current spike ringing beyond 160ns leading-edge blanking,
thereby producing light load instability in certain applications. So any 1:N turns ratio should be fully evaluated
before its use with the LT8303.
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer specifies turns ratio accuracy within ±1%.
Transformer leakage inductance on either the primary or
secondary causes a voltage spike to appear on the primary
after the power switch turns off. This spike is increasingly
prominent at higher load currents where more stored energy must be dissipated. It is very important to minimize
transformer leakage inductance.
When designing an application, adequate margin should be
kept for the worst-case leakage voltage spikes even under
overload conditions. In most cases shown in Figure 5, the
reflected output voltage on the primary plus VIN should
be kept below 120V. This leaves at least 30V margin for
the leakage spike across line and load conditions. A larger
voltage margin will be required for poorly wound transformers or for excessive leakage inductance.
In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trigger boundary mode detector, the LT8303 internally blanks
the boundary mode detector for approximately 250ns. Any
remaining voltage ringing after 250ns may turn the power
switch back on again before the secondary current falls
to zero. So the leakage inductance spike ringing should
be limited to less than 250ns.
8303f
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13
LT8303
Applications Information
VSW
VSW
<150V
VSW
<150V
<150V
VLEAKAGE
VLEAKAGE
<120V
VLEAKAGE
<120V
<120V
tOFF > 350ns
tOFF > 350ns
tOFF > 350ns
tSP < 250ns
tSP < 250ns
tSP < 250ns
TIME
TIME
No Snubber
TIME
with DZ Snubber
with RC Snubber
8303 F05
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
Lℓ
Lℓ
•
Z
D
•
C
•
•
R
8303 F06
DZ Snubber
RC Snubber
Figure 6. Snubber Circuits
A snubber circuit is recommended for most applications.
Two types of snubber circuits shown in Figure 6 that can
protect the internal power switch include the DZ (diodeZener) snubber and the RC (resistor-capacitor) snubber. The
DZ snubber ensures well defined and consistent clamping
voltage and has slightly higher power efficiency, while the
RC snubber quickly damps the voltage spike ringing and
provides better load regulation and EMI performance.
Figure 5 shows the flyback waveforms with the DZ and
RC snubbers.
For the DZ snubber, proper care must be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leakage inductance spike. Choose a diode that has a reversevoltage rating higher than the maximum SW pin voltage.
The Zener diode breakdown voltage should be chosen to
balance power loss and switch voltage protection. The best
compromise is to choose the largest voltage breakdown.
Use the following equation to make the proper choice:
VZENER(MAX) ≤ 150V – VIN(MAX)
For an application with a maximum input voltage of 80V,
choose a 62V Zener diode, the VZENER(MAX) of which is
around 65V and below the 70V maximum.
The power loss in the clamp will determine the power rating of the Zener diode. Power loss in the clamp is highest
at maximum load and minimum input voltage. The switch
current is highest at this point along with the energy stored
in the leakage inductance. A 0.5W Zener will satisfy most
applications when the highest VZENER is chosen.
8303f
14
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LT8303
Applications Information
Tables 2 and 3 show some recommended diodes and
Zener diodes.
Table 2. Recommended Zener Diodes
VZENER
(V)
POWER
(W)
CASE
VENDOR
MMSZ5266BT1G
68
0.5
SOD-123
On Semi
MMSZ5270BT1G
91
0.5
SOD-123
CMHZ5266B
68
0.5
SOD-123
CMHZ5267B
75
0.5
SOD-123
BZX84J-68
68
0.5
SOD323F NXP
BZX100A
100
0.5
SOD323F
PART
Central
Semiconductor
Table 3. Recommended Diodes
PART
I (A)
VREVERSE
(V)
BAV21W
0.625
200
SOD-123 Diodes Inc.
BAV20W
0.625
150
SOD-123
CASE
VENDOR
The recommended approach for designing an RC snubber
is to measure the period of the ringing on the SW pin when
the power switch turns off without the snubber and then
add capacitance (starting with 100pF) until the period of
the ringing is 1.5 to 2 times longer. The change in period
will determine the value of the parasitic capacitance, from
which the parasitic inductance can be determined from
the initial period, as well. Once the value of the SW node
capacitance and inductance is known, a series resistor can
be added to the snubber capacitance to dissipate power
and critically dampen the ringing. The equation for deriving
the optimal series resistance using the observed periods
( tPERIOD and tPERIOD(SNUBBED)) and snubber capacitance
(CSNUBBER) is:
CPAR =
CSNUBBER
2
Note that energy absorbed by the RC snubber will be
converted to heat and will not be delivered to the load.
In high voltage or high current applications, the snubber
may need to be sized for thermal dissipation.
Undervoltage Lockout (UVLO)
A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). The EN/UVLO pin
falling threshold is set at 1.223V with 16mV hysteresis.
In addition, the EN/UVLO pin sinks 2.5µA when the voltage at the pin is below 1.223V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:
1.239V • (R1+ R2)
+ 2.5µA • R1
R2
1.223V • (R1+ R2)
VIN(UVLO−) =
R2
VIN(UVLO+) =
Figure 7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8303 in shutdown with quiescent current less than 2.5µA.
VIN
R1
EN/UVLO
LT8303
R2
RUN/STOP
CONTROL
(OPTIONAL)
GND
8303 F07
Figure 7. Undervoltage Lockout (UVLO)
 tPERIOD(SNUBBED) 

 − 1
t
PERIOD
L PAR =
tPERIOD 2
CPAR • 4π 2
RSNUBBER =
LPAR
CPAR
8303f
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15
LT8303
Applications Information
Minimum Load Requirement
Design Example
The LT8303 samples the isolated output voltage from
the primary-side flyback pulse waveform. The flyback
pulse occurs once the primary switch turns off and the
secondary winding conducts current. In order to sample
the output voltage, the LT8303 has to turn on and off at
least for a minimum amount of time and with a minimum
frequency. The LT8303 delivers a minimum amount of
energy even during light load conditions to ensure accurate output voltage information. The minimum energy
delivery creates a minimum load requirement, which can
be approximately estimated as:
Use the following design example as a guide to design
applications for the LT8303. The design example involves
designing a 12V output with a 200mA load current and an
input range from 30V to 80V.
ILOAD(MIN) =
LPRI •ISW(MIN)2 • f MIN
VIN(MIN) = 30V, VIN(NOM) = 48V, VIN(MAX) = 80V,
VOUT = 12V, IOUT = 200mA
Step 1: Select the Transformer Turns Ratio.
NPS <
150V − VIN(MAX) − VLEAKAGE
VOUT + VF
VLEAKAGE = Margin for transformer leakage spike = 30V
VF = Output diode forward voltage = ~0.3V
2 • VOUT
Example:
LPRI = Transformer primary inductance
ISW(MIN) = Minimum switch current limit = 140mA (Max)
fMIN = Minimum switching frequency = 9kHz (Max)
The LT8303 typically needs less than 0.5% of its full output
power as minimum load. Alternatively, a Zener diode with its
breakdown of 20% higher than the output voltage can serve
as a minimum load if pre-loading is not acceptable. For a 5V
output, use a 6V Zener with cathode connected to the output.
Output Short Protection
When the output is heavily overloaded or shorted, the
reflected SW pin waveform rings longer than the internal
blanking time. After the 350ns minimum switch-off time,
the excessive ring falsely trigger the boundary mode
detector and turn the power switch back on again before
the secondary current falls to zero. Under this condition,
the LT8303 runs into continuous conduction mode at
350kHz maximum switching frequency. Depending on
the VIN supply voltage, the switch current may run away
and exceed 450mA maximum current limit. Once the
switch current hits 1A over current limit, a soft-start cycle
initiates and throttles back both switch current limit and
switch frequency. This output short protection prevents the
switch current from running away and limits the average
output diode current.
NPS <
150V − 80V − 30V
= 3.3
12V + 0.3V
The choice of transformer turns ratio is critical in determining output current capability of the converter. Table 4
shows the switch voltage stress and output current capability at different transformer turns ratio.
Table 4. Switch Voltage Stress and Output Current Capability
vs Turns Ratio
NPS
VSW(MAX) at
VIN(MAX) (V)
IOUT(MAX) at
VIN(MIN) (mA)
DUTY CYCLE (%)
1:1
92.3
139
13 to 29
2:1
104.6
215
24 to 45
3:1
116.9
264
32 to 55
Since both NPS = 2 and NPS = 3 can meet the 200mA output
current requirement, NPS = 2 is chosen in this example
to allow more margin for transformer leakage inductance
voltage spike.
8303f
16
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LT8303
Applications Information
Step 2: Determine the Primary Inductance.
Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:
LPRI ≥
LPRI ≥
(
tOFF(MIN) • NPS • VOUT + VF
ISW(MIN)
)
Example:
IDIODE(MAX) = 1.07A
Next calculate reverse voltage requirement using maximum VIN:
VREVERSE = VOUT +
tON(MIN) • VIN(MAX)
VIN(MAX)
NPS
Example:
ISW(MIN)
VREVERSE = 12V +
tOFF(MIN) = 350ns
72V
= 48V
2
tON(MIN) = 160ns
The DFLS2100 (2A, 100V diode) from Diodes Inc. is chosen.
ISW(MIN) = 105mA
Step 4: Choose the Output Capacitor.
Example:
350ns • 2 • (12V + 0.3V)
= 82µH
105mA
160ns • 80V
LPRI ≥
= 122µH
105mA
LPRI ≥
Most transformers specify primary inductance with a tolerance of ±20%. With other component tolerance considered,
choose a transformer with its primary inductance 40% to
60% larger than the minimum values calculated above.
LPRI = 150µH is then chosen in this example.
The transformer also needs to be rated for the correct
saturation current level across line and load conditions. A
saturation current rating larger than 620mA is necessary
to work with the LT8303. The PS15-111 from Sumida is
chosen as the flyback transformer.
Step 3: Choose the Output Diode.
Two main criteria for choosing the output diode include
forward current rating and reverse voltage rating. The
maximum load requirement is a good first-order guess
as the average current requirement for the output diode.
A conservative metric is the maximum switch current limit
multiplied by the turns ratio,
IDIODE(MAX) = ISW(MAX) • NPS
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
and cost of a larger capacitor. Use the equation below to
calculate the output capacitance:
COUT =
LPRI •ISW 2
2 • VOUT • ∆VOUT
Example:
Design for output voltage ripple less than 1% of VOUT,
i.e., 120mV.
COUT =
150µH • (0.535A)2
= 14.9µF
2 • 12V • 0.12V
Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop to 40% of quoted
capacitance at the maximum voltage rating. So a 22µF,
25V rating X5R or X7R ceramic capacitor is chosen.
Step 5: Design Snubber Circuit.
The snubber circuit protects the power switch from leakage
inductance voltage spike. A DZ snubber is recommended
for this application because of lower leakage inductance
and larger voltage margin. The Zener and the diode need
to be selected.
8303f
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17
LT8303
Applications Information
The maximum Zener breakdown voltage is set according
to the maximum VIN:
VZENER(MAX) ≤ 150V – VIN(MAX)
Example:
Example:
A 62V Zener with a maximum of 65V will provide optimal
protection and minimize power loss. So a 62V, 0.5W Zener
from Central Semiconductor (CMHZ5265B) is chosen.
Choose a diode that is fast and has sufficient reverse
voltage breakdown:
Choose 2.5V of hysteresis,
R1 = 1M
Determine the UVLO thresholds and calculate R2 resistor
value:
VIN(UVLO+) =
VREVERSE > VSW(MAX)
VSW(MAX) = VIN(MAX) + VZENER(MAX)
1.239V • (R1+ R2)
+ 2.5µA • R1
R2
Example:
Example:
Set VIN UVLO rising threshold to 34.5V,
VREVERSE > 144V
A 200V, 1A diode from Central Semiconductor
(CMMRIU-02) is chosen.
R2 = 49.9k
VIN(UVLO+) = 28.6V
VIN(UVLO–) = 25.7V
Step 6: Select the RFB Resistor.
Use the following equation to calculate the starting value
for RFB:
NPS • (VOUT + VF )
100µA
Step 8: Ensure minimum load.
The theoretical minimum load can be approximately
estimated as:
ILOAD(MIN) =
Example:
RFB =
Determine the amount of hysteresis required and calculate
R1 resistor value:
VIN(HYS) = 2.5µA • R1
VZENER(MAX) ≤ 150V – 80V = 70V
RFB =
Step 7: Select the EN/UVLO Resistors.
2 • (12V + 0.3V)
= 246k
100µA
Depending on the tolerance of standard resistor values,
the precise resistor value may not exist. For 1% standard
values, a 243k resistor in series with a 3.01k resistor
should be close enough. As discussed in the Application
Information section, the final RFB value should be adjusted
on the measured output voltage.
150µH • (140mA)2 • 9kHz
= 1.1mA
2 • 12V
Remember to check the minimum load requirement in
real application. The minimum load occurs at the point
where the output voltage begins to climb up as the converter delivers more energy than what is consumed at
the output. The real minimum load for this application is
about 1mA. In this example, a 12.1k resistor is selected
as the minimum load.
8303f
18
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LT8303
Typical Applications
30V to 80VIN, 3.3VOUT Isolated Flyback Converter
4.7µF
100V
Z1
VIN
1M
D1
LT8303
EN/UVLO
•
150µH
2.3µH
•
330µF
6.3V
VOUT–
SW
287k
49.9k
VOUT+
3.3V
4mA TO 0.9A (VIN = 36V)
4mA TO 1A (VIN = 48V)
4mA TO 1.1A (VIN = 72V)
D2
T1
8:1
VIN
30V TO 80V
D1: CENTRAL CMMR1U-02
D2: DIODES SBR3U30P1-7
T1: SUMIDA PS15-108
Z1: CENTRAL CMHZ5265B
RFB
GND
8303 TA02a
Efficiency vs Load Current
Output Load and Line Regulation
100
3.50
3.45
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
90
80
70
60
VIN = 36V
VIN = 48V
VIN = 72V
50
40
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
1.0
3.40
3.35
3.30
3.25
3.20
VIN = 36V
VIN = 48V
VIN = 72V
3.15
1.2
3.10
0
8303 TA02b
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
1.0
1.2
8303 TA02c
8303f
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19
LT8303
Typical Applications
30V to 80VIN, 5VOUT Isolated Flyback Converter
VIN
30V TO 80V
T1
6:1
4.7µF
100V
1M
D1
LT8303
EN/UVLO
•
Z1
150µH
VIN
D2
4.2µH
•
100µF
10V
VOUT–
SW
49.9k
316k
RFB
GND
VOUT+
5V
2.5mA TO 0.65A (VIN = 36V)
2.5mA TO 0.73A (VIN = 48V)
2.5mA TO 0.84A (VIN = 72V)
D1: CENTRAL CMMR1U-02
D2: DIODES SBR3U30P1-7
T1: SUMIDA PS15-109
Z1: CENTRAL CMHZ5265B
8303 TA03a
Output Load and Line Regulation
100
5.3
90
5.2
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
Efficiency vs Load Current
80
70
60
VIN = 36V
VIN = 48V
VIN = 72V
50
40
0
100 200 300 400 500 600 700 800 900
LOAD CURRENT (mA)
5.1
5.0
4.9
VIN = 36V
VIN = 48V
VIN = 72V
4.8
4.7
0
8303 TA03b
100 200 300 400 500 600 700 800 900
LOAD CURRENT (mA)
8303 TA03c
8303f
20
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LT8303
Typical Applications
30V to 80VIN, 12VOUT Isolated Flyback Converter
VIN
30V TO 80V
•
Z1
4.7µF
100V
150µH
VIN
1M
D1
LT8303
EN/UVLO
37.5µH
•
22µF
25V
SW
249k
49.9k
RFB
GND
8303 TA04a
12.6
90
12.4
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
Output Load and Line Regulation
100
80
70
60
40
VIN = 36V
VIN = 48V
VIN = 72V
0
40
VOUT–
D1: CENTRAL CMMR1U-02
D2: DIODES DFLS2100-7
T1: SUMIDA PS15-111
Z1: CENTRAL CMHZ5265B
Efficiency vs Load Current
50
VOUT+
12V
1mA TO 250mA (VIN = 36V)
1mA TO 270mA (VIN = 48V)
1mA TO 310mA (VIN = 72V)
D2
T1
2:1
80 120 160 200 240 280 320
LOAD CURRENT (mA)
12.2
12.0
11.8
VIN = 36V
VIN = 48V
VIN = 72V
11.6
11.4
0
8303 TA04b
40
80 120 160 200 240 280 320
LOAD CURRENT (mA)
8303 TA04c
8303f
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21
LT8303
Typical Applications
30V to 80VIN, 24VOUT Isolated Flyback Converter
VIN
30V TO 80V
T1
1:1
4.7µF
100V
VIN
1M
LT8303
EN/UVLO
•
Z1
150µH
D1
VOUT+
24V
0.6mA TO 120mA (VIN = 36V)
0.6mA TO 140mA (VIN = 48V)
0.6mA TO 150mA (VIN = 72V)
D2
150µH
•
22µF
50V
VOUT–
SW
249k
49.9k
RFB
GND
D1: CENTRAL CMMR1U-02
D2: DIODES DFLS1200-7
T1: SUMIDA PS15-112
Z1: CENTRAL CMHZ5265B
8303 TA05a
Output Load and Line Regulation
25.2
90
24.8
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
Efficiency vs Load Current
100
80
70
60
VIN = 36V
VIN = 48V
VIN = 72V
50
40
0
20
40 60 80 100 120 140 160
LOAD CURRENT (mA)
24.4
24.0
23.6
VIN = 36V
VIN = 48V
VIN = 72V
23.2
22.8
8303 TA05b
0
20
40 60 80 100 120 140 160
LOAD CURRENT (mA)
8303 TA05c
8303f
22
For more information www.linear.com/LT8303
LT8303
Package Description
Please refer to http://www.linear.com/product/LT8303#packaging for the most recent package drawings.
S5 Package
5-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1635)
0.62
MAX
0.95
REF
2.90 BSC
(NOTE 4)
1.22 REF
1.4 MIN
3.85 MAX 2.62 REF
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45 TYP
5 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1.90 BSC
S5 TSOT-23 0302
8303f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LT8303
23
LT8303
Typical Application
30V to 80VIN, 48VOUT Isolated Flyback Converter
VIN
30V TO 80V
T1
1:2
4.7µF
100V
1M
•
Z1
150µH
VIN
LT8303
D1
EN/UVLO
VOUT+
48V
0.3mA TO 60mA (VIN = 36V)
0.3mA TO 70mA (VIN = 48V)
0.3mA TO 75mA (VIN = 72V)
D2
600µH
•
4.7µF
100V
VOUT–
SW
243k
49.9k
D1: CENTRAL CMMR1U-02
D2: DIODES SBR1U400P1-7
T1: SUMIDA PS15-113
Z1: CENTRAL CMHZ5265B
RFB
GND
8303 TA06a
Output Load and Line Regulation
100
50.4
90
49.6
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
Efficiency vs Load Current
80
70
60
VIN = 36V
VIN = 48V
VIN = 72V
50
40
0
10
20 30 40 50 60
LOAD CURRENT (mA)
70
80
48.8
48.0
47.2
VIN = 36V
VIN = 48V
VIN = 72V
46.4
45.6
0
10
20 30 40 50 60
LOAD CURRENT (mA)
8303 TA06b
70
80
8303 TA06c
Related Parts
PART NUMBER
DESCRIPTION
LT8300
LT8304
100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23
Low IQ Monolithic No-Opto Flyback, 8-Lead SO-8E
100VIN Micropower Isolated Flyback Converter with 150V/2A Switch
COMMENTS
LT8301
42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch
Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23
LT8302
42VIN Micropower Isolated Flyback Converter with 65V/3.6mA Switch
Low IQ Monolithic No-Opto Flyback, 8-Lead SO-8E
LT8309
Secondary-Side Synchronous Rectifier Driver
4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead
TSOT-23
LT3748
100V Isolated Flyback Controller
5V ≤ VIN ≤ 100V, No-Opto Flyback, MSOP-16(12)
LT3798
Off-Line Isolated No-Opto Flyback Controller with Active PFC
VIN and VOUT Limited Only by External Components
LT3757/LT3759/
LT3758
40V/100V Flyback/Boost Controller
Universal Controllers with Small Package and Powerful
Gate Drive
LT3957/LT3958
40V/80V Boost/Flyback Converter
Monolithic with Integrated 5A/3.3A Switch
LTC3803/LTC3803-3/ 200kHz/300kHz Flyback Controller in SOT-23
LTC3803-5
VIN and VOUT Limited Only by External Components
LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers
VIN and VOUT Limited Only by External Components
8303f
24 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LT8303
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LT8303
LT 0816 • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2016