AD ADP1611 20 v,1.2 mhz step-up dc-to-dc switching converter Datasheet

20 V,1.2 MHz Step-Up
DC-to-DC Switching Converter
ADP1611
FEATURES
GENERAL DESCRIPTION
Fully integrated 1.2 A , 0.23 Ω power switch
Pin-selectable 700 kHz or 1.2 MHz PWM frequency
90% efficiency
Adjustable output voltage up to 20 V
3% output regulation accuracy
Adjustable soft start
Input undervoltage lockout
MSOP 8-lead package
The ADP1611 is a step-up dc-to-dc switching converter with an
integrated 1.2 A, 0.23 Ω power switch capable of providing an
output voltage as high as 20 V. With a package height of less
than 1.1 mm, the ADP1611 is optimal for space-constrained
applications such as portable devices or thin film transistor
(TFT) liquid crystal displays (LCDs).
The ADP1611 operates in pulse-width modulation (PWM)
current mode with up to 90% efficiency. Adjustable soft start
prevents inrush currents at startup. The pin-selectable switching
frequency and PWM current-mode architecture allow excellent
transient response, easy noise filtering, and the use of small,
cost-saving external inductors and capacitors.
APPLICATIONS
TFT LC bias supplies
Portable applications
Industrial/instrumentation equipment
The ADP1611 is offered in the Pb-free 8-lead MSOP and
operates over the temperature range of −40°C to +85°C.
FUNCTIONAL BLOCK DIAGRAM
COMP
IN
1
6
ERROR
AMP
REF
ADP1611
gm
BIAS
FB 2
5
F/F
SW
R Q
S
RAMP
GEN
DRIVER
COMPARATOR
SS 8
OSC
SOFT START
CURRENTSENSE
AMPLIFIER
SD 3
4
GND
04906-001
RT 7
Figure 1.
Rev. 0
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infringements of patents or other rights of third parties that may result from its use.
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registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.326.8703
© 2005 Analog Devices, Inc. All rights reserved.
ADP1611
TABLE OF CONTENTS
Specifications..................................................................................... 3
Choosing the Input and Output Capacitors ........................... 11
Absolute Maximum Ratings............................................................ 4
Diode Selection........................................................................... 12
ESD Caution.................................................................................. 4
Loop Compensation .................................................................. 12
Pin Configuration and Function Descriptions............................. 5
Soft-Start Capacitor ................................................................... 13
Typical Performance Characteristics ............................................. 6
Application Circuits ................................................................... 14
Theory of Operation ...................................................................... 10
Step-Up DC-to-DC Converter with True Shutdown ............ 14
Current-Mode PWM Operation .............................................. 10
TFT LCD Bias Supply ................................................................ 14
Frequency Selection ................................................................... 10
SEPIC Power Supply .................................................................. 15
Soft Start ...................................................................................... 10
Layout Procedure ........................................................................... 16
On/Off Control........................................................................... 10
Outline Dimensions ....................................................................... 18
Setting the Output Voltage ........................................................ 10
Ordering Guide .......................................................................... 18
REVISION HISTORY
2/05—Revision 0: Initial Version
Rev. 0 | Page 2 of 20
ADP1611
SPECIFICATIONS
VIN = 3.3 V, TA = −40°C to +85°C, unless otherwise noted. All limits at temperature extremes are guaranteed by correlation and
characterization using standard statistical quality control (SQC), unless otherwise noted.
Table 1.
Parameter
SUPPLY
Input Voltage
Quiescent Current
Nonswitching State
Shutdown
Switching State1
OUTPUT
Output Voltage
Load Regulation
Overall Regulation
REFERENCE
Feedback Voltage
Line Regulation
ERROR AMPLIFIER
Transconductance
Voltage Gain
FB Input Bias Current
SWITCH
SW On Resistance
SW Leakage Current
Peak Current Limit2
OSCILLATOR
Oscillator Frequency
Maximum Duty Cycle
SHUTDOWN
Shutdown Input Voltage Low
Shutdown Input Voltage High
Shutdown Input Bias Current
SOFT START
SS Charging Current
UNDERVOLTAGE LOCKOUT3
UVLO Threshold
UVLO Hysteresis
Symbol
Conditions
VIN
Min
Typ
2.5
Max
Unit
5.5
V
IQ
IQSD
VFB = 1.3 V, RT = VIN
VSD = 0 V
390
0.01
600
10
µA
µA
IQSW
fSW = 1.23 MHz, no load
1
2
mA
20
V
mV/mA
%
1.248
+0.15
V
%/V
VOUT
VIN
ILOAD = 10 mA to 150 mA, VOUT = 10 V
Line, load, temperature
VFB
VIN = 2.5 V to 5.5 V
gm
AV
0.05
±3
1.212
−0.15
∆I = 1 µA
100
60
10
VFB = 1.23 V
RON
ISW = 1.0 A
VSW = 20 V
ICLSET
fOSC
DMAX
VIL
VIH
ISD
RT = GND
RT = IN
COMP = open, VFB = 1 V, RT = GND
1.230
0.49
0.89
78
µA/V
dB
nA
230
0.01
2.0
600
20
mΩ
µA
A
0.7
1.23
83
0.885
1.6
90
MHz
MHz
%
0.6
V
V
µA
2.2
VSD = 3.3 V
0.01
VSS = 0 V
3
VIN rising
1
This parameter specifies the average current while switching internally and with SW (Pin 5) floating.
Guaranteed by design and not fully production tested.
3
Guaranteed by characterization.
2
Rev. 0 | Page 3 of 20
2.2
2.4
220
1
µA
2.5
V
mV
ADP1611
ABSOLUTE MAXIMUM RATINGS
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability. Absolute maximum ratings apply individually
only, not in combination. Unless otherwise specified, all other
voltages are referenced to GND.
Table 2.
Parameter
Rating
IN, COMP, SD, SS, RT, FB to GND
SW to GND
RMS SW Pin Current
Operating Ambient Temperature Range
Operating Junction Temperature Range
Storage Temperature Range
θJA, Two Layers
θJA, Four Layers
Lead Temperature Range (Soldering, 60 sec)
−0.3 V to +6 V
22 V
1.2 A
−40°C to +85°C
−40°C to +125°C
−65°C to +150°C
206°C/W
142°C/W
300°C
IN
RC
CC
VOUT
COMP
1
6
ERROR
AMP
R1
REF
L1
ADP1611
BIAS
gm
FB
CIN
IN
2
R2
SW
COMPARATOR
RAMP
GEN
VOUT
COUT
R Q
S
DRIVER
VIN
1.2MHz
D1
5
F/F
RT
7
OSC
700kHz
SD 3
SS
8
SOFT START
CURRENTSENSE
AMPLIFIER
4
GND
Figure 2. Block Diagram and Typical Application Circuit
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 4 of 20
04906-002
CSS
ADP1611
COMP 1
FB 2
ADP1611
SD 3
TOP VIEW
(Not to Scale)
GND 4
8
SS
7
RT
6
IN
5
SW
04906-0-003
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
Figure 3. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1
Mnemonic
COMP
2
FB
3
4
5
SD
GND
SW
6
IN
7
RT
8
SS
Description
Compensation Input. Connect a series resistor-capacitor network from COMP to GND to compensate the
regulator.
Output Voltage Feedback Input. Connect a resistive voltage divider from the output voltage to FB to set the
regulator output voltage.
Shutdown Input. Drive SD low to shut down the regulator; drive SD high to turn it on.
Ground.
Switching Output. Connect the power inductor from the input voltage to SW and connect the external rectifier
from SW to the output voltage to complete the step-up converter.
Main Power Supply Input. IN powers the ADP1611 internal circuitry. Connect IN to the input source voltage.
Bypass IN to GND with a 10 µF or greater capacitor as close to the ADP1611 as possible.
Frequency Setting Input. RT controls the switching frequency. Connect RT to GND to program the oscillator to
700 kHz, or connect RT to IN to program it to 1.2 MHz.
Soft-Start Timing Capacitor Input. A capacitor from SS to GND brings up the output slowly at power-up.
Rev. 0 | Page 5 of 20
ADP1611
TYPICAL PERFORMANCE CHARACTERISTICS
100
100
VIN = 5V
FSW = 700kHz
L = 10µH
90
VIN = 3.3V
FSW = 1.2MHz
L = 4.7µH
VOUT = 10V
90
VOUT = 13V
80
VOUT = 15V
EFFICIENCY (%)
EFFICIENCY (%)
VOUT = 20V
80
VOUT = 5V
70
60
VOUT = 8.5V
70
60
50
50
40
1
10
100
LOAD CURRENT (mA)
04906-007
04906-004
40
30
1
1000
Figure 4. Output Efficiency vs. Load Current
2.8
VIN = 5V
FSW = 1.2MHz
L = 6.8µH
VOUT = 10V
VOUT = 10V
2.6
VIN = 5.5V
VOUT = 20V
2.4
VOUT = 15V
CURRENT LIMIT (A)
80
EFFICIENCY (%)
1000
Figure 7. Output Efficiency vs. Load Current
100
90
10
100
LOAD CURRENT (mA)
70
60
2.2
VIN = 3.3V
2.0
VIN = 2.5V
1.8
50
1.6
30
10
100
LOAD CURRENT (mA)
1.2
–40
1000
Figure 5. Output Efficiency vs. Load Current
80
VOUT = 8.5V
75
70
65
60
04906-006
EFFICIENCY (%)
85
RT = VIN
VOUT = 13V
85
55
50
1
60
1.4
VOUT = 5V
OSCILLATORY FREQUENCY (MHz)
90
10
35
AMBIENT TEMPERATURE (°C)
Figure 8. Current Limit vs. Ambient Temperature, VOUT = 10 V
95
VIN = 3.3V
FSW = 700kHz
L = 10µH
–15
10
100
LOAD CURRENT (mA)
1.2
1.0
0.8
RT = GND
0.6
0.4
0.2
0
–40
1000
Figure 6. Output Efficiency vs. Load Current
04906-009
1
04906-008
1.4
04906-005
40
VOUT = 10V
VIN = 3.3V
–15
10
35
AMBIENT TEMPERATURE (°C)
60
Figure 9. Oscillatory Frequency vs. Ambient Temperature
Rev. 0 | Page 6 of 20
85
ADP1611
1.4
0.50
FSW = 700kHz
VFB = 1.3V
1.2
QUIESCENT CURRENT (mA)
0.45
1.0
0.8
RT = GND
0.6
0.4
VIN = 5.5V
0.35
VIN = 3.3V
0.30
VIN = 2.5V
VOUT = 10V
3.0
3.5
4.0
4.5
SUPPLY VOLTAGE (V)
5.0
0.20
–40
5.5
Figure 10. Oscillatory Frequency vs. Supply Voltage
04906-013
0.2
0
2.5
–15
10
35
AMBIENT TEMPERATURE (°C)
60
85
Figure 13. Quiescent Current vs. Ambient Temperature
350
0.60
FSW = 1.23kHz
VFB = 1.3V
VIN = 5.5V
0.55
300
QUIESCENT CURRENT (mA)
SWITCH RESISTANCE (mΩ)
0.40
0.25
04906-010
OSCILLATORY FREQUENCY (MHz)
RT = VIN
VIN = 3.3V
250
VIN = 2.5V
200
150
0.50
VIN = 5.5V
0.45
VIN = 3.3V
0.40
VIN = 2.5V
–15
10
35
AMBIENT TEMPERATURE (°C)
60
0.30
–40
85
Figure 11. Switch Resistance vs. Ambient Temperature
–15
10
35
AMBIENT TEMPERATURE (°C)
60
85
Figure 14. Quiescent Current vs. Ambient Temperature
1.4
VIN = 3.3V
FSW = 700kHz
VFB = 1V
1.3
1.242
SUPPLY CURRENT (mA)
1.2
1.232
1.222
VIN = 5.5V
1.1
1.0
0.9
0.8
VIN = 3.3V
0.7
0.6
–15
10
35
AMBIENT TEMPERATURE (°C)
60
VIN = 2.5V
04906-015
1.212
–40
04906-012
REGULATION FB VOLTAGE (V)
04906-014
100
–40
04906-011
0.35
0.5
0.4
–40
85
Figure 12. Regulation FB Voltage vs. Ambient Temperature
–15
10
35
AMBIENT TEMPERATURE (°C)
60
Figure 15. Supply Current vs. Ambient Temperature
Rev. 0 | Page 7 of 20
85
ADP1611
2.0
300
FSW = 1.23kHz
VFB = 1V
250
1.6
200
UVLO HYS (mV)
VIN = 5.5V
1.4
1.2
VIN = 3.3V
150
100
1.0
0.6
–40
–15
10
35
AMBIENT TEMPERATURE (°C)
60
50
0
–40
85
Figure 16. Supply Current vs. Ambient Temperature
04906-019
VIN = 2.5V
0.8
04906-016
SUPPLY CURRENT (mA)
1.8
–15
10
35
AMBIENT TEMPERATURE (°C)
60
85
Figure 19. UVLO Hysteresis vs. Ambient Temperature
1.0
VIN = 3.3V
SD = 0.4V
VSW = 20V
0.8
3
0.7
0.6
0.5
0.4
1
0.3
CH1 = IL 500mA/DIV
CH2 = OUTPUT RIPPLE 100mV/DIV
CH3 = SW 10V/DIV
VIN = 5V, VOUT = 20V,
ILOAD = 200mA, FSW = 700kHz,
L = 10µH, COUT = 10µF
0.1
0
–40
15
70
AMBIENT TEMPERATURE (°C)
04906-020
2
0.2
04906-017
SWITCH LEAKAGE CURRENT (µA)
0.9
CH1 10.0mVΩ CH2 100mV
CH3 10.0V
125
M2.00µs
T
Figure 17. Switch Leakage Current vs. Ambient Temperature
A CH3
12.4V
0.00000s
Figure 20. Switching Waveform in Continuous Conduction
1.2
VIN = 3.5V
VIH
3
0.8
VIL
CH1 = IL 500mA/DIV
CH2 = OUTPUT RIPPLE 100mV/DIV
CH3 = SW 10V/DIV
0.6
VIN = 5V, VOUT = 20V,
ILOAD = 20mA, FSW = 700kHz,
L = 10µH, COUT = 10µF
0.4
1
0.2
15
70
AMBIENT TEMPERATURE (°C)
04906-021
0
–40
2
04906-018
SHUTDOWN THRESHOLD (V)
1.0
CH1 10.0mVΩ CH2 100mV
CH3 10.0V
125
Figure 18. Shutdown Threshold vs. Ambient Temperature
M2.00µs
A CH3
12.2V
Figure 21. Switching Waveform in Discontinuous Conduction
Rev. 0 | Page 8 of 20
ADP1611
4
2
VIN = 5V
VOUT = 20V
COUT = 10µF
L = 10µH
FSW = 700kHz
RC = 400kΩ
CC = 100pF
CH1 = ILOAD 200mA/DIV
CH2 = VOUT 200mV/DIV
CH1 = IL 2A/DIV
CH2 = VOUT 10V/DIV
CH3 = SD 1V/DIV
CH4 = COMP 2V/DIV
2
VIN = 5V
VOUT = 20V
IOUT = 200mA
CSS = 0F
1
CH1 10.0mVΩ CH2 200mV
M2.00µs
T
A CH1
04906-024
04906-022
1
3
4.8mV
CH1 10.0mVΩ CH2 10.0V
CH4 2.00V
CH3 1.00V
571.200µs
M200µs
A CH3
680mV
Figure 24. Start-Up Response from Shutdown, CSS = 0 F
Figure 22. Load Transient Response, 700 kHz, VOUT = 20 V
4
2
VIN = 5V
VOUT = 20V
COUT = 10µF
L = 10µH
FSW = 1.2MHz
RC = 400kΩ
CC = 100pF
CH1 = ILOAD 200mA/DIV
CH2 = VOUT 200mV/DIV
2
CH1 = IL 2A/DIV
CH2 = VOUT 10V/DIV
CH3 = SD 1V/DIV
CH4 = COMP 2V/DIV
VIN = 5V
VOUT = 20V
IOUT = 200mA
CSS = 100nF
1
CH1 10.0mVΩ CH2 200mV
M200µs
T
A CH1
7.20mV
04906-025
04906-023
1
3
CH1 10.0mVΩ CH2 10.0V
CH4 2.00V
CH3 1.00V
488.000µs
M400µs
A CH3
680mV
Figure 25. Start-Up Response from Shutdown, CSS = 100 nF
Figure 23. Load Transient Response, 1.2 MHz, VOUT = 20 V
Rev. 0 | Page 9 of 20
ADP1611
THEORY OF OPERATION
The ADP1611 current-mode step-up switching converter
converts a 2.5 V to 5.5 V input voltage up to an output voltage
as high as 20 V. The 1.2 A internal switch allows a high output
current, and the high 1.2 MHz switching frequency allows tiny
external components. The switch current is monitored on a
pulse-by-pulse basis to limit it to 2 A.
CURRENT-MODE PWM OPERATION
The ADP1611 uses current-mode architecture to regulate the
output voltage. The output voltage is monitored at FB through a
resistive voltage divider. The voltage at FB is compared to the
internal 1.23 V reference by the internal transconductance error
amplifier to create an error current at COMP. A series resistorcapacitor at COMP converts the error current to a voltage.
The switch current is internally measured and added to the
stabilizing ramp, and the resulting sum is compared to the error
voltage at COMP to control the PWM modulator. This currentmode regulation system allows fast transient response, while
maintaining a stable output voltage. By selecting the proper
resistor-capacitor network from COMP to GND, the regulator
response is optimized for a wide range of input voltages, output
voltages, and load conditions.
FREQUENCY SELECTION
The ADP1611 frequency is user-selectable and operates at
either 700 kHz to optimize the regulator for high efficiency
or at 1.2 MHz for small external components. Connect RT to
IN for 1.2 MHz operation, or connect RT to GND for 700 kHz
operation. To achieve the maximum duty cycle, which might
be required for converting a low input voltage to a high output
voltage, use the lower 700 kHz switching frequency.
SOFT START
To prevent input inrush current at startup, connect a capacitor
from SS to GND to set the soft-start period. When the device is
in shutdown (SD is at GND) or the input voltage is below the
2.4 V undervoltage lockout voltage, SS is internally shorted to
GND to discharge the soft start capacitor. Once the ADP1611 is
turned on, SS sources 3 µA to the soft-start capacitor at startup.
As the soft-start capacitor charges, it limits the voltage at
COMP. Because of the current-mode regulator, the voltage at
COMP is proportional to the switch peak current, and,
therefore, the input current. By slowly charging the soft-start
capacitor, the input current ramps slowly to prevent it from
overshooting excessively at startup.
ON/OFF CONTROL
The SD input turns the ADP1611 regulator on or off. Drive SD
low to turn off the regulator and reduce the input current to
10 nA. Drive SD high to turn on the regulator.
When the step-up dc-to-dc switching converter is turned off,
there is a dc path from the input to the output through the
inductor and output rectifier. This causes the output voltage to
remain slightly below the input voltage by the forward voltage
of the rectifier, preventing the output voltage from dropping to
0 when the regulator is shut down. Figure 28 shows the application circuit to disconnect the output voltage from the input
voltage at shutdown.
SETTING THE OUTPUT VOLTAGE
The ADP1611 features an adjustable output voltage range of VIN
to 20 V. The output voltage is set by the resistive voltage divider
(R1 and R2 in Figure 2) from the output voltage (VOUT) to the
1.230 V feedback input at FB. Use the following formula to
determine the output voltage:
VOUT = 1.23 × (1 + R1/R2)
(1)
Use an R2 resistance of 10 kΩ or less to prevent output voltage
errors due to the 10 nA FB input bias current. Choose R1 based
on the following formula:
− 1.23 ⎞
⎛V
R1 = R2 × ⎜ OUT
⎟
1.23
⎠
⎝
(2)
INDUCTOR SELECTION
The inductor is an essential part of the step-up switching
converter. It stores energy during the on time, and transfers that
energy to the output through the output rectifier during the off
time. Use inductance in the range of 1 µH to 22 µH. In general,
lower inductance values have higher saturation current and
lower series resistance for a given physical size. However, lower
inductance results in higher peak current that can lead to
reduced efficiency and greater input and/or output ripple and
noise. Peak-to-peak inductor ripple current at close to 30% of
the maximum dc input current typically yields an optimal
compromise.
For determining the inductor ripple current, the input (VIN) and
output (VOUT) voltages determine the switch duty cycle (D) by
the following equation:
D=
Rev. 0 | Page 10 of 20
VOUT − VIN
VOUT
(3)
ADP1611
Table 4. Inductor Manufacturers
Vendor
Sumida
847-956-0666
www.sumida.com
Coilcraft 847-639-6400
www.coilcraft.com
Toko 847-297-0070
www.tokoam.com
Part
CMD4D11-2R2MC
CMD4D11-4R7MC
CDRH4D28-100
CDRH5D18-220
CR43-4R7
CR43-100
DS1608-472
DS1608-103
D52LC-4R7M
D52LC-100M
L (µH)
2.2
4.7
10
22
4.7
10
4.7
10
4.7
10
Using the duty cycle and switching frequency, fSW, determine
the on time by the following equation:
tON =
D
f SW
(4)
Max DC Current
0.95
0.75
1.00
0.80
1.15
1.04
1.40
1.00
1.14
0.76
Max DCR (mΩ)
116
216
128
290
109
182
60
75
87
150
Height (mm)
1.2
1.2
3.0
2.0
3.5
3.5
2.9
2.9
2.0
2.0
The output capacitor maintains the output voltage and supplies
current to the load while the ADP1611 switch is on. The value
and characteristics of the output capacitor greatly affect the
output voltage ripple and stability of the regulator. Use a low
ESR output capacitor; ceramic dielectric capacitors are
preferred.
The inductor ripple current (∆IL) in steady state is
V ×t
∆ IL = IN ON
L
(5)
Solving for the inductance value, L,
L=
VIN × tON
∆ IL
(6)
∆VOUT =
Make sure that the peak inductor current (the maximum input
current plus half the inductor ripple current) is below the rated
saturation current of the inductor. Likewise, make sure that the
maximum rated rms current of the inductor is greater than the
maximum dc input current to the regulator.
For duty cycles greater than 50%, which occur with input
voltages greater than one-half the output voltage, slope
compensation is required to maintain stability of the currentmode regulator. For stable current-mode operation, ensure that
the selected inductance is equal to or greater than LMIN
L > L MIN =
For very low ESR capacitors, such as ceramic capacitors, the
ripple current due to the capacitance is calculated as follows.
Because the capacitor discharges during the on time, tON, the
charge removed from the capacitor, QC, is the load current
multiplied by the on time. Therefore, the output voltage ripple
(∆VOUT) is
VOUT − V IN
1.8 A × f SW
QC
I ×t
= L ON
COUT
COUT
(8)
where:
COUT is the output capacitance.
IL is the average inductor current.
D
V
− VIN
tON =
and D = OUT
f SW
VOUT
Choose the output capacitor based on the following equation:
C OUT ≥
I L × (VOUT − V IN )
(9)
f SW × VOUT × ∆VOUT
(7)
Table 5. Capacitor Manufacturers
CHOOSING THE INPUT AND OUTPUT CAPACITORS
The ADP1611 requires input and output bypass capacitors to
supply transient currents while maintaining constant input and
output voltage. Use a low equivalent series resistance (ESR)
input capacitor, 10 µF or greater, to prevent noise at the
ADP1611 input. Place the capacitor between IN and GND as
close to the ADP1611 as possible. Ceramic capacitors are
preferred because of their low ESR characteristics. Alternatively,
use a high value, medium ESR capacitor in parallel with a 0.1 µF
low ESR capacitor as close to the ADP1611 as possible.
Vendor
AVX
Murata
Sanyo
Taiyo–Yuden
Rev. 0 | Page 11 of 20
Phone No.
408-573-4150
714-852-2001
408-749-9714
408-573-4150
Web Address
www.avxcorp.com
www.murata.com
www.sanyovideo.com
www.t-yuden.com
ADP1611
The regulator loop gain is
DIODE SELECTION
The output rectifier conducts the inductor current to the output
capacitor and load while the switch is off. For high efficiency,
minimize the forward voltage drop of the diode. For this reason,
Schottky rectifiers are recommended. However, for high
voltage, high temperature applications where the Schottky
rectifier reverse leakage current becomes significant and can
degrade efficiency, use an ultrafast junction diode.
Make sure that the diode is rated to handle the average output
load current. Many diode manufacturers derate the current
capability of the diode as a function of the duty cycle. Verify
that the output diode is rated to handle the average output load
current with the minimum duty cycle. The minimum duty cycle
of the ADP1611 is
DMIN =
VOUT − VIN − MAX
VOUT
Table 6. Schottky Diode Manufacturers
Phone No.
602-244-6600
805-446-4800
631-435-1110
310-322-3331
Web Address
www.onsemi.com
www.diodes.com
www.centralsemi.com
www.sanyo.com
(12)
where:
AVL is the loop gain.
VFB is the feedback regulation voltage, 1.230 V.
VOUT is the regulated output voltage.
VIN is the input voltage.
GMEA is the error amplifier transconductance gain.
ZCOMP is the impedance of the series RC network from COMP to
GND.
GCS is the current-sense transconductance gain (the inductor
current divided by the voltage at COMP), which is internally set
by the ADP1611.
ZOUT is the impedance of the load and output capacitor.
The ADP1611 uses external components to compensate the
regulator loop, allowing optimization of the loop dynamics for a
given application.
The step-up converter produces an undesirable right-half plane
zero in the regulation feedback loop. This requires compensating the regulator such that the crossover frequency occurs
well below the frequency of the right-half plane zero. The righthalf plane zero is determined by the following equation:
2
⎞
R
⎟ × LOAD
⎟ 2π × L
⎠
To determine the crossover frequency, it is important to note
that, at that frequency, the compensation impedance (ZCOMP) is
dominated by the resistor, and the output impedance (ZOUT) is
dominated by the impedance of the output capacitor. So, when
solving for the crossover frequency, the equation (by definition
of the crossover frequency) is simplified to
V
V
1
| A | = FB × IN × G
× R
× G ×
=1
VL V
MEA
COMP
CS 2π × f × C
V
OUT OUT
C OUT
(13)
where fC is the crossover frequency and RCOMP is the
compensation resistor.
LOOP COMPENSATION
⎛ V
FZ ( RHP ) = ⎜⎜ IN
⎝ VOUT
VFB
V
× IN × GMEA × Z COMP × GCS × Z OUT
VOUT VOUT
(10)
where VIN-MAX is the maximum input voltage.
Vendor
On Semiconductor
Diodes, Inc.
Central Semiconductor
Sanyo
AVL =
(11)
where:
FZ(RHP) is the right-half plane zero.
RLOAD is the equivalent load resistance or the output voltage
divided by the load current.
To stabilize the regulator, ensure that the regulator crossover
frequency is less than or equal to one-fifth of the right-half
plane zero and less than or equal to one-fifteenth of the
switching frequency.
Solving for RCOMP
R COMP =
2π × f C × C OUT × VOUT × VOUT
V FB × V IN × G MEA × GCS
(14)
For VFB = 1.23, GMEA = 100 µS, and GCS = 2 S
RCOMP =
2.55 × 10 4 × f C × COUT × VOUT × VOUT
VIN
(15)
Once the compensation resistor is known, set the zero formed
by the compensation capacitor and resistor to one-fourth of the
crossover frequency, or
CCOMP =
2
π × f C × RCOMP
(16)
where CCOMP is the compensation capacitor.
The capacitor, C2, is chosen to cancel the zero introduced by
output capacitance ESR.
Solving for C2,
C2 =
Rev. 0 | Page 12 of 20
ESR × COUT
RCOMP
(17)
ADP1611
For low ESR output capacitance, such as with a ceramic capacitor, C2 is optional. For optimal transient performance, the
RCOMP and CCOMP might need to be adjusted by observing the
load transient response of the ADP1611. For most applications,
the compensation resistor should be in the range of 30 kΩ to
400 kΩ, and the compensation capacitor should be in the range
of 100 pF to 1.2 nF. Table 7 shows external component values
for several applications.
ERROR AMP
REF
gm
COMP
1
The voltage at SS ramps up slowly by charging the soft-start
capacitor (CSS) with an internal 3 µA current source. Table 8
lists the values for the soft-start period, based on maximum
output current and maximum switching frequency.
The soft-start capacitor limits the rate of voltage rise on the
COMP pin, which in turn limits the peak switch current at
startup. Table 8 shows a typical soft-start period, tSS, at
maximum output current, IOUT_MAX, for several conditions.
A 20 nF soft-start capacitor results in negligible input current
overshoot at startup, and so is suitable for most applications.
However, if an unusually large output capacitor is used, a longer
soft-start period is required to prevent input inrush current.
FB 2
RC
SOFT-START CAPACITOR
C2
04906-026
CC
Conversely, if fast startup is a requirement, the soft-start
capacitor can be reduced or even removed, allowing the
ADP1611 to start quickly, but allowing greater peak switch
current (see Figure 24 and Figure 25).
Figure 26. Compensation Components
Table 7. Recommended External Components for Popular Input/Output Voltage Conditions
VIN (V)
3.3
5
VOUT (V)
5
5
9
9
12
12
9
9
12
12
20
20
fSW (MHz)
0.70
1.23
0.70
1.23
0.70
1.23
0.70
1.23
0.70
1.23
0.70
1.23
L (µH)
4.7
2.7
10
4.7
10
4.7
10
4.7
10
4.7
10
6.8
COUT (µF)
10
10
10
10
10
10
10
10
10
10
10
10
CIN (µF)
10
10
10
10
10
10
10
10
10
10
10
10
R1 (kΩ)
30.9
30.9
63.4
63.4
88.7
88.7
63.4
63.4
88.7
88.7
154
154
R2 (kΩ)
10
10
10
10
10
10
10
10
10
10
10
10
RCOMP (kΩ)
50
90.9
71.5
150
130
280
84.5
178
140
300
400
400
CCOMP (pF)
520
150
820
180
420
100
390
100
220
100
100
100
IOUT_MAX (mA)
600
600
350
350
250
250
450
450
350
350
250
250
Table 8. Typical Soft Start Period
VIN (V)
3.3
VOUT (V)
5
5
9
9
12
12
COUT (µF)
10
10
10
10
10
10
CSS (nF)
20
100
20
100
20
100
tSS (ms)
0.3
2
2.5
8.2
3.5
15
VIN (V)
5
Rev. 0 | Page 13 of 20
VOUT (V)
9
9
12
12
20
20
COUT (µF)
10
10
10
10
10
10
CSS (nF)
20
100
20
100
20
100
tSS (ms)
0.4
1.5
0.62
2
1.1
4.1
ADP1611
APPLICATION CIRCUITS
TFT LCD BIAS SUPPLY
The circuit in Figure 27 shows the ADP1611 in a step-up
configuration. The ADP1611 is used here to generate a 15 V
regulator with the following specifications:
Figure 29 shows a power supply circuit for TFT LCD module
applications. This circuit has +10 V, −5 V, and +22 V outputs.
The +10 V is generated in the step-up configuration. The −5 V
and +22 V are generated by the charge-pump circuit. During
step-up , the SW node switches between 10 V and ground
(neglecting forward drop of the diode and on resistance of the
switch). When the SW node is high, C5 charges up to 10 V. C5
holds its charge and forward-biases D8 to charge C6 to −10 V.
The Zener diode, D9, clamps and regulates the output to −5 V.
VIN = 3.5 V to 5.5 V
VOUT = 15 V
IOUT ≤ 400 mA
The output can be set to the desired voltage using Equation 2.
Use Equations 16 and 17 to change the compensation network.
L1
4.7µH
6
IN
3
SD
7
RT
BAV99 C5
10nF
C6
D8
10µF
VGL
–5V
15V
D9
BZT52C5VIS
SW 5
D7
R1
112kΩ
ON
C4
10nF
R3
200Ω
D5
D4
C3
10µF
COMP 1
SS
CSS
22nF
RCOMP
220kΩ
CCOMP
150pF
GND
4
CSS
22nF
ADP1611
IN
3
SHDN
15V
SW 5
10kΩ
112kΩ
FB
Q1 B
7
RT
8
SS
10µF
ON
22nF
2
10kΩ
10µF
COMP 1
GND
4
220kΩ
150pF
SD
7
RT
8
SS
R1
71.3kΩ
R2
10kΩ
COMP 1
GND
4
RCOMP
220kΩ
CCOMP
150pF
COUT
10µF
The VGH output is generated in a similar manner by the
charge-pump capacitors, C1, C2, and C4. The output voltage is
tripled and regulated down to 22 V by the Zener diode, D5.
04906-028
6
3
10V
SW 5
Figure 29. TFT LCD Bias Supply
4.7µH
FDC6331
IN
C2
1µF
FB 2
CIN
10µF
Some battery-powered applications require very low standby
current. The ADP1611 typically consumes 10 nA from the
input, which makes it suitable for these applications. However,
the output is connected to the input through the inductor and
the rectifying diode, allowing load current draw from the input
while shut down. The circuit in Figure 28 enables the ADP1611
to achieve output load disconnect at shutdown. To shut down
the ADP1611 and disconnect the output from the input, drive
the SD pin below 0.4 V.
A
6
ON
STEP-UP DC-TO-DC CONVERTER WITH TRUE
SHUTDOWN
Q1
D1
ADP1611
3.3V
Figure 27. 5 V to 15 V Step-Up Regulator
5V
C1
10nF D2
BAV99
L1
4.7µH
COUT
10µF
04906-027
8
R2
10kΩ
D5
BZT52C22
BAV99
D3
FB 2
CIN
10µF
VGH
22V
04906-029
D1
ADP1611
5V
R4
200Ω
Figure 28. Step-Up Regulator with True Shutdown
Rev. 0 | Page 14 of 20
ADP1611
SEPIC POWER SUPPLY
The input and the output are dc-isolated by a coupling capacitor, C1. In steady state, the average voltage of C1 is the input
voltage. When the ADP1611 switch turns on and the diode
turns off, the input voltage provides energy to L1, and C1
provides energy to L2. When the ADP1611 switch turns off and
the diode turns on, the energy in L1 and L2 is released to charge
the output capacitor, COUT, and the coupling capacitor, C1, and
to supply current to the load.
L1
4.7µH
C1
10µF
ADP1611
2.5V–5.5V
6
IN
3
SD
ON
CIN
10µF
Rev. 0 | Page 15 of 20
CSS
22nF
7
RT
8
SS
3.3V
SW 5
L2
4.7µH
R1
16.8kΩ
FB 2
COMP 1
GND
4
RCOMP
60kΩ
CCOMP
1nF
Figure 30. 3.3 V DC-to-DC Converter
COUT
10µF
R2
10kΩ
04906-030
The circuit in Figure 30 shows the ADP1611 in a single-ended
primary inductance converter (SEPIC) topology. This topology
is useful for an unregulated input voltage, such as a batterypowered application in which the input voltage can vary
between 2.7 V to 5 V, and the regulated output voltage falls
within the input voltage range.
ADP1611
LAYOUT PROCEDURE
To achieve high efficiency, good regulation, and stability, a welldesigned printed circuit board layout is required. Where
possible, use the sample application board layout as a model.
Follow these guidelines when designing printed circuit boards
(see Figure 1):
•
Keep the low ESR input capacitor, CIN, close to IN and
GND.
Keep the high current path from CIN through the inductor,
L1, to SW and PGND as short as possible.
•
Keep the high current path from CIN through L1, the
rectifier, D1, and the output capacitor, COUT, as short as
possible.
Keep high current traces as short and as wide as possible.
•
Place the feedback resistors as close to FB as possible to
prevent noise pickup.
•
Place the compensation components as close as possible to
COMP.
•
Avoid routing high impedance traces near any node
connected to SW or near the inductor to prevent radiated
noise injection.
04472-027
•
•
Figure 31. Sample Application Board (Bottom Layer)
Rev. 0 | Page 16 of 20
04472-028
ADP1611
04906-033
Figure 32. Sample Application Board (Top Layer)
Figure 33. Sample Application Board (Silkscreen Layer)
Rev. 0 | Page 17 of 20
ADP1611
OUTLINE DIMENSIONS
3.00
BSC
8
5
4.90
BSC
3.00
BSC
4
PIN 1
0.65 BSC
1.10 MAX
0.15
0.00
0.38
0.22
COPLANARITY
0.10
0.23
0.08
8°
0°
0.80
0.60
0.40
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-187AA
Figure 34. 8-Lead Mini Small Outline Package [MSOP]
(RM-8)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADP1611ARMZ-R71
ADP1611-EVAL
1
Temperature Range
−40°C to +85°C
Package Description
8-Lead Mini Small Outline Package [MSOP]
Evaluation Board
Z = Pb-free part.
Rev. 0 | Page 18 of 20
Package Option
RM-8
Branding
P11
ADP1611
NOTES
Rev. 0 | Page 19 of 20
ADP1611
NOTES
© 2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D04906–0–2/05(0)
Rev. 0 | Page 20 of 20