LINER LT1934ES6-1 Micropower step-down switching regulators in thinsot Datasheet

LT1934/LT1934-1
Micropower Step-Down
Switching Regulators
in ThinSOT
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FEATURES
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DESCRIPTIO
The LT ®1934 is a micropower step-down DC/DC converter with internal 400mA power switch, packaged in a
low profile (1mm) ThinSOT. With its wide input range of
3.2V to 34V, the LT1934 can regulate a wide variety of
power sources, from 4-cell alkaline batteries and 5V logic
rails to unregulated wall transformers and lead-acid batteries. Quiescent current is just 12µA and a zero current
shutdown mode disconnects the load from the input
source, simplifying power management in battery-powered systems. Burst Mode® operation and the low drop
internal power switch result in high efficiency over a broad
range of load current.
Wide Input Voltage Range: 3.2V to 34V
Micropower Operation: IQ = 12µA
5V at 250mA from 6.5V to 34V Input (LT1934)
5V at 60mA from 6.5V to 34V Input (LT1934-1)
3.3V at 250mA from 4.5V to 34V Input (LT1934)
3.3V at 60mA from 4.5V to 34V Input (LT1934-1)
Low Shutdown Current: <1µA
Low VCESAT Switch: 200mV at 300mA
Low Profile (1mm) SOT-23 (ThinSOTTM) Package
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APPLICATIO S
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Wall Transformer Regulation
Automotive Battery Regulation
Standby Power for Portable Products
Distributed Supply Regulation
Industrial Control Supplies
The LT1934 provides up to 300mA of output current. The
LT1934-1 has a lower current limit, allowing optimum
choice of external components when the required output
current is less than 60mA. Fast current limiting protects
the LT1934 and external components against shorted
outputs, even at 34V input.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
ThinSOT is a trademark of Linear Technology Corporation.
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TYPICAL APPLICATIO
3.3V Step-Down Converter
Efficiency
100
D2
VIN
4.5V TO 34V
C2
2.2µF
ON OFF
90
L1
47µH
VOUT
3.3V
250mA
SW
VIN
D1
LT1934
SHDN
10pF
1M
+
FB
GND
C1: SANYO 4TPB100M
C2: TAIYO YUDEN GMK325BJ225MN
D1: ON SEMICONDUCTOR MBR0540
D2: CENTRAL CMDSH-3
L1: SUMIDA CDRH4D28-470
C1
100µF
604k
EFFICIENCY (%)
0.22µF
BOOST
LT1934
VIN = 12V
VOUT = 5V
80
VOUT = 3.3V
70
60
1934 TA01
50
0.1
1
10
100
LOAD CURRENT (mA)
1934 TA02
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LT1934/LT1934-1
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PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Voltage (VIN) ................................................. 34V
BOOST Pin Voltage ................................................. 40V
BOOST Pin Above SW Pin ...................................... 20V
SHDN Pin ............................................................... 34V
FB Voltage ................................................................ 6V
SW Voltage ............................................................... VIN
Operating Temperature Range (Note 2) ..........................
LT1934E/LT1934E-1 ......................... – 40°C to 85°C
LT1934I/LT1934I-1 ......................... – 40°C to 125°C
Maximum Junction Temperature .......................... 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
LT1934ES6
LT1934ES6-1
LT1934IS6
LT1934IS6-1
TOP VIEW
BOOST 1
6 SW
GND 2
5 VIN
4 SHDN
FB 3
S6 PACKAGE
6-LEAD PLASTIC SOT-23
S6 PART MARKING
TJMAX = 125°C, θJA = 250°C/ W, θJC = 102°C/ W
LTXP
LTF8
LTAJB
LTAJC
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VBOOST = 15V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
Undervoltage Lockout
Quiescent Current
V
V
V
22
26
26
µA
µA
µA
0.01
2
µA
1.25
1.25
1.27
1.27
V
V
●
●
–40°C ≤ TA ≤ 85°C
–40°C ≤ TA ≤ 125°C
●
●
12
12
12
VFB Falling
–40°C ≤ TA ≤ 85°C
–40°C ≤ TA ≤ 125°C
●
●
VFB = 1.25V
–40°C ≤ TA ≤ 85°C
–40°C ≤ TA ≤ 125°C
●
●
VFB = 1.3V
1.22
1.21
10
FB Voltage Line Regulation
4V < VIN < 34V
Switch Off Time
VFB > 1V
VFB = 0V
Maximum Duty Cycle
VFB = 1V
Switch VCESAT
ISW = 300mA (LT1934)
ISW = 75mA (LT1934-1)
Switch Current Limit
LT1934
LT1934-1
BOOST Pin Current
Minimum Boost Voltage (Note 3)
Switch Leakage Current
UNITS
3.2
3.6
3.6
–40°C ≤ TA ≤ 85°C
–40°C ≤ TA ≤ 125°C
FB Comparator Hysteresis
FB Pin Bias Current
MAX
3
3
3
VSHDN = 0V
FB Comparator Trip Voltage
TYP
2
2
mV
±15
±60
0.007
–40°C ≤ TA ≤ 85°C
–40°C ≤ TA ≤ 125°C
●
●
1.4
1.8
12
85
83
88
88
nA
nA
%/V
2.3
µs
µs
%
%
200
65
300
120
mV
mV
400
120
490
160
mA
mA
ISW = 300mA (LT1934)
ISW = 75mA (LT1934-1)
8.5
6.0
12
10
mA
mA
ISW = 300mA (LT1934)
ISW = 75mA (LT1934-1)
1.8
1.7
2.5
2.5
V
V
2
µA
350
90
1934f
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LT1934/LT1934-1
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VBOOST = 15V, unless otherwise noted.
PARAMETER
CONDITIONS
SHDN Pin Current
VSHDN = 2.3V
VSHDN = 34V
MIN
SHDN Input Voltage High
TYP
MAX
0.5
1.5
5
UNITS
µA
µA
2.3
V
SHDN Input Voltage Low
0.25
Note 1: Absolute Maximum Ratings are those values beyond which the life
of the device may be impaired.
Note 2: The LT1934E and LT1934E-1 are guaranteed to meet performance
specifications from 0°C to 70°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization and
V
correlation with statistical process controls. The LT1934I and LT1934I-1
specifications are guaranteed over the –40°C to 125°C temperature range.
Note 3: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the internal power switch.
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TYPICAL PERFOR A CE CHARACTERISTICS
LT1934 Efficiency, VOUT = 5V
LT1934 Efficiency, VOUT = 3.3V
LT1934
VOUT = 3.3V
L = 47µH
90
TA = 25°C
VIN = 12V
80
EFFICIENCY (%)
EFFICIENCY (%)
LT1934
VOUT = 5V
L = 47µH
90
TA = 25°C
VIN = 24V
70
60
90
80
VIN = 24V
VIN = 12V
70
50
1
10
100
LOAD CURRENT (mA)
1934 G03
Current Limit vs Temperature
Off Time vs Temperature
3.0
LT1934
VIN = 12V
80
VIN = 24V
70
60
50
1
10
100
LOAD CURRENT (mA)
1934 G04
2.5
400
OFF TIME (µs)
SWITCH CURRENT LIMIT (mA)
EFFICIENCY (%)
0.1
500
0.1
70
1934 G02
LT1934-1 Efficiency, VOUT = 3.3V
90
VIN = 24V
1
10
100
LOAD CURRENT (mA)
1934 G01
LT1934-1
VOUT = 3.3V
L = 100µH
TA = 25°C
VIN = 12V
80
60
50
0.1
1
10
100
LOAD CURRENT (mA)
LT1934-1
VOUT = 5V
L = 150µH
TA = 25°C
VIN = 5V
60
50
0.1
100
LT1934-1 Efficiency, VOUT = 5V
100
100
EFFICIENCY (%)
100
300
200
1.5
1.0
LT1934-1
100
0
–50
2.0
0.5
–25
50
25
0
75
TEMPERATURE (°C)
100
125
1934 G05
0
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1934 G06
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LT1934/LT1934-1
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TYPICAL PERFOR A CE CHARACTERISTICS
Frequency Foldback
16
SHDN Bias Current
vs SHDN Voltage
VFB vs Temperature
1.27
TA = 25°C
2.0
TA = 25°C
14
10
8
6
4
SHDN PIN CURRENT (µA)
FEEDBACK VOLTAGE (V)
SWITCH OFF TIME (µs)
1.26
12
1.25
1.24
1.5
1.0
0.5
1.23
2
0
0
0.2
0.4
0.8
1.0
0.6
FEEDBACK PIN VOLTAGE (V)
1.2
1.22
–50 –25
0
50
25
0
75
TEMPERATURE (°C)
100
2
4
6
8
SHDN PIN VOLTAGE (V)
Quiescent Current
vs Temperature
12
Undervoltage Lockout
vs Temperature
4.0
15
3.5
UVLO (V)
20
10
5
3.0
2.5
0
–50 –25
75
0
25
50
TEMPERATURE (°C)
100
2.0
–50 –25
125
75
0
25
50
TEMPERATURE (°C)
100
1934 G10
Minimum Input Voltage
VOUT = 5V
6.0
8
LT1934
VOUT = 5V
TA = 25°C
7 BOOST DIODE TIED TO OUTPUT
INPUT VOLTAGE (V)
LT1934
VOUT = 3.3V
5.5 TA = 25°C
BOOST DIODE TIED TO OUTPUT
5.0
125
1934 G11
Minimum Input Voltage
VOUT = 3.3V
INPUT VOLTAGE (V)
10
1934 G09
1934 G08
1934 G07
QUIESCENT CURRENT (µA)
0
125
VIN TO START
4.5
4.0
VIN TO RUN
VIN TO START
6
VIN TO RUN
5
3.5
3.0
0.1
4
0.1
1
10
100
LOAD CURRENT (mA)
1934 G12
1
10
100
LOAD CURRENT (mA)
1934 G13
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LT1934/LT1934-1
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BOOST (Pin 1): The BOOST pin is used to provide a drive
voltage, higher than the input voltage, to the internal
bipolar NPN power switch.
SHDN (Pin 4): The SHDN pin is used to put the LT1934 in
shutdown mode. Tie to ground to shut down the LT1934.
Apply 2.3V or more for normal operation. If the shutdown
feature is not used, tie this pin to the VIN pin.
GND (Pin 2): Tie the GND pin to a local ground plane below
the LT1934 and the circuit components. Return the feedback divider to this pin.
VIN (Pin 5): The VIN pin supplies current to the LT1934’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
FB (Pin 3): The LT1934 regulates its feedback pin to 1.25V.
Connect the feedback resistor divider tap to this pin. Set
the output voltage according to VOUT = 1.25V (1 + R1/R2)
or R1 = R2 (VOUT/1.25 – 1).
SW (Pin 6): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
W
BLOCK DIAGRA
5
VIN
VIN
+
C2
+
–
BOOST
D2
1
ON TIME
R Q′
12µs DELAY
S
OFF TIME
SW
1.8µs DELAY
ON OFF
4
SHDN
C3
Q
L1
VOUT
6
D1
C1
+
VREF 1.25V
ENABLE
–
2
GND
3
R2
FEEDBACK
COMPARATOR
FB
R1
1934 BD
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LT1934/LT1934-1
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OPERATIO (Refer to Block Diagram)
The LT1934 uses Burst Mode control, combining both low
quiescent current operation and high switching frequency,
which result in high efficiency across a wide range of load
currents and a small total circuit size.
A comparator monitors the voltage at the FB pin of the
LT1934. If this voltage is higher than the internal 1.25V
reference, the comparator disables the oscillator and power
switch. In this state, only the comparator, reference and
undervoltage lockout circuits are active, and the current
into the VIN pin is just 12µA. As the load current discharges
the output capacitor, the voltage at the FB pin falls below
1.25V and the comparator enables the oscillator. The
LT1934 begins to switch, delivering current to the output
capacitor. The output voltage rises, and when it overcomes
the feedback comparator’s hysteresis, the oscillator is
disabled and the LT1934 returns to its micropower state.
The oscillator consists of two one-shots and a flip-flop.
A rising edge from the off-time one-shot sets the flipflop, which turns on the internal NPN power switch. The
switch remains on until either the on-time one-shot trips
or the current limit is reached. A sense resistor and
amplifier monitor the current through the switch and resets
the flip-flop when this current reaches 400mA (120mA
for the LT1934-1). After the 1.8µs delay of the off-time
one-shot, the cycle repeats. Generally, the LT1934 will
reach current limit on every cycle—the off time is fixed
and the on time is regulated so that the LT1934 operates
at the correct duty cycle. The 1.8µs off time is lengthened
when the FB pin voltage falls below 0.8V; this foldback
behavior helps control the output current during start-up
and overload. Figure 1 shows several waveforms of an
LT1934 producing 3.3V from a 10V input. When the switch
is on, the SW pin voltage is at 10V. When the switch is off,
the inductor current pulls the SW pin down until it is
clamped near ground by the external catch diode.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate the
bipolar switch for efficient operation.
If the SHDN pin is grounded, all internal circuits are turned
off and VIN current reduces to the device leakage current,
typically a few nA.
VOUT
50mV/DIV
VSW
10V/DIV
ISW
0.5A/DIV
ILI
0.5A/DIV
5µs/DIV
1934 F01a
Figure 1. Operating Waveforms of the LT1934 Converting
10V to 3.3V at 180mA (Front Page Schematic)
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LT1934/LT1934-1
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Which One to Use: LT1934 or LT1934-1?
The only difference between the LT1934 and LT1934-1 is
the peak current through the internal switch and the
inductor. If your maximum load current is less than 60mA,
use the LT1934-1. If your maximum load is higher, use the
LT1934; it can supply up to ~300mA.
While the LT1934-1 can’t deliver as much output current,
it has other advantages. The lower peak switch current
allows the use of smaller components (input capacitor,
inductor and output capacitor). The ripple current at the
input of the LT1934-1 circuit will be smaller and may be an
important consideration if the input supply is current
limited or has high impedance. The LT1934-1’s current
draw during faults (output overload or short) and start-up
is lower.
The maximum load current that the LT1934 or LT1934-1
can deliver depends on the value of the inductor used.
Table 1 lists inductor value, minimum output capacitor
and maximum load for 3.3V and 5V circuits. Increasing the
value of the capacitor will lower the output voltage ripple.
Component selection is covered in more detail in the
following sections.
Minimum Input Voltage
The minimum input voltage required to generate a particular output voltage is determined by either the LT1934’s
undervoltage lockout of ~3V or by its maximum duty
Table 1
cycle. The duty cycle is the fraction of time that the internal
switch is on and is determined by the input and output
voltages:
DC = (VOUT + VD)/(VIN – VSW + VD)
where VD is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.3V at maximum load for the LT1934, ~0.1V for the
LT1934-1). This leads to a minimum input voltage of:
VIN(MIN) = (VOUT + VD)/DCMAX – VD + VSW
with DCMAX = 0.85.
Inductor Selection
A good first choice for the inductor value is:
L = 2.5 • (VOUT + VD) • 1.8µs/ILIM
where ILIM is the switch current limit (400mA for the
LT1934 and 120mA for the LT1934-1). This choice provides a worst-case maximum load current of 250mA
(60mA for the LT1934-1). The inductor’s RMS current
rating must be greater than the load current and its
saturation current should be greater than ILIM. To keep
efficiency high, the series resistance (DCR) should be less
than 0.3Ω (1Ω for the LT1934-1). Table 2 lists several
vendors and types that are suitable.
This simple rule may not provide the optimum value for
your application. If the load current is less, then you can
relax the value of the inductor and operate with higher
ripple current. This allows you to use a physically smaller
inductor, or one with a lower DCR resulting in higher
efficiency. The following provides more details to guide
inductor selection. First, the value must be chosen so that
the LT1934 can supply the maximum load current drawn
from the output. Second, the inductor must be rated
appropriately so that the LT1934 will function reliably and
the inductor itself will not be overly stressed.
PART
VOUT
L
MINIMUM
COUT
MAXIMUM
LOAD
LT1934
3.3V
100µH
47µH
33µH
100µF
47µF
33µF
300mA
250mA
200mA
5V
150µH
68µH
47µH
47µF
33µF
22µF
300mA
250mA
200mA
3.3V
150µH
100µH
68µH
15µF
10µF
10µF
60mA
45mA
20mA
Detailed Inductor Selection and
Maximum Load Current
5V
220µH
150µH
100µH
10µF
4.7µF
4.7µF
60mA
45mA
20mA
The square wave that the LT1934 produces at its switch
pin results in a triangle wave of current in the inductor. The
LT1934 limits the peak inductor current to ILIM. Because
LT1934-1
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LT1934/LT1934-1
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Table 2. Inductor Vendors
Vendor
Phone
URL
Part Series
Comments
Murata
(404) 426-1300
www.murata.com
LQH3C
Small, Low Cost, 2mm Height
Sumida
(847) 956-0666
www.sumida.com
CR43
CDRH4D28
CDRH5D28
Coilcraft
(847) 639-6400
www.coilcraft.com
DO1607C
DO1608C
DT1608C
Wurth
Electronics
(866) 362-6673
www.we-online.com
WE-PD1, 2, 3, 4
the average inductor current equals the load current, the
maximum load current is:
IOUT(MAX) = IPK – ∆IL /2
where IPK is the peak inductor current and ∆IL is the peakto-peak ripple current in the inductor. The ripple current is
determined by the off time, tOFF = 1.8µs, and the inductor
value:
∆IL = (VOUT + VD) • tOFF /L
IPK is nominally equal to ILIM. However, there is a slight
delay in the control circuitry that results in a higher peak
current and a more accurate value is:
IPK = ILIM + 150ns • (VIN – VOUT)/L
These expressions are combined to give the maximum
load current that the LT1934 will deliver:
IOUT(MAX) = 350mA + 150ns • (VIN – VOUT)/L – 1.8µs
• (VOUT + VD)/2L (LT1934)
IOUT(MAX) = 90mA + 150ns • (VIN – VOUT)/L – 1.8µs
• (VOUT + VD)/2L (LT1934-1)
The minimum current limit is used here to be conservative. The third term is generally larger than the second
term, so that increasing the inductor value results in a
higher output current. This equation can be used to evaluate a chosen inductor or it can be used to choose L for a
given maximum load current. The simple, single equation rule given above for choosing L was found by setting
∆IL = ILIM /2.5. This results in IOUT(MAX) ~0.8ILIM (ignoring the delay term). Note that this analysis assumes that
the inductor current is continuous, which is true if the
ripple current is less than the peak current or ∆IL < IPK.
The inductor must carry the peak current without saturating excessively. When an inductor carries too much
current, its core material can no longer generate additional magnetic flux (it saturates) and the inductance
drops, sometimes very rapidly with increasing current.
This condition allows the inductor current to increase at
a very high rate, leading to high ripple current and
decreased overload protection.
Inductor vendors provide current ratings for power inductors. These are based on either the saturation current or on
the RMS current that the inductor can carry without dissipating too much power. In some cases it is not clear which
of these two determine the current rating. Some data
sheets are more thorough and show two current ratings,
one for saturation and one for dissipation. For LT1934
applications, the RMS current rating should be higher than
the load current, while the saturation current should be
higher than the peak inductor current calculated above.
Input Capacitor
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT1934 and to force this switching current into a
tight local loop, minimizing EMI. The input capacitor must
have low impedance at the switching frequency to do this
effectively. A 2.2µF ceramic capacitor (1µF for the
LT1934-1) satisfies these requirements.
If the input source impedance is high, a larger value
capacitor may be required to keep input ripple low. In this
case, an electrolytic of 10µF or more in parallel with a 1µF
ceramic is a good combination. Be aware that the input
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LT1934/LT1934-1
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capacitor is subject to large surge currents if the LT1934
circuit is connected to a low impedance supply, and that
some electrolytic capacitors (in particular tantalum) must
be specified for such use.
Output Capacitor and Output Ripple
The output capacitor filters the inductor’s ripple current
and stores energy to satisfy the load current when the
LT1934 is quiescent. In order to keep output voltage ripple
low, the impedance of the capacitor must be low at the
LT1934’s switching frequency. The capacitor’s equivalent
series resistance (ESR) determines this impedance. Choose
one with low ESR intended for use in switching regulators.
The contribution to ripple voltage due to the ESR is
approximately ILIM • ESR. ESR should be less than ~150mΩ
for the LT1934 and less than ~500mΩ for the LT1934-1.
The value of the output capacitor must be large enough to
accept the energy stored in the inductor without a large
change in output voltage. Setting this voltage step equal to
1% of the output voltage, the output capacitor must be:
COUT > 50 • L • (ILIM /VOUT)2
For example, an LT1934 producing 3.3V with L = 47µH
requires 33µF. This value can be relaxed if small circuit size
is more important than low output ripple.
Sanyo’s POSCAP series in B-case and C-case sizes provides very good performance in a small package for the
LT1934. Similar performance in traditional tantalum capacitors requires a larger package (C- or D-case). The
LT1934-1, with its lower switch current, can use a B-case
tantalum capacitor.
With a high quality capacitor filtering the ripple current
from the inductor, the output voltage ripple is determined
by the hysteresis and delay in the LT1934’s feedback
comparator. This ripple can be reduced further by adding
a small (typically 10pF) phase lead capacitor between the
output and the feedback pin.
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT1934.
Not all ceramic capacitors are suitable. X5R and X7R types
are stable over temperature and applied voltage and give
dependable service. Other types (Y5V and Z5U) have very
large temperature and voltage coefficients of capacitance.
In the application circuit they may have only a small
fraction of their nominal capacitance and voltage ripple
may be much larger than expected.
Ceramic capacitors are piezoelectric. The LT1934’s switching frequency depends on the load current, and at light
loads the LT1934 can excite the ceramic capacitor at audio
frequencies, generating audible noise. If this is unacceptable, use a high performance electrolytic capacitor at the
output. The input capacitor can be a parallel combination
of a 2.2µF ceramic capacitor and a low cost electrolytic
capacitor. The level of noise produced by the LT1934-1
Table 3. Capacitor Vendors
Vendor
Phone
URL
Part Series
Comments
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
Polymer,
Tantalum
EEF Series
Kemet
(864) 963-6300
www.kemet.com
Ceramic,
Tantalum
Sanyo
(408) 749-9714
www.sanyovideo.com
Ceramic,
Polymer,
Tantalum
Murata
(404) 436-1300
AVX
Taiyo Yuden
(864) 963-6300
www.murata.com
Ceramic
www.avxcorp.com
Ceramic,
Tantalum
www.taiyo-yuden.com
Ceramic
T494, T495
POSCAP
TPS Series
1934f
9
LT1934/LT1934-1
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APPLICATIO S I FOR ATIO
D2
when used with ceramic capacitors will be lower and may
be acceptable.
C3
BOOST
A final precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT1934. A
ceramic input capacitor combined with trace or cable
inductance forms a high quality (under damped) tank
circuit. If the LT1934 circuit is plugged into a live supply,
the input voltage can ring to twice its nominal value,
possibly exceeding the LT1934’s rating. This situation is
easily avoided; see the Hot Plugging Safely section.
LT1934
VIN
VIN
GND
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
(2a)
D2
Schottky diodes with lower reverse voltage ratings usually
have a lower forward drop and may result in higher
efficiency with moderate to high load currents. However,
these diodes also have higher leakage currents. This
leakage current mimics a load current at the output and
can raise the quiescent current of the LT1934 circuit,
especially at elevated temperatures.
BOOST Pin Considerations
Capacitor C3 and diode D2 are used to generate a boost
voltage that is higher than the input voltage. In most cases
a 0.1µF capacitor and fast switching diode (such as the
1N4148 or 1N914) will work well. Figure 2 shows two
ways to arrange the boost circuit. The BOOST pin must be
more than 2.5V above the SW pin for best efficiency. For
outputs of 3.3V and above, the standard circuit (Figure 2a)
is best. For outputs between 2.8V and 3V, use a 0.22µF
capacitor and a small Schottky diode (such as the
BAT-54). For lower output voltages the boost diode can be
tied to the input (Figure 2b). The circuit in Figure 2a is more
efficient because the BOOST pin current comes from a
lower voltage source. You must also be sure that the
maximum voltage rating of the BOOST pin is not exceeded.
The minimum operating voltage of an LT1934 application
is limited by the undervoltage lockout (~3V) and by the
C3
BOOST
Catch Diode
A 0.5A Schottky diode is recommended for the catch
diode, D1. The diode must have a reverse voltage rating
equal to or greater than the maximum input voltage. The
ON Semiconductor MBR0540 is a good choice; it is rated
for 0.5A forward current and a maximum reverse voltage
of 40V.
VOUT
SW
LT1934
VIN
VIN
SW
VOUT
GND
1934 F02
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
(2b)
Figure 2. Two Circuits for Generating the Boost Voltage
maximum duty cycle as outlined above. For proper startup, the minimum input voltage is also limited by the boost
circuit. If the input voltage is ramped slowly, or the LT1934
is turned on with its SHDN pin when the output is already
in regulation, then the boost capacitor may not be fully
charged. Because the boost capacitor is charged with the
energy stored in the inductor, the circuit will rely on some
minimum load current to get the boost circuit running
properly. This minimum load will depend on input and
output voltages, and on the arrangement of the boost
circuit. The minimum load generally goes to zero once the
circuit has started. Figure 3 shows a plot of minimum load
to start and to run as a function of input voltage. In many
cases the discharged output capacitor will present a load
to the switcher which will allow it to start. The plots show
the worst-case situation where VIN is ramping very slowly.
Use a Schottky diode (such as the BAT-54) for the lowest
start-up voltage.
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above VOUT. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
1934f
10
LT1934/LT1934-1
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APPLICATIO S I FOR ATIO
Minimum Input Voltage VOUT = 3.3V
6.0
INPUT VOLTAGE (V)
LT1934
VOUT = 3.3V
5.5 TA = 25°C
BOOST DIODE TIED TO OUTPUT
5.0
VIN TO START
4.5
4.0
VIN TO RUN
3.5
3.0
0.1
VIN), then the LT1934’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, then parasitic diodes inside the LT1934 can
pull large currents from the output through the SW pin and
the VIN pin. Figure 4 shows a circuit that will run only when
the input voltage is present and that protects against a
shorted or reversed input.
D4
1
10
100
LOAD CURRENT (mA)
5
VIN
VIN
1934 G12
Minimum Input Voltage VOUT = 5V
INPUT VOLTAGE (V)
GND
2
SW
6
VOUT
FB
3
BACKUP
VIN TO START
D4: MBR0530
6
1934 F07
Figure 4. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output; It Also Protects the Circuit
from a Reversed Input. The LT1934 Runs Only When the Input
is Present
VIN TO RUN
5
4
0.1
SHDN
1M
LT1934
VOUT = 5V
TA = 25°C
7 BOOST DIODE TIED TO OUTPUT
1
LT1934
100k 4
8
BOOST
PCB Layout
1
10
100
LOAD CURRENT (mA)
1934 G13
Figure 3. The Minimum Input Voltage Depends
on Output Voltage, Load Current and Boost Circuit
maximum duty cycle of the LT1934, requiring a higher
input voltage to maintain regulation.
Shorted Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT1934 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT1934 is absent. This may occur in battery charging
applications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT1934’s
output. If the VIN pin is allowed to float and the SHDN pin
is held high (either by a logic signal or because it is tied to
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 5 shows
the high current paths in the buck regulator circuit. Note
that large, switched currents flow in the power switch, the
catch diode (D1) and the input capacitor (C2). The loop
formed by these components should be as small as
possible. Furthermore, the system ground should be tied
to the regulator ground in only one place; this prevents the
switched current from injecting noise into the system
ground. These components, along with the inductor and
output capacitor, should be placed on the same side of the
circuit board, and their connections should be made on
that layer. Place a local, unbroken ground plane below
these components, and tie this ground plane to system
ground at one location, ideally at the ground terminal of the
output capacitor C1. Additionally, the SW and BOOST
nodes should be kept as small as possible. Finally, keep
the FB node as small as possible so that the ground pin and
1934f
11
LT1934/LT1934-1
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APPLICATIO S I FOR ATIO
VIN
VIN
SW
GND
SW
GND
(5a)
(5b)
IC1
VSW
VIN
C2
L1
SW
D1
GND
C1
1934 F05
(5c)
Figure 5. Subtracting the Current When the Switch is On (a) from the Current When the Switch is Off (b) Reveals the Path of the High
Frequency Switching Current (c). Keep This Loop Small. The Voltage on the SW and BOOST Nodes Will Also be Switched; Keep These
Nodes as Small as Possible. Finally, Make Sure the Circuit is Shielded with a Local Ground Plane
SHUTDOWN
VIN
VOUT
SYSTEM
GROUND
VIAS TO LOCAL GROUND PLANE
OUTLINE OF LOCAL GROUND PLANE
1934 F06
Figure 6. A Good PCB Layout Ensures Proper, Low EMI Operation
ground traces will shield it from the SW and BOOST nodes.
Figure 6 shows component placement with trace, ground
plane and via locations. Include two vias near the GND pin
of the LT1934 to help remove heat from the LT1934 to the
ground plane.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT1934 and LT1934-1 circuits. However, these capacitors can cause problems if the LT1934
is plugged into a live supply (see Linear Technology
Application Note 88 for a complete discussion). The low
loss ceramic capacitor combined with stray inductance in
series with the power source forms an under damped tank
circuit, and the voltage at the VIN pin of the LT1934 can ring
to twice the nominal input voltage, possibly exceeding the
LT1934’s rating and damaging the part. If the input supply
is poorly controlled or the user will be plugging the LT1934
into an energized supply, the input network should be
designed to prevent this overshoot.
1934f
12
LT1934/LT1934-1
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APPLICATIO S I FOR ATIO
Figure 7 shows the waveforms that result when an LT1934
circuit is connected to a 24V supply through six feet of
24-gauge twisted pair. The first plot is the response with
a 2.2µF ceramic capacitor at the input. The input voltage
rings as high as 35V and the input current peaks at 20A.
One method of damping the tank circuit is to add another
capacitor with a series resistor to the circuit. In Figure 7b
an aluminum electrolytic capacitor has been added. This
capacitor’s high equivalent series resistance damps the
circuit and eliminates the voltage overshoot. The extra
capacitor improves low frequency ripple filtering and can
slightly improve the efficiency of the circuit, though it is
likely to be the largest component in the circuit. An
alternative solution is shown in Figure 7c. A 1Ω resistor is
CLOSING SWITCH
SIMULATES HOT PLUG
IIN
VIN
LT1934
+
VIN
10V/DIV
2.2µF
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
IIN
10A/DIV
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
10µs/DIV
(7a)
LT1934
10µF
35V
AI.EI.
+
2.2µF
(7b)
1Ω
LT1934
0.1µF
2.2µF
(7c)
LT1934-1
1µF
(7d)
4.7Ω
LT1934-1
0.1µF
1µF
(7e)
1934 F07
Figure 7. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation When the LT1934 is Connected to a Live Supply
1934f
13
LT1934/LT1934-1
U
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APPLICATIO S I FOR ATIO
added in series with the input to eliminate the voltage
overshoot (it also reduces the peak input current). A 0.1µF
capacitor improves high frequency filtering. This solution
is smaller and less expensive than the electrolytic capacitor. For high input voltages its impact on efficiency is
minor, reducing efficiency less than one half percent for a
5V output at full load operating from 24V.
Voltage overshoot gets worse with reduced input capacitance. Figure 7d shows the hot plug response with a 1µF
ceramic input capacitor, with the input ringing above 40V.
The LT1934-1 can tolerate a larger input resistance, such
as shown in Figure 7e where a 4.7Ω resistor damps the
voltage transient and greatly reduces the input current
glitch on the 24V supply.
High Temperature Considerations
The die temperature of the LT1934 must be lower than the
maximum rating of 125°C. This is generally not a concern
unless the ambient temperature is above 85°C. For higher
temperatures, care should be taken in the layout of the
circuit to ensure good heat sinking of the LT1934. The
maximum load current should be derated as the ambient
temperature approaches 125°C.
The die temperature is calculated by multiplying the LT1934
power dissipation by the thermal resistance from junction
to ambient. Power dissipation within the LT1934 can be
estimated by calculating the total power loss from an
efficiency measurement and subtracting the catch diode
loss. The resulting temperature rise at full load is nearly
independent of input voltage. Thermal resistance depends
on the layout of the circuit board, but a value of 150°C/W
is typical.
The temperature rise for an LT1934 producing 5V at
250mA is approximately 25°C, allowing it to deliver full
load to 100°C ambient. Above this temperature the load
current should be reduced. For 3.3V at 250mA the temperature rise is 15°C.
Finally, be aware that at high ambient temperatures the
external Schottky diode, D1, is likely to have significant
leakage current, increasing the quiescent current of the
LT1934 converter.
Outputs Greater Than 6V
For outputs greater than 6V, tie a diode (such as a 1N4148)
from the SW pin to VIN to prevent the SW pin from ringing
above VIN during discontinuous mode operation. The 12V
output circuit in Typical Applications shows the location of
this diode. Also note that for outputs above 6V, the input
voltage range will be limited by the maximum rating of the
BOOST pin. The 12V circuit shows how to overcome this
limitation using an additional Zener diode.
1934f
14
LT1934/LT1934-1
U
TYPICAL APPLICATIO S
3.3V Step-Down Converter
D2
0.1µF
BOOST
VIN
4.5V TO 34V
ON OFF
VOUT
3.3V
45mA
SW
VIN
C2
1µF
L1
100µH
D1
LT1934-1
SHDN
10pF
1M
+
FB
C1
22µF
604k
GND
C1: TAIYO YUDEN JMK316BJ226ML
C2: TAIYO YUDEN GMK316BJ105ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMDSH-3
L1: COILCRAFT DO1608C-104 OR
WURTH ELECTRONICS WE-PD4 TYPE S
1934 TA04
5V Step-Down Converter
D2
0.1µF
BOOST
VIN
6.5V TO 34V
ON OFF
VOUT
5V
45mA
SW
VIN
C2
1µF
L1
150µH
D1
LT1934-1
SHDN
10pF
1M
+
FB
GND
C1: TAIYO YUDEN JMK316BJ226ML
C2: TAIYO YUDEN GMK316BJ105ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMPD914
L1: COILCRAFT DO1608C-154 OR
WURTH ELECTRONICS WE-PD4 TYPE S
C1
22µF
332k
1934 TA05
1934f
15
LT1934/LT1934-1
U
TYPICAL APPLICATIO S
1.8V Step-Down Converter
D2
0.1µF
BOOST
VIN
3.6V TO 16V
VIN
C2
2.2µF
VOUT
1.8V
250mA
SW
D1
LT1934
ON OFF
L1
33µH
147k
+
C1
100µF
FB
SHDN
332k
GND
C1: SANYO 2R5TPB100M
C2: TAIYO YUDEN EMK316BJ225ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMPD914
L1: SUMIDA CR43-330
1934 TA06
Loop Powered 3.3V Supply with Additional Isolated Output
D3
L1B
50µH •
VIN
D4
10V
SW
390k
•
D1
LT1934-1
1M
10pF
VOUT
3V
9mA
+
SHDN
1µF
D2
C1 L1A
50µH
BOOST
VIN
14V TO 32V
<3.6mA
+
ISOLATED
OUT
3V
10µF 3mA
GND
FB
33µF
715k
D1: ON SEMICONDUCTOR MBR0540
D2, D3: BAT54
D4: CENTRAL CMPZ5240B
L1: COILTRONICS CTX50-1
ZENER DIODE D4 PROVIDES AN UNDERVOLTAGE LOCKOUT,
REDUCING THE INPUT CURRENT REQUIRED AT START-UP
1934 TA08
1934f
16
LT1934/LT1934-1
U
TYPICAL APPLICATIO S
Standalone 350mA Li-Ion Battery Charger
D2
0.1µF
L1
47µH
BOOST
D3
VIN
7V TO 28V
1k
CHRG
SHDN
332k
GATE
LTC4052
FB
GND
0.047µF
VIN
D1 1M
LT1934
+
10k
SW
VIN
C2
1µF
1k
+
ACPR
SENSE
TIMER
GND
BAT
350mA
C1
47µF
CTIMER
0.1µF
C5
10µF
C1: SANYO 6TPB47M
C2: TAIYO YUDEN GMK316BJ105ML
D1, D3: ON SEMICONDUCTOR MBR0540
D2: CENTRAL CMDSH-3
L1: SUMIDA CR43-470
0.022µF
(619) 661-6835
(408) 573-4150
(602) 244-6600
(516) 435-1110
(847) 956-0667
CHARGE STATUS
AC PRESENT
1-CELL 4.2V
Li-Ion
BATTERY
1934 TA07a
500
VIN = 24V
CHARGE CURRENT (mA)
400
300
VIN = 8V
VIN = 12V
200
100
0
2.5
3.5
3
4
BATTERY VOLTAGE (V)
4.5
1934 TA07b
1934f
17
LT1934/LT1934-1
U
TYPICAL APPLICATIO S
12V Step-Down Converter
D2
D4
0.1µF
D3
L1
100µH
BOOST
VIN
15V TO 32V
C2
2.2µF
ON OFF
VOUT
12V
170mA
SW
VIN
LT1934
SHDN
D1
866k
FB
GND
C1: KEMET T495D226K020AS
C2: TAIYO YUDEN GMK325BJ225MN
D1: ON SEMI MBR0540
D2, D4: CENTRAL CMPD914
D3: CENTRAL CMPZ5234B 6.2V ZENER
L1: TDK SLF6028T-101MR42
+
C1
22µF
100k
1934 TA09
1934f
18
LT1934/LT1934-1
U
PACKAGE DESCRIPTION
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
0.62
MAX
2.90 BSC
(NOTE 4)
0.95
REF
1.22 REF
3.85 MAX 2.62 REF
1.4 MIN
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
1.90 BSC
S6 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1934f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LT1934/LT1934-1
U
TYPICAL APPLICATIO
5V Step-Down Converter
D2
0.1µF
BOOST
VIN
6.5V TO 34V
D1
LT1934
ON OFF
VOUT
5V
250mA
SW
VIN
C2
2.2µF
L1
68µH
SHDN
10pF
1M
+
FB
C1
68µF
332k
GND
C1: SANYO 6TPB68M
C2: TAIYO YUDEN GMK325BJ225MN
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMPD914
L1: SUMIDA CDRH5D28-680
1934 TA03
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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Step-Down DC/DC Converter
VIN = 3.6V to 25V, VOUT = 1.25V, IQ = 1.9mA, ISD = <1µA,
ThinSOT Package
LT1676
60V, 440mA (IOUT), 100kHz, High Efficiency
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VIN = 7.4V to 60V, VOUT = 1.24V, IQ = 3.2mA, ISD = 2.5µA,
S8 Package
LT1765
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LT1766
60V, 1.2A (IOUT), 200kHz, High Efficiency
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VIN = 5.5V to 60V, VOUT = 1.2V, IQ = 2.5mA, ISD = 25µA,
TSSOP16/E Package
LT1767
25V, 1.2A (IOUT), 1.25MHz, High Efficiency
Step-Down DC/DC Converter
VIN = 3V to 25V; VOUT = 1.2V, IQ = 1mA, ISD = 6µA,
MS8/E Packages
LT1776
40V, 550mA (IOUT), 200kHz, High Efficiency
Step-Down DC/DC Converter
VIN = 7.4V to 40V; VOUT = 1.24V, IQ = 3.2mA, ISD = 30µA,
N8, S8 Packages
LTC®1877
600mA (IOUT), 550kHz, Synchronous
Step-Down DC/DC Converter
VIN = 2.7V to 10V; VOUT = 0.8V, IQ = 10µA, ISD = <1µA,
MS8 Package
LTC1879
1.2A (IOUT), 550kHz, Synchronous
Step-Down DC/DC Converter
VIN = 2.7V to 10V; VOUT = 0.8V, IQ = 15µA, ISD = <1µA,
TSSOP16 Package
LT1956
60V, 1.2A (IOUT), 500kHz, High Efficiency
Step-Down DC/DC Converter
VIN = 5.5V to 60V, VOUT = 1.2V, IQ = 2.5mA, ISD = 25µA,
TSSOP16/E Package
LTC3405/LTC3405A
300mA (IOUT), 1.5MHz, Synchronous
Step-Down DC/DC Converter
VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 20µA, ISD = <1µA,
ThinSOT Package
LTC3406/LTC3406B
600mA (IOUT), 1.5MHz, Synchronous
Step-Down DC/DC Converter
VIN = 2.5V to 5.5V, VOUT = 0.6V, IQ = 20µA, ISD = <1µA,
ThinSOT Package
LTC3411
1.25A (IOUT), 4MHz, Synchronous
Step-Down DC/DC Converter
VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD = <1µA,
MS Package
LTC3412
2.5A (IOUT), 4MHz, Synchronous
Step-Down DC/DC Converter
VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD = <1µA,
TSSOP16E Package
LTC3430
60V, 2.75A (IOUT), 200kHz, High Efficiency
Step-Down DC/DC Converter
VIN = 5.5V to 60V, VOUT = 1.2V, IQ = 2.5mA, ISD = 30µA,
TSSOP16E Package
1934f
20
Linear Technology Corporation
LT/TP 0703 1K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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