AD AD5443YRMZ-REEL 8-/10-/12-bit high bandwidth multiplying dacs with serial interface Datasheet

8-/10-/12-Bit High Bandwidth
Multiplying DACs with Serial Interface
AD5426/AD5432/AD5443
Data Sheet
FEATURES
GENERAL DESCRIPTION
2.5 V to 5.5 V supply operation
50 MHz serial interface
10 MHz multiplying bandwidth
2.5 MSPS update rate
INL of ±1 LSB for 12-bit DAC
±10 V reference input
Low glitch energy < 2 nV-s
Extended temperature range −40°C to +125°C
10-lead MSOP package
Pin-compatible 8-, 10-, and 12-bit current output DACs
Guaranteed monotonic
4-quadrant multiplication
Power-on reset with brownout detection
Daisy-chain mode
Readback function
0.4 µA typical power consumption
The AD5426/AD5432/AD54431 are CMOS 8-, 10-, and 12-bit
current output digital-to-analog converters (DACs), respectively.
These devices operate from a 2.5 V to 5.5 V power supply,
making them suitable for battery-powered applications and
many other applications.
These DACs use a double buffered, 3-wire serial interface that is
compatible with SPI, QSPI™, MICROWIRE™, and most DSP
interface standards. In addition, a serial data out pin (SDO)
allows for daisy-chaining when multiple packages are used.
Data readback allows the user to read the contents of the DAC
register via the SDO pin. On power-up, the internal shift register
and latches are filled with 0s and the DAC outputs are at zero scale.
As a result of manufacturing on a CMOS submicron process,
the parts offer excellent 4-quadrant multiplication characteristics
with large signal multiplying bandwidths of 10 MHz. The applied
external reference input voltage, VREF, determines the full-scale
output current. An integrated feedback resistor, RFB, provides
temperature tracking and full-scale voltage output when combined
with an external current to voltage precision amplifier.
APPLICATIONS
Portable battery-powered applications
Waveform generators
Analog processing
Instrumentation
Programmable amplifiers and attenuators
Digitally controlled calibration
Programmable filters and oscillators
Composite video
Ultrasound
Gain, offset, and voltage trimming
The AD5426/AD5432/AD5443 DACs are available in small,
10-lead MSOP packages.
The EV-AD5443/46/53SDZ evaluation board is available for
evaluating DAC performance. For more information, see the
UG-327 evaluation board user guide.
FUNCTIONAL BLOCK DIAGRAM
VREF
VDD
AD5426/
AD5432/
AD5443
R
RFB
IOUT1
IOUT2
8-/10-/12-BIT
R-2R DAC
DAC REGISTER
POWER-ON
RESET
INPUT LATCH
CONTROL LOGIC AND
INPUT SHIFT REGISTER
GND
SDO
03162-001
SYNC
SCLK
SDIN
Figure 1.
1
U.S. Patent No. 5,689,257.
Rev. G
Document Feedback
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Tel: 781.329.4700 ©2004–2013 Analog Devices, Inc. All rights reserved.
Technical Support
www.analog.com
AD5426/AD5432/AD5443
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Circuit Operation ....................................................................... 15
Applications ....................................................................................... 1
Single-Supply Applications ....................................................... 17
General Description ......................................................................... 1
Positive Output Voltage ............................................................. 17
Functional Block Diagram .............................................................. 1
Adding Gain ................................................................................ 18
Revision History ............................................................................... 2
DACs Used as a Divider or Programmable Gain Element ... 18
Specifications..................................................................................... 3
Reference Selection .................................................................... 18
Timing Characteristics ................................................................ 5
Amplifier Selection .................................................................... 18
Absolute Maximum Ratings ............................................................ 6
Serial Interface ............................................................................ 20
ESD Caution .................................................................................. 6
PCB Layout and Power Supply Decoupling................................ 22
Pin Configuration and Function Descriptions ............................. 7
Overview of AD54xx and AD55xx Devices ............................... 23
Typical Performance Characteristics ............................................. 8
Outline Dimensions ....................................................................... 24
Terminology .................................................................................... 14
Ordering Guide .......................................................................... 24
Theory of Operation ...................................................................... 15
REVISION HISTORY
6/13—Rev. F to Rev. G
Change to General Description Section ........................................ 1
Changes to Ordering Guide .......................................................... 24
7/12—Rev. E to Rev. F
No Change to Content, Changed VDD Values in 7/12 Revision
History Only ...................................................................................... 2
7/12—Rev. D to Rev. E
Changed VDD = 3 V to VDD = 2.5 V ............................. Throughout
Changes to Table 2 ............................................................................ 4
Changes to Table 4 ............................................................................ 7
Change to Daisy-Chain Mode Section ........................................ 20
Change to Ordering Guide ............................................................ 24
4/12—Rev. C to Rev. D
Changed VDD = 2.5 V to VDD = 3 V ............................. Throughout
Changes to General Description Section ...................................... 1
Deleted Microprocessor Interface Section, ADSP-21xx to
AD5426/AD5432/AD5443 Interface Section, Figure 51,
Figure 52, Table 11, ADSP-BF5x to AD5426/AD5432/AD5443
Interface Section, Figure 53 and Figure 54; Renumbered
Sequentially ..................................................................................... 21
Deleted 80C51/80L51 to AD5426/AD5432/AD5443 Interface
Section, Figure 55, MC68HC11 Interface to AD5426/AD5432/
AD5443 Interface Section, Figure 56, MICROWIRE to
AD5426/AD5432/AD5443 Interface Section, Figure 57,
PIC16C6x/7x to AD5426/AD5432/AD5443, and Figure 58 .... 22
Deleted Evaluation Board for the AD5426/AD5432/AD5443
Series of DACs Section, Operating the Evaluation Board
Section, and Power Supplies Section ........................................... 23
Deleted Figure 59 and Figure 60................................................... 24
Updated Outline Dimensions ....................................................... 24
Changes to Ordering Guide .......................................................... 24
Deleted Figure 61............................................................................ 25
Deleted Figure 62............................................................................ 26
2/09—Rev. B to Rev. C
Changes to Low Power Serial Interface Section and DaisyChain Mode Section....................................................................... 20
Updated Outline Dimensions ....................................................... 28
11/08—Rev. A to Rev. B
Changes to Ordering Guide .......................................................... 28
5/05—Rev. 0 to Rev. A
Updated Format .................................................................. Universal
Changes to Specifications .................................................................3
Changes to Figure 42...................................................................... 16
Change to Figure 45 ....................................................................... 17
Change to Figure 46 ....................................................................... 18
Changes to Table 7, Table 8, and Table 9 ..................................... 19
Additions to Microprocessor Interface Section.......................... 21
2/04—Revision 0: Initial Version
Rev. G | Page 2 of 24
Data Sheet
AD5426/AD5432/AD5443
SPECIFICATIONS
VDD = 2.5 V to 5.5 V, VREF = 10 V, IOUT2 = 0 V; temperature range for Y version: −40°C to +125°C; all specifications TMIN to TMAX, unless
otherwise noted; dc performance measured with OP177; ac performance with AD8038, unless otherwise noted.
Table 1.
Parameter
STATIC PERFORMANCE
AD5426
Resolution
Relative Accuracy
Differential Nonlinearity
AD5432
Resolution
Relative Accuracy
Differential Nonlinearity
AD5443
Resolution
Relative Accuracy
Differential Nonlinearity
Gain Error
Gain Error Temperature Coefficient 1
Output Leakage Current
REFERENCE INPUT1
Reference Input Range
VREF Input Resistance
RFB Resistance
Input Capacitance
Code Zero Scale
Code Full Scale
DIGITAL INPUT/OUTPUT1
Input High Voltage, VIH
Input Low Voltage, VIL
Output High Voltage, VOH
Min
Typ
Max
Unit
Test Conditions/Comments
8
±0.25
±0.5
Bits
LSB
LSB
Guaranteed monotonic
10
±0.5
±1
Bits
LSB
LSB
Guaranteed monotonic
12
±1
−1/+2
±10
±10
±20
Bits
LSB
LSB
mV
ppm FSR/°C
nA
nA
Data = 0x0000, TA = 25°C, IOUT1
Data = 0x0000, T = −40°C to 125°C, IOUT1
±10
10
10
12
12
V
kΩ
kΩ
Input resistance TC = −50 ppm/°C
Input resistance TC = −50 ppm/°C
3
5
6
8
pF
pF
±5
8
8
1.7
0.6
VDD − 1
VDD − 0.5
Output Low Voltage, VOL
Input Leakage Current, IIL
Input Capacitance
DYNAMIC PERFORMANCE1
Reference Multiplying Bandwidth
Output Voltage Settling Time
Measured to ±16 mV of FS
Measured to ±4 mV of FS
Measured to ±1 mV of FS
Digital Delay
10% to 90% Rise/Fall Time
Digital-to-Analog Glitch Impulse
Multiplying Feedthrough Error
4
0.4
0.4
1
10
10
50
55
90
40
15
2
70
48
V
V
V
V
V
V
µA
pF
MHz
100
110
160
75
30
ns
ns
ns
ns
ns
nV-s
dB
dB
Rev. G | Page 3 of 24
Guaranteed monotonic
VDD = 4.5 V to 5 V, ISOURCE = 200 µA
VDD = 2.5 V to 3.6 V, ISOURCE = 200 µA
VDD = 4.5 V to 5 V, ISINK = 200 µA
VDD = 2.5 V to 3.6 V, ISINK = 200 µA
VREF = ±3.5 V; DAC loaded all 1s
VREF = 10 V; RLOAD = 100 Ω, DAC latch alternately
loaded with 0s and 1s
Interface delay time
Rise and fall time, VREF = 10 V, RLOAD = 100 Ω
1 LSB change around major carry, VREF = 0 V
DAC latch loaded with all 0s, VREF = ±3.5
1 MHz
10 MHz
AD5426/AD5432/AD5443
Parameter
Output Capacitance
IOUT1
Data Sheet
Min
IOUT2
Digital Feedthrough
Analog THD
Digital THD
50 kHz fOUT
20 kHz fOUT
Output Noise Spectral Density
SFDR Performance (Wide Band)
50 kHz fOUT
20 kHz fOUT
SFDR Performance (Narrow Band)
50 kHz fOUT
20 kHz fOUT
Intermodulation Distortion
POWER REQUIREMENTS
Power Supply Range
IDD
Typ
Max
Unit
Test Conditions/Comments
12
10
22
10
0.1
17
12
25
12
pF
pF
pF
pF
nV-s
All 0s loaded
All 1s loaded
All 0s loaded
All 1s loaded
Feedthrough to DAC output with SYNC high and
alternate loading of all 0s and all 1s
VREF = 3.5 V p-p, all 1s loaded, f = 1 kHz
Clock = 1 MHz, VREF = 3.5 V, CCOMP = 1.8 pF
81
dB
73
74
25
dB
dB
nV/√Hz
75
76
dB
dB
87
87
78
dB
dB
dB
Clock = 1 MHz, f1 = 20 kHz, f2 = 25 kHz, VREF = 3.5 V
V
µA
µA
%/%
TA = 25°C, logic inputs = 0 V or VDD
T = −40°C to +125°C , logic inputs = 0 V or VDD
∆VDD = ±5%
Clock = 1 MHz, VREF = 3.5 V
2.5
0.4
Power Supply Sensitivity1
1
@ 1 kHz
Clock = 1 MHz, VREF = 3.5 V
5.5
0.6
5
0.001
Guaranteed by design and characterization, not subject to production testing.
Rev. G | Page 4 of 24
Data Sheet
AD5426/AD5432/AD5443
TIMING CHARACTERISTICS
All input signals are specified with tr = tf = 1 ns (10% to 90% of VDD) and timed from a voltage level of (VIL + VIH)/2. VDD = 2.5 V to 5.5 V,
VREF = 10 V, IOUT2 = 0 V; temperature range for Y version: −40°C to +125°C; all specifications TMIN to TMAX, unless otherwise noted.
Table 2.
Parameter
fSCLK
t1
t2
t3
t41
t5
t6
t7
t8
t92, 3
2.5 V to 5.5 V
50
20
8
8
13
5
3
5
30
80
120
4.5 V to 5.5 V
50
20
8
8
13
5
3
5
30
45
65
Unit
MHz max
ns min
ns min
ns min
ns min
ns min
ns min
ns min
ns min
ns typ
ns max
Test Conditions/Comments
Max clock frequency
SCLK cycle time
SCLK high time
SCLK low time
SYNC falling edge to SCLK active edge setup time
Data setup time
Data hold time
SYNC rising edge to SCLK active edge
Minimum SYNC high time
SCLK active edge to SDO valid
1
Falling or rising edge as determined by control bits of serial word.
Daisy-chain and readback modes cannot operate at maximum clock frequency. SDO timing specifications measured with load circuit, as shown in Figure 4.
3
SDO operates with a VDD of 3.0 V to 5.5 V.
2
t1
SCLK
t8
t2
t4
t3
t7
SYNC
t6
t5
DIN
DB0
03162-002
DB15
ALTERNATIVELY, DATA MAY BE CLOCKED INTO INPUT SHIFT REGISTER ON RISING EDGE OF
SCLK AS DETERMINED BY CONTROL BITS. TIMING AS PER ABOVE, WITH SCLK INVERTED.
Figure 2. Standalone Mode Timing Diagram
t1
SCLK
t2
t3
t7
t8
t4
t6
SYNC
t5
SDIN
DB15 (N)
t6
DB0 (N)
DB15
(N + 1)
DB0
(N + 1)
DB15(N)
DB0(N)
SDO
ALTERNATIVELY, DATA MAY BE CLOCKED INTO INPUT SHIFT REGISTER ON RISING EDGE OF SCLK AS
DETERMINED BY CONTROL BITS. IN THIS CASE, DATA WOULD BE CLOCKED OUT OF SDO ON FALLING
EDGE OF SCLK. TIMING AS PER ABOVE, WITH SCLK INVERTED.
Figure 3. Daisy-Chain and Readback Modes Timing Diagram
Rev. G | Page 5 of 24
03162-003
t9
AD5426/AD5432/AD5443
Data Sheet
ABSOLUTE MAXIMUM RATINGS
Table 3.
Parameter
VDD to GND
VREF, RFB to GND
IOUT1, IOUT2 to GND
Logic Inputs and Output 1
Operating Temperature Range
Extended Industrial (Y Version)
Storage Temperature Range
Junction Temperature
10-lead MSOP θJA Thermal Impedance
Lead Temperature, Soldering (10 sec)
IR Reflow, Peak Temperature (<20 sec)
1
Rating
−0.3 V to +7 V
−12 V to +12 V
−0.3 V to VDD + 0.3 V
−0.3 V to VDD + 0.3 V
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
200µA
−40°C to +125°C
−65°C to +150°C
150°C
206°C/W
300°C
235°C
IOL
VOH (MIN) + VOL (MAX)
2
TO OUTPUT PIN
CL
20pF
200µA
IOH
Figure 4. Load Circuit for SDO Timing Specifications
Overvoltages at SCLK, SYNC, and DIN are clamped by internal diodes.
ESD CAUTION
Rev. G | Page 6 of 24
03162-004
Transient currents of up to 100 mA do not cause SCR latch-up.
TA = 25°C, unless otherwise noted.
Data Sheet
AD5426/AD5432/AD5443
IOUT1 1
IOUT2 2
GND 3
SCLK 4
SDIN 5
AD5426/
AD5432/
AD5443
TOP VIEW
(Not to Scale)
10
RFB
9
VREF
8
VDD
7
SDO
6
SYNC
03162-005
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
Figure 5. Pin Configuration
Table 4. Pin Function Descriptions
Pin No.
1
2
3
4
Mnemonic
IOUT1
IOUT2
GND
SCLK
5
SDIN
6
SYNC
7
SDO
8
9
10
VDD
VREF
RFB
Description
DAC Current Output.
DAC Analog Ground. This pin should normally be tied to the analog ground of the system.
Digital Ground Pin.
Serial Clock Input. By default, data is clocked into the input shift register on the falling edge of the serial clock
input. Alternatively, by means of the serial control bits, the device may be configured such that data is clocked into
the shift register on the rising edge of SCLK. The device can accommodate clock rates up to 50 MHz.
Serial Data Input. Data is clocked into the 16-bit input register on the active edge of the serial clock input. By
default, on power-up, data is clocked into the shift register on the falling edge of SCLK. The control bits allow the
user to change the active edge to rising edge.
Active Low Control Input. This is the frame synchronization signal for the input data. When SYNC goes low, it
powers on the SCLK and DIN buffers, and the input shift register is enabled. Data is loaded to the mode, the serial
interface counts clocks, and data is latched to the shift register on the 16th active clock edge.
Serial Data Output. This allows a number of parts to be daisy-chained. By default, data is clocked into the shift
register on the falling edge and out via SDO on the rising edge of SCLK. Data is always clocked out on the
alternate edge to loading data to the shift register. Writing the readback control word to the shift register makes
the DAC register contents available for readback on the SDO pin, clocked out on the opposite edges to the active
clock edge. SDO operates with a VDD of 3.0 V to 5.5 V.
Positive Power Supply Input. These parts can be operated from a supply of 2.5 V to 5.5 V.
DAC Reference Voltage Input.
DAC Feedback Resistor Pin. Establish voltage output for the DAC by connecting to external amplifier output.
Rev. G | Page 7 of 24
AD5426/AD5432/AD5443
Data Sheet
TYPICAL PERFORMANCE CHARACTERISTICS
0.20
0.20
0.10
0.10
0.05
0.05
TA = 25°C
VREF = 10V
0.15 VDD = 5V
–0.05
0
–0.05
–0.10
–0.10
–0.15
–0.15
–0.20
50
100
150
200
250
CODE
–0.20
0
50
0.3
0.2
0.2
0.1
0.1
DNL (LSB)
0.3
–0.1
0
–0.1
–0.2
–0.2
–0.3
–0.3
–0.4
–0.4
200
400
600
800
10000
CODE
–0.5
0
200
TA = 25°C
0.8 VREF = 10V
VDD = 5V
0.6
0.6
0.4
0.4
0.2
0.2
DNL (LSB)
1.0
TA = 25°C
0.8 VREF = 10V
VDD = 5V
0
–0.2
4000
–0.2
–0.6
–0.6
–0.8
–0.8
–1.0
3500
4000
03162-008
–0.4
3000
1000
0
–0.4
2000 2500
CODE
800
Figure 10. DNL vs. Code (10-Bit DAC)
1.0
1500
600
CODE
Figure 7. INL vs. Code (10-Bit DAC)
1000
400
03162-010
0
03162-007
iNL (LSB)
TA = 25°C
0.4 VREF = 10V
VDD = 5V
–0.5
INL (LSB)
250
0.5
TA = 25°C
0.4 VREF = 10V
VDD = 5V
500
200
Figure 9. DNL vs. Code (8-Bit DAC)
0.5
0
150
CODE
Figure 6. INL vs. Code (8-Bit DAC)
0
100
03162-011
0
03162-009
INL (LSB)
0
03162-006
INL (LSB)
TA = 25°C
VREF = 10V
0.15 VDD = 5V
Figure 8. INL vs. Code (12-Bit DAC)
–1.0
0
500
1000
1500
2000 2500
CODE
3000
Figure 11. DNL vs. Code (12-Bit DAC)
Rev. G | Page 8 of 24
3500
Data Sheet
AD5426/AD5432/AD5443
0.6
2.0
0.5
1.5
MAX INL
0.4
0.2
TA = 25°C
VDD = 5V
AD5443
LSB
INL (LSB)
0.3
0.1
MAX INL
1.0
TA = 25°C
0.5 VREF = 0V
VDD = 3V
AD5443
0
MAX DNL
–0.5
0
MIN INL
–1.0
MIN INL
MIN DNL
–1.5
–0.2
2
3
4
5
6
7
REFERENCE VOLTAGE
8
9
10
–2.0
0.5
03162-012
–0.3
Figure 12. INL vs. Reference Voltage
0.7
0.8
0.9
1.0
1.1
VBIAS (V)
1.2
1.3
1.4
1.5
Figure 15. Linearity vs. VBIAS Voltage Applied to IOUT2
–0.40
4
TA = 25°C
VDD = 5V
AD5443
–0.45
0.6
03162-015
–0.1
TA = 25°C
VREF = 2.5V
VDD = 3V
AD5443
3
2
MAX DNL
MAX INL
1
0
–0.55
LSB
DNL (LSB)
–0.50
–0.60
–1
MIN DNL
–2
MIN INL
–3
MIN DNL
–0.65
–4
4
5
6
7
REFERENCE VOLTAGE
8
9
10
–5
0
Figure 13. DNL vs. Reference Voltage
0.4
0.6
0.8
1.0
1.2
VBIAS (V)
1.4
1.6
1.8
2.0
1.5
Figure 16. Linearity vs. VBIAS Voltage Applied to IOUT2
5
0.5
TA = 25°C
0.4 VREF = 0V
VDD = 3V AND 5V
VREF = 10V
4
3
0.3
VDD = 5V
0.2
VOLTAGE (mV)
2
1
0
VDD = 3V
–1
–2
0.1
GAIN ERROR
0
OFFSET ERROR
–0.1
–0.2
–3
–0.3
–4
–0.4
–5
–60
–40
–20
0
20
40
60
80
TEMPERATURE (°C)
100
120
140
03162-014
ERROR (mV)
0.2
03162-016
3
03162-017
2
03162-013
–0.70
Figure 14. Gain Error vs. Temperature
–0.5
0.5
0.6
0.7
0.8
0.9
1.0
1.1
VBIAS (V)
1.2
1.3
1.4
Figure 17. Gain and Offset Errors vs. VBIAS Voltage Applied to IOUT2
Rev. G | Page 9 of 24
AD5426/AD5432/AD5443
Data Sheet
0.7
0.5
TA = 25°C
0.4 VREF = 2.5V
VDD = 3V AND 5V
0.6
GAIN ERROR
0.3
0.5
CURRENT (mA)
0.2
VOLTAGE (mV)
TA = 25°C
0.1
OFFSET ERROR
0
–0.1
–0.2
VDD = 5V
0.4
0.3
0.2
–0.3
0.1
–0.4
0.4
0.2
0.8
0.6
1.0
1.2
VBIAS (V)
1.4
1.6
1.8
2.0
03162-018
0
0
Figure 18. Gain and Offset Errors vs. VBIAS Voltage Applied to IOUT2
0
1
2
3
INPUT VOLTAGE (V)
4
5
03162-021
VDD = 3V
–0.5
Figure 21. Supply Current vs. Logic Input Voltage, SYNC (SCLK), DATA = 0
1.6
3
TA = 25°C
VREF = 0V
VDD = 5V
2 AD5443
MAX INL
1.4
1.2
IOUT LEAKAGE (nA)
1
LSB
MAX DNL
0
–1
MIN INL
IOUT1 VDD 5V
1.0
0.8
0.6
0.4
IOUT1 VDD 3V
–2
MIN DNL
1.0
1.5
VBIAS (V)
2.0
2.5
0
–40
03162-019
–3
0.5
80
100
120
Figure 22. IOUT1 Leakage Current vs. Temperature
Figure 19. Linearity vs. VBIAS Voltage Applied to IOUT2
0.50
4
TA = 25°C
= 2.5V
V
3 REF
VDD = 5V
AD5443
2
0.45
0.40
CURRENT (µA)
MAX INL
MIN DNL
–2
ALL 0s
0.30
ALL 1s
0.25
0.20
VDD = 3V
0.15
–3
0.10
MIN INL
–5
0.5
1.0
1.5
VBIAS (V)
ALL 0s
0.05
2.0
03162-020
–4
ALL 1s
0
–60
–40
–20
0
20
40
60
80
TEMPERATURE (°C)
100
Figure 23. Supply Current vs. Temperature
Figure 20. Linearity vs. VBIAS Voltage Applied to IOUT2
Rev. G | Page 10 of 24
120
140
03162-023
0
–1
VDD = 5V
0.35
MAX DNL
1
LSB
60
20
40
TEMPERATURE (°C)
0
–20
03162-022
0.2
Data Sheet
AD5426/AD5432/AD5443
3.5
3
TA = 25°C
AD5443
3.0 LOADING 010101010101
VREF = ±0.15V, AD8038 CC 1pF
VREF = ±2V, AD8038 CC 1pF
0
2.5
1.5
GAIN (dB)
VCC = 5V
1
VREF = ±0.15V, AD8038 CC 1.47pF
–6
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
100M
–9
10k
Figure 24. Supply Current vs. Update Rate
0.060
ALL ON
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
ALL OFF
100
1k
10k
100k
FREQUENCY (Hz)
0.050
OUTPUT VOLTAGE (V)
0.040
TA = 25°C
VDD = 5V
VREF = ±3.5V
CCOMP = 1.8pF
AD8038 AMPLIFIER
1M
10M
100M
0.020
VDD 3V, 0V REF
NRG = 1.877nVs
0x7FF TO 0x800
0.010
0
VDD 5V, 0V REF
NRG = 0.119nVs,
0x800 TO 0x7FF
–0.020
0
OUTPUT VOLTAGE (V)
–0.4
TA = 25°C
VDD = 5V
VREF = ±3.5V
CCOMP = 1.8pF
AD8038 AMPLIFIER
100
200
250
300
TA = 25°C
VREF = 3.5V
AD8038 AMPLIFIER
CCOMP = 1.8pF
AD5443
–1.72
VDD 3V, 3.5V REF
NRG = 1.433nVs
0x7FF TO 0x800
–1.73
VDD 3V, 3.5V REF
NRG = 0.647nVs
0x800 TO 0x7FF
–1.74
–1.75
1k
10k
100k
FREQUENCY (Hz)
1M
10M
100M
03162-026
10
150
TIME (ns)
VDD 5V, 3.5V REF
NRG = 1.184nVs
0x7FF TO 0x800
–1.71
–0.2
100
Figure 28. Midscale Transition VREF = 0 V
0
GAIN (dB)
50
–1.70
1
100M
VDD 3V, 0V REF
NRG = 0.088nVs
0x800 TO 0x7FF
0.030
–0.010
0.2
–0.8
10M
TA = 25°C
VREF = 0V
AD8038 AMPLIFIER
CCOMP = 1.8pF
AD5443
VDD 5V, 0V REF
NRG = 2.049nVs
0x7FF TO 0x800
Figure 25. Reference Multiplying Bandwidth vs. Frequency and Code
–0.6
1M
FREQUENCY (Hz)
Figure 27. Reference Multiplying Bandwidth vs. Frequency and
Compensation Capacitor
03162-025
6
LOADING
0
ZS TO FS
–6
–12
–18
–24
–30
–36
–42
–48
–54
–60
–66
–72
–78
–84
–90
–96
–102
1
10
100k
03162-028
1
03162-024
0
TA = 25°C
VDD = 5V
AD8038 AMPLIFIER
03162-027
VCC = 3V
0.5
GAIN (dB)
VREF = ±2V, AD8038 CC 1.47pF
–3
Figure 26. Reference Multiplying Bandwidth—All 1s Loaded
VDD 5V, 3.5V REF, NRG = 0.364nVs,
0x800 TO 0x7FF
–1.76
0
50
100
150
TIME (ns)
200
250
Figure 29. Midscale Transition VREF = 3.5 V
Rev. G | Page 11 of 24
300
03162-029
IDD (mA)
VREF = ±3.51V, AD8038 CC 1.8pF
2.0
AD5426/AD5432/AD5443
Data Sheet
20
100
TA = 25°C
VDD = 3V
AMPLIFIER
= AD8038
0
MCLK = 200kHz
80
MCLK = 500kHz
MCLK = 1MHz
–40
–60
SFDR (dB)
FULL SCALE
–80
60
40
ZERO SCALE
20
TA = 25°C
VREF = 3.5V
AD8038 AMP
AD5443
–100
1
10
100
1k
10k
FREQUENCY (Hz)
100k
1M
10M
0
03162-030
–120
0
50
Figure 33. Wideband SFDR vs. fOUT Frequency (AD5443)
80
TA = 25°C
VDD = 3V
VREF = 3.5V p-p
–65
40
fOUT (kHz)
Figure 30. Power Supply Rejection vs. Frequency
–60
30
20
10
03162-034
PSRR (dB)
–20
MCLK = 500kHz
MCLK = 1MHz
60
MCLK = 200kHz
SFDR (dB)
THD + N (dB)
–70
–75
40
–80
1
10
100
1k
10k
FREQUENCY (Hz)
100k
1M
03162-031
–90
TA = 25°C
VREF = 3.5V
AD8038 AMP
AD5426
0
0
20
30
40
50
fOUT (kHz)
Figure 34. Wideband SFDR vs. fOUT Frequency (AD5426)
Figure 31. THD and Noise vs. Frequency
1.8
10
03162-035
20
–85
0
TA = 25°C
TA = 25°C
VREF = 3.5V
AD8038 AMPLIFIER
AD5443
–10
1.6
–20
VIH
–30
1.0
VIL
0.8
–40
–50
–60
0.6
–70
0.4
–80
0.2
–90
0
2.5
3.0
3.5
4.0
VOLTAGE (V)
4.5
5.0
5.5
–100
0
50
100
150
200 250 300 350
FREQUENCY (Hz)
400
450
500
Figure 35. Wideband SFDR fOUT = 50 kHz, Update = 1 MHz
Figure 32. Threshold Voltages vs. Supply Voltage
Rev. G | Page 12 of 24
03162-036
SFDR (dB)
1.2
03162-033
THRESHOLD VOLTAGE (V)
1.4
Data Sheet
AD5426/AD5432/AD5443
–20
–20
–30
–40
–40
SFDR (dB)
–30
–50
–60
–50
–60
–70
–70
–80
–80
–90
–100
10
50
100
150
200 250 300 350
FREQUENCY (Hz)
400
450
500
03162-037
–90
–100
0
Figure 36. Wideband SFDR fOUT = 20 kHz, Update = 1 MHz
0
–20
0
–10
–20
–30
–40
–40
–50
–50
dB
–30
–60
–70
–70
–80
–80
–90
–90
30
35
40
45
50
55
FREQUENCY (Hz)
60
65
70
75
03162-038
–60
–100
25
12
14
16
18
20
22
FREQUENCY (Hz)
24
26
28
30
Figure 38. Narrowband (±50%) SFDR fOUT = 20 kHz, Update = 1 MHz
TA = 25°C
VREF = 3.5V
AD8038 AMPLIFIER
AD5443
–10
SFDR (dB)
TA = 25°C
VREF = 3.5V
AD8038 AMPLIFIER
AD5443
–10
03162-039
–10
SFDR (dB)
0
TA = 25°C
VREF = 3.5V
AD8038 AMPLIFIER
AD5443
Figure 37. Narrowband (±50%) SFDR fOUT = 50 kHz, Update = 1 MHz
TA = 25°C
VREF = 3.5V
AD8038 AMPLIFIER
AD5443
–100
10
15
20
25
FREQUENCY (Hz)
30
35
03162-040
0
Figure 39. Narrowband (±50%) IMD fOUT = 20 kHz, 25 kHz, Update = 1 MHz
Rev. G | Page 13 of 24
AD5426/AD5432/AD5443
Data Sheet
TERMINOLOGY
Relative Accuracy
Relative accuracy or endpoint nonlinearity is a measure of the
maximum deviation from a straight line passing through the
endpoints of the DAC transfer function. It is measured after
adjusting for 0 and full scale and is normally expressed in LSBs
or as a percentage of full-scale reading.
Differential Nonlinearity
Differential nonlinearity is the difference between the measured
change and the ideal 1 LSB change between any two adjacent
codes. A specified differential nonlinearity of −1 LSB maximum
over the operating temperature range ensures monotonicity.
Gain Error
Gain error or full-scale error is a measure of the output error
between an ideal DAC and the actual device output. For these
DACs, ideal maximum output is VREF − 1 LSB. Gain error of the
DACs is adjustable to 0 with external resistance.
Output Leakage Current
Output leakage current is current that flows in the DAC ladder
switches when these are turned off. For the IOUT1 terminal, it
can be measured by loading all 0s to the DAC and measuring
the IOUT1 current. Minimum current flows in the IOUT2 line
when the DAC is loaded with all 1s.
Output Capacitance
Capacitance from IOUT1 or IOUT2 to AGND.
Output Current Settling Time
This is the amount of time it takes for the output to settle to a
specified level for a full-scale input change. For these devices, it
is specified with a 100 Ω resistor to ground.
The settling time specification includes the digital delay from
SYNC rising edge to the full-scale output charge.
Digital-to-Analog Glitch Impulse
The amount of charge injected from the digital inputs to the
analog output when the inputs change state. This is normally
specified as the area of the glitch in either pA-s or nV-s
depending upon whether the glitch is measured as a current
or voltage signal.
Digital Feedthrough
When the device is not selected, high frequency logic activity
on the device digital inputs may be capacitively coupled to show
up as noise on the IOUT pins and subsequently into the following
circuitry. This noise is digital feedthrough.
Multiplying Feedthrough Error
This is the error due to capacitive feedthrough from the DAC
reference input to the DAC IOUT1 terminal, when all 0s are
loaded to the DAC.
Total Harmonic Distortion (THD)
The DAC is driven by an ac reference. The ratio of the rms sum
of the harmonics of the DAC output to the fundamental value is
the THD. Usually only the lower order harmonics are included,
such as second to fifth.
THD = 20 log
(V
2
2
+ V3 2 + V4 2 + V5 2 )
V1
Digital Intermodulation Distortion
Second-order intermodulation distortion (IMD) measurements
are the relative magnitude of the fa and fb tones generated
digitally by the DAC and the second-order products at 2fa − fb
and 2fb − fa.
Spurious-Free Dynamic Range (SFDR)
SFDR is the usable dynamic range of a DAC before spurious
noise interferes or distorts the fundamental signal. It is the measure of the difference in amplitude between the fundamental
and the largest harmonically or nonharmonically related spur
from dc to full Nyquist bandwidth (half the DAC sampling rate,
or fS/2). Narrow band SFDR is a measure of SFDR over an
arbitrary window size, in this case 50% of the fundamental.
Digital SFDR is a measure of the usable dynamic range of the
DAC when the signal is a digitally generated sine wave.
Rev. G | Page 14 of 24
Data Sheet
AD5426/AD5432/AD5443
THEORY OF OPERATION
The AD5426, AD5432, and AD5443 are 8-, 10-, and 12-bit
current output DACs consisting of a standard inverting R-2R
ladder configuration. A simplified diagram for the 8-bit AD5426 is
shown in Figure 40. The matching feedback resistor, RFB, has a
value of R. The value of R is typically 10 kΩ (minimum 8 kΩ
and maximum 12 kΩ). If IOUT1 and IOUT2 are kept at the same
potential, a constant current flows in each ladder leg, regardless
of digital input code. Therefore, the input resistance presented
at VREF is always constant and nominally of value R. The DAC
output (IOUT) is code-dependent, producing various resistances
and capacitances. External amplifier choice should take into
account the variation in impedance generated by the DAC on
the amplifiers inverting input node.
R
R
R
These DACs are designed to operate with either negative or
positive reference voltages. The VDD power pin is used by only the
internal digital logic to drive the DAC switches’ on and off states.
These DACs are also designed to accommodate ac reference
input signals in the range of −10 V to +10 V.
With a fixed 10 V reference, the circuit shown in Figure 41 gives
a unipolar 0 V to −10 V output voltage swing. When VIN is an ac
signal, the circuit performs 2-quadrant multiplication. Table 5
shows the relationship between digital code and expected output
voltage for unipolar operation (AD5426, 8-bit device).
Table 5. Unipolar Code Table
2R
2R
2R
2R
S1
S2
S3
S8
Digital Input
1111 1111
1000 0000
0000 0001
0000 0000
2R
DAC DATA LATCHES
AND DRIVERS
R
RFBA
IOUT1
IOUT2
03162-041
VREF
Note that the output voltage polarity is opposite to the VREF
polarity for dc reference voltages.
Analog Output (V)
−VREF (255/256)
−VREF (128/256) = −VREF/2
−VREF (1/256)
−VREF (0/256) = 0
Figure 40. Simplified Ladder
VDD
VDD
R1
C1
RFB
IOUT1
A1
A1
IOUT2
VOUT = 0
TO –VREF
SYNC SCLK SDIN GND
MICROCONTROLLER
AGND
NOTES
1. R1 AND R2 USED ONLY IF GAIN ADJUSTMENT IS REQUIRED.
2. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE REQUIRED
IF A1 IS A HIGH SPEED AMPLIFIER.
CIRCUIT OPERATION
Unipolar Mode
Using a single op amp, these devices can easily be configured to
provide 2-quadrant multiplying operation or a unipolar output
voltage swing, as shown in Figure 41.
When an output amplifier is connected in unipolar mode, the
output voltage is given by
VOUT   VREF 
VREF
VREF
AD5426/
AD5432/
AD5443
R2
D
2n
where D is the fractional representation of the digital word
loaded to the DAC, and n is the number of bits.
D = 0 to 255 (8-bit AD5426)
= 0 to 1023 (10-bit AD5432)
= 0 to 4095 (12-bit AD5443)
Rev. G | Page 15 of 24
Figure 41. Unipolar Operation
03162-042
Access is provided to the VREF, RFB, IOUT1, and IOUT2 terminals of
the DAC, making the device extremely versatile and allowing it
to be configured in several different operating modes. For example,
it can be configured to provide a unipolar output, 4-quadrant
multiplication in bipolar or single-supply modes of operation.
Note that a matching switch is used in series with the internal
RFB feedback resistor. If users attempt to measure RFB, power
must be applied to VDD to achieve continuity.
AD5426/AD5432/AD5443
Data Sheet
Bipolar Operation
Table 6. Bipolar Code Table
In some applications, it may be necessary to generate full
4-quadrant multiplying operation or a bipolar output swing.
This can easily be accomplished by using another external
amplifier and some external resistors, as shown in Figure 42.
In this circuit, the second amplifier, A2, provides a gain of 2.
Biasing the external amplifier with an offset from the reference
voltage results in full 4-quadrant multiplying operation. The
transfer function of this circuit shows that both negative and
positive output voltages are created as the input data, D, which
is incremented from code zero (VOUT = −VREF) to midscale
(VOUT = 0 V) to full scale (VOUT = +VREF).
Digital Input
1111 1111
1000 0000
0000 0001
0000 0000
Analog Output (V)
+VREF (127/128)
0
−VREF (127/128)
−VREF (128/128)
Stability
In the I-to-V configuration, the IOUT of the DAC and the inverting
node of the op amp must be connected as close as possible and
proper PCB layout techniques must be employed. Since every
code change corresponds to a step function, gain peaking may
occur if the op amp has limited gain bandwidth product (GBP)
and there is excessive parasitic capacitance at the inverting node.
This parasitic capacitance introduces a pole into the open-loop
response that can cause ringing or instability in closed-loop
applications.
D
VOUT =  VREF × n − 1  − VREF
2


where D is the fractional representation of the digital word
loaded to the DAC and n is the resolution of the DAC.
An optional compensation capacitor, C1, can be added in parallel
with RFB for stability, as shown in Figure 41 and Figure 42. Too
small a value of C1 can produce ringing at the output, while
too large a value can adversely affect the settling time. C1 should
be found empirically, but 1 pF to 2 pF is generally adequate for
compensation.
D = 0 to 255 (8-bit AD5426)
= 0 to 1023 (10-bit AD5432)
= 0 to 4095 (12-bit AD5443)
When VIN is an ac signal, the circuit performs 4-quadrant
multiplication.
Table 6 shows the relationship between digital code and the
expected output voltage for bipolar operation (AD5426,
8-bit device).
R3
20kΩ
VDD
±10V
R1
VREF
RFB
AD5426/
AD5432/
AD5443
IOUT1
IOUT2
C1
A1
A1
A2
VOUT = –VREF
TO +VREF
SYNC SCLK SDIN GND
MICROCONTROLLER
R4
10kΩ
AGND
NOTES
1. R1 AND R2 ARE USED ONLY IF GAIN ADJUSTMENT IS REQUIRED. ADJUST R1 FOR
VOUT = 0V WITH CODE 10000000 LOADED TO DAC.
2. MATCHING AND TRACKING IS ESSENTIAL FOR RESISTOR PAIRS R3 AND R4.
3. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE REQUIRED IF A1/A2 IS A HIGH
SPEED AMPLIFIER.
Figure 42. Bipolar Operation
Rev. G | Page 16 of 24
03162-043
VDD
VREF
R5
20kΩ
R2
Data Sheet
AD5426/AD5432/AD5443
SINGLE-SUPPLY APPLICATIONS
VDD
R1
R2
Current Mode Operation
These DACs are specified and tested to guarantee operation
in single-supply applications. In the current mode circuit of
Figure 43, IOUT2 and hence IOUT1 is biased positive by an amount
applied to VBIAS.
RFB
VDD
A1
VIN
IOUT1
VOUT
VREF
GND
VDD
IOUT1
VIN
VREF
A1
A1
VOUT
Figure 44. Single-Supply Voltage Switching Mode Operation
IOUT2
It is important to note that VIN is limited to low voltages because
the switches in the DAC ladder no longer have the same source
drain drive voltage. As a result, their on resistance differs, which
degrades the linearity of the DAC.
GND
A2
Also, VIN must not go negative by more than 0.3 V or an
internal diode turns on, exceeding the maximum ratings of the
device. In this type of application, the full range of multiplying
capability of the DAC is lost.
03162-044
VBIAS
NOTES
1. ADDITIONAL PINS OMITTED FOR CLARITY.
2. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE
REQUIRED IF A1 IS A HIGH SPEED AMPLIFIER.
03162-045
NOTES
1. ADDITIONAL PINS OMITTED FOR CLARITY.
2. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE REQUIRED
IF A1 IS A HIGH SPEED AMPLIFIER.
C1
RFB
Figure 43. Single-Supply Current Mode Operation
POSITIVE OUTPUT VOLTAGE
In this configuration, the output voltage is given by
VOUT = {D × (RFB/RDAC) × (VBIAS − VIN)} + VBIAS
As D varies from 0 to 255 (AD5426), 1023 (AD5432) or 4095
(AD5443), the output voltage varies from
VOUT = VBIAS to VOUT = 2 VBIAS − VIN
VBIAS should be a low impedance source capable of sinking and
sourcing all possible variations in current at the IOUT2 terminal
without any problems.
Note that the output voltage polarity is opposite to the VREF
polarity for dc reference voltages. To achieve a positive voltage
output, an applied negative reference to the input of the DAC
is preferred over the output inversion through an inverting
amplifier because of the resistor’s tolerance errors. To generate
a negative reference, the reference can be level shifted by an
op amp such that the VOUT and GND pins of the reference
become the virtual ground and −2.5 V, respectively, as shown
in Figure 45.
It is important to note that VIN is limited to low voltages because
the switches in the DAC ladder no longer have the same source
drain drive voltage. As a result, their on resistance differs, which
degrades the linearity of the DAC. See Figure 15 to Figure 20.
VDD = 5V
ADR03
VOUT VIN
GND
C1
+5V
VDD
Voltage Switching Mode of Operation
RFB
IOUT1
Figure 44 shows these DACs operating in the voltage switching
mode. The reference voltage, VIN, is applied to the IOUT1 pin,
IOUT2 is connected to AGND, and the output voltage is available
at the VREF terminal. In this configuration, a positive reference
voltage results in a positive output voltage, making singlesupply operation possible. The output from the DAC is voltage
at a constant impedance (the DAC ladder resistance), thus an
op amp is necessary to buffer the output voltage. The reference
input no longer sees a constant input impedance, but one that
varies with code. Therefore, the voltage input should be driven
from a low impedance source.
VREF
IOUT2
–2.5V
–5V
GND
A1
VOUT = 0V
TO +2.5V
NOTES
1. ADDITIONAL PINS OMITTED FOR CLARITY.
2. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE REQUIRED IF A1 IS
A HIGH SPEED AMPLIFIER.
Rev. G | Page 17 of 24
Figure 45. Positive Voltage Output with Minimum of Components
03162-046
VDD
AD5426/AD5432/AD5443
Data Sheet
ADDING GAIN
In applications where the output voltage is required to be greater
than VIN, gain can be added with an additional external amplifier or
it can be achieved in a single stage. It is important to consider the
effect of temperature coefficients of the thin film resistors of the
DAC. Simply placing a resistor in series with the RFB resistor causes
mismatches in the temperature coefficients, resulting in larger
gain temperature coefficient errors. Instead, the circuit shown
in Figure 46 is a recommended method of increasing the gain of
the circuit. R1, R2, and R3 should all have similar temperature
coefficients, but they need not match the temperature coefficients
of the DAC. This approach is recommended in circuits where
gains of greater than 1 are required.
DAC leakage current is also a potential error source in divider
circuits. The leakage current must be counterbalanced by an
opposite current supplied from the op amp through the DAC.
Since only a fraction D of the current into the VREF terminal is
routed to the IOUT1 terminal, the output voltage has to change
as follows:
Output Error Voltage due to DAC Leakage = (Leakage × R)/D
where R is the DAC resistance at the VREF terminal. For a DAC
leakage current of 10 nA, R = 10 kΩ, and a gain (that is, 1/D) of 16,
the error voltage is 1.6 mV.
VDD
VIN
RFB
VDD
VDD
IOUT1
VREF
VDD
VIN
R1
RFB
IOUT1
VREF
IOUT2
IOUT2
C1
GND
A1
VOUT
R3
GND
VOUT
GAIN = R2 + R3
R2
R1 = R2R3
ADDITIONAL PINS OMITTED FOR CLARITY.
03162-047
R2 + R3
NOTES
1. ADDITIONAL PINS OMITTED FOR CLARITY.
2. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE REQUIRED
IF A1 IS A HIGH SPEED AMPLIFIER.
Figure 46. Increasing Gain of Current Output DAC
DACS USED AS A DIVIDER OR PROGRAMMABLE
GAIN ELEMENT
Current-steering DACs are very flexible and lend themselves to
many different applications. If this type of DAC is connected as
the feedback element of an op amp and RFB is used as the input
resistor as shown in Figure 47, then the output voltage is inversely
proportional to the digital input fraction, D.
For D = 1 − 2−n the output voltage is
VOUT = −VIN/D = −VIN/(1 − 2−N)
As D is reduced, the output voltage increases. For small values
of D, it is important to ensure that the amplifier does not saturate
and also that the required accuracy is met. For example, an 8-bit
DAC driven with the binary code 0x10 (00010000), that is, 16
decimal, in the circuit of Figure 47 should cause the output
voltage to be 16 × VIN. However, if the DAC has a linearity
specification of ±0.5 LSB, then D can in fact have the weight
anywhere in the range 15.5/256 to 16.5/256 so that the possible
output voltage will be in the range 15.5 VIN to 16.5 VIN—an error of
+3% even though the DAC itself has a maximum error of 0.2%.
03162-048
R2
Figure 47. Current Steering DAC as a Divider or Programmable Gain Element
REFERENCE SELECTION
When selecting a reference for use with the AD5426 series of
current output DACs, pay attention to the references output
voltage temperature coefficient specification. This parameter not
only affects the full-scale error, but can also affect the linearity (INL
and DNL) performance. The reference temperature coefficient
should be consistent with the system accuracy specifications. For
example, an 8-bit system required to hold its overall specification to
within 1 LSB over the temperature range 0°C to 50°C dictates
that the maximum system drift with temperature should be less
than 78 ppm/°C. A 12-bit system with the same temperature
range to overall specification within 2 LSBs requires a maximum
drift of 10 ppm/°C. By choosing a precision reference with low
output temperature coefficient this error source can be minimized.
Table 7 suggests some references available from Analog Devices
that are suitable for use with this range of current output DACs.
AMPLIFIER SELECTION
The primary requirement for the current-steering mode is an
amplifier with low input bias currents and low input offset
voltage. The input offset voltage of an op amp is multiplied by
the variable gain (due to the code-dependent output resistance
of the DAC) of the circuit. A change in this noise gain between
two adjacent digital fractions produces a step change in the
output voltage due to the amplifier’s input offset voltage. This
output voltage change is superimposed on the desired change in
output between the two codes and gives rise to a differential
linearity error, which, if large enough, could cause the DAC to
be nonmonotonic. In general, the input offset voltage should be
Rev. G | Page 18 of 24
Data Sheet
AD5426/AD5432/AD5443
a fraction (~ <1/4) of an LSB to ensure monotonic behavior
when stepping through codes.
The input bias current of an op amp also generates an offset at
the voltage output as a result of the bias current flowing in the
feedback resistor, RFB. Most op amps have input bias currents low
enough to prevent any significant errors in 12-bit applications.
Common-mode rejection of the op amp is important in voltage
switching circuits since it produces a code-dependent error at
the voltage output of the circuit. Most op amps have adequate
common-mode rejection at an 8-, 10-, or 12-bit resolution.
Provided the DAC switches are driven from true wideband low
impedance sources (VIN and AGND), they settle quickly.
Consequently, the slew rate and settling time of a voltage switching
DAC circuit is determined largely by the output op amp.
To obtain minimum settling time in this configuration, it is
important to minimize capacitance at the VREF of the DAC. This
is done by using low input capacitance buffer amplifiers and
careful board design. Most single-supply circuits include ground as
part of the analog signal range, which in turn requires an amplifier
that can handle rail-to-rail signals,. There is a large range of
single-supply amplifiers available from Analog Devices.
Table 7. Suitable ADI Precision References
Part No.
ADR01
ADR01
ADR02
ADR02
ADR03
ADR03
ADR06
ADR06
ADR431
ADR435
ADR391
ADR395
Output Voltage (V)
10
10
5
5
2.5
2.5
3
3
2.5
5
2.5
5
Initial Tolerance (%)
0.05
0.05
0.06
0.06
0.10
0.10
0.10
0.10
0.04
0.04
0.16
0.10
Temp Drift (ppm/°C)
3
9
3
9
3
9
3
9
3
3
9
9
ISS (mA)
1
1
1
1
1
1
1
1
0.8
0.8
0.12
0.12
Output Noise µV p-p
20
20
10
10
6
6
10
10
3.5
8
5
8
Package
SOIC-8
TSOT-23, SC70
SOIC-8
TSOT-23, SC70
SOIC-8
TSOT-23, SC70
SOIC-8
TSOT-23, SC70
SOIC-8
SOIC-8
TSOT-23
TSOT-23
IB (Max) (nA)
0.1
2
0.05
0.001
0.1
0.1 Hz to 10 Hz
Noise (µV p-p)
0.5
0.4
1
2.3
0.5
Supply Current (µA)
600
500
975
50
850
Package
SOIC-8
MSOP, SOIC-8
MSOP, SOIC-8
TSOT
TSOT, SOIC-8
Slew Rate (V/µs)
180
100
425
1,300
VOS (Max) (µV)
1,500
1,000
3,000
10,000
IB (Max) (nA)
6,000
10,500
750
7,000
Table 8. Suitable ADI Precision Op Amps
Part No.
OP97
OP1177
AD8551
AD8603
AD8628
Supply Voltage (V)
±2 to ±20
±2.5 to ±15
2.7 to 5
1.8 to 6
2.7 to 6
VOS (Max) (µV)
25
60
5
50
5
Table 9. Suitable ADI High Speed Op Amps
Part No.
AD8065
AD8021
AD8038
AD9631
Supply Voltage (V)
5 to 24
±2.5 to ±12
3 to 12
±2 to ±6
BW @ ACL (MHz)
145
490
350
320
Rev. G | Page 19 of 24
Package
SOIC-8, SOT-23, MSOP
SOIC-8, MSOP
SOIC-8, SC70-5
SOIC-8
AD5426/AD5432/AD5443
Data Sheet
Low Power Serial Interface
DAC Control Bits C3 to C0
Control Bits C3 to C0 allow control of various functions of
the DAC, as seen in Table 10. Default settings of the DAC on
power-on are as follows: Data is clocked into the shift register
on falling clock edges and daisy-chain mode is enabled.
Device powers on with zero-scale load to the DAC register
and IOUT lines.
The DAC control bits allow the user to adjust certain features
on power-on, for example, daisy-chaining may be disabled if
not in use, active clock edge may be changed to rising edge, and
DAC output may be cleared to either zero scale or midscale.
The user may also initiate a readback of the DAC register
contents for verification purposes.
Table 10. DAC Control Bits
C2
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
C1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
C0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
DB0 (LSB)
C1
C0
DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0
Function Implemented
No operation (power-on default)
Load and update
Initiate readback
Reserved
Reserved
Reserved
Reserved
Reserved
Reserved
Daisy-chain disable
Clock data to shift register on rising edge
Clear DAC output to zero scale
Clear DAC output to midscale
Reserved
Reserved
Reserved
X
X
X
X
DATA BITS
CONTROL BITS
Figure 48. AD5426 8-Bit Input Shift Register Contents
DB15 (MSB)
C3
C2
DB0 (LSB)
C1
C0
DB9 DB8 DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0
CONTROL BITS
X
X
DATA BITS
Figure 49. AD5432 10-Bit Input Shift Register Contents
DB15 (MSB)
C3
C2
DB0 (LSB)
C1
C0 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0
CONTROL BITS
To minimize the power consumption of the device, the interface
powers up fully only when the device is being written to, that is,
on the falling edge of SYNC. The SCLK and DIN input buffers
are powered down on the rising edge of SYNC. The SYNC of
the AD5426/AD5432/AD5443 needs to be synchronous with
the microprocessor control. Unfinished data frames are latched
into the part and will affect the output.
C3
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
C2
03162-050
The AD5426/AD5432/AD5443 have an easy to use 3-wire interface that is compatible with SPI/QSPI/MICROWIRE and DSP
interface standards. Data is written to the device in 16 bit words.
This 16-bit word consists of 4 control bits and either 8 , 10 , or 12
data bits as shown in Figure 48, Figure 49, and Figure 50. The
AD5443 uses all 12 bits of DAC data. The AD5432 uses 10 bits
and ignores the 2 LSBs, while the AD5426 uses 8 bits and ignores
the last 4 bits.
C3
03162-049
DB15 (MSB)
DATA BITS
03162-051
SERIAL INTERFACE
Figure 50. AD5443 12-Bit Input Shift Register Contents
SYNC Function
SYNC is an edge-triggered input that acts as a frame synchronization signal and chip enable. Data can be transferred into the
device only while SYNC is low. To start the serial data transfer,
SYNC should be taken low observing the minimum SYNC
falling to SCLK falling edge setup time, t4.
Daisy-Chain Mode
Daisy-chain is the default power-on mode. Note that the SDO
line operates with a VDD of 3.0 V to 5.5 V. To disable the daisy
chain function, write 1001 to the control word. In daisy-chain
mode, the internal gating on SCLK is disabled. The SCLK is
continuously applied to the input shift register when SYNC is
low. If more than 16 clock pulses are applied, the data ripples
out of the shift register and appears on the SDO line. This data
is clocked out on the rising edge of SCLK (this is the default, use
the control word to change the active edge) and is valid for the
next device on the falling edge (default). By connecting this line
to the DIN input on the next device in the chain, a multidevice
interface is constructed. Sixteen clock pulses are required for
each device in the system. Therefore, the total number of clock
cycles must equal 16 N where N is the total number of devices
in the chain. See the timing diagram in Figure 4.
When the serial transfer to all devices is complete, SYNC
should be taken high. This prevents any further data being
clocked into the input shift register. A burst clock containing
the exact number of clock cycles may be used and SYNC taken
high some time later. After the rising edge of SYNC, data is
automatically transferred from each device’s input shift register
to the addressed DAC.
When control bits = 0000, the device is in no operation mode.
This may be useful in daisy-chain applications where the user
does not want to change the settings of a particular DAC in the
chain. Simply write 0000 to the control bits for that DAC and
the following data bits will be ignored. To re-enable the daisychain mode, if disabled, a power recycle is required.
Rev. G | Page 20 of 24
Data Sheet
AD5426/AD5432/AD5443
Standalone Mode
After power-on, write 1001 to the control word to disable daisychain mode. The first falling edge of SYNC resets a counter that
counts the number of serial clocks, ensuring the correct number
of bits are shifted in and out of the serial shift registers. A rising
edge on SYNC during a write causes the write cycle to be aborted.
After the falling edge of the 16th SCLK pulse, data is automatically transferred from the input shift register to the DAC.
For another serial transfer to take place, the counter must be
reset by the falling edge of SYNC.
Rev. G | Page 21 of 24
AD5426/AD5432/AD5443
Data Sheet
PCB LAYOUT AND POWER SUPPLY DECOUPLING
In any circuit where accuracy is important, careful consideration of
the power supply and ground return layout helps to ensure the
rated performance. The printed circuit board on which the
AD5426/AD5432/AD5443 is mounted should be designed so
that the analog and digital sections are separated and confined
to certain areas of the board. If the DAC is in a system where
multiple devices require an AGND-to-DGND connection, the
connection should be made at one point only. The star ground
point should be established as close to the device as possible.
The DAC should have ample supply bypassing of 10 µF in parallel
with 0.1 µF on the supply located as close to the package as
possible, ideally right up against the device. The 0.1 µF capacitor
should have low effective series resistance (ESR) and effective
series inductance (ESI), like the common ceramic types that
provide a low impedance path to ground at high frequencies to
handle transient currents due to internal logic switching. Low
ESR, 1 µF to 10 µF tantalum or electrolytic capacitors should
also be applied at the supplies to minimize transient disturbance
and filter out low frequency ripple.
Fast switching signals such as clocks should be shielded with
digital ground to avoid radiating noise to other parts of the
board and should never be run near the reference inputs.
Avoid crossover of digital and analog signals. Traces on opposite
sides of the board should run at right angles to each other. This
reduces the effects of feedthrough through the board. A microstrip technique is by far the best, but not always possible with a
double-sided board. In this technique, the component side of the
board is dedicated to ground plane while signal traces are placed
on the solder side.
It is good practice to employ compact, minimum lead length
PCB layout design. Leads to the input should be as short as
possible to minimize IR drops and stray inductance.
The PCB metal traces between VREF and RFB should also be
matched to minimize gain error. To maximize on high frequency
performance, the I-to-V amplifier should be located as close to
the device as possible.
Rev. G | Page 22 of 24
Data Sheet
AD5426/AD5432/AD5443
OVERVIEW OF AD54xx AND AD55xx DEVICES
Table 11.
Part No.
AD5424
AD5426
AD5428
AD5429
AD5450
AD5432
AD5433
AD5439
AD5440
AD5451
AD5443
AD5444
AD5415
AD5405
AD5445
AD5447
AD5449
AD5452
AD5446
AD5453
AD5553
AD5556
AD5555
AD5557
AD5543
AD5546
AD5545
AD5547
Resolution
8
8
8
8
8
10
10
10
10
10
12
12
12
12
12
12
12
12
14
14
14
14
14
14
16
16
16
16
No. DACs
1
1
2
2
1
1
1
2
2
1
1
1
2
2
2
2
2
1
1
1
1
1
2
2
1
1
2
2
INL (LSB)
±0.25
±0.25
±0.25
±0.25
±0.25
±0.5
±0.5
±0.5
±0.5
±0.25
±1
±0.5
±1
±1
±1
±1
±1
±0.5
±1
±2
±1
±1
±1
±1
±2
±2
±2
±2
Interface
Parallel
Serial
Parallel
Serial
Serial
Serial
Parallel
Serial
Parallel
Serial
Serial
Serial
Serial
Parallel
Parallel
Parallel
Serial
Serial
Serial
Serial
Serial
Parallel
Serial
Parallel
Serial
Parallel
Serial
Parallel
Package
RU-16, CP-20
RM-10
RU-20
RU-10
RJ-8
RM-10
RU-20, CP-20
RU-16
RU-24
RJ-8
RM-10
RM-8
RU-24
CP-40
RU-20, CP-20
RU-24
RU-16
RJ-8, RM-8
RM-8
UJ-8, RM-8
RM-8
RU-28
RM-8
RU-38
RM-8
RU-28
RU-16
RU-38
Rev. G | Page 23 of 24
Features
10 MHz BW, 17 ns CS pulse width
10 MHz BW, 50 MHz serial
10 MHz BW, 17 ns CS pulse width
10 MHz BW, 50 MHz serial
10 MHz BW, 50 MHz serial
10 MHz BW, 50 MHz serial
10 MHz BW, 17 ns CS pulse width
10 MHz BW, 50 MHz serial
10 MHz BW, 17 ns CS pulse width
10 MHz BW, 50 MHz serial
10 MHz BW, 50 MHz serial
50 MHz serial interface
10 MHz BW, 50 MHz serial
10 MHz BW, 17 ns CS pulse width
10 MHz BW, 17 ns CS pulse width
10 MHz BW, 17 ns CS pulse width
10 MHz BW, 50 MHz serial
10 MHz BW, 50 MHz serial
10 MHz BW, 50 MHz serial
10 MHz BW, 50 MHz serial
4 MHz BW, 50 MHz serial clock
4 MHz BW, 20 ns WR pulse width
4 MHz BW, 50 MHz serial clock
4 MHz BW, 20 ns WR pulse width
4 MHz BW, 50 MHz serial clock
4 MHz BW, 20 ns WR pulse width
4 MHz BW, 50 MHz serial clock
4 MHz BW, 20 ns WR pulse width
AD5426/AD5432/AD5443
Data Sheet
OUTLINE DIMENSIONS
3.10
3.00
2.90
10
3.10
3.00
2.90
1
5.15
4.90
4.65
6
5
PIN 1
IDENTIFIER
0.50 BSC
0.95
0.85
0.75
15° MAX
1.10 MAX
0.30
0.15
6°
0°
0.70
0.55
0.40
0.23
0.13
COMPLIANT TO JEDEC STANDARDS MO-187-BA
091709-A
0.15
0.05
COPLANARITY
0.10
Figure 51. 10-Lead Mini Small Outline Package [MSOP]
(RM-10)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1
AD5426YRM
AD5426YRM-REEL
AD5426YRM-REEL7
AD5426YRMZ
AD5426YRMZ-REEL
AD5426YRMZ-REEL7
AD5432YRM
AD5432YRMZ
AD5432YRMZ-REEL
AD5432YRMZ-REEL7
AD5443YRM
AD5443YRM-REEL
AD5443YRM-REEL7
AD5443YRMZ
AD5443YRMZ-REEL
AD5443YRMZ-REEL7
EV-AD5443/46/53SDZ
1
Resolution (Bit)
8
8
8
8
8
8
10
10
10
10
12
12
12
12
12
12
INL (LSB)
±0.25
±0.25
±0.25
±0.25
±0.25
±0.25
±0.5
±0.5
±0.5
±0.5
±1
±1
±1
±1
±1
±1
Temperature Range
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
Z = RoHS Compliant Part, # denotes RoHS compliant product may be top or bottom marked.
©2004–2013 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D03162-0-6/13(G)
Rev. G | Page 24 of 24
Package Description
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
Evaluation Board
Package Option
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
RM-10
Branding
D1Q
D1Q
D1Q
D6W
D6W
D6W
D1R
D1R#
D1R#
D1R#
D1S
D1S
D1S
D1S#
D1S#
D1S#