ISL6520 ® Data Sheet October 4, 2005 Single Synchronous Buck Pulse-Width Modulation (PWM) Controller Features • Operates from +5V Input The ISL6520 makes simple work out of implementing a complete control and protection scheme for a DC/DC stepdown converter. Designed to drive N-channel MOSFETs in a synchronous buck topology, the ISL6520 integrates the control, output adjustment, monitoring and protection functions into a single 8-pin package. The ISL6520 provides simple, single feedback loop, voltagemode control with fast transient response. The output voltage can be precisely regulated to as low as 0.8V, with a maximum tolerance of ±1.5% over temperature and line voltage variations. A fixed frequency oscillator reduces design complexity, while balancing typical application cost and efficiency. The error amplifier features a 15MHz gain-bandwidth product and 8V/µs slew rate which enables high converter bandwidth for fast transient performance. The resulting PWM duty cycles range from 0% to 100%. Protection from over-current conditions is provided by monitoring the rDS(ON) of the upper MOSFET to inhibit PWM operation appropriately. This approach simplifies the implementation and improves efficiency by eliminating the need for a current sense resistor. Ordering Information PART NUMBER ISL6520CB PART TEMP. MARKING RANGE (°C) 6520CB ISL6520CBZ 6520CBZ (Note) ISL6520IB 6520IB ISL6520IBZ 6520IBZ (Note) PKG. DWG. # PACKAGE FN9009.4 0 to 70 8 Ld SOIC M8.15 0 to 70 8 Ld SOIC (Pb-free) M8.15 -40 to 85 8 Ld SOIC M8.15 -40 to 85 8 Ld SOIC (Pb-free) M8.15 • 0.8V to VIN Output Range - 0.8V Internal Reference - ±1.5% Over Line Voltage and Temperature • Drives N-Channel MOSFETs • Simple Single-Loop Control Design - Voltage-Mode PWM Control • Fast Transient Response - High-Bandwidth Error Amplifier - Full 0% to 100% Duty Cycle • Lossless, Programmable Over-Current Protection - Uses Upper MOSFET’s rDS(on) • Small Converter Size - 300kHz Fixed Frequency Oscillator - Internal Soft Start - 8 Ld SOIC or 16Ld 4x4mm QFN • QFN Package: - Compliant to JEDEC PUB95 MO-220 QFN - Quad Flat No Leads - Package Outline - Near Chip Scale Package footprint, which improves PCB efficiency and has a thinner profile • Pb-Free Plus Anneal Available (RoHS Compliant) Applications • Power Supplies for Microprocessors - PCs - Embedded Controllers • Subsystem Power Supplies - PCI/AGP/GTL+ Buses - ACPI Power Control • Cable Modems, Set Top Boxes, and DSL Modems ISL6520CR ISL6520CR 0 to 70 16 Ld 4x4mm QFN L16.4x4 • DSP and Core Communications Processor Supplies ISL6520IR ISL6520IR -40 to 85 16 Ld 4x4mm QFN L16.4x4 • Memory Supplies ISL6520EVAL1 Evaluation Board • Personal Computer Peripherals * Add “-T” suffix for tape and reel. NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 1 • Industrial Power Supplies • 5V-Input DC/DC Regulators • Low-Voltage Distributed Power Supplies CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2003, 2005. All Rights Reserved All other trademarks mentioned are the property of their respective owners. ISL6520 Pinouts 7 COMP/SD 16 15 14 13 6 FB GND 3 BOOT 1 12 NC UGATE 2 5 VCC LGATE 4 NC NC 8 PHASE BOOT 1 UGATE 2 PHASE QFN TOP VIEW NC SOIC TOP VIEW 11 COMP/OCSET GND GND 3 10 NC NC 4 6 7 8 LGATE NC VCC NC 9 FB 5 Block Diagram VCC POR AND SOFTSTART + SAMPLE AND HOLD - BOOT OC COMPARATOR UGATE + 0.8V PWM COMPARATOR ERROR AMP + - INHIBIT PHASE GATE CONTROL LOGIC PWM + - - VCC FB LGATE COMP/OCSET 20µA OSCILLATOR FIXED 300kHz GND Typical Application VCC CBULK CDCPL CHF DBOOT VCC ROCSET 5 1 ISL6520 COMP/OCSET 2 7 8 RF CI CF 6 FB 4 3 BOOT CBOOT UGATE LOUT PHASE LGATE +VO COUT GND ROFFSET RS 2 FN9009.4 October 4, 2005 ISL6520 Absolute Maximum Ratings Thermal Information Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +6.0V Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . +15.0V Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . . . . +6.0V Input, Output or I/O Voltage . . . . . . . . . . . GND -0.3V to VCC +0.3V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2 Thermal Resistance Recommended Operating Conditions Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±10% Ambient Temperature Range - ISL6520C . . . . . . . . . . . 0oC to 70oC Ambient Temperature Range - ISL6520I . . . . . . . . . . -40oC to 85oC Junction Temperature Range. . . . . . . . . . . . . . . . . . -40oC to 125oC θJA (oC/W) θJC (oC/W) SOIC Package (Note 1) . . . . . . . . . . . . . . 95 . . N/A QFN Package (Note 2, 3). . . . . . . . . . . . . . 45 . . 7 Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150oC Maximum Storage Temperature Range . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300oC (SOIC - Lead Tips Only) CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTES: 1. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 2. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379. 3. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS 2.6 3.2 3.8 mA 4.19 4.30 4.5 V - 0.25 - V VCC SUPPLY CURRENT Nominal Supply IVCC UGATE and LGATE Open POWER-ON RESET Rising VCC POR Threshold POR VCC POR Threshold Hysteresis OSCILLATOR Frequency fOSC ISL6520C, VCC = 5V 250 300 340 kHz ISL6520I, VCC = 5V 230 300 340 kHz - 1.5 - VP-P ISL6520C -1.5 - +1.5 % ISL6520I -2.5 +2.5 % - V ∆VOSC Ramp Amplitude REFERENCE Reference Voltage Tolerance Nominal Reference Voltage - VREF 0.800 ERROR AMPLIFIER DC Gain - 88 - dB GBWP Guaranteed By Design - 15 - MHz SR - 8 - V/µs Upper Gate Source Current IUGATE-SRC - -1 - A Upper Gate Sink Current IUGATE-SNK - 1 - A Lower Gate Source Current ILGATE-SRC - -1 - A Lower Gate Sink Current ILGATE-SNK - 2 - A ISL6520C 17 20 22 µA ISL6520I 14 20 24 µA - 0.8 - V Gain-Bandwidth Product Slew Rate GATE DRIVERS PROTECTION / DISABLE OCSET Current Source IOCSET Disable Threshold VDISABLE 3 FN9009.4 October 4, 2005 ISL6520 Functional Pin Description An over-current trip cycles the soft-start function. VCC During soft-start, and all the time during normal converter operation, this pin represents the output of the error amplifier. Use this pin, in combination with the FB pin, to compensate the voltage-control feedback loop of the converter. This is the main bias supply for the ISL6520, as well as the lower MOSFET’s gate. Connect a well-decoupled 5V supply to this pin. FB This pin is the inverting input of the internal error amplifier. Use this pin, in combination with the COMP/OCSET pin, to compensate the voltage-control feedback loop of the converter. GND This pin represents the signal and power ground for the IC. Tie this pin to the ground island/plane through the lowest impedance connection available. PHASE Connect this pin to the upper MOSFET source. This pin is used to monitor the voltage drop across the upper MOSFET for over-current protection. This pin is also monitored by the continuously adaptive shoot-through protection circuitry to determine when the upper MOSFET has turned off. UGATE Connect this pin to the upper MOSFET’s gate. This pin provides the PWM-controlled gate drive for the upper MOSFET. This pin is also monitored by the adaptive shootthrough protection circuitry to determine when the upper MOSFET has turned off. Do not insert any circuitry between this pin and the gate of the upper MOSFET, as it may interfere with the internal adaptive shoot-through protection circuitry and render it ineffective. BOOT This pin provides ground referenced bias voltage to the upper MOSFET driver. A bootstrap circuit is used to create a voltage suitable to drive a logic-level N-channel MOSFET. COMP/OCSET This is a multiplexed pin. During a short period of time following power-on reset (POR), this pin is used to determine the overcurrent threshold of the converter. Connect a resistor (ROCSET) from this pin to the drain of the upper MOSFET (VCC). ROCSET, an internal 20µA current source (IOCSET), and the upper MOSFET on-resistance (rDS(ON)) set the converter overcurrent (OC) trip point according to the following equation: I OCSET xR OCSET I PEAK = -----------------------------------------------r DS ( ON ) Internal circuitry of the ISL6520 will not recognize a voltage drop across ROCSET larger than 0.5V. Any voltage drop across ROCSET that is greater than 0.5V will set the overcurrent trip point to: 0.5V I PEAK = ---------------------r DS ( ON ) 4 Pulling OCSET to a level below 0.8V will disable the controller. Disabling the ISL6520 causes the oscillator to stop, the LGATE and UGATE outputs to be held low, and the softstart circuitry to re-arm. LGATE Connect this pin to the lower MOSFET’s gate. This pin provides the PWM-controlled gate drive for the lower MOSFET. This pin is also monitored by the adaptive shoot-through protection circuitry to determine when the lower MOSFET has turned off. Do not insert any circuitry between this pin and the gate of the lower MOSFET, as it may interfere with the internal adaptive shoot-through protection circuitry and render it ineffective. Functional Description Initialization The ISL6520 automatically initializes upon receipt of power. The Power-On Reset (POR) function continually monitors the bias voltage at the VCC pin. The POR function initiates the Over-Current Protection (OCP) sampling and hold operation after the supply voltage exceeds its POR threshold. Upon completion of the OCP sampling and hold operation, the POR function initiates the Soft Start operation. Over Current Protection The over-current function protects the converter from a shorted output by using the upper MOSFET’s on-resistance, rDS(ON), to monitor the current. This method enhances the converter’s efficiency and reduces cost by eliminating a current sensing resistor. The over-current function cycles the soft-start function in a hiccup mode to provide fault protection. A resistor (ROCSET) programs the over-current trip level (see Typical Application diagram). Immediately following POR, the ISL6520 initiates the OverCurrent Protection sampling and hold operation. First, the internal error amplifier is disabled. This allows an internal 20µA current sink to develop a voltage across ROCSET. The ISL6520 then samples this voltage at the COMP pin. This sampled voltage, which is referenced to the VCC pin, is held internally as the Over-Current Set Point. When the voltage across the upper MOSFET, which is also referenced to the VCC pin, exceeds the Over-Current Set Point, the over-current function initiates a soft-start sequence. Figure 1 shows the inductor current after a fault is introduced while running at 15A. The continuous fault causes the ISL6520 to go into a hiccup mode with a typical period of 25ms. The inductor current increases to 18A during the Soft FN9009.4 October 4, 2005 ISL6520 Start interval and causes an over-current trip. The converter dissipates very little power with this method. The measured input power for the conditions of Figure 1 is only 1.5W. increasing width that charge the output capacitor(s). When the internally generated Soft Start voltage exceeds the feedback (FB pin) voltage, the output voltage is in regulation. This method provides a rapid and controlled output voltage rise. The entire startup sequence typically take about 11ms. OUTPUT INDUCTOR CURRENT 5A/DIV. VOUT 500mV/DIV. COMP/OCSET 1V/DIV. TIME (5ms/DIV.) FIGURE 1. OVERCURRENT OPERATION TIME (2ms/DIV.) The over-current function will trip at a peak inductor current (IPEAK) determined by: I OCSET x R OCSET I PEAK = ---------------------------------------------------r DS ( ON ) Application Guidelines 1. The maximum rDS(ON) at the highest junction temperature. 2. The minimum IOCSET from the specification table. Layout Considerations As in any high frequency switching converter, layout is very important. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. The critical components should be located as close together as possible, using ground plane construction or single point grounding. ( ∆I ) I PEAK > I OUT ( MAX ) + ---------- , 2 where ∆I is the output inductor ripple current. For an equation for the ripple current see the section under component guidelines titled ‘Output Inductor Selection’. VIN ISL6520 UGATE Q1 PHASE Soft Start The POR function initiates the soft start sequence after the overcurrent set point has been sampled. Soft start clamps the error amplifier output (COMP pin) and reference input (noninverting terminal of the error amp) to the internally generated Soft Start voltage. Figure 2 shows a typical start up interval where the COMP/OCSET pin has been released from a grounded (system shutdown) state. Initially, the COMP/OCSET is used to sample the oversurrent setpoint by disabling the error amplifier and drawing 20µA through ROCSET. Once the overcurrent level has been sampled, the soft start function is initiated. The clamp on the error amplifier (COMP/OCSET pin) initially controls the converter’s output voltage during soft start. The oscillator’s triangular waveform is compared to the ramping error amplifier voltage. This generates PHASE pulses of 5 LGATE Q2 LO CIN VOUT LOAD where IOCSET is the internal OCSET current source (20µA typical). The OC trip point varies mainly due to the MOSFET’s rDS(ON) variations. To avoid over-current tripping in the normal operating load range, find the ROCSET resistor from the equation above with: 3. Determine IPEAK for FIGURE 2. START UP SEQUENCE CO RETURN FIGURE 3. PRINTED CIRCUIT BOARD POWER AND GROUND PLANES OR ISLANDS Figure 3 shows the critical power components of the converter. To minimize the voltage overshoot, the interconnecting wires indicated by heavy lines should be part of a ground or power plane in a printed circuit board. The components shown in Figure 3 should be located as close together as possible. FN9009.4 October 4, 2005 ISL6520 Please note that the capacitors CIN and CO may each represent numerous physical capacitors. Locate the ISL6520 within 3 inches of the MOSFETs, Q1 and Q2 . The circuit traces for the MOSFETs’ gate and source connections from the ISL6520 must be sized to handle up to 1A peak current. Figure 4 shows the circuit traces that require additional layout consideration. Use single point and ground plane construction for the circuits shown. Minimize any leakage current paths on the COMP/OCSET pin and locate the resistor, ROSCET close to the COMP/OCSET pin because the internal current source is only 20µA. Provide local VCC decoupling between VCC and GND pins. Locate the capacitor, CBOOT as close as practical to the BOOT and PHASE pins. All components used for feedback compensation should be located as close to the IC a practical. BOOT D1 CBOOT ISL6520 Q1 5. Place 2ND Pole at Half the Switching Frequency. 6. Check Gain against Error Amplifier’s Open-Loop Gain. 7. Estimate Phase Margin - Repeat if Necessary. VIN DRIVER OSC PWM COMPARATOR LO - ∆VOSC DRIVER + PHASE VOUT CO ESR (PARASITIC) - LO VOUT +5V 4. Place 1ST Pole at the ESR Zero. VE/A PHASE VCC 3. Place 2ND Zero at Filter’s Double Pole. ZFB +VIN Q2 CO LOAD ROCSET +5V 2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC). ZIN + ERROR AMP REFERENCE DETAILED COMPENSATION COMPONENTS COMP/OCSET ZFB C2 CVCC GND C1 C3 R2 R3 R1 COMP FIGURE 4. PRINTED CIRCUIT BOARD SMALL SIGNAL LAYOUT GUIDELINES VOUT ZIN FB - + ISL6520 Feedback Compensation REFERENCE Figure 5 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage (VOUT) is regulated to the Reference voltage level. The error amplifier (Error Amp) output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulsewidth modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). Modulator Break Frequency Equations 1 F LC = ------------------------------------------2π x L O x C O 1 F ESR = -------------------------------------------2π x ESR x C O The compensation network consists of the error amplifier (internal to the ISL6520) and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. The equations below relate the compensation network’s poles, zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in Figure 7. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick Gain (R2/R1) for desired converter bandwidth. 6 FIGURE 5. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN The modulator transfer function is the small-signal transfer function of VOUT/VE/A . This function is dominated by a DC Gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR . The DC Gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage ∆VOSC . Compensation Break Frequency Equations 1 F Z1 = -----------------------------------2π x R 2 x C 1 1 F P1 = -------------------------------------------------------- C 1 x C 2 2π x R 2 x ---------------------- C1 + C2 1 F Z2 = ------------------------------------------------------2π x ( R 1 + R 3 ) x C 3 1 F P2 = -----------------------------------2π x R 3 x C 3 Figure 6 shows an asymptotic plot of the DC/DC converter’s gain vs frequency. The actual Modulator Gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 6. Using the above guidelines should give a Compensation Gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. FN9009.4 October 4, 2005 ISL6520 Check the compensation gain at FP2 with the capabilities of the error amplifier. The Closed Loop Gain is constructed on the graph of Figure 6 by adding the Modulator Gain (in dB) to the Compensation Gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. 100 FZ1 FZ2 FP1 FP2 Output Inductor Selection 80 OPEN LOOP ERROR AMP GAIN GAIN (dB) 60 40 20 20LOG (R2/R1) 0 20LOG (VIN/DVOSC) MODULATOR GAIN -20 COMPENSATION GAIN -40 -60 Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the Equivalent Series Inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select a suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. CLOSED LOOP GAIN FLC 10 100 1K The output inductor is selected to meet the output voltage ripple requirements and minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations: ∆I = VIN - VOUT FESR 10K 100K 1M 10M FREQUENCY (Hz) FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN Component Selection Guidelines Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern components and loads are capable of producing transient load rates above 1A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (Effective Series Resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. 7 Fs x L x VOUT VIN ∆VOUT = ∆I x ESR Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the ISL6520 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: tRISE = L x ITRAN VIN - VOUT tFALL = L x ITRAN VOUT where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. The worst case response time can be either at the application or removal of load. Be sure to check both of these equations at the FN9009.4 October 4, 2005 ISL6520 minimum and maximum output levels for the worst case response time. Input Capacitor Selection Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the source of Q2 . The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. to package thermal-resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. PUPPER = Io2 x rDS(ON) x D + PLOWER = Io2 x rDS(ON) x (1 - D) Where: D is the duty cycle = VOUT / VIN , tSW is the switching interval, and FS is the switching frequency. Given the reduced available gate bias voltage (5V), logic-level or sub-logic-level transistors should be used for both N-MOSFETs. Caution should be exercised with devices exhibiting very low VGS(ON) characteristics. The shoot-through protection present aboard the ISL6520 may be circumvented by these MOSFETs if they have large parasitic impedences and/or capacitances that would inhibit the gate of the MOSFET from being discharged below its threshold level before the complementary MOSFET is turned on. For a through hole design, several electrolytic capacitors may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. Some capacitor series available from reputable manufacturers are surge current tested. +5V VCC DBOOT BOOT CBOOT ISL6520 MOSFET Selection/Considerations The ISL6520 requires two N-Channel power MOSFETs. These should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. PHASE 8 +5V + VD - UGATE In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. The conduction losses are the largest component of power dissipation for both the upper and the lower MOSFETs. These losses are distributed between the two MOSFETs according to duty factor (see the equations below). Only the upper MOSFET has switching losses, since the lower MOSFETs body diode or an external Schottky rectifier across the lower MOSFET clamps the switching node before the synchronous rectifier turns on. These equations assume linear voltage-current transitions and do not adequately model power loss due the reverse-recovery of the lower MOSFET’s body diode. The gate-charge losses are dissipated by the ISL6520 and don't heat the MOSFETs. However, large gate-charge increases the switching interval, tSW which increases the upper MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according 1 Io x V x t IN SW x FS 2 - + LGATE Q1 NOTE: VG-S ≈ VCC -VD Q2 NOTE: VG-S ≈ VCC GND FIGURE 7. UPPER GATE DRIVE BOOTSTRAP Figure 7 shows the upper gate drive (BOOT pin) supplied by a bootstrap circuit from VCC . The boot capacitor, CBOOT, develops a floating supply voltage referenced to the PHASE pin. The supply is refreshed to a voltage of VCC less the boot diode drop (VD) each time the lower MOSFET, Q2, turns on. FN9009.4 October 4, 2005 ISL6520 ISL6520 DC/DC Converter Application Circuit Figure 8 shows an application circuit of a DC/DC Converter. Detailed information on the circuit, including a complete Bill- of-Materials and circuit board description, can be found in Application Note AN9932. +5V + CIN 2 x 330µF 0.1µF 2 x 1µF VCC 5 ISL6520 6.19kΩ D1 MONITOR AND PROTECTION 1 2 UGATE COMP/OCSET 7 REF 10.0kΩ 8 PHASE 0.1µF Q1 L1 + 470pF - 8200pF 4 + - FB 6 OSC 1.00kΩ BOOT U1 VOUT LGATE Q2 3 + COUT 3 x 330µF 0.1µF GND 3.16kΩ 60.4Ω 18000pF Component Selection Notes: CIN - Each 330mF 6.3WVDC, Sanyo 6TPB330M or Equivalent. COUT - Each 330mF 6.3WVDC, Sanyo 6TPB330M or Equivalent. D1 - 30mA Schottky Diode, MA732 or Equivalent L1 - 3.1µH Inductor, Panasonic P/N ETQ-P6F2ROLFA or Equivalent. Q1 , Q2 - Intersil MOSFET; HUF76143. FIGURE 8. 5V to 3.3V 15A DC/DC CONVERTER 9 FN9009.4 October 4, 2005 ISL6520 Small Outline Plastic Packages (SOIC) M8.15 (JEDEC MS-012-AA ISSUE C) N INDEX AREA 8 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE H 0.25(0.010) M B M INCHES E SYMBOL -B1 2 3 L SEATING PLANE -A- A D h x 45° -C- e A1 B 0.25(0.010) M C 0.10(0.004) C A M MIN MAX MIN MAX NOTES A 0.0532 0.0688 1.35 1.75 - A1 0.0040 0.0098 0.10 0.25 - B 0.013 0.020 0.33 0.51 9 C 0.0075 0.0098 0.19 0.25 - D 0.1890 0.1968 4.80 5.00 3 E 0.1497 0.1574 3.80 4.00 4 e α B S 0.050 BSC - 0.2284 0.2440 5.80 6.20 - h 0.0099 0.0196 0.25 0.50 5 L 0.016 0.050 0.40 1.27 6 α 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. 1.27 BSC H N NOTES: MILLIMETERS 8 0° 8 8° 0° 7 8° Rev. 1 6/05 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. 10 FN9009.4 October 4, 2005 ISL6520 Quad Flat No-Lead Plastic Package (QFN) Micro Lead Frame Plastic Package (MLFP) L16.4x4 16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE (COMPLIANT TO JEDEC MO-220-VGGC ISSUE C) MILLIMETERS SYMBOL MIN NOMINAL MAX NOTES A 0.80 0.90 1.00 - A1 - - 0.05 - A2 - - 1.00 A3 b 0.23 D 0.28 9 0.35 5, 8 4.00 BSC D1 D2 9 0.20 REF - 3.75 BSC 1.95 2.10 9 2.25 7, 8 E 4.00 BSC - E1 3.75 BSC 9 E2 1.95 e 2.10 2.25 7, 8 0.65 BSC - k 0.25 - - - L 0.50 0.60 0.75 8 L1 - - 0.15 10 N 16 2 Nd 4 3 Ne 4 3 P - - 0.60 9 θ - - 12 9 Rev. 5 5/04 NOTES: 1. Dimensioning and tolerancing conform to ASME Y14.5-1994. 2. N is the number of terminals. 3. Nd and Ne refer to the number of terminals on each D and E. 4. All dimensions are in millimeters. Angles are in degrees. 5. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. 7. Dimensions D2 and E2 are for the exposed pads which provide improved electrical and thermal performance. 8. Nominal dimensions are provided to assist with PCB Land Pattern Design efforts, see Intersil Technical Brief TB389. 9. Features and dimensions A2, A3, D1, E1, P & θ are present when Anvil singulation method is used and not present for saw singulation. 10. Depending on the method of lead termination at the edge of the package, a maximum 0.15mm pull back (L1) maybe present. L minus L1 to be equal to or greater than 0.3mm. All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 11 FN9009.4 October 4, 2005