LM22671
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SNVS589M – SEPTEMBER 2008 – REVISED APRIL 2013
LM22671/LM22671-Q1 42V, 500mA SIMPLE SWITCHER® Step-Down Voltage Regulator
with Features
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FEATURES
DESCRIPTION
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The LM22671 switching regulator provides all of the
functions necessary to implement an efficient high
voltage step-down (buck) regulator using a minimum
of external components. This easy to use regulator
incorporates a 42V N-channel MOSFET switch
capable of providing up to 500mA of load current.
Excellent line and load regulation along with high
efficiency (>90%) are featured. Voltage mode control
offers short minimum on-time, allowing the widest
ratio between input and output voltages. Internal loop
compensation means that the user is free from the
tedious task of calculating the loop compensation
components. Fixed 5V output and adjustable output
voltage options are available. The default switching
frequency is set at 500 kHz allowing for small
external components and good transient response. In
addition, the frequency can be adjusted over a range
of 200 kHz to 1MHz with a single external resistor.
The internal oscillator can be synchronized to a
system clock or to the oscillator of another regulator.
A precision enable input allows simplification of
regulator control and system power sequencing. In
shutdown mode the regulator draws only 25 µA (typ.).
An adjustable soft-start feature is provided through
the selection of a single external capacitor. The
LM22671 also has built in thermal shutdown, and
current limiting to protect against accidental
overloads.
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Wide Input Voltage Range: 4.5V to 42V
Internally Compensated Voltage Mode Control
Stable with Low ESR Ceramic capacitors
200 mΩ N-Channel MOSFET
Output Voltage Options:
– ADJ (Outputs as Low as 1.285V)
– 5.0 (Output Fixed to 5V)
±1.5% Feedback Reference Accuracy
500 kHz Default Switching Frequency
Adjustable Switching Frequency and
Synchronization
–40°C to 125°C Operating Junction
Temperature Range
Precision Enable Pin
Integrated Boot-Strap Diode
Adjustable Soft-Start
Fully WEBENCH® Enabled
LM22671-Q1 is an Automotive Grade Product
that is AEC-Q100 Grade 1 Qualified (-40°C to
+125°C Operating Junction Temperature)
PACKAGE
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SO PowerPAD (Exposed Pad)
APPLICATIONS
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Industrial Control
Telecom and Datacom Systems
Embedded Systems
Conversions from Standard 24V, 12V and 5V
Input Rails
The LM22671 is a member of Texas Instruments'
SIMPLE SWITCHER™ family. The SIMPLE
SWITCHER™ concept provides for an easy to use
complete design using a minimum number of external
components and the TI WEBENCH® design tool. TI's
WEBENCH® tool includes features such as external
component calculation, electrical simulation, thermal
simulation, and Build-It boards for easy design-in.
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Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008–2013, Texas Instruments Incorporated
LM22671
SNVS589M – SEPTEMBER 2008 – REVISED APRIL 2013
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Simplified Application Schematic
VIN
FB
VIN
LM22671-ADJ
BOOT
VOUT
SW
RT/SYNC SS
EN
BOOT
1
8
SW
SS
2
7
VIN
RT/SYNC
3
6
GND
FB
4
5
EN
GND
Connection Diagram
Exposed Pad
Connect to GND
Figure 1. 8-Lead Plastic SO PowerPAD Package
Package Number DDA0008B
PIN DESCRIPTIONS
Pin
Name
Description
Application Information
1
BOOT
Bootstrap input
Provides the gate voltage for the high side NFET.
2
SS
Soft-start pin
Used to increase soft-start time. See Soft-Start section of data sheet.
3
RT/SYNC
Oscillator frequency adjust pin or
frequency synchronization
Used to control oscillator mode of regulator. See Switching Frequency
Adjustment and Synchronization section of data sheet.
4
FB
Feedback pin
Feedback input to regulator.
5
EN
Precision enable pin
Used to control regulator start-up and shut-down. See Precision Enable and
UVLO section of data sheet.
6
GND
System ground
System ground pin.
7
VIN
Source input voltage
Supply input to the regulator.
8
SW
Switch pin
Switching output of regulator.
EP
EP
Exposed Pad
Connect to ground. Provides thermal connection to PCB. See Application
Information.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
2
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Absolute Maximum Ratings (1) (2)
VIN to GND
43V
EN Pin Voltage
-0.5V to 6V
SS, RT/SYNC Pin Voltage
-0.5V to 7V
SW to GND (3)
-5V to VIN
BOOT Pin Voltage
VSW + 7V
FB Pin Voltage
-0.5V to 7V
Power Dissipation
Internally Limited
Junction Temperature
150°C
For soldering specifications, refer to the following document: www.ti.com/lit/snoa549
ESD Rating (4)
Human Body Model
Storage Temperature Range
(1)
(2)
(3)
(4)
±2 kV
-65°C to +150°C
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of
device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or
other conditions beyond those indicated in the recommended Operating Ratings is not implied. The recommended Operating Ratings
indicate conditions at which the device is functional and should not be operated beyond such conditions.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
The absolute maximum specification of the ‘SW to GND’ applies to DC voltage. An extended negative voltage limit of -10V applies to a
pulse of up to 50 ns.
ESD was applied using the human body model, a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Operating Ratings (1)
Supply Voltage (VIN)
4.5V to 42V
Junction Temperature Range
(1)
-40°C to +125°C
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of
device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or
other conditions beyond those indicated in the recommended Operating Ratings is not implied. The recommended Operating Ratings
indicate conditions at which the device is functional and should not be operated beyond such conditions.
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Electrical Characteristics
Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C
to +125°C. Minimum and Maximum limits are ensured through test, design, or statistical correlation. Typical values represent
the most likely parametric norm at TA = TJ = 25°C, and are provided for reference purposes only. Unless otherwise specified:
VIN = 12V.
Symbol
Parameter
Conditions
Feedback Voltage
VIN = 8V to 42V
Feedback Voltage
VIN = 4.7V to 42V
Min (1)
Typ (2)
Max (1)
Units
4.925/4.9
5.0
5.075/5.1
V
1.266/1.259
1.285
1.304/1.311
V
3.4
6
mA
µA
LM22671-5.0
VFB
LM22671-ADJ
VFB
All Output Voltage Versions
IQ
ISTDBY
EN Pin = 0V
IL
Output Leakage Current
25
40
0.7
0.84/0.9
A
VIN = 42V, EN Pin = 0V, VSW = 0V
0.2
2
µA
VSW = -1V
0.1
3
µA
0.62/0.56
0.2
0.24/0.32
Ω
Oscillator Frequency
400
500
600
kHz
TOFFMIN
Minimum Off-time
100
200
300
ns
TONMIN
Minimum On-time
Switch On-Resistance
IBIAS
Feedback Bias Current
VFB = 1.3V (ADJ Version Only)
VEN
Enable Threshold Voltage
Falling
VENHYST
Enable Voltage Hysteresis
1.3
100
ns
230
nA
1.6
1.9
V
0.6
V
Enable Input Current
EN Input = 0V
6
µA
FSYNC
Maximum Synchronization
Frequency
VSYNC = 3.5V, 50% duty-cycle
1
MHz
VSYNC
Synchronization Threshold
Voltage
1.75
V
IEN
4
Standby Quiescent Current
Current Limit
fO
(2)
(3)
VFB = 5V
ICL
RDS(ON)
(1)
Quiescent Current
ISS
Soft-Start Current
TSD
Thermal Shutdown Threshold
θJA
Thermal Resistance
30
MR package, Junction to ambient
thermal resistance (3)
50
70
µA
150
°C
60
°C/W
Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are ensured through correlation
using Statistical Quality Control (SQC) methods. Limits are used to calculate TI's Average Outgoing Quality Level (AOQL).
Typical values represent most likely parametric norms at the conditions specified and are not ensured.
The value of θJA for the SO PowerPAD exposed pad (MR) package of 60°C/W is valid if package is mounted to 1 square inch of copper.
The θJA value can range from 42 to 115°C/W depending on the amount of PCB copper dedicated to heat transfer.
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Typical Performance Characteristics
Unless otherwise specified the following conditions apply: Vin = 12V, TJ = 25°C.
Efficiency vs IOUT and VIN
VOUT = 3.3V
Normalized Switching Frequency vs Temperature
Figure 2.
Figure 3.
Current Limit vs Temperature
Normalized RDS(ON) vs Temperature
Figure 4.
Figure 5.
Feedback Bias Current vs Temperature
Normalized Enable Threshold Voltage vs Temperature
Figure 6.
Figure 7.
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Typical Performance Characteristics (continued)
Unless otherwise specified the following conditions apply: Vin = 12V, TJ = 25°C.
Standby Quiescent Current vs Input Voltage
Normalized Feedback Voltage vs Temperature
Figure 8.
Figure 9.
Normalized Feedback Voltage vs Input Voltage
Switching Frequency vs RT/SYNC Resistor
Figure 10.
Figure 11.
Soft-start Current vs Temperature
Figure 12.
6
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SIMPLIFIED BLOCK DIAGRAM
VIN
VIN
Vcc
BOOT
INT REG, EN,UVLO
EN
ILimit
PWM Cmp.
FB
TYPE III
COMP
+
+
-
LOGIC
Error Amp.
VOUT
SW
OSC
SS
1.285V/SS
50 µA
RT/SYNC
GND
Figure 13. Simplified Block Diagram
Detailed Operating Description
The LM22671 incorporates a voltage mode constant frequency PWM architecture. In addition, input voltage feedforward is used to stabilize the loop gain against variations in input voltage. This allows the loop compensation to
be optimized for transient performance. The power MOSFET, in conjunction with the diode, produce a
rectangular waveform at the switch pin, that swings from about zero volts to VIN. The inductor and output
capacitor average this waveform to become the regulator output voltage. By adjusting the duty cycle of this
waveform, the output voltage can be controlled. The error amplifier compares the output voltage with the internal
reference and adjusts the duty cycle to regulate the output at the desired value.
The internal loop compensation of the -ADJ option is optimized for outputs of 5V and below. If an output voltage
of 5V or greater is required, the -5.0 option can be used with an external voltage divider. The minimum output
voltage is equal to the reference voltage; 1.285V (typ.).
The functional block diagram of the LM22671 is shown in Figure 13 .
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Precision Enable and UVLO
The precision enable input (EN) is used to control the regulator. The precision feature allows simple sequencing
of multiple power supplies with a resistor divider from another supply. Connecting this pin to ground or to a
voltage less than 1.6V (typ.) will turn off the regulator. The current drain from the input supply, in this state, is 25
µA (typ.) at an input voltage of 12V. The EN input has an internal pull-up of about 6 µA. Therefore this pin can be
left floating or pulled to a voltage greater than 2.2V (typ.) to turn the regulator on. The hysteresis on this input is
about 0.6V (typ.) above the 1.6V (typ.) threshold. When driving the enable input, the voltage must never exceed
the 6V absolute maximum specification for this pin.
Although an internal pull-up is provided on the EN pin, it is good practice to pull the input high, when this feature
is not used, especially in noisy environments. This can most easily be done by connecting a resistor between
VIN and the EN pin. The resistor is required, since the internal zener diode, at the EN pin, will conduct for
voltages above about 6V. The current in this zener must be limited to less than 100 µA. A resistor of 470 kΩ will
limit the current to a safe value for input voltages as high 42V. Smaller values of resistor can be used at lower
input voltages.
The LM22671 also incorporates an input under voltage lock-out (UVLO) feature. This prevents the regulator from
turning on when the input voltage is not great enough to properly bias the internal circuitry. The rising threshold is
4.3V (typ.) while the falling threshold is 3.9V (typ.). In some cases these thresholds may be too low to provide
good system performance. The solution is to use the EN input as an external UVLO to disable the part when the
input voltage falls below a lower boundary. This is often used to prevent excessive battery discharge or early
turn-on during start-up. This method is also recommended to prevent abnormal device operation in applications
where the input voltage falls below the minimum of 4.5V. Figure 14 shows the connections to implement this
method of UVLO. The following equations can be used to determine the correct resistor values:
(1)
(2)
Where Voff is the input voltage where the regulator shuts off, and Von is the voltage where the regulator turns on.
Due to the 6 µA pull-up, the current in the divider should be much larger than this. A value of 20 kΩ, for RENB is a
good first choice. Also, a zener diode may be needed between the EN pin and ground, in order to comply with
the absolute maximum ratings on this pin.
Vin
RENT
EN
RENB
Figure 14. External UVLO Connections
Duty-Cycle Limits
Ideally the regulator would control the duty cycle over the full range of zero to one. However due to inherent
delays in the circuitry, there are limits on both the maximum and minimum duty cycles that can be reliably
controlled. This in turn places limits on the maximum and minimum input and output voltages that can be
converted by the LM22671. A minimum on-time is imposed by the regulator in order to correctly measure the
switch current during a current limit event. A minimum off-time is imposed in order the re-charge the bootstrap
capacitor. The following equation can be used to determine the approximate maximum input voltage for a given
output voltage:
(3)
8
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Where Fsw is the switching frequency and TON is the minimum on-time; both found in the Electrical
Characteristics table. If the frequency adjust feature is used, that value should be used for Fsw. Nominal values
should be used. The worst case is lowest output voltage, and highest switching frequency. If this input voltage is
exceeded, the regulator will skip cycles, effectively lowering the switching frequency. The consequences of this
are higher output voltage ripple and a degradation of the output voltage accuracy.
The second limitation is the maximum duty cycle before the output voltage will "dropout" of regulation. The
following equation can be used to approximate the minimum input voltage before dropout occurs:
(4)
The values of TOFF and RDS(ON) are found in the Electrical Characteristics table. The worst case here is highest
switching frequency and highest load. In this equation, RL is the D.C. inductor resistance. Of course, the lowest
input voltage to the regulator must not be less than 4.5V (typ.).
Current Limit
The LM22671 has current limiting to prevent the switch current from exceeding safe values during an accidental
overload on the output. This peak current limit is found in the Electrical Characteristics table under the heading of
ICL. The maximum load current that can be provided, before current limit is reached, is determined from the
following equation:
(5)
Where L is the value of the power inductor.
When the LM22671 enters current limit, the output voltage will drop and the peak inductor current will be fixed at
ICL at the end of each cycle. The switching frequency will remain constant while the duty cycle drops. The load
current will not remain constant, but will depend on the severity of the overload and the output voltage.
For very severe overloads ("short-circuit"), the regulator changes to a low frequency current foldback mode of
operation. The frequency foldback is about 1/5 of the nominal switching frequency. This will occur when the
current limit trips before the minimum on-time has elapsed. This mode of operation is used to prevent inductor
current "run-away", and is associated with very low output voltages when in overload. The following equation can
be used to determine what level of output voltage will cause the part to change to low frequency current foldback:
(6)
Where Fsw is the normal switching frequency and Vin is the maximum for the application. If the overload drives
the output voltage to less than or equal to Vx, the part will enter current foldback mode. If a given application can
drive the output voltage to ≤Vx, during an overload, then a second criterion must be checked. The next equation
gives the maximum input voltage, when in this mode, before damage occurs:
(7)
Where Vsc is the value of output voltage during the overload and Fsw is the normal switching frequency. If the
input voltage should exceed this value, while in foldback mode, the regulator and/or the diode may be
damaged. It is important to note that the voltages in these equations are measured at the inductor. Normal trace
and wiring resistance will cause the voltage at the inductor to be higher than that at a remote load. Therefore,
even if the load is shorted with zero volts across its terminals, the inductor will still see a finite voltage. It is this
value that should be used for Vx and Vsc in the calculations. In order to return from foldback mode, the load must
be reduced to a value much lower than that required to initiate foldback. This load "hysteresis" is a normal aspect
of any type of current limit foldback associated with voltage regulators.
If the frequency synchronization feature is used, the current limit frequency fold-back is not operational, and the
system may not survive a hard short-circuit at the output.
The safe operating areas, when in short circuit mode, are shown in Figure 15 through Figure 17, for different
switching frequencies. Operating points below and to the right of the curve represent safe operation. Note that
these curves are not valid when the LM22671 is in frequency synchronization mode.
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45
45
40
40
35
35
INPUT VOLTAGE (v)
INPUT VOLTAGE (v)
SNVS589M – SEPTEMBER 2008 – REVISED APRIL 2013
30
25
SAFE OPERATING AREA
20
15
30
25
SAFE OPERATING AREA
20
15
10
10
5
5
0.0
0.2
0.4
0.6
0.8
1.0
SHORT CIRCUIT VOLTAGE (v)
1.2
0.0
Figure 15. SOA at 300 kHz
0.2
0.4
0.6
0.8
1.0
SHORT CIRCUIT VOLTAGE (v)
1.2
Figure 16. SOA at 500 kHz
45
INPUT VOLTAGE (v)
40
35
30
25
20
SAFE OPERATING AREA
15
10
5
0.0
0.2
0.4
0.6
0.8
1.0
SHORT CIRCUIT VOLTAGE (v)
1.2
Figure 17. SOA at 800 kHz
Soft-Start
The soft-start feature allows the regulator to gradually reach steady-state operation, thus reducing start-up
stresses. The internal soft-start feature brings the output voltage up in about 500 µs. This time can be extended
by using an external capacitor connected to the SS pin. Values in the range of 100 nF to 1 µF are recommended.
The approximate soft-start time can be estimated from the following equation:
(8)
Soft-start is reset any time the part is shut down or a thermal overload event occurs.
Switching Frequency Adjustment and Synchronization
The LM22671 will operate in three different modes, depending on the condition of the RT/SYNC pin. With the
RT/SYNC pin floating, the regulator will switch at the internally set frequency of 500 kHz (typ.). With a resistor in
the range of 25 kΩ to 200 kΩ, connected from RT/SYNC to ground, the internal switching frequency can be
adjusted from 1MHz to 200 kHz. Figure 18 shows the typical curve for switching frequency vs. the external
resistance connected to the RT/SYNC pin. The accuracy of the switching frequency, in this mode, is slightly
worse than that of the internal oscillator; about +/- 25% is to be expected. Finally, an external clock can be
applied to the RT/SYNC pin to allow the regulator to synchronize to a system clock or another LM22671. The
mode is set during start-up of the regulator. When the LM22671 is enabled, or after VIN is applied, a weak pull-up
is connected to the RT/SYNC pin and, after approximately 100 µs, the voltage on the pin is checked against a
threshold of about 0.8V. With the RT/SYNC pin open, the voltage floats above this threshold, and the mode is set
to run with the internal clock. With a frequency set resistor present, an internal reference holds the pin voltage at
0.8V; the resulting current sets the mode to allow the resistor to control the clock frequency. If the external circuit
forces the RT/SYNC pin to a voltage much greater or less than 0.8v, the mode is set to allow external
synchronization. The mode is latched until either the EN or the input supply is cycled.
10
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The choice of switching frequency is governed by several considerations. As an example, lower frequencies may
be desirable to reduce switching losses or improve duty cycle limits. Higher frequencies, or a specific frequency,
may be desirable to avoid problems with EMI or reduce the physical size of external components. The flexibility
of increasing the switching frequency above 500 kHz can also be used to operate outside a critical signal
frequency band for a given application. Keep in mind that the values of inductor and output capacitor cannot be
reduced dramatically, by operating above 500 kHz. This is true because the design of the internal loop
compensation restricts the range of these components.
Frequency synchronization requires some care. First the external clock frequency must be greater than the
internal clock frequency, and less than 1 MHz. The maximum internal switching frequency is ensured in the
Electrical Characteristics table. Note that the frequency adjust feature and the synchronization feature can not be
used simultaneously. The synchronizing frequency must always be greater than the internal clock frequency.
Secondly, the RT/SYNC pin must see a valid high or low voltage, during start-up, in order for the regulator to go
into the synchronizing mode (see above). Also, the amplitude of the synchronizing pulses must comport with
VSYNC levels found in the Electrical Characteristics table. The regulator will synchronize on the rising edge of the
external clock. If the external clock is lost during normal operation, the regulator will revert to the 500 kHz (typ.)
internal clock.
If the frequency synchronization feature is used, current limit foldback is not operational; see the Current Limit
section for details.
Figure 18. Switching Frequency vs RT/SYNC Resistor
Self Synchronization
It is possible to synchronize multiple LM22671 regulators together to share the same switching frequency. This
can be done by tieing the RT/SYNC pins together through a MOSFET and connecting a 1 KΩ resistor to ground
at each pin. Figure 19 shows this connection. The gate of the MOSFET should be connected to the regulator
with the highest output voltage. Also, the EN pins of both regulators should be tied to the common system
enable, in order to properly initialize both regulators. The operation is as follows: When the regulators are
enabled, the outputs are low and the MOSFET is off. The 1 kΩ resistors pull the RT/SYNC pins low, thus
enabling the synchronization mode. These resistors are small enough to pull the RT/SYNC pin low, rather than
activate the frequency adjust mode. Once the output voltage of one of the regulators is sufficient to turn on the
MOSFET, the two RT/SYNC pins are tied together and the regulators will run in synchronized mode. The two
regulators will be clocked at the same frequency but slightly phase shifted according to the minimum off-time of
the regulator with the fastest internal oscillator. The slight phase shift helps to reduce stress on the input
capacitors of the regulator. It is important to choose a MOSFET with a low gate threshold voltage so that the
MOSFET will be fully enhanced. Also, a MOSFET with low inter-electrode capacitance is required. The 2N7002
is a good choice.
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ENABLE
EN
EN
LM22671
LM22671
RT/SYNC
RT/SYNC
2N7002
1k
1k
Vout
Figure 19. Self Synchronizing Setup
Boot-Strap Supply
The LM22671 incorporates a floating high-side gate driver to control the power MOSFET. The supply for this
driver is the external boot-strap capacitor connected between the BOOT pin and SW. A good quality 10 nF
ceramic capacitor must be connected to these pins with short, wide PCB traces. One reason the regulator
imposes a minimum off-time is to ensure that this capacitor recharges every switching cycle. A minimum load of
about 5 mA is required to fully recharge the boot-strap capacitor in the minimum off-time. Some of this load can
be provided by the output voltage divider, if used.
Thermal Protection
Internal thermal shutdown circuitry protects the LM22671 should the maximum junction temperature be
exceeded. This protection is activated at about 150°C, with the result that the regulator will shutdown until the
temperature drops below about 135°C.
Internal Loop Compensation
The LM22671 has internal loop compensation designed to provide a stable regulator over a wide range of
external power stage components. The internal compensation of the -ADJ option is optimized for output voltages
below 5V. If an output voltage of 5V or greater is needed, the -5.0 option with an external resistor divider can be
used.
Ensuring stability of a design with a specific power stage (inductor and output capacitor) can be tricky. The
LM22671 stability can be verified using the WEBENCH® Designer online circuit simulation tool at www.ti.com. A
quick start spreadsheet can also be downloaded from the online product folder.
The complete transfer function for the regulator loop is found by combining the compensation and power stage
transfer functions. The LM22671 has internal type III loop compensation, as detailed in Figure 20 . This is the
approximate "straight line" function from the FB pin to the input of the PWM modulator. The power stage transfer
function consists of a D.C. gain and a second order pole created by the inductor and output capacitor(s). Due to
the input voltage feedforward employed in the LM22671, the power stage D.C. gain is fixed at 20dB. The second
order pole is characterized by its resonant frequency and its quality factor (Q). For a first pass design, the
product of inductance and output capacitance should conform to the following equation:
(9)
Alternatively, this pole should be placed between 1.5kHz and 15kHz and is given by the equation shown below:
(10)
The Q factor depends on the parasitic resistance of the power stage components and is not typically in the
control of the designer. Of course, loop compensation is only one consideration when selecting power stage
components; see the Application Information section for more details.
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COMPENSATOR GAIN (dB)
40
-ADJ
-5.0
35
30
25
20
15
10
5
0
100
1k
10k
100k
1M
FREQUENCY (Hz)
10M
Figure 20. Compensator Gain
In general, hand calculations or simulations can only aid in selecting good power stage components. Good
design practice dictates that load and line transient testing should be done to verify the stability of the application.
Also, Bode plot measurements should be made to determine stability margins. Application note AN-1889
SNVA364 shows how to perform a loop transfer function measurement with only an oscilloscope and function
generator.
Application Information
TYPICAL BUCK REGULATOR APPLICATION
Figure 21 shows an example of converting an input voltage range of 5.5V to 42V, to an output of 3.3v at 500mA.
See AN-1895 (SNVA368) for more information.
RFBB
976:
VIN 4.5V to 42V
FB
VIN
EN
C2
22 PF
+
C3
10 nF
LM22671-ADJ
EN
BOOT
SYNC
C1
2.2 PF
RFBT
1.54 k:
L1
39 PH
RT/SYNC
SW
SS
GND
R3
C6
1 PF
D1
60V, 1A
VOUT 3.3V
C4
22 PF
+
GND
GND
Figure 21. Typical Buck Regulator Application
EXTERNAL COMPONENTS
The following guidelines should be used when designing a step-down (buck) converter with the LM22671.
INDUCTOR
The inductor value is determined based on the load current, ripple current, and the minimum and maximum input
voltages. To keep the application in continuous conduction mode (CCM), the maximum ripple current, IRIPPLE ,
should be less than twice the minimum load current.
The general rule of keeping the inductor current peak-to-peak ripple around 30% of the nominal output current is
a good compromise between excessive output voltage ripple and excessive component size and cost. Using this
value of ripple current, the value of inductor, L, is calculated using the following formula:
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(11)
where Fsw is the switching frequency and Vin should be taken at its maximum value, for the given application.
The above formula provides a guide to select the value of the inductor L; the nearest standard value will then be
used in the circuit.
Once the inductor is selected, the actual ripple current can be found from the equation shown below:
(12)
Increasing the inductance will generally slow down the transient response but reduce the output voltage ripple.
Reducing the inductance will generally improve the transient response but increase the output voltage ripple.
The inductor must be rated for the peak current, IPK, in a given application, to prevent saturation. During normal
loading conditions, the peak current is equal to the load current plus 1/2 of the inductor ripple current.
During an overload condition, as well as during certain load transients, the controller may trip current limit. In this
case the peak inductor current is given by ICL, found in the Electrical Characteristics table. Good design practice
requires that the inductor rating be adequate for this overload condition. If the inductor is not rated for the
maximum expected current, it can saturate resulting in damage to the LM22671 and/or the power diode.
INPUT CAPACITOR
The input capacitor selection is based on both input voltage ripple and RMS current. Good quality input
capacitors are necessary to limit the ripple voltage at the VIN pin while supplying most of the regulator current
during switch on-time. Low ESR ceramic capacitors are preferred. Larger values of input capacitance are
desirable to reduce voltage ripple and noise on the input supply. This noise may find its way into other circuitry,
sharing the same input supply, unless adequate bypassing is provided. A very approximate formula for
determining the input voltage ripple is shown below:
(13)
Where Vri is the peak-to-peak ripple voltage at the switching frequency. Another concern is the RMS current
passing through this capacitor. The following equation gives an approximation to this current:
(14)
The capacitor must be rated for at least this level of RMS current at the switching frequency.
All ceramic capacitors have large voltage coefficients, in addition to normal tolerances and temperature
coefficients. To help mitigate these effects, multiple capacitors can be used in parallel to bring the minimum
capacitance up to the desired value. This may also help with RMS current constraints by sharing the current
among several capacitors. Many times it is desirable to use an electrolytic capacitor on the input, in parallel with
the ceramics. The moderate ESR of this capacitor can help to damp any ringing on the input supply caused by
long power leads. This method can also help to reduce voltage spikes that may exceed the maximum input
voltage rating of the LM22671.
It is good practice to include a high frequency bypass capacitor as close as possible to the LM22671. This small
case size, low ESR, ceramic capacitor should be connected directly to the VIN and GND pins with the shortest
possible PCB traces. Values in the range of 0.47 µF to 1 µF are appropriate. This capacitor helps to provide a
low impedance supply to sensitive internal circuitry. It also helps to suppress any fast noise spikes on the input
supply that may lead to increased EMI.
14
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OUTPUT CAPACITOR
The output capacitor is responsible for filtering the output voltage and supplying load current during transients.
Capacitor selection depends on application conditions as well as ripple and transient requirements. Best
performance is achieved with a parallel combination of ceramic capacitors and a low ESR SP™ or POSCAP™
type. Very low ESR capacitors such as ceramics reduce the output ripple and noise spikes, while higher value
electrolytics or polymer provide large bulk capacitance to supply transients. Assuming very low ESR, the
following equation gives an approximation to the output voltage ripple:
(15)
Typically, a total value of 100 µF, or greater, is recommended for output capacitance.
In applications with Vout less than 3.3V, it is critical that low ESR output capacitors are selected. This will limit
potential output voltage overshoots as the input voltage falls below the device normal operating range.
If the switching frequency is set higher than 500 kHz, the capacitance value may not be reduced proportionally
due to stability requirements. The internal compensation is optimized for circuits with a 500 kHz switching
frequency. See the Internal Loop Compensation section for more details.
BOOT-STRAP CAPACITOR
The bootstrap capacitor between the BOOT pin and the SW pin supplies the gate current to turn on the Nchannel MOSFET. The recommended value of this capacitor is 10 nF and should be a good quality, low ESR
ceramic capacitor.In some cases it may be desirable to slow down the turn-on of the internal power MOSFET, in
order to reduce EMI. This can be done by placing a small resistor in series with the Cboot capacitor. Resistors in
the range of 10Ω to 50Ω can be used. This technique should only be used when absolutely necessary, since it
will increase switching losses and thereby reduce efficiency.
OUTPUT VOLTAGE DIVIDER SELECTION
For output voltages between about 1.285V and 5V, the -ADJ option should be used, with an appropriate voltage
divider as shown in Figure 22. The following equation can be used to calculate the resistor values of this divider:
(16)
A good value for RFBB is 1k Ω. This will help to provide some of the minimum load current requirement and
reduce susceptibility to noise pick-up. The top of RFBT should be connected directly to the output capacitor or to
the load for remote sensing. If the divider is connected to the load, a local high-frequency bypass should be
provided at that location.
For output voltages of 5V, the -5.0 option should be used. In this case no divider is needed and the FB pin is
connected to the output. The approximate values of the internal voltage divider are as follows: 7.38kΩ from the
FB pin to the input of the error amplifier and 2.55kΩ from there to ground.
Both the -ADJ and -5.0 options can be used for output voltages greater than 5V, by using the correct output
divider. As mentioned in the Internal Loop Compensation section, the -5.0 option is optimized for output voltages
of 5V. However, for output voltages greater than 5V, this option may provide better loop bandwidth than the -ADJ
option, in some applications. If the -5.0 option is to be used at output voltages greater than 5V, the following
equation should be used to determine the resistor values in the output divider:
(17)
Again a value of RFBB of about 1k Ω is a good first choice.
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Vout
RFBT
FB
RFBB
Figure 22. Resistive Feedback Divider
A maximum value of 10 kΩ is recommended for the sum of RFBB and RFBT to maintain good output voltage
accuracy for the -ADJ option. A maximum of 2 kΩ is recommended for the -5.0 option. For the -5.0 option, the
total internal divider resistance is typically 9.93 kΩ.
In all cases the output voltage divider should be placed as close as possible to the FB pin of the LM22671; since
this is a high impedance input and is susceptible to noise pick-up.
POWER DIODE
A Schottky type power diode is required for all LM22671 applications. Ultra-fast diodes are not recommended
and may result in damage to the IC due to reverse recovery current transients. The near ideal reverse recovery
characteristics and low forward voltage drop of Schottky diodes are particularly important for high input voltage
and low output voltage applications common to the LM22671. The reverse breakdown rating of the diode should
be selected for the maximum VIN, plus some safety margin. A good rule of thumb is to select a diode with a
reverse voltage rating of 1.3 times the maximum input voltage.
Select a diode with an average current rating at least equal to the maximum load current that will be seen in the
application.
Circuit Board Layout
Board layout is critical for the proper operation of switching power supplies. First, the ground plane area must be
sufficient for thermal dissipation purposes. Second, appropriate guidelines must be followed to reduce the effects
of switching noise. Switch mode converters are very fast switching devices. In such cases, the rapid increase of
input current combined with the parasitic trace inductance generates unwanted L di/dt noise spikes. The
magnitude of this noise tends to increase as the output current increases. This noise may turn into
electromagnetic interference (EMI) and can also cause problems in device performance. Therefore, care must be
taken in layout to minimize the effect of this switching noise.
The most important layout rule is to keep the AC current loops as small as possible. Figure 23 shows the current
flow in a buck converter. The top schematic shows a dotted line which represents the current flow during the FET
switch on-state. The middle schematic shows the current flow during the FET switch off-state.
The bottom schematic shows the currents referred to as AC currents. These AC currents are the most critical
since they are changing in a very short time period. The dotted lines of the bottom schematic are the traces to
keep as short and wide as possible. This will also yield a small loop area reducing the loop inductance. To avoid
functional problems due to layout, review the PCB layout example. Best results are achieved if the placement of
the LM22671, the bypass capacitor, the Schottky diode, RFBT, RFBB, and the inductor are placed as shown in the
example. Note that, in the layout shown, R1 = RFBB and R2 = RFBT. It is also recommended to use 2oz copper
boards or heavier to help thermal dissipation and to reduce the parasitic inductances of board traces. See
application note AN-1229 (SNVA054) for more information.
16
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Figure 23. Current Flow in a Buck Application
Thermal Considerations
The components with the highest power dissipation are the power diode and the power MOSFET internal to the
LM22671 regulator. The easiest method to determine the power dissipation within the LM22671 is to measure
the total conversion losses then subtract the power losses in the diode and inductor. The total conversion loss is
the difference between the input power and the output power. An approximation for the power diode loss is:
(18)
Where VD is the diode voltage drop. An approximation for the inductor power is:
(19)
where RL is the DC resistance of the inductor and the 1.1 factor is an approximation for the AC losses.
The regulator has an exposed thermal pad to aid power dissipation. Adding several vias under the device to the
ground plane will greatly reduce the regulator junction temperature. Selecting a diode with an exposed pad will
also aid the power dissipation of the diode. The most significant variables that affect the power dissipation of the
regulator are output current, input voltage and operating frequency. The power dissipated while operating near
the maximum output current and maximum input voltage can be appreciable. The junction-to-ambient thermal
resistance of the LM22671 will vary with the application. The most significant variables are the area of copper in
the PC board, the number of vias under the IC exposed pad and the amount of forced air cooling provided. A
large continuous ground plane on the top or bottom PCB layer will provide the most effective heat dissipation.
The integrity of the solder connection from the IC exposed pad to the PC board is critical. Excessive voids will
greatly diminish the thermal dissipation capacity. The junction-to-ambient thermal resistance of the LM22671 SO
PowerPAD package is specified in the Electrical Characteristics table. See application note AN-2020 (SNVA419)
for more information.
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LM22671
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PCB Layout Example
RFBB
2.55 k:
VIN 5.5V to 35V
EN
C2
22 PF
+
C1
2.2 PF
C3
10 nF
BOOT
L1
10 PH
RFBT
7.32 k:
SW
RT/SYNC
C6
4.7 PF
VOUT
FB
VIN
SS LM22671-ADJ
GND
D1
60V 3A
C4
120 PF
VOUT -5V
Figure 24. Inverting Regulator Application
18
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SNVS589M – SEPTEMBER 2008 – REVISED APRIL 2013
REVISION HISTORY
Changes from Revision L (April 2013) to Revision M
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 18
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19
PACKAGE OPTION ADDENDUM
www.ti.com
18-Oct-2013
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
LM22671MR-5.0/NOPB
ACTIVE SO PowerPAD
DDA
8
95
Green (RoHS
& no Sb/Br)
SN | CU SN
Level-3-260C-168 HR
-40 to 125
L22671
5.0
LM22671MR-ADJ/NOPB
ACTIVE SO PowerPAD
DDA
8
95
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
ADJ
LM22671MRE-5.0/NOPB
ACTIVE SO PowerPAD
DDA
8
250
Green (RoHS
& no Sb/Br)
SN | CU SN
Level-3-260C-168 HR
-40 to 125
L22671
5.0
LM22671MRE-ADJ/NOPB
ACTIVE SO PowerPAD
DDA
8
250
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
ADJ
LM22671MRX-5.0/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
5.0
LM22671MRX-ADJ/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
ADJ
LM22671QMR-5.0/NOPB
ACTIVE SO PowerPAD
DDA
8
95
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
Q5.0
LM22671QMR-ADJ/NOPB
ACTIVE SO PowerPAD
DDA
8
95
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
QADJ
LM22671QMRE-5.0/NOPB
ACTIVE SO PowerPAD
DDA
8
250
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
Q5.0
LM22671QMRE-ADJ/NOPB
ACTIVE SO PowerPAD
DDA
8
250
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
QADJ
LM22671QMRX-5.0/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
Q5.0
LM22671QMRX-ADJ/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
L22671
QADJ
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
18-Oct-2013
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM22671, LM22671-Q1 :
• Catalog: LM22671
• Automotive: LM22671-Q1
NOTE: Qualified Version Definitions:
• Catalog - TI's standard catalog product
• Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
8-Apr-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
LM22671MRE-5.0/NOPB
SO
Power
PAD
DDA
8
250
178.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM22671MRE-ADJ/NOPB
SO
Power
PAD
DDA
8
250
178.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM22671MRX-5.0/NOPB
SO
Power
PAD
DDA
8
2500
330.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM22671MRX-ADJ/NOPB
SO
Power
PAD
DDA
8
2500
330.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM22671QMRE-5.0/NOP
B
SO
Power
PAD
DDA
8
250
178.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM22671QMRE-ADJ/NOP
B
SO
Power
PAD
DDA
8
250
178.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM22671QMRX-5.0/NOP
B
SO
Power
PAD
DDA
8
2500
330.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM22671QMRX-ADJ/NOP
SO
DDA
8
2500
330.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
8-Apr-2013
Device
B
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
Power
PAD
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM22671MRE-5.0/NOPB
SO PowerPAD
DDA
8
250
213.0
191.0
55.0
LM22671MRE-ADJ/NOPB
SO PowerPAD
DDA
8
250
213.0
191.0
55.0
LM22671MRX-5.0/NOPB
SO PowerPAD
DDA
8
2500
367.0
367.0
35.0
LM22671MRX-ADJ/NOPB
SO PowerPAD
DDA
8
2500
367.0
367.0
35.0
LM22671QMRE-5.0/NOPB
SO PowerPAD
DDA
8
250
213.0
191.0
55.0
LM22671QMRE-ADJ/NOP
B
SO PowerPAD
DDA
8
250
213.0
191.0
55.0
LM22671QMRX-5.0/NOPB
SO PowerPAD
DDA
8
2500
367.0
367.0
35.0
LM22671QMRX-ADJ/NOP
B
SO PowerPAD
DDA
8
2500
367.0
367.0
35.0
Pack Materials-Page 2
MECHANICAL DATA
DDA0008B
MRA08B (Rev B)
www.ti.com
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