AD AD8340-EVAL 700 mhz to 1000 mhz rf vector modulator Datasheet

700 MHz to 1000 MHz
RF Vector Modulator
AD8340
FUNCTIONAL BLOCK DIAGRAM
FEATURES
VPRF
QBBP OBBM
VPS2
90°
RFIP
RFOP
RFIM
RFOM
0°
CMOP
IBBP IBBM
DSOP
04699-0-001
Cartesian amplitude and phase modulation
700 MHz to 1.0 GHz frequency range
Continuous magnitude control of −2 dB to −32 dB
Continuous phase control of 0° to 360°
Output third-order intercept 24 dBm
Output 1 dB compression point 11 dBm
Output noise floor −149 dBm/Hz @ full gain
Adjustable modulation bandwidth up to 230 MHz
Fast output power disable
4.75 V to 5.25 V single-supply voltage
Figure 1.
APPLICATIONS
RF PA linearization/RF predistortion
Amplitude and phase modulation
Variable attenuators and phase shifters
CDMA2000, GSM/EDGE linear power amplifiers
Smart antennas
GENERAL DESCRIPTION
The AD8340 vector modulator performs arbitrary amplitude
and phase modulation of an RF signal. Since the RF signal path
is linear, the original modulation is preserved. This part can be
used as a general-purpose RF modulator, a variable attenuator/phase shifter, or a remodulator. The amplitude can be
controlled from a maximum of −2 dB to less than −32 dB, and
the phase can be shifted continuously over the entire 360° range.
For maximum gain, the AD8340 delivers an OP1dB of 11 dBm,
an OIP3 of 24 dBm, and an output noise floor of −149 dBm/Hz,
independent of phase. It operates over a frequency range of
700 MHz to 1.0 GHz.
The baseband inputs in Cartesian I and Q format control the
amplitude and phase modulation imposed on the RF input
signal. Both I and Q inputs are dc-coupled with a ±500 mV
differential full-scale range. The maximum modulation bandwidth is 230 MHz, which can be reduced by adding external
capacitors to limit the noise bandwidth on the control lines.
Both the RF inputs and outputs can be used differentially or
single-ended and must be ac-coupled. The RF input and output
impedances are nominally 50 Ω over the operating frequency
range. The DSOP pin allows the output stage to be disabled
quickly in order to protect subsequent stages from overdrive.
The AD8340 operates off supply voltages from 4.75 V to 5.25 V
while consuming approximately 130 mA.
The AD8340 is fabricated on Analog Devices’ proprietary, high
performance 25 GHz SOI complementary bipolar IC process. It
is available in a 24-lead Pb-free LFCSP package and operates
over a −40°C to +85°C temperature range. Evaluation boards
are available.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.326.8703
© 2004 Analog Devices, Inc. All rights reserved.
AD8340
TABLE OF CONTENTS
Specifications..................................................................................... 3
RF Frequency Range .................................................................. 11
Absolute Maximum Ratings............................................................ 4
Applications..................................................................................... 12
ESD Caution.................................................................................. 4
Using the AD8340 ...................................................................... 12
Pin Configuration and Function Descriptions............................. 5
RF Input and Matching ............................................................. 12
Typical Performance Characteristics ............................................. 6
RF Output and Matching .......................................................... 13
Theory of Operation ...................................................................... 10
Driving the I-Q Baseband Controls......................................... 13
RF Quadrature Generator ......................................................... 10
Interfacing to High Speed DACs.............................................. 14
I-Q Attenuators and Baseband Amplifiers.............................. 11
CDMA2000 Application............................................................ 14
Output Amplifier ........................................................................ 11
Evaluation Board ............................................................................ 16
Noise and Distortion.................................................................. 11
Outline Dimensions ....................................................................... 20
Gain and Phase Accuracy.......................................................... 11
Ordering Guide .......................................................................... 20
REVISION HISTORY
6/04—Revision 0: Initial Version
Rev. 0 | Page 2 of 20
AD8340
SPECIFICATIONS
VS = 5 V, TA = 25°C, ZO = 50 Ω, f = 880 MHz, single-ended, ac-coupled source drive to RFIP through 5.6 nH series inductor, RFIM
ac-coupled through 5.6 nH series inductor to common, differential-to-single-ended conversion at output using 1:1 balun.
Table 1.
Parameter
OVERALL FUNCTION
Frequency Range
Maximum Gain
Minimum Gain
Gain Control Range
Phase Control Range
Gain Flatness
Group Delay Flatness
RF INPUT STAGE
Input Return Loss
CARTESIAN CONTROL INTERFACE (I & Q)
Gain Scaling
Modulation Bandwidth
Second Harmonic Distortion
Third Harmonic Distortion
Step Response
RF OUTPUT STAGE
Output Return Loss
f = 880 MHz
Gain
Output Noise Floor
Output IP3
ACPR
Output 1 dB Compression Point
POWER SUPPLY
Positive Supply Voltage
Total Supply Current
OUTPUT DISABLE
Disable Threshold
Maximum Attenuation
Enable Response Time
Disable Response Time
Conditions
Min
Typ
Max
Unit
1000
−2
−32
30
360
0.25
10
MHz
dB
dB
dB
°
dB
ps
20
dB
2
230
47
45
45
1/V
MHz
dBc
dBc
ns
47
ns
7.5
dB
−2
−149
−147
24
62
dB
dBm/Hz
dBm/Hz
dBm
dBc
11
dBm
700
Maximum gain setpoint for all phase setpoints
VBBI = VBBQ = 0 V
Relative to maximum gain
Over 30 dB control range
Over any 60 MHz bandwidth
Over any 60 MHz bandwidth
RFIM, RFIP (Pins 21 and 22)
From RFIP to CMRF (with 5.6 nH series inductors)
IBBP, IBBM, QBBP, QBBM (Pins 16, 15, 3, 4)
250 mV p-p sinusoidal baseband input single-ended
250 mV p-p, 1 MHz, sinusoidal baseband input differential
250 mV p-p, 1 MHz, sinusoidal baseband input differential
For gain setpoint from 0.1 to 0.9
(VBBP = 0.5 V, VBBM = 0.55 V to 0.95 V)
For gain setpoint from 0.9 to 0.1
(VBBP = 0.5 V, VBBM = 0.95 V to 0.55 V)
RFOP, RFOM (Pins 9, 10)
Measured through balun
Maximum gain setpoint
Maximum gain setpoint, no input
PIN = 0 dBm, frequency offset = 20 MHz
f1 = 880 MHz, f2 = 877.5 MHz, maximum gain setpoint
IS-95, single carrier, POUT = 0 dBm, maximum gain,
phase setpoint = 45°
Maximum gain
VPS2 (Pin 5, 6, 14); RFOP, RFOM (Pins 9, 10)
Includes load current
DSOP (Pin 13)
DSOP = 5 V
Delay following high-to-low transition until device
meets full specifications
Delay following low-to-high transition until device
produces full attenuation
Rev. 0 | Page 3 of 20
4.75
110
5
130
5.25
150
V
mA
2.5
40
15
V
dB
ns
10
ns
AD8340
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameters
Supply Voltage VPRF, VPS2
DSOP
IBBP, IBBM, QBBP, QBBM
RFOP, RFOM
RF Input Power at Maximum Gain
(RFIP or RFIM, Single-Ended Drive)
Equivalent Voltage
Internal Power Dissipation
θJA (With Pad Soldered to Board)
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Lead Temperature Range (Soldering 60 sec)
Rating
5.5 V
5.5 V
2.5 V
5.5V
13 dBm, re: 50 Ω
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
2.8 V p-p
825 mW
59 °C/W
125°C
−40°C to +85°C
−65°C to +150°C
300°C
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic
discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of
functionality.
Rev. 0 | Page 4 of 20
AD8340
24 VPRF
23 CMRF
22 RFIP
21 RFIM
20 CMRF
19 VPRF
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
AD8340
TOP VIEW
(Not to Scale)
18 IFLP
17 IFLM
16 IBBP
15 IBBM
14 VPS2
13 DSOP
04699-0-002
1
2
3
4
5
6
CMOP 7
CMOP 8
RFOP 9
RFOM 10
CMOP 11
CMOP 12
QFLP
QFLM
QBBP
QBBM
VPS2
VPS2
Figure 2. 24-Lead Lead Frame Chip Scale Package (LFCSP)
Table 3. Pin Function Descriptions
Pin No.
1, 2
Mnemonic
QFLP, QFLM
3, 4
5, 6, 14, 19, 24
7, 8, 11, 12, 20, 23
9, 10
13
15, 16
17, 18
QBBP, QBBM
VPS2, VPRF
CMOP, CMRF
RFOP, RFOM
DSOP
IBBM, IBBP
IFLM, IFLP
21, 22
RFIM, RFIP
Function
Q Baseband Input Filter Pins. Connect optional capacitor to reduce Q baseband channel low-pass
corner frequency.
Q Channel Differential Baseband Inputs.
Positive Supply Voltage. 4.75 V − 5.25 V.
Device Common. Connect via lowest possible impedance to external circuit common.
Differential RF Outputs. Must be ac-coupled. Differential impedance 50 Ω nominal.
Output disable. Pull high to disable output stage.
I Channel Differential Baseband Inputs.
I Baseband Input Filter Pins. Connect optional capacitor to reduce I baseband channel low-pass
corner frequency.
Differential RF Inputs. Must be ac-coupled. Differential impedance 50 Ω nominal.
Rev. 0 | Page 5 of 20
AD8340
TYPICAL PERFORMANCE CHARACTERISTICS
0
0.4
PHASE SETPOINT = 0°
–10
PHASE SETPOINT = 270°
–20
–25
04699-0-003
–30
–35
PHASE SETPOINT = 180°
–40
0
0.1
0.2
0.3
0.4
0.5
0.6
GAIN SETPOINT
0.7
0.8
0.9
–0.6
–0.8
–1.2
–1.4
–1.6
–2.0
0
1
PHASE SETPOINT = 0°
180
225
270
315
360
330
300
270
–2
PHASE SETPOINT = 270°
–3
GAIN SETPOINT = 1.0
240
210
180
GAIN SETPOINT = 0.1
150
GAIN SETPOINT = 0.5
120
90
PHASE SETPOINT = 180°
–5
PHASE SETPOINT = 225°
–7
0
0.1
0.2
0.3
0.4
0.5 0.6
GAIN SETPOINT
0.7
0.8
0.9
04699-0-007
04699-0-004
60
–6
30
0
0
1.0
Figure 4. Gain Conformance Error vs. Gain Setpoint at
Different Phase Setpoints
30
60
90 120 150 180 210 240 270 300 330 360
PHASE SETPOINT (Degrees)
Figure 7. Phase vs. Phase Setpoint at Different Gain Setpoints
0
6
–2
4
GAIN SETPOINT = 1.0
–4
PHASE ERROR (Degrees)
–6
–8
GAIN SETPOINT = 0.5
–10
–12
–14
–16
–18
GAIN SETPOINT = 0.1
2
0
–2
GAIN SETPOINT = 1.0
–4
GAIN SETPOINT = 0.5
–6
–8
–20
GAIN SETPOINT = 0.1
04699-0-005
GAIN (dB)
135
360
PHASE SETPOINT = 315°
–4
90
Figure 6. Gain Conformance Error vs. Phase Setpoint at Different Gain Setpoints
0
–1
45
PHASE SETPOINT (Degrees)
PHASE (Degrees)
PHASE SETPOINT = 90°
GAIN SETPOINT = 0.1
–1.8
PHASE SETPOINT = 45°
2
GAIN SETPOINT = 0.5
–1.0
PHASE SETPOINT = 135°
3
GAIN CONFORMANCE ERROR (dB)
–0.4
1.0
Figure 3. Gain Magnitude vs. Gain Setpoint at Different Phase Setpoints,
RF Frequency = 880 MHz
4
–0.2
–22
–24
0
45
90
135
180
225
270
315
04699-0-008
GAIN (dB)
–15
GAIN SETPOINT = 1.0
0.0
04699-0-006
PHASE SETPOINT = 90°
GAIN CONFORMANCE ERROR (dB)
0.2
–5
–10
–12
0
360
PHASE SETPOINT (Degrees)
45
90
135
180
225
270
PHASE SETPOINT (Degrees)
315
Figure 8. Phase Error vs. Phase Setpoint at Different Gain Setpoints
Figure 5. Gain Magnitude vs. Phase Setpoint at Different Gain Setpoints
Rev. 0 | Page 6 of 20
360
AD8340
–142
0
–143
–0.5
–145
GAIN FLATNESS (dB)
RF PIN = +5dBm
–146
–147
RF PIN = –5dBm
–148
RF PIN = 0dBm
–149
–1.0
–1.5
–150
–2.0
04699-0-009
NO RF INPUT
–151
–152
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
04699-0-012
NOISE FLOOR (dBm/Hz)
–144
–2.5
700
1.0
750
800
850
900
FREQUENCY (MHz)
GAIN SETPOINT
Figure 9. Output Noise Floor vs. Gain, Noise in dBm/Hz, No Carrier,
With Carrier (20 MHz Offset) Pin = −5, 0, and +5 dBm
1000
Figure 12. Gain Flatness vs. Frequency, Maximum Gain, Phase Setpoint = 0°
0
–6
–8
GAIN SETPOINT = 0.5
–10
–12
–14
–16
–18
GAIN SETPOINT = 0.1
04699-0-010
–20
–22
–24
700
750
800
850
900
950
–10
–20
–30
–50
SECOND BASEBAND HARMONIC PRODUCT,
878MHz, 882MHz
–60
–70
THIRD BASEBAND HARMONIC PRODUCT,
877MHz, 883MHz
–80
–90
–100
100
1000
FUNDAMENTAL POWER, 879MHz, 881MHz
–40
04699-0-013
GAIN SETPOINT = 1.0
–4
RF OUTPUT AM SIDEBAND POWER (dBm)
0
–2
GAIN (dB)
950
200
300
400
500
600
700
800
900
1000
DIFFERENTIAL BASEBAND INPUT LEVEL (mV p-p)
(I OR Q CHANNEL DRIVEN AT 1MHz)
FREQUENCY (MHz)
Figure 10. Gain vs. Frequency at Different Gain Setpoints (700 MHz to 1000 MHz),
Phase Setpoint = 0°
Figure 13. Baseband Harmonic Distortion (I and Q Channel, RF Input = 0 dBm,
Balun and Cable Losses of Approximately 2 dB Not Accounted for in Plot)
–145
14
–146
12
–147
10
TEMP = +25°C
OP1dB (dBm)
–148
–149
TEMP = +85°C
8
6
4
–150
04699-0-011
–151
–152
700
750
800
850
900
RF FREQUENCY (MHz)
950
Figure 11. Output Noise Floor vs. Frequency, Maximum Gain,
No RF Carrier, Phase Setpoint = 0°
2
0
700
1000
04699-0-014
NOISE (dBm/Hz)
TEMP = –40°C
750
800
850
900
FREQUENCY (MHz)
950
1000
Figure 14. Output 1dB Compression Point vs. Frequency and Temperature,
Maximum Gain, Phase Setpoint = 0°
Rev. 0 | Page 7 of 20
AD8340
30
30
28
TEMP = –40°C
24
GAIN SETPOINT = 0.5
20
OIP3 (dBm)
22
TEMP = +85°C
20
TEMP = +25°C
18
15
10
16
14
10
700
750
800
850
900
FREQUENCY (MHz)
950
0
0
1000
Figure 15. Output IP3 vs. Frequency and Temperature, Maximum Gain, I Only
04699-0-018
04699-0-015
12
GAIN SETPOINT = 0.1
5
45
90
135
180
225
270
PHASE SETPOINT (Degrees)
RBW 30kHz
VBW 30kHz
SWT 100ms
0
0
–5
360
REF LVL
0 dBm
RF ATT
20dB
MIXER –10dBm
UNIT
dBm
A
200mV p-p BB INPUT
–30
0
50
100
150
200
250
FREQUENCY (MHz)
300
350
–40
–50
–60
–70
–80
–90
400
–100
Figure 16. I/Q Modulation Bandwidth vs. Baseband Magnitude
CENTER 880 MHz
500 kHz/
1 RM
04699-0-019
–25
–30
RF FEEDTHROUGH
–20
SECOND BASEBAND HARMONIC
–15
OUTPUT POWER (dBm)
–20
500mV p-p BB INPUT
DESIRED SIDEBAND
–10
SECOND BASEBAND HARMONIC
–10
1V p-p BB INPUT
04699-0-016
RF OUTPUT AM SIDEBAND POWER (dBm)
315
Figure 18. Output IP3 vs. Gain and Phase Setpoints, 2.5 MHz Carrier Spacing
UNDESIRED SIDEBAND
OIP3 (dBm)
GAIN SETPOINT = 1.0
25
26
SPAN 5 MHz
FREQUENCY (MHz)
Figure 19. Single-Sideband Performance, 880 MHz, −10 dBm RF Input;
1 MHz, 500 mV p-p Differential BB Drive
90
14
GAIN SETPOINT = 1.0
12
60
120
10
8
GAIN SETPOINT = 0.5
30
150
4
2
0
180
–2
0
1.5GHz
500MHz
–4
1.5GHz
–6
–10
–12
0
45
90
225
180
270
135
PHASE SETPOINT (Degrees)
210
315
330
500MHz
360
240
300
270
Figure 17. Output 1dB Compression Point vs. Gain and Phase Setpoints
IMPEDANCE CIRCLE
S11 RF PORT WITH 5.6nH INDUCTORS
S11 RF PORT WITHOUT INDUCTORS
04699-0-020
GAIN SETPOINT = 0.1
–8
04699-0-017
OP1dB (dBm)
6
Figure 20. Input and Output Impedance Smith Chart (with Frequency Markers)
Rev. 0 | Page 8 of 20
AD8340
0
90
–5
60
120
1.5GHz
30
500MHz
0
180
500MHz
1.5GHz
210
–15
–20
–25
–30
–35
–40
330
04699-0-024
150
RF OUTPUT POWER (dBm)
–10
–45
–50
0
240
0.5
1.0
1.5
300
2.0
2.5
3.0
3.5
DSOP VOLTAGE (V)
4.0
4.5
5.0
270
Figure 24. Power Shutdown Attenuation
04699-0-021
IMPEDANCE CIRCLE
S22 PORT WITH 1 TO 1 TRANSFORMER
SDD22 PORT DIFFERENTIAL
Figure 21. Output Impedance Smith Chart (with Frequency Markers)
6
TEK FAST ACQ SAMPLE
4
PHASE SETPOINT = 45°
PHASE SETPOINT = 0°
0
2V/DIV
–2
CHAN 1/3 (V)
PHASE ERROR (Degrees)
2
–4
–6
–8
DSOP
200mV/DIV
3
PHASE SETPOINT = 90°
04699-0-022
–10
–14
0
0.1
0.2
0.3
0.4
0.5
0.6
GAIN SETPOINT
0.7
0.8
0.9
RF OUTPUT
1.0
CH1 200mV Ω
CH3 2.0V
04699-0-025
–12
M 10.0ns 5.0GS/s
A CH2 160mV
ET 200ps/pt 74.0ns
TIME (10ns/DIV)
Figure 22. Phase Error vs. Gain Setpoint by Phase Setpoint, 5 V dc, 25°C, 880 MHz
Figure 25. Power Shutdown Response Time
135
134
132
5.25V
131
130
129
5V
128
127
04699-0-023
SUPPLY CURRENT (mA)
133
4.75V
126
125
–40 –30 –20 –10
0
10
20
30
40
50
60
70
80
TEMPERATURE (°C)
Figure 23. Supply Current vs. Temperature
Rev. 0 | Page 9 of 20
AD8340
THEORY OF OPERATION
By controlling the relative amounts of I and Q components that
are summed, continuous magnitude and phase control of the
gain is possible. Consider the vector gain representation of the
AD8340 expressed in polar form in Figure 27. The attenuation
factors for the I and Q signal components are represented on
the x- and y-axis, respectively, by the baseband inputs, VBBI and
VBBQ. The resultant of their vector sum represents the vector
gain, which can also be expressed as a magnitude and phase. By
applying different combinations of baseband inputs, any vector
gain within the unit circle can be programmed.
Pure amplitude modulation is represented by radial movement
of the gain vector tip at a fixed angle, while pure phase modulation is represented by rotation of the tip around the circle at a
fixed radius. Unlike traditional I-Q modulators, the AD8340 is
designed to have a linear RF signal path from input to output.
Traditional I-Q modulators provide a limited LO carrier path
through which any amplitude information is removed.
VBBI
I CHANNEL INPUT
LINEAR
ATTENUATOR
V-I
SINGLE-ENDED OR
DIFFERENTIAL
50Ω INPUT Z
0°/90°
SINGLE-ENDED OR
DIFFERENTIAL
50Ω OUTPUT
I-V
V-I
OUTPUT
DISABLE
LINEAR
ATTENUATOR
04699-0-026
The AD8340 is a linear RF vector modulator with Cartesian
baseband controls. In the simplified block diagram given in
Figure 26, the RF signal propagates from the left to the right
while baseband controls are placed above and below. The RF
input is first split into in-phase (I) and quadrature (Q) components. The variable attenuators independently scale the I and Q
components of the RF input. The attenuator outputs are then
summed and buffered to the output.
Q CHANNEL INPUT
VBBQ
Figure 26. Simplified Architecture of the AD8340
A change in sign of VBBI or VBBQ can be viewed as a change in
sign of the gain or as a 180° phase change. The outermost
circle represents the maximum gain magnitude of unity. The
circle origin implies, in theory, a gain of 0. In practice, circuit
mismatches and unavoidable signal feedthrough limit the
minimum gain to approximately −40 dB. The phase angle
between the resultant gain vector and the positive x-axis is defined as the phase shift. Note that there is a nominal, systematic
insertion phase through the AD8340 to which the phase shift is
added. In the following discussions, the systematic insertion
phase is normalized to 0°.
Vq
MAX GAIN = 0dB
+0.5
A
|A|
+0.5
MIN GAIN < –30dB
–0.5
The correspondence between the desired gain and phase setpoints, GainSP and PhaseSP, and the Cartesian inputs, VBBI and
VBBQ, is given by simple trigonometric identities
GainSP =
[(V
BBI
/ VO )2 + (VBBQ /VO )2
]
PhaseSP = arctan(VBBQ /VBBI )
where:
VO is the baseband scaling constant (500 mV).
VBBI and VBBQ are the differential I and Q baseband voltages,
respectively.
Note that when evaluating the arctangent function, the proper
phase quadrant must be selected. For example, if the principal
value of the arctangent (known as the Arctangent(x)) is used,
quadrants 2 and 3 would be interpreted mistakenly as quadrants
4 and 1, respectively. In general, both VBBI and VBBQ are needed
in concert to modulate the gain and the phase.
Vi
04699-0-027
θ
–0.5
Figure 27. Vector Gain Representation
RF QUADRATURE GENERATOR
The RF input is directly coupled differentially or single-ended
to the quadrature generator, which consists of a multistage RC
polyphase network tuned over the operating frequency range of
700 MHz to 1000 MHz. The recycling nature of the polyphase
network generates two replicas of the input signal, which are in
precise quadrature, i.e., 90°, to each other. Since the passive
network is perfectly linear, the amplitude and phase information
contained in the RF input is transmitted faithfully to both channels. The quadrature outputs are then separately buffered to
drive the respective attenuators. The characteristic impedance
of the polyphase network is used to set the input
impedance to the AD8340.
Rev. 0 | Page 10 of 20
AD8340
I-Q ATTENUATORS AND BASEBAND AMPLIFIERS
GAIN AND PHASE ACCURACY
The proprietary linear-responding attenuator structure is an
active solution with differential inputs and outputs that offer
excellent linearity, low noise, and greater immunity from mismatches than other variable attenuator methods. The gain, in
linear terms, of the I and Q channels is proportional to its control
voltage with a scaling factor designed to be 2/V, i.e., a full-scale
gain setpoint of 1.0 (−2 dB) for VBBI (Q) of 500 mV. The control
voltages can be driven differentially or single-ended. The combination of the baseband amplifiers and attenuators allows for
maximum modulation bandwidths in excess of 200 MHz.
There are numerous ways to express the accuracy of the
AD8340. Ideally, the gain and phase should precisely follow the
setpoints. Figure 3 illustrates the gain error in dB from a best fit
line, normalized to the gain measured at the gain setpoint = 1.0,
for the different phase setpoints. Figure 6 shows the gain error
in a different form; the phase setpoint is swept from 0° to 360°
for different gain setpoints. Figure 8 and Figure 22 show analogous errors for the phase error as a function of gain and phase
setpoints. The accuracy clearly depends on the region of operation within the vector gain unit circle. Operation very close to
the origin generally results in larger errors as the relative
accuracy of the I and Q vectors degrades.
OUTPUT AMPLIFIER
The output amplifier accepts the sum of the attenuator outputs
and delivers a differential output signal into the external load.
The output pins must be pulled up to an external supply,
preferably through RF chokes. When the 50 Ω load is taken
differentially, an output P1dB and IP3 of 11 dBm and 24 dBm is
achieved, respectively, at 880 MHz. The output can be taken in
single-ended fashion, albeit at lower performance levels.
NOISE AND DISTORTION
The output noise floor and distortion levels vary with the gain
magnitude but do not vary significantly with the phase. At the
higher gain magnitude setpoints, the OIP3 and the noise floor
vary in direct proportion with the gain. At lower gain magnitude setpoints, the noise floor levels off while the OIP3
continues to vary with the gain.
RF FREQUENCY RANGE
The frequency range on the RF input is limited by the internal
polyphase quadrature phase-splitter. The phase-splitter splits
the incoming RF input into two signals, 90° out of phase, as
previously described in the RF Quadrature Generator section.
This polyphase network has been designed to ensure robust
quadrature accuracy over standard fabrication process parameter variations for the 700 MHz to 1 GHz specified RF frequency
range. Using the AD8340 as a single-sideband modulator and
measuring the resulting sideband suppression is a good gauge
of how the quadrature accuracy is maintained over RF
frequency. A typical plot of sideband suppression from
500 MHz to 1.5 GHz is shown in Figure 28. The level of sideband suppression degradation outside the 700 MHz to 1 GHz
specified range will be subject to manufacturing process
variations.
0
SB SUPPRESSION (dBc)
–5
–10
–15
–20
–25
–35
500
04699-0-028
–30
600
700
800
900
1000 1100 1200 1300 1400 1500
FREQUENCY (MHz)
Figure 28. Sideband Suppression vs. Frequency
Rev. 0 | Page 11 of 20
AD8340
APPLICATIONS
loss of >10 dB over the operating frequency range. Different
matching inductors can improve matching over a narrower
frequency range. The single-ended and differential input
impedances are exactly the same.
USING THE AD8340
The AD8340 is designed to operate in a 50 Ω impedance
system. Figure 30 illustrates an example where the RF input is
driven in a single-ended fashion while the differential RF output is converted to a single-ended output with a RF balun. The
baseband controls for the I and Q channels are typically driven
from differential DAC outputs. The power supplies, VPRF and
VPS2, should be bypassed appropriately with 0.1 µF and 100 pF
capacitors. Low inductance grounding of the CMOP and CMRF
common pins is essential to prevent unintentional peaking of
the gain.
100pF 5.6nH
RFIM
~1VDC
RC
PHASE
RF
04699-0-029
100pF 5.6nH
RFIP
50Ω
Figure 29. RF Input Interface to the AD8340 Showing
Coupling Capacitors and Matching Inductors
RF INPUT AND MATCHING
The RFIP and RFIM should be ac-coupled through low loss
series capacitors as shown in Figure 29. The internal dc levels
are at approximately 1 V. For single-ended operation, one input
is driven by the RF signal while the other input is ac grounded.
The input impedance of the AD8340 is defined by the characteristics of the polyphase network. The capacitive component of
the network causes its impedance to roll-off with frequency
albeit at a slower rate than 6 dB/octave. By using matching
inductors on the order of 5.6 nH in series with each of the RF
inputs, RFIP and RFIM, a 50 Ω match is achieved with a return
VP
C2
100pF
C1
0.1µF
IBBM
VP
IBBP
C12
(SEE TEXT)
C6
100pF
VPS2
OUTPUT
DISABLE
DSOP
CMOP
CMRF
CMOP
RFIM
RFOM
C17
100pF
AD8340
RFOP
CMRF
CMOP
VPRF
C3
0.1µF
C4
100pF
QFLP
VPS2
VP
RFIP
QBBM
L4
5.6nH
QBBP
C5
100pF
QFLM
RF
INPUT
L3
5.6nH
IFLP
VPRF
IBBM
VP
IBBP
C7
100pF
B
IFLM
C8
0.1µF
A
L1
120nH
ETC1-1-13
RF
OUTPUT
C18
L2
100pF
120nH
CMOP
VPS2
C14
0.1µF
C10
0.1µF
QBBP
QBBM
C9
100pF
Figure 30. Basic Connections
Rev. 0 | Page 12 of 20
04699-0-030
VP
C11
(SEE TEXT)
AD8340
–0.5
RF OUTPUT AND MATCHING
–1.0
The RF outputs of the AD8340, RFOP and RFOM, are open
collectors of a transimpedance amplifier which need to be
pulled up to the positive supply, preferably with RF chokes as
shown in Figure 31. The nominal output impedance looking
into each individual output pin is 25 Ω. Consequently, the
differential output impedance is 50 Ω.
RL2 = SHORT
–1.5
–2.0
GAIN (dB)
–2.5
–3.0
RL2 = 50Ω
–3.5
–4.0
VP
–5.0
120nH
RT
RFOM
±ISIG
GM
–5.5
–6.0
700
100pF
1:1
100pF
RF
OUTPUT
04699-0-031
RT
1000
Figure 32. Gain of the AD8340 Using a Single-Ended Output with Different Dummy
Loads, RL2 on the Unused Output
RFOP
50Ω
DIFFERENTIAL
RL2 = OPEN
800
900
FREQUENCY (MHz)
04699-0-032
–4.5
Figure 31. RF Output Interface to the AD8340 Showing
Coupling Capacitors, Pull-Up RF Chokes, and Balun
Since the output dc levels are at the positive supply, ac coupling
capacitors will usually be needed between the AD8340 outputs
and the next stage in the system.
A 1:1 RF broadband output balun, such as the ETC1-1-13
(M/A-COM), converts the differential output of the AD8340
into a single-ended signal. Note that the loss and balance of the
balun directly impact the apparent output power, noise floor,
and gain/phase errors of the AD8340. In critical applications,
narrow-band baluns with low loss and superior balance are
recommended.
If the output is taken in a single-ended fashion directly into a
50 Ω load through a coupling capacitor, there will be an impedance mismatch. This can be resolved with a 1:2 balun to convert
the single-ended 25 Ω output impedance to 50 Ω. If loss of
signal swing is not critical, a 25 Ω back termination in series
with the output pin can also be used. The unused output pin
must still be pulled up to the positive supply. The user may load
it through a coupling capacitor with a dummy load to preserve
balance. The gain of the AD8340 when the output is singleended varies slightly with dummy load value as shown in Figure 32.
The RF output signal can be disabled by raising the DSOP pin
to the positive supply. The shutdown function provides >40 dB
attenuation of the input signal even at full gain. The interface
to DSOP is high impedance and the shutdown and turn-on
response times are <100 ns. If the disable function is not
needed, the DSOP should be tied to ground.
DRIVING THE I-Q BASEBAND CONTROLS
The I and Q inputs to the AD8340 set the gain and phase between input and output. These inputs are differential and should
normally have a common-mode level of 0.5 V. However, when
differentially driven, the common mode can vary from 250 mV
to 750 mV while still allowing full gain control. Each input pair
has a nominal input swing of ±0.5 V differential around the
common-mode level. The maximum gain of unity is achieved if
the differential voltage is equal to +500 mV or −500 mV. So
with a common-mode level of 500 mV, IBBP and IBBM will
each swing between 250 mV and 750 mV.
The I and Q inputs can also be driven with a single-ended
signal. In this case, one side of each input should be tied to a
low noise 0.5 V voltage source (a 0.1 µF decoupling capacitor
located close to the pin is recommended), while the other input
swings from 0 V to 1 V. Differential drive generally offers superior even-order distortion and lower noise than single-ended
drive.
The bandwidth of the baseband controls exceeds 200 MHz even
at full-scale baseband drive. This allows for very fast gain and
phase modulation of the RF input signal. In cases where lower
modulation bandwidths are acceptable or desired, external filter
capacitors can be connected across Pins IFLP to IFLM and
QFLP to QFLM to reduce the ingress of baseband noise and
spurious signal into the control path.
Rev. 0 | Page 13 of 20
f3dB ≈
45 kHz × 10 nF
C external + 0.5 pF
This equation has been verified for values of CFLT from 10 pF to
0.1 µF (bandwidth settings of approximately 4.5 kHz to 43 MHz).
INTERFACING TO HIGH SPEED DACs
The AD977x family of dual DACs is well suited to driving the I
and Q vector controls of the AD8340. While these inputs can in
general be driven by any DAC, the differential outputs and bias
level of the ADI TxDAC® family allows for a direct connection
between DAC and modulator.
The AD977x family of dual DACs have differential current outputs. The full-scale current is user programmable and is usually
set to 20 mA, that is, each output swings from 0 mA to 20 mA.
The basic interface between the AD9777 DAC outputs and the
AD8340 I and Q inputs is shown in Figure 33. The Resistors R1
and R2 set the dc bias level according to the equation:
Bias Level = Average Output Current × R1
For example, if the full-scale current from each output is 20 mA,
each output will have an average current of 10 mA. Therefore to
set the bias level to the recommended 0.5 V, R1 and R2 should
be set to 50 Ω each. R1 and R2 should always be equal.
If R3 is omitted, this will result in an available swing from
the DAC of 2 V p-p differential, which is twice the maximum
voltage range required by the AD8340. DAC resolution can be
maximized by adding R3, which scales down this voltage
according to the following equation:
Full Scale Swing =
Figure 34. Peak-to-Peak DAC Output Swing vs.
Swing Scaling Resistor R3 (R1 = R2 = 50 Ω)
Figure 34 shows the relationship between the value of R3 and
the peak baseband voltage with R1 and R2 equal to 50 Ω.
From Figure 34, it can be seen that a value of 100 Ω for R3 will
provide a peak-to-peak swing of 1 V p-p differential into the
AD8340’s I and Q inputs.
When using a DAC, low-pass image reject filters are typically
used to eliminate the Nyquist images produced by the DAC.
They also provide the added benefit of eliminating broadband
noise that might feed into the modulator from the DAC.
CDMA2000 APPLICATION
To test the compliance to the CDMA2000 base station standard,
a single-carrier CDMA2000 test model signal (forward pilot,
sync, paging, and six traffic as per 3GPP2 C.S0010-B, Table
6.5.2.1) was applied to the AD8340. A cavity tuned filter was
used to reduce noise from the signal source being applied to the
device. The 4.6 MHz pass band of this filter is apparent in the
subsequent spectral plots.
Figure 35 shows a plot of the spectrum of the output signal under nominal conditions. POUT is equal to −5 dBm and VI = VQ =
0.353 V, i.e., VIBBP − VIBBM = VQBBP − VQBBM = 0.353 V.
Adjacent channel power is measured in 30 kHz resolution
bandwidth at 750 KHz and 1.98 MHz carrier offset. Noise floor
is measured at ±4 MHz carrier offset.
R2 ⎤
⎡
2 × I MAX (R1 || (R2 + R3)) × ⎢1 −
R2
+ R3 ⎥⎦
⎣
AD9777
1.15
1.13
1.10
1.08
1.05
1.02
1.00
0.97
0.95
0.92
0.90
0.88
0.85
0.82
0.80
0.77
0.75
0.72
0.70
50 55 60 65 70 75 80 85 90 95 100 105 110 115 120 125 130
R3
AD8340
IBBP
IOUTA1
R1
R2
OPTIONAL
LOW-PASS
FILTER
R3
IOUTB1
IBBM
IOUTA2
QBBP
R2
OPTIONAL
LOW-PASS
FILTER
R3
IOUTB2
QBBM
04699-0-033
R1
04699-0-034
The 3 dB bandwidth is set by choosing CFLT according to the
following equation:
DIFFERENTIAL PEAK-TO-PEAK SWING (R3)
AD8340
Figure 33. Basic AD9777 to AD8340 Interface
Rev. 0 | Page 14 of 20
*RBW 30kHz
*VBW 30kHz
*SWT 100ms
OFFSET 0.5 dB
CH PWR
ACP LOW
ACP UP
ALT1 LOW
ALT1 UP
–20
–30
10
MARKER 2 [T1 NOI]
–148.76dBm/Hz
876.009615385MHz
0
–5.17dBm
–60.94dB
–60.08dB
–86.40dB
–86.80dB
LVL
MARKER 1 [T1 NOI]
–148.89dBm/Hz
884.006410256MHz
–40
–50
NOR
–60
SWP 50 OF 50
–70
04699-0-035
1
2
–110
–60
ACP – 750kHz OFFSET, 30kHz RBW
–20
CENTER 880MHz
1MHz/
–40
–75
ACP – 1.98MHz OFFSET, 30kHz RBW
–80
NOISE – 4MHz OFFSET, 1MHz RBW
–40
–40
–50
–50
–60
–60
ACP – 750kHz OFFSET, 30kHz RBW
–70
–70
ACP – 1.98MHz OFFSET, 30kHz RBW
–80
–90
–90
NOISE – 4MHz OFFSET, 1MHz RBW
–20
–15
–10
–5
OUTPUT POWER (dBm)
50
100
150
200 250 300
VI = VQ =VIN (mV)
350
400
450
–90
500
Figure 37. Output Power, Noise, and ACP vs. I and Q Control Voltages, CDMA2000 Test
Model, VI = VQ, ACP Measured in 30 kHz RBW at ±750 kHz and ±1.98 kHz Carrier Offset,
Noise Measured at ±4 MHz Carrier Offset
0
5
–100
04699-0-036
–30
NOISE – dBm @ 4MHz CARRIER OFFSET (1MHz RBW)
–30
–25
0
–85
SPAN 10MHz
Holding the I and Q control voltages steady at 0.353 V, input
power was swept. Figure 36 shows the resulting output power,
noise floor, and adjacent channel power ratio. Noise floor is
presented as noise in a 1 MHz bandwidth as defined by the
3GPP2 specification.
–80
–65
–70
–30
–70
Figure 35. Output Spectrum, Single-Carrier CDMA2000 Test Model at −5 dBm,
VI = VQ = 0.353 V, ACP Measured at 750 kHz and 1.98 KHz Carrier Offset,
Noise Measured at ±4 MHz Carrier Offset, Input Signal Filtered Using a
Cavity Tuned Filter (Pass Band = 4.6 MHz)
ACP – dBc (30kHz RBW)
–10
–60
–90
–100
–30
–55
–50
–80
–100
–50
POUT vs. VIN
04699-0-037
*ATT 5dB
In contrast to Figure 36, Figure 37 shows that for a fixed input
power, ACP remains fairly constant as gain and phase are
changed (this is not true for very high input powers). The noise
floor still drops with decreasing gain, but it never reaches the
−90 dBm level in Figure 37.
Figure 38 shows the output spectrum for a 3-carrier
CDMA2000 spectrum. Again, the signal being applied to the
AD8340 is filtered by a cavity-tuned filter with a −3 dB bandwidth of 4.6 MHz. To reduce distortion, the total output carrier
power has been reduced to approximately −8 dBm (per-carrier
power = −12.6 dBm). Adjacent channel power ratios of −61 dBc
(2 MHz from center of spectrum) and −82 dBc (3.23 MHz from
center of spectrum) were measured. The noise floor, measured
at 5.25 MHz carrier offset, is approximately −149 dBm/Hz
(−89 dBm in a 1 MHz bandwidth). So while some dynamic
range has been lost due to output power back-off, ACP stays
approximately equal and noise floor improves slightly.
REF –15 dBm
–20
Figure 36. Noise and ACP vs. Output Power, Single-Carrier CDMA2000 Test Model,
VI = VQ = 0.353, ACP Measured in 30 kHz RBW at ±750 kHz and ±1.98 KHz Carrier Offset,
Noise Measured at ±4 MHz Carrier Offset
*ATT 5dB
*RBW 30kHz
*VBW 300kHz
*SWT 5s
OFFSET 0.5dB
CH1
CH2
CH3
TOTAL
ACP LOW
ACP UP
ALT1 LOW
ALT1 UP
–30
–40
The results show that at an output power of +3 dBm, ACP is still
in compliance with the standard (<−45 dBc @ 750 MHz and
<−60 dBc @ 1.98 MHz). At low output power levels, ACP at
1.98 MHz carrier offset degrades as the noise floor of the
AD8340 becomes the dominant contributor to measured ACP.
Measured noise at 4 MHz carrier offset begins to increase
sharply above 0 dBm output power. This increase is not due to noise
but results from increased carrier-induced distortion. As output
power drops below 0 dBm, the noise floor drops towards −90 dBm.
With a fixed input power of 2.4 dBm, the output power was
again swept by exercising the I and Q inputs. VI and VQ were
kept equal and were swept from 10 mV to 500 mV. The resulting output power, ACP, and noise floor are shown in Figure 37.
MARKER 1 [T1 NOI]
–148.83dBm/Hz
885.252403846MHz
–50
–12.65dBm
–12.58dB
A
–12.87dB
SOL
–7.93dB
–61.41dB
–61.87dB
LVL
–82.36dB
–81.92dB
NOR
–60
–70
–80
–90
1
–100
–110
CENTER 880MHz
1.5MHz/
04699-0-038
REF –12dBm
OUTPUT POWER (dBm)
BS, 1X, C0 : ADJ CHANNEL
ACP – dBc (30kHz RBW)
NOISE – 4MHz CARRIER OFFSET – dBm (1MHz RBW)
AD8340
SPAN 15MHz
Figure 38. Output Spectrum, 3-Carrier CDMA2000 Test Model at −12.5 dBm/Carrier,
VI = VQ = 0.353 V, ACP Measured at 2 MHz and 3.23 KHz Offset from Center of
Spectrum, Noise Measured at 5.25 MHz Carrier Offset, Input Signal Filtered
Using a Cavity Tuned Filter (Pass Band = 4.6 MHz)
Rev. 0 | Page 15 of 20
AD8340
EVALUATION BOARD
The evaluation board circuit schematic for the AD8340 is
shown in Figure 39.
The evaluation board is configured to be driven from a
single-ended 50 Ω source. Although the input of the AD8340 is
differential, it may be driven single-ended, with no loss of
performance.
The low-pass corner frequency of the baseband I and Q channels can be reduced by installing capacitors in the C11 and C12
positions. The low-pass corner frequency for either channel is
approximated by
f 3dB
45 kHz × 10 nF
≈
C external + 0.5 pF
On this evaluation board, the I and Q baseband circuits are
identical to each other, so the following description applies
equally to each. The connections and circuit configuration for
the Q baseband inputs are described in Table 4.
The baseband input of the AD8340 requires a differential voltage drive. The evaluation board is set up to allow such a drive by
connecting the differential voltage source to QBBP and QBBM.
The common-mode voltage should be maintained at approximately 0.5 V. For this configuration, Jumpers W1 to W4 should
be removed.
The baseband input of the evaluation board may also be driven
with a single-ended voltage. In this case, a bias level is provided
to the unused input from Potentiometer R10 by installing either
W1 or W2.
Setting SW1 in Position B disables the AD8340 output amplifier.
With SW1 set to Position A, the output amplifier is enabled.
With SW1 set to Position A, an external voltage signal, such as a
pulse, can be applied to the DSOP SMA connector to exercise
the output amplifier enable/disable function.
Rev. 0 | Page 16 of 20
AD8340
Table 4. Evaluation Board Configuration Options
Components
R7, R9, R11,
R14, R15, R19,
R20, R21, C15,
C19, W3, W4
Function
I Channel Baseband Interface. Resistors R7 and R9 may be installed to accommodate a
baseband source that requires a specific terminating impedance. Capacitors C15 and C19
are bypass capacitors.
For single-ended baseband drive, the Potentiometer R11 can be used to provide a bias level
to the unused input (install either W3 or W4).
R1, R3, R10,
R12, R13, R16,
R17, R18, C16,
C20, W1, W2
Q Channel Baseband Interface. See the I Channel Baseband Interface section.
C11, C12
Baseband Low-Pass Filtering. By adding Capacitor C11 between QFLP and QFLM, and C12
between IFLP and IFLM, the 3 dB low-pass corner frequency of the baseband interface can
be reduced from 230 MHz (nominal). See equation in text.
Output Interface. The 1:1 balun transformer, T1, converts the 50 Ω differential output to
50 Ω single-ended. C17 and C18 are dc blocks. L1 and L2 provide dc bias for the output.
T1, C17, C18,
L1, L2
L3, L4, C5, C6
C2, C4, C7,
C9, C14, C1,
C3, C8, C10,
R2, R4, R5, R6
R8, SW1
Input Interface. The input impedance of the AD8340 requires 5.6 nH inductors in series
with RFIP and RFIM for optimum return loss when driven by a single-ended 50 Ω line. C5
and C6 are dc blocks.
Supply Decoupling.
Output Disable Interface. The output stage of the AD8340 is disabled by applying a high
voltage to the DSOP pin by moving SW1 to Position B. The output stage is enabled moving
SW1 to Position A. The output disable function can also be exercised by applying an external high or low voltage to the DSOP SMA connector with SW1 in Position A.
Rev. 0 | Page 17 of 20
Default Conditions
R7, R9 = Not Installed
R11 = Potentiometer, 2 kΩ,
10 Turn (Bourns)
R14 = 4 kΩ (Size 0603)
R15 = 44 kΩ (Size 0603)
R19, R20, R21 = 0 Ω
(Size 0603)
C15, C19 = 0.1 µF
(Size 0603)
W3 = Jumper (Installed)
W4 = Jumper (Open)
R1, R3 = Not Installed
R10 = Potentiometer, 2 kΩ,
10 Turn (Bourns)
R12 = 4 kΩ (Size 0603)
R13 = 44 kΩ (Size 0603)
R16, R17, R18 = 0 Ω
(Size 0603)
C16, C20 = 0.1 µF
(Size 0603)
W1 = Jumper (Installed)
W2 = Jumper (Open)
C11, C12 = Not Installed
C17, C18 = 100 pF
(Size 0603)
T1 = ETC1-1-13 (M/A-COM)
L1, L2 = 120 nH
(Size 0603)
L3, L4 = 5.6 nH (Size 0402)
C5, C6 = 100 pF (Size 0603)
C2, C4, C7, C9, C14 = Open
(Size 0603)
C1, C3, C8, C10 = 0.1 µF
(Size 0603)
R2, R4, R5, R6 = 0 Ω
(Size 0603)
R8 = 10 kΩ (Size 0603)
SW1 = SPDT (Position A,
Output Enabled)
AD8340
IBBP
IBBM
C19
R7
0.1µF (OPEN)
R9
(OPEN)
W4
R21
0Ω
VP
TEST POINT
C2
(OPEN)
R20
0Ω
C15
0.1µF
R2
0Ω
VS
R14
4kΩ
GND
TEST POINT
R19
0Ω
W3
R11
2kΩ
R15
44kΩ
C1
0.1µF
C12
(OPEN)
C6
100pF
VPS2
IBBM
IFLP
VPRF
IBBP
VP
C8
0.1µF
R5
0Ω
IFLM
C7
(OPEN)
SW1
CMRF
L3
5.6nH
R8
10kΩ
DSOP
CMOP
CMOP
RFIM
B
A
DSOP
C17
100pF
T1
ETC1-1-13
M/A-COM
RFOP
RFOM
AD8340
RFOP
CMRF
CMOP
VPRF
R4
0Ω
C3
0.1µF
QFLP
L1
120nH
C18
L2
100pF
120nH
CMOP
C14
0.1µF
VPS2
VP
C11
(OPEN)
R6
0Ω
C10
0.1µF
R12
4kΩ
R10
2kΩ
C9
(OPEN)
R13
44kΩ
VS
C16
0.1µF
W2
R17
0Ω
R1
OPEN
QBBP
W1
R16
0Ω
R18
0Ω
R3
C20 OPEN
0.1µF
QBBM
Figure 39. Evaluation Board Schematic
Rev. 0 | Page 18 of 20
04699-0-039
C4
(OPEN)
VPS2
VP
RFIP
QBBM
L4
5.6nH
QBBP
C5
100pF
QFLM
RFIN
04699-0-041
04699-0-040
AD8340
Figure 41. Component Side Silkscreen
Figure 40. Component Side Layout
Rev. 0 | Page 19 of 20
AD8340
OUTLINE DIMENSIONS
0.60 MAX
4.00
BSC SQ
PIN 1
INDICATOR
0.60 MAX
TOP
VIEW
3.75
BSC SQ
0.50
BSC
0.50
0.40
0.30
1.00
0.85
0.80
12° MAX
PIN 1
INDICATOR
19
18
24 1
2.25
2.10 SQ
1.95
BOTTOM
VIEW
13
12
7
6
0.25 MIN
2.50 REF
0.80 MAX
0.65 TYP
0.05 MAX
0.02 NOM
SEATING
PLANE
0.30
0.23
0.18
0.20 REF
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VGGD-2
Figure 42. 24-Lead Lead Frame Chip Scale Package [LFCSP]
(CP-24)
Dimensions shown in millimeters
ORDERING GUIDE
Models
AD8340ACPZ-WP1, 2
AD8340ACPZ-REEL71
AD8340-EVAL
1
2
Temperature Range
−40°C to +85°C
−40°C to +85°C
Package Description
24-Lead Lead Frame Chip Scale Package (LFCSP)
24-Lead Lead Frame Chip Scale Package (LFCSP)
Evaluation Board
Z = Pb-free part.
WP = Waffle pack.
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners.
D04699–0–6/04(0)
Rev. 0 | Page 20 of 20
Package Option
CP-24
CP-24
Order Multiple
64
1,500
1