LTC3863 60V Low IQ Inverting DC/DC Controller DESCRIPTION FEATURES n n n n n n n n n n n n n n Wide Operating VIN Range: 3.5V to 60V Wide Negative VOUT Range: –0.4V to Beyond –150V Low Operating IQ = 70µA Strong High Voltage MOSFET Gate Driver Constant Frequency Current Mode Architecture Verified FMEA for Adjacent Pin Open/Short Selectable High Efficiency Burst Mode® Operation or Pulse-Skipping Mode at Light Loads Programmable Fixed Frequency: 50kHz to 850kHz Phase-Lockable Frequency: 75kHz to 750kHz Accurate Current Limit Programmable Soft-Start or Voltage Tracking Internal Soft-Start Guarantees Smooth Start-Up Low Shutdown IQ = 7µA Available in Small 12-Lead Thermally Enhanced MSOP and DFN Packages The LTC®3863 is a robust, inverting DC/DC PMOS controller optimized for automotive and industrial applications. It drives a P-channel power MOSFET to generate a negative output and requires just a single inductor to complete the circuit. Output voltages from –0.4V to –150V are typically achievable with higher voltages possible, only limited by external components. The LTC3863 offers excellent light load efficiency, drawing only 70μA quiescent current in a user programmable Burst Mode operation. Its peak current mode, constant frequency PWM architecture provides for good control of switching frequency and output current limit. The switching frequency can be programmed from 50kHz to 850kHz with an external resistor and can be synchronized to an external clock from 75kHz to 750kHz. The LTC3863 offers programmable soft-start or output tracking. Safety features include overvoltage, overcurrent and short-circuit protection including frequency foldback. APPLICATIONS n n n Industrial and Automotive Power Supplies Telecom Power Supplies Distributed Power Systems The LTC3863 is available in thermally enhanced 12-lead MSOP and 3mm × 4mm DFN packages. L, LT, LTC, LTM, OPTI-LOOP, Linear Technology, Burst Mode and the Linear logo are registered trademarks and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5731694. TYPICAL APPLICATION 4.5V to 16V Input, –5V/1.7A Output, 350kHz Inverting Converter CAP VIN SENSE SS 14.7k 61.9k GATE ITH Si7129 LTC3863 FREQ 10µH B540C 511k 33µF ×2 VOUT –5V 150µF 1.7A 16V ×2 PGND VFB 9 70 7 VIN = 12V 80 VOUT = –5V SGND VFBN 90 60 80.6k 3863 TA01a EFFICIENCY 6 50 40 30 5 4 PULSE-SKIPPING MODE Burst Mode OPERATION 3 POWER LOSS 1 20 10 68pF 8 0 0.002 POWER LOSS (W) 16mΩ PLLIN/MODE 27nF 100µF 20V + 350kHz RUN + EFFICIENCY (%) 10µF 25V ×2 0.47µF Efficiency VIN 4.5V TO 16V 2 0.02 0.2 LOAD CURRENT (A) 2 0 3863 TA01b 3863f For more information www.linear.com/3863 1 LTC3863 ABSOLUTE MAXIMUM RATINGS (Note 1) Input Supply Voltage (VIN).......................... –0.3V to 65V VIN-VSENSE Voltage....................................... –0.3V to 6V VIN-VCAP Voltage......................................... –0.3V to 10V RUN Voltage............................................... –0.3V to 65V VFBN, PLLIN/MODE Voltages........................ –0.3V to 6V SS, ITH, FREQ, VFB Voltages......................... –0.3V to 5V Operating Junction Temperature Range (Notes 2, 3, 4) LTC3863E,I ........................................ –40°C to 125°C LTC3863H........................................... –40°C to 150°C LTC3863MP........................................ –55°C to 150°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) MSOP Package.................................................. 300°C PIN CONFIGURATION TOP VIEW PLLIN/MODE 1 12 GATE FREQ 2 11 VIN SGND 3 10 SENSE SS 4 VFB ITH 13 PGND 9 CAP 5 8 RUN 6 7 VFBN TOP VIEW PLLIN/MODE FREQ SGND SS VFB ITH 1 2 3 4 5 6 13 PGND 12 11 10 9 8 7 GATE VIN SENSE CAP RUN VFBN MSE PACKAGE 12-LEAD PLASTIC MSOP DE PACKAGE 12-LEAD (4mm × 3mm) PLASTIC DFN TJMAX = 150°C, θJA = 43°C/W, θJC = 5.5°C/W EXPOSED PAD (PIN 13) IS PGND, MUST BE SOLDERED TO PCB TJMAX = 150°C, θJA = 40°C/W, θJC = 10°C/W EXPOSED PAD (PIN 13) IS PGND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3863EMSE#PBF LTC3863EMSE#TRPBF 3863 12-Lead Plastic MSOP –40°C to 125°C LTC3863IMSE#PBF LTC3863IMSE#TRPBF 3863 12-Lead Plastic MSOP –40°C to 125°C LTC3863HMSE#PBF LTC3863HMSE#TRPBF 3863 12-Lead Plastic MSOP –40°C to 150°C LTC3863MPMSE#PBF LTC3863MPMSE#TRPBF 3863 12-Lead Plastic MSOP –55°C to 150°C LTC3863EDE#PBF LTC3863EDE#TRPBF 3863 12-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C LTC3863IDE#PBF LTC3863IDE#TRPBF 3863 12-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C LTC3863HDE#PBF LTC3863HDE#TRPBF 3863 12-Lead (4mm × 3mm) Plastic DFN –40°C to 150°C LTC3863MPDE#PBF LTC3863MPDE#TRPBF 3863 12-Lead (4mm × 3mm) Plastic DFN –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3863f 2 For more information www.linear.com/3863 LTC3863 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, unless otherwise noted. (Note 4) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Input Supply VIN Input Voltage Operating Range VUVLO Undervoltage Lockout IQ Input DC Supply Current 3.5 (VIN-VCAP) Ramping Up Threshold (VIN-VCAP) Ramping Down Threshold Hysteresis l l 3.25 3.00 60 V 3.50 3.25 0.25 3.8 3.50 V V V 0.77 1.2 mA Pulse-Skipping Mode PLLIN/MODE = 0V, FREQ = 0V, VFB = 0.83V (No Load) Burst Mode Operation PLLIN/MODE = Open, FREQ = 0V, VFB = 0.83V (No Load) 50 70 µA Shutdown Supply Current RUN = 0V 7 12 µA 0.800 0.809 V 0.005 %/V Output Sensing VREG Regulated Feedback Voltage VREG = (VFB – VFBN) VITH = 1.2V (Note 5) ∆VREG ∆VIN Feedback Voltage Line Regulation VIN = 3.8V to 60V (Note 5) ∆VREG ∆VITH Feedback Voltage Load Regulation VITH = 0.6V to 1.8V (Note 5) gm(EA) Error Amplifier Transconductance VITH = 1.2V, ∆IITH = ±5µA (Note 5) IFBN Feedback Negative Input Bias Current l 0.791 –0.005 –0.1 –0.015 0.1 1.8 % mS –50 –10 50 nA 85 95 103 mV 0.1 2 µA 1.26 1.32 V Current Sensing VILIM Current Limit Threshold (VIN-VSENSE) VFB = 0.77V ISENSE SENSE Pin Input Current VSENSE = VIN l Start-Up and Shutdown VRUN RUN Pin Enable Threshold VRUNHYS RUN Pin Hysteresis ISS Soft-Start Pin Charging Current VRUN Rising l 1.22 VSS = 0V 150 mV 10 µA Switching Frequency and Clock Synchronization f Programmable Switching Frequency RFREQ = 24.9kΩ RFREQ = 64.9kΩ RFREQ = 105kΩ 375 105 440 810 505 kHz kHz kHz fLO Low Switching Frequency FREQ = 0V 320 350 380 kHz fHI High Switching Frequency FREQ = Open 485 535 585 kHz 750 kHz fSYNC Synchronization Frequency l VCLK(IH) Clock Input High Level into PLLIN/MODE l VCLK(LO) Clock Input Low Level into PLLIN/MODE l fFOLD Foldback Frequency as Percentage of Programmable Frequency tON(MIN) Minimum On-Time VFB = 0V, VFREQ = 0V 75 2 V 0.5 V 18 % 220 ns 3863f For more information www.linear.com/3863 3 LTC3863 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, unless otherwise noted. (Note 4) SYMBOL PARAMETER CONDITIONS VCAP Gate Bias LDO Output Voltage (VIN-VCAP) IGATE = 0mA VCAPDROP Gate Bias LDO Dropout Voltage VIN = 5V, IGATE = 15mA MIN TYP MAX UNITS 7.6 8.0 8.5 V 0.2 0.5 V 0.002 0.03 %/V Gate Driver l ∆VCAP(LINE) Gate Bias LDO Line Regulation 9V ≤ VIN ≤ 60V, IGATE = 0mA ∆VCAP(LOAD) Gate Bias LDO Load Regulation Load = 0mA to 20mA –3.5 % RUP Gate Pull-Up Resistance Gate High 2 Ω RDN Gate Pull-Down Resistance Gate Low 0.9 Ω VFB Overvoltage Lockout Threshold GATE Going High without Delay, VFB(OV)-VFB(NOM) in Percent 10 % Overvoltage VFBOV Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Continuous operation above the specified maximum operating junction temperature may impair device reliability or permanently damage the device. Note 3: The junction temperature (TJ in °C) is calculated from the ambient temperature (TA in °C) and power dissipation (PD in Watts) as follows: TJ = TA + (PD • θJA) where θJA (in °C/W) is the package thermal impedance provided in the Pin Configuration section for the corresponding package. Note 4: The LTC3863 is tested under pulsed load conditions such that TJ ≈ TA. The LTC3863E is guaranteed to meet performance specifications from 0°C to 85°C operating junction temperature range. The LTC3863E specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3863I is guaranteed to meet performance specifications over the –40°C to 125°C operating junction temperature range, the LTC3863H is guaranteed over the –40°C to 150°C operating junction temperature range, and the LTC3863MP is guaranteed and tested over the full –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes; operating lifetime is derated for junction temperatures greater than 125°C. The maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. Note 5: The LTC3863 is tested in a feedback loop that adjust VREG or (VFB – VFBN) to achieve a specified error amplifier output voltage (on ITH pin). 3863f 4 For more information www.linear.com/3863 LTC3863 TYPICAL PERFORMANCE CHARACTERISTICS Pulse-Skipping Mode Operation Waveforms TA = 25°C, unless otherwise noted. Burst Mode Operation Waveforms Transient Response: Burst Mode Operation VOUT 50mV/DIV VOUT 50mV/DIV ILOAD 1A/DIV VSW 10V/DIV VSW 10V/DIV VOUT 200mV/DIV IL 500mA/DIV IL 500mA/DIV IL 1A/DIV 2µs/DIV VIN = 12V VOUT = –5V ILOAD = 100mA FIGURE 7 CIRCUIT 20µs/DIV VIN = 12V VOUT = –5V ILOAD = 100mA FIGURE 7 CIRCUIT 3863 G01 Transient Response: Pulse-Skipping Mode Operation Transient Response: Rising Edge Pulse-Skipping Mode Operation ILOAD 1A/DIV ILOAD 1A/DIV VOUT 200mV/DIV VOUT 200mV/DIV IL 1A/DIV IL 1A/DIV 100µs/DIV VIN = 12V VOUT = –5V TRANSIENT = 100mA TO 1.6A FIGURE 7 CIRCUIT Transient Response: Falling Edge Pulse-Skipping Mode Operation VOUT 200mV/DIV IL 1A/DIV 10µs/DIV VIN = 12V VOUT = –5V TRANSIENT = 1.6A TO 100mA FIGURE 7 CIRCUIT 3863 G05 Soft-Start into a Prebiased Output VOUT1 2V/DIV VOUT2 2V/DIV TRACK/SS 200mV/DIV VOUT2 2V/DIV VOUT2 2V/DIV TRACK/SS 200mV/DIV 3863 G07 3863 G06 Output Tracking RUN 5V/DIV VOUT1 2V/DIV VIN 5V/DIV 3863 G03 ILOAD 1A/DIV 10µs/DIV VIN = 12V VOUT = –5V TRANSIENT = 100mA TO 1.6A FIGURE 7 CIRCUIT 3863 G04 Normal Soft-Start 1ms/DIV VIN = 12V VOUT1 = 5V VOUT2 = –5V ILOAD1 = ILOAD2 = 100mA FIGURE 11 CIRCUIT 100µs/DIV VIN = 12V VOUT = –5V TRANSIENT = 100mA TO 1.6A FIGURE 7 CIRCUIT 3863 G02 TRACK/SS 200mV/DIV VOUT1 2V/DIV 1ms/DIV VIN = 12V VOUT1 = 5V, VOUT2 = –5V PRE-BIAS1 = 2V, PRE-BIAS2 = –2V ILOAD = 50mA FIGURE 11 CIRCUIT 3863 G08 20ms/DIV VIN = 12V VOUT1 = 5V VOUT2 = –5V ILOAD1 = ILOAD2 = 100mA FIGURE 11 CIRCUIT 3863 G09 3863f For more information www.linear.com/3863 5 LTC3863 TYPICAL PERFORMANCE CHARACTERISTICS Overcurrent Protection VIN Line Transient Behavior Short-Circuit Protection SHORTCIRCUIT TRIGGER ILOAD 1A/DIV VOUT 500mV/DIV SOFT-START FOLDBACK COUT DISCHARGE 500µs/DIV VIN = 12V VOUT = –5V ILOAD2 = 1A TO SHORT-CIRCUIT FIGURE 7 CIRCUIT 3863 G10 1100 VIN = 12V VOUT = –5V ILOAD = 0A FIGURE 7 CIRCUIT 135 VOUT 50mV/DIV 3863 G11 10ms/DIV VIN = 12V, SURGE TO 48V VOUT = –5V ILOAD = 500mA FIGURE 7 CIRCUIT Pulse-Skipping Mode Input Current Over Input Voltage (No Load) Burst Mode Input Current Over Input Voltage (No Load) 140 RECOVERY 1000 130 30 20 900 850 120 VIN = 12V FIGURE 7 CIRCUIT 25 950 IVIN (µA) 125 3863 G12 Shutdown Current Over Input Voltage VIN = 12V VOUT = –5V ILOAD = 0A FIGURE 7 CIRCUIT 1050 12V GATE 20V/DIV IVIN (µA) 20ms/DIV VIN = 12V VOUT = –5V ILOAD2 = 1A TO 3.2A FIGURE 7 CIRCUIT 48V VIN 20V/DIV VOUT 5V/DIV IL 1A/DIV IL 1A/DIV IVIN (µA) TA = 25°C, unless otherwise noted. 15 10 800 115 110 5 750 0 10 20 30 VIN (V) 40 50 700 60 10 0 20 30 40 50 –0.5 1.0 VIN = 12V, VOUT = –5V ILOAD NORMALIZED AT ILOAD = 1A FIGURE 7 CIRCUIT 0.8 0.5 0 –0.5 PULSE-SKIPPING MODE Burst Mode OPERATION 0 10 20 30 VIN (V) 40 50 60 30 VIN (V) 40 50 –1.0 –0.5 60 Output Regulation Over Temperature NORMALIZED ∆VOUT (%) NORMALIZED ∆VOUT (%) NORMALIZED ∆VOUT (%) 1.0 VOUT = –5V ILOAD = 100mA NOMALIZED AT VIN = 12V FIGURE 7 CIRCUIT 20 3863 G15 Output Regulation Over Load Current 0 –1.0 10 3863 G14 Output Regulation Over Input Voltage 0.5 0 VIN (V) 3863 G13 1.0 0 60 0.6 VIN = 12V, VOUT = –5V ILOAD = 200mA VOUT NOMALIZED TO TA = 25°C FIGURE 7 CIRCUIT 0.4 0.2 0 –0.2 –0.4 –0.6 PULSE-SKIPPING MODE Burst Mode OPERATION 0 0.5 1.0 1.5 ILOAD (A) 2.0 3863 G16 2.5 3863 G17 PULSE-SKIPPING MODE Burst Mode OPERATION –0.8 –1.0 –75 –25 75 125 25 TEMPERATURE (°C) 175 3963 G18 3863f 6 For more information www.linear.com/3863 LTC3863 TYPICAL PERFORMANCE CHARACTERISTICS Free Running Frequency Over Input Voltage Free Running Frequency Over Temperature FREQ = OPEN 550 500 f (kHz) 500 f (kHz) 120 FREQUENCY FOLDBACK (%) FREQ = OPEN 550 450 450 400 400 FREQ = 0V 350 0 20 10 30 VIN (V) FREQ = 0V 350 40 50 300 –75 60 25 75 125 TEMPERATURE (°C) –25 GATE Bias LDO (VIN - VCAP) Load Regulation 0.1 –0.3 –0.4 –3.0 0 5 10 IGATE (mA) 15 –0.5 20 90 –75 –25 25 75 125 TEMPERATURE (°C) 175 3863 G25 800 80 70 60 50 40 30 20 10 0 5 10 IGATE (mA) 15 –10 20 0 0.4 0.8 1.2 ITH VOLTAGE (V) RUN Pin Pull-Up Current Over Temperature 0.65 VSS = 0V 12 10 8 6 –75 –25 2 1.6 3863 G24 RUN PULL-UP CURRENT (µA) 14 SS PULL-UP CURRENT (µA) CURRENT LIMIT SENSE VOLTAGE (mV) 100 92 600 Burst Mode OPERATION PULSE-SKIPPING 90 SS Pin Pull-Up Current Over Temperature 94 400 VFB (mV) 3863 G23 Current Sense Voltage Over Temperature 96 200 0 3863 G22 98 0 100 CURRENT SENSE VOLTAGE (mV) –0.2 –2.5 20 3863 G21 VIN = 5V –0.1 –2.0 40 Current Sense Voltage Over ITH Voltage 0.0 –1.5 60 0 175 (VIN - VCAP) DROPOUT (V) (VIN - VCAP) REGULATION (%) 0.0 –1.0 80 GATE Bias LDO (VIN - VCAP) Dropout Behavior 0.5 –0.5 100 3864 G20 3863 G19 –3.5 Frequency Foldback % Over Feedback Voltage 600 600 300 TA = 25°C, unless otherwise noted. 175 25 75 125 TEMPERATURE (°C) 3863 G26 VRUN = 0V 0.55 0.45 0.35 0.25 –75 –25 25 75 125 TEMPERATURE (°C) 175 3863 G27 3863f For more information www.linear.com/3863 7 LTC3863 PIN FUNCTIONS PLLIN/MODE (Pin 1): External Reference Clock Input and Burst Mode Enable/Disable. When an external clock is applied to this pin, the internal phase-locked loop will synchronize the turn-on edge of the gate drive signal with the rising edge of the external clock. When no external clock is applied, this input determines the operation during light loading. Floating this pin selects low IQ (40μA) Burst Mode operation. Pulling to ground selects pulse-skipping mode operation. FREQ (Pin 2): Switching Frequency Setpoint Input. The switching frequency is programmed by an external setpoint resistor RFREQ connected between the FREQ pin and signal ground. An internal 20µA current source creates a voltage across the external setpoint resistor to set the internal oscillator frequency. Alternatively, this pin can be driven directly by a DC voltage to set the oscillator frequency. Grounding selects a fixed operating frequency of 350kHz. Floating selects a fixed operating frequency of 535kHz. SGND (Pin 3): Ground Reference for Small-Signal Analog Component (Signal Ground). Signal ground should be used as the common ground for all small-signal analog inputs and compensation components. Connect the signal ground to the power ground (ground reference for power components) only at one point using a single PCB trace. SS (Pin 4): Soft-Start and External Tracking Input. The LTC3863 regulates the feedback voltage to the smaller of 0.8V or the voltage on the SS pin. An internal 10μA pull-up current source is connected to this pin. A capacitor to ground at this pin sets the ramp time to the final regulated output voltage. Alternatively, another voltage supply connected through a resistor divider to this pin allows the output to track the other supply during start-up. VFB (Pin 5): Output Feedback Sense. A resistor divider from the regulated output point to this pin sets the output voltage. The LTC3863 will nominally regulate VFB to the internal reference value of 0.8V. If VFB is less than 0.4V, the switching frequency will linearly decrease and fold back to about one-fifth of the internal oscillator frequency to reduce the minimum duty cycle. ITH (Pin 6): Current Control Threshold and Controller Compensation Point. This pin is the output of the error amplifier and the switching regulator’s compensation 8 point. The voltage ranges from 0V to 2.9V, with 0.8V corresponding to zero sense voltage (zero current). VFBN (Pin 7): Feedback Input for an Inverting PWM Controller. Connect VFBN to the center of a resistor divider between the output and VFB. The VFBN threshold is 0V. To defeat the inverting amplifier and use the LTC3863 as an LTC3864 (noninverting buck), tie VFBN > 2V. RUN (Pin 8): Digital Run Control Input. A RUN voltage above the 1.26V threshold enables normal operation, while a voltage below the threshold shuts down the controller. An internal 0.4µA current source pulls the RUN pin up to about 3.3V. The RUN pin can be connected to an external power supply up to 60V. CAP (Pin 9): Gate Driver (–) Supply. A low ESR ceramic bypass capacitor of at least 0.1µF or 10X the effective CMILLER of the P-channel power MOSFET, is required from VIN to this pin to serve as a bypass capacitor for the internal regulator. To ensure stable low noise operation, the bypass capacitor should be placed adjacent to the VIN and CAP pins and connected using the same PCB metal layer. SENSE (Pin 10): Current Sense Input. A sense resistor, RSENSE, from the VIN pin to the SENSE pin sets the maximum current limit. The peak inductor current limit is equal to 95mV/RSENSE. For accuracy, it is important that the VIN pin and the SENSE pin route directly to the current sense resistor and make a Kelvin (4-wire) connection. VIN (Pin 11): Chip Power Supply. A minimum bypass capacitor of 0.1µF is required from the VIN pin to power ground. For best performance use a low ESR ceramic capacitor placed near the VIN pin. GATE (Pin 12): Gate Drive Output for External P-Channel MOSFET. The gate driver bias supply voltage (VIN-VCAP) is regulated to 8V when VIN is greater than 8V. The gate driver is disabled when (VIN-VCAP) is less than 3.5V (typical), 3.8V maximum in start-up and 3.25V (typical) 3.5V maximum in normal operation. PGND (Exposed Pad Pin 13): Ground Reference for Power Components (Power Ground). The PGND exposed pad must be soldered to the circuit board for electrical contact and for rated thermal performance of the package. Connect signal ground to power ground only at one point using a single PCB trace. For more information www.linear.com/3863 3863f LTC3863 FUNCTIONAL DIAGRAM VIN UVLO + – 0.4µA RUN 1.26V CIN VIN 3.25V RSENSE SENSE RUN + – – LOGIC CONTROL DRV GATE MP D1 Q R PLLIN/MODE 20µA FREQ MODE/CLOCK DETECT + PLL SYSTEM – VCO RFREQ ICMP OUT VIN – 8V O.425V – CAP + + 10µA + + – SGND SLOPE COMPENSATION OV + EA (Gm = 1.8mS) 0.88V – PGND L LDO SS 0.8V EN CSS 0.5µA ITH RFB1 VFBN + CLOCK VOUT COUT CCAP IN Burst Mode OPERATION – S CFB2 RFB2 VFB 3863 FD RITH CITH1 3863f For more information www.linear.com/3863 9 LTC3863 OPERATION LTC3863 Main Control Loop The LTC3863 is a nonsynchronous inverting PMOS controller, where an inverting amplifier is used to sense the negative output voltage below ground. The LTC3863 uses a peak current mode control architecture to regulate the output. A feedback resistor, RFB1, is placed between VOUT and VFBN and a second resistor, RFB2, is placed between VFBN and and VFB. The LTC3863 has a trimmed internal reference, VREF, that is equal to (VFB – VFBN). The output voltage is equal to –(RFB1/RFB2) • VREF where VREF is equal to 800mV in normal regulation. The LTC3863 can also be configured as a noninverting step-down buck regulator when the VFBN node is pulled greater than 2V but held less than 5V, which disables the internal inverting amplifier. A feedback resistor, RFB1, is placed between VOUT and VFB and a second resistor, RFB2, is placed between VFB and SGND. In the noninverting buck mode the VFB input is compared to the internal reference, VREF, by a transconductance error amplifier (EA). The internal reference can be either a fixed 0.8V reference, VREF, or the voltage input on the SS pin. In normal operation VFB regulates to the internal 0.8V reference voltage. The output voltage in normal regulation is equal to (RFB1 + RFB2)/RFB2 • 800mV. In soft-start or tracking mode when the SS pin voltage is less than the internal 0.8V reference voltage, VFB will regulate to the SS pin voltage. The error amplifier output connects to the ITH (current [I] threshold [TH]) pin. The voltage level on the ITH pin is then summed with a slope compensation ramp to create the peak inductor current set point. The peak inductor current is measured through a sense resistor, RSENSE, placed across the VIN and SENSE pins. The resultant differential voltage from VIN to SENSE is proportional to the inductor current and is compared to the peak inductor current setpoint. During normal operation the P-channel power MOSFET is turned on when the clock leading edge sets the SR latch through the S input. The P-channel MOSFET is turned off through the SR latch R input when the differential voltage from VIN to SENSE is greater than the peak inductor current setpoint and the current comparator, ICMP, trips high. Power CAP and VIN Undervoltage Lockout (UVLO) Power for the P-channel MOSFET gate driver is derived from the CAP pin. The CAP pin is regulated to 8V below VIN in order to provide efficient P-channel operation. The power for the VCAP supply comes from an internal LDO, which regulates the VIN-CAP differential voltage. A minimum capacitance of 0.1µF (low ESR ceramic) is required between VIN and CAP to assure stability. For VIN ≤ 8V, the LDO will be in dropout and the CAP voltage will be at ground, i.e., the VIN-CAP differential voltage will equal VIN. If VIN-CAP is less than 3.25V (typical), the LTC3863 enters a UVLO state where the GATE is prevented from switching and most internal circuitry is shut down. In order to exit UVLO, the VIN-CAP voltage would have to exceed 3.5V (typical). Shutdown and Soft-Start When the RUN pin is below 0.7V, the controller and most internal circuits are disabled. In this micropower shutdown state, the LTC3863 draws only 7µA. Releasing the RUN pin allows a small internal pull-up current to pull the RUN pin above 1.26V and enable the controller. The RUN pin can be pulled up to an external supply of up to 60V or it can be driven directly by logic levels. 3863f 10 For more information www.linear.com/3863 LTC3863 OPERATION The start-up of the output voltage VOUT is controlled by the voltage on the SS pin. When the voltage on the SS pin is less than the 0.8V internal reference, the VFB pin is regulated to the voltage on the SS pin. This allows the SS pin to be used to program a soft-start by connecting an external capacitor from the SS pin to signal ground. An internal 10µA pull-up current charges this capacitor, creating a voltage ramp on the SS pin. As the SS voltage rises from 0V to 0.8V, the output voltage VOUT rises smoothly from zero to its final value. Alternatively, the SS pin can be used to cause the start-up of VOUT to track that of another supply. Typically, this requires connecting the SS pin to an external resistor divider from the other supply to ground (see Applications Information). Under shutdown or UVLO, the SS pin is pulled to ground and prevented from ramping up. If the slew rate of the SS pin is greater than 1.2V/ms, the output will track an internal soft-start ramp instead of the SS pin. The internal soft-start will guarantee a smooth start-up of the output under all conditions, including in the case of a short-circuit recovery where the output voltage will recover from near ground. Light Load Current Operation (Burst Mode Operation or Pulse-Skipping Mode) The LTC3863 can be enabled to enter high efficiency Burst Mode operation or pulse-skipping mode at light loads. To select pulse-skipping operation, tie the PLLIN/MODE pin to signal ground. To select Burst Mode operation, float the PLLIN/MODE pin. In Burst Mode operation, if the VFB is higher than the reference voltage, the error amplifier will decrease the voltage on the ITH pin. When the ITH voltage drops below 0.425V, the internal sleep signal goes high, enabling sleep mode. The ITH pin is then disconnected from the output of the error amplifier and held at 0.45V. In sleep mode, much of the internal circuitry is turned off, reducing the quiescent current to 70µA while the load current is supplied by the output capacitor. As the output voltage and hence the feedback voltage decreases, the error amplifier’s output will rise. When the output voltage drops enough, the ITH pin is reconnected to the output of the error amplifier, the sleep signal goes low, and the controller resumes normal operation by turning on the external P-channel MOSFET on the next cycle of the internal oscillator. In Burst Mode operation, the peak inductor current has to reach at least 25% of current limit for the current comparator, ICMP, to trip and turn the P-channel MOSFET back off, even though the ITH voltage may indicate a lower current setpoint value. When the PLLIN/MODE pin is connected for pulse-skipping mode, the LTC3863 will skip pulses during light loads. In this mode, ICMP may remain tripped for several cycles and force the external MOSFET to stay off, thereby skipping pulses. This mode offers the benefits of smaller output ripple, lower audible noise, and reduced RF interference, at the expense of lower efficiency when compared to Burst Mode operation. Frequency Selection and Clock Synchronization The switching frequency of the LTC3863 can be selected using the FREQ pin. If the PLLIN/MODE pin is not being driven by an external clock source, the FREQ pin can be tied to signal ground, floated, or programmed through an external resistor. Tying FREQ to signal ground selects 350kHz, while floating selects 535kHz. Placing a resistor between FREQ and signal ground allows the frequency to be programmed between 50kHz and 850kHz. The phase-locked loop (PLL) on the LTC3863 will synchronize the internal oscillator to an external clock source when connected to the PLLIN/MODE pin. The PLL forces the turn-on edge of the external P-channel MOSFET to be aligned with the rising edge of the synchronizing signal. 3863f For more information www.linear.com/3863 11 LTC3863 OPERATION The oscillator’s default frequency is based on the operating frequency set by the FREQ pin. If the oscillator’s default frequency is near the external clock frequency, only slight adjustments are needed for the PLL to synchronize the external P-channel MOSFET’s turn-on edge to the rising edge of the external clock. This allows the PLL to lock rapidly without deviating far from the desired frequency. The PLL is guaranteed from 75kHz to 750kHz. The clock input levels should be greater than 2V for HI and less than 0.5V for LO. Fault Protection When the VFB voltage is above +10% of the regulated voltage of 0.8V, this is considered as an overvoltage condition and the external P-MOSFET is immediately turned off and prevented from ever turning on until VFB returns below +7.5%. In the event of an output short circuit or overcurrent condition that causes the output voltage to drop significantly while in current limit, the LTC3863 operating frequency will fold back. Anytime the output feedback VFB voltage is less than 50% of the 0.8V internal reference (i.e., 0.4V), frequency foldback is active. The frequency will continue to drop as VFB drops until reaching a minimum foldback frequency of about 18% of the setpoint frequency. Frequency foldback is designed, in combination with peak current limit, to limit current in start-up and short-circuit conditions. Setting the foldback frequency as a percentage of operating frequency assures that start-up characteristics scale appropriately with operating frequency. 3863f 12 For more information www.linear.com/3863 LTC3863 APPLICATIONS INFORMATION The LTC3863 is a nonsynchronous inverting, current mode, constant frequency PWM controller. It drives an external P-channel power MOSFET which connects to a Schottky power diode acting as the commutating catch diode. The input range extends from 3.5V to 60V. The output range has no theoretical minimum or maximum, but the duty factor and external components practically limit the output to one-tenth and ten times the input voltage. Higher output ratios can be obtained with transformers and more efficient external components. The LTC3863 offers a highly efficient Burst Mode operation with 70µA quiescent current, which delivers outstanding efficiency in light load operation. The LTC3863 is a low pin count, robust and easy-to-use inverting power supply solution in applications which require high efficiency and operate with widely varying input and output voltages. The typical application on the front page is a basic LTC3863 application circuit. The LTC3863 can sense the inductor current through a high side series sense resistor, RSENSE, placed between VIN and the source of the external P-channel MOSFET. Once the required output voltage and operating frequency have been determined, external component selection is driven by load requirements, and begins with the selection of inductor and RSENSE. Next, the power MOSFET and catch diode are selected. Finally, input and output capacitors are selected. Output Voltage Programming The output voltage is programmed by connecting a feedback resistor divider from the output to the VFB pin as shown in Figure 1. The output voltage in steady-state operation is set by the feedback resistors according to the equation: VOUT = –0.8V • LTC3863 VFB RFB2 VFBN CFB2 3863 F01 RFB1 VOUT Figure 1. Setting the Output Voltage Great care should be taken to route the VFB and VFBN lines away from noise sources, such as the inductor or SW node or the GATE signal that drives the external Pchannel MOSFET. The integrator capacitor, CFB2, should be sized to ensure the negative sense amplifier gain rolls off and limits high frequency gain peaking in the DC/DC control loop. The integrator capacitor pole can be safely set to be two times the switching frequency without affecting the DC/DC phase margin according to the following equation. It is highly recommended that CFB2 be used in most applications. CFB2 = 1 2 • π • 2 •FREQ SW Switching Frequency and Clock Synchronization The choice of operating frequency is a trade-off between efficiency and component size. Lowering the operating frequency improves efficiency by reducing MOSFET switching losses but requires larger inductance and/or capacitance to maintain low output ripple voltage. Conversely, raising the operating frequency degrades efficiency but reduces component size. RFB1 RFB2 3863f For more information www.linear.com/3863 13 LTC3863 APPLICATIONS INFORMATION The LTC3863 can free-run at a user programmed switching frequency, or it can synchronize with an external clock to run at the clock frequency. When the LTC3863 is synchronized, the GATE pin will synchronize in phase with the rising edge of the applied clock in order to turn the external P-channel MOSFET on. The switching frequency of the LTC3863 is programmed with the FREQ pin, and the external clock is applied at the PLLIN/MODE pin. Table 1 highlights the different states in which the FREQ pin can be used in conjunction with the PLLIN/MODE pin. Table 1 FREQ PIN PLLIN/MODE PIN FREQUENCY OV DC Voltage 350kHz Floating DC Voltage 535kHz Resistor to GND DC Voltage 50kHz to 850kHz Either of the Above External Clock Phase Locked to External Clock The free-running switching frequency can be programmed from 50kHz to 850kHz by connecting a resistor from FREQ to signal ground. The resulting switching frequency as a function of resistance on the FREQ pin is shown in Figure 2. Set the free-running frequency to the desired synchronization frequency using the FREQ pin so that the internal oscillator is prebiased approximately to the synchronization frequency. While it is not required that the free-running frequency be near the external clock frequency, doing so will minimize synchronization time. 1000 Inductor Selection Operating frequency, inductor selection, capacitor selection and efficiency are interrelated. Higher operating frequencies allow the use of smaller inductors, smaller capacitors, but result in lower efficiency because of higher MOSFET gate charge and transition losses. In addition to this basic trade-off, the selection of inductor value is also influenced by other factors. Small inductor values result in large inductor ripple currents, large output voltage ripples and low efficiency due to higher core and conduction loss. Large inductor ripple currents result in high inductor peak currents, which require physically large inductors with large magnetic cross sections and higher saturation current ratings. The value of the inductor can also impact the stability of the feedback loop. In continuous mode, the buck-boost converter transfer function has a right-half plane zero at a frequency that is inversely proportional to the value of the inductor. As a result, large inductor values can move this zero to a frequency that is low enough to degrade the phase margin of the feedback loop. Large inductor values also tend to degrade stability due to low noise margin caused from low ripple current. Additionally, large value inductors can lead to slow transient response due to slow inductor current ramping time. For an inverting buck-boost converter operating in continuous conduction mode (CCM), given the desired input, output voltages and switching frequency, the peak-to-peak inductor ripple current is determined by the inductor value: 900 ∆IL(CCM) = FREQUENCY (kHz) 800 700 600 where VD is the diode forward conduction voltage. In cases where VOUT >> VD, VD can be ignored. D is the duty factor and is given as: 500 400 300 200 D= 100 0 VIN • (| VOUT | +VD ) VIN •D = L • f L • f • ( VIN +| VOUT | +VD ) 15 25 35 45 55 65 75 85 95 105 115 125 FREQ PIN RESISTOR (kΩ) | VOUT | +VD VIN +| VOUT | +VD (0 <D < 1) 3863 F02 Figure 2. Switching Frequency vs Resistor on FREQ 3863f 14 For more information www.linear.com/3863 LTC3863 APPLICATIONS INFORMATION The duty factor increases with increasing VOUT and decreasing VIN. For a given VOUT, the maximum duty factor occurs at minimum VIN. A typical starting point for selecting an inductor is to choose the inductance such that the maximum peak-to-peak inductor ripple current, ∆IL(MAX), is set to 40% ~ 50% of the inductor average current, IL(AVG), at maximum load current. Since ∆IL(MAX) occurs at maximum VIN in continuous mode, the inductance is calculated at maximum VIN: L= VIN(MAX)2 • (| VOUT | +VD ) ( 0.4 •IOUT(MAX) • f • VIN(MAX) +| VOUT | +VD )2 The inductance can be further adjusted to achieve specific design optimization of efficiency, output ripple, component size and loop response. Once the inductance value has been determined, the type of inductor must be selected. Core loss is independent of core size for a given inductor value, but it is very dependent on the inductance selected. As inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of wire and therefore, copper losses will increase. High efficiency converters generally cannot tolerate the core loss of low cost powdered iron cores, forcing the use of more expensive ferrite materials. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates hard, which means that inductance collapses abruptly when the peak design current is exceeded. This will result in an abrupt increase in inductor ripple current and output voltage ripple. Do not allow the core to saturate! Current Sensing and Current Limit Programming The LTC3863 senses the inductor current through a current sense resistor, RSENSE, placed across the VIN and SENSE pins. The voltage across the resistor, VSENSE, is proportional to inductor current and in normal operation is compared to the peak inductor current setpoint. An inductor current limit condition is detected when VSENSE exceeds 95mV. When the current limit threshold is exceeded, the P-channel MOSFET is immediately turned off by pulling the GATE voltage to VIN regardless of the controller input. The peak inductor current limit is equal to: 95mV IL(PEAK) ≅ RSENSE This inductor current limit would translate to an output current limit based on the inductor ripple and duty factor: 95mV ∆IL IOUT(LIMIT) = – • (1–D) RSENSE 2 The SENSE pin is a high impedance input with a maximum leakage of ±2µA. Since the LTC3863 is a peak current mode controller, noise on the SENSE pin can create pulse width jitter. Careful attention must be paid to the layout of RSENSE. To ensure the integrity of the current sense signal, VSENSE, the traces from VIN and SENSE pins should be short and run together as a differential pair and Kelvin (4-wire) connected across RSENSE (Figure 3). VIN VIN LTC3863 A variety of inductors are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft, Toko, Vishay, Pulse and Würth. SENSE OPTIONAL FILTERING CF RSENSE RF MP 3863 F03 Figure 3. Inductor Current Sensing 3863f For more information www.linear.com/3863 15 LTC3863 APPLICATIONS INFORMATION The LTC3863 has internal filtering of the current sense voltage which should be adequate in most applications. However, adding a provision for an external filter offers added flexibility and noise immunity, should it be necessary. The filter can be created by placing a resistor from the RSENSE resistor to the SENSE pin and a capacitor across the VIN and SENSE pins. Power MOSFET Selection The LTC3863 drives a P-channel power MOSFET that serves as the main switch for the nonsynchronous inverting converter. Important P-channel power MOSFET parameters include drain-to-source breakdown voltage BVDSS, threshold voltage VGS(TH), on-resistance RDS(ON), gate-to-drain reverse transfer capacitance CRSS, maximum drain current ID(MAX), and the MOSFET’s thermal resistance θJC(MOSFET) and θJA(MOSFET). The drain-to-source breakdown voltage must meet the following condition: BVDSS > VIN(MAX) + |VOUT| + VD The gate driver bias voltage VIN-VCAP is set by an internal LDO regulator. In normal operation, the CAP pin will be regulated to 8V below VIN. A minimum 0.1µF capacitor is required across the VIN and CAP pins to ensure LDO stability. If required, additional capacitance can be added to accommodate higher gate currents without voltage droop. In shutdown and Burst Mode operation, the CAP LDO is turned off. In the event of CAP leakage to ground, the CAP voltage is limited to 9V by a weak internal clamp from VIN to CAP. As a result, a minimum 10V VGS rated MOSFET is required. VSG MILLER EFFECT The power dissipated by the P-channel MOSFET when the LTC3863 is in continuous conduction mode is given by: 2 I PPMOS ≈D• OUT • ρT •RDS(ON) 1–D f •CMILLER • ( VIN +| VOUT | +VD ) IOUT + • 2 1–D RDN RUP • + ( VIN – VCAP – VMILLER ) VMILLER 2 where D is duty factor, RDS(ON) is on-resistance of P-channel MOSFET, ρT is temperature coefficient of onresistance, RDN is the pull-down driver resistance specified at 0.9Ω typical and RUP is the pull-up driver resistance specified at 2Ω typical. VMILLER is the Miller effective VGS voltage and is taken graphically from the power MOSFET data sheet. The power MOSFET input capacitance, CMILLER, is the most important selection criteria for determining the transition loss term in the P-channel MOSFET but is not directly specified on MOSFET data sheets. CMILLER is a combination of several components, but it can be derived from the typical gate charge curve included on most data sheets (Figure 4). The curve is generated by forcing a constant current out of the gate of a common-source connected P-channel MOSFET that is loaded with a resistor, and then plotting the gate voltage versus time. The initial slope is the effect of the gate-to-source and gate-to-drain capacitances. The flat portion of the curve is the result of the Miller multiplication effect of the drain-to-gate capacitance as the drain G S D a b IGATE RLOAD + V – SD(TEST) 3863 F04 QIN CMILLER = (QB – QA)/VSD(TEST) (4a) (4b) Figure 4. (4a) Typical P-Channel MOSFET Gate Charge Characteristics and (4b) Test Set-Up to Generate Gate Charge Curve 3863f 16 For more information www.linear.com/3863 LTC3863 APPLICATIONS INFORMATION voltage rises across the resistor load. The Miller charge (the increase in coulombs on the horizontal axis from a to b while the curve is flat) is specified for a given VSD test voltage, but can be adjusted for different VSD voltages by multiplying by the ratio of the adjusted VSD to the curve specified VSD value. A way to estimate the CMILLER term is to take the change in gate charge from points a and b (or the parameter QGD on a manufacturer’s data sheet) and dividing it by the specified VSD test voltage, VSD(TEST). CMILLER ≅ QGD Once the average forward diode current is calculated, the power dissipation can be determined. Refer to the Schottky diode data sheet for the power dissipation, PDIODE, as a function of average forward current, IF(AVG). PDIODE can also be iteratively determined by the two equations below, where VF(IOUT,TJ) is a function of both IF(AVG) and junction temperature TJ. Note that the thermal resistance, θJA(DIODE), given in the data sheet is typical and can be highly layout dependent. It is therefore important to make sure that the Schottky diode has adequate heat sinking. TJ ≅ PDIODE • θJA(DIODE) VSD(TEST) The term with CMILLER accounts for transition loss, which is highest at high input voltages. For VIN < 20V, the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V, the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. Schottky Diode Selection When the P-channel MOSFET is turned off, a power Schottky diode is required to function as a commutating diode to carry the inductor current. The average forward diode current is independent of duty factor and is described as: IF(AVG) = IOUT PDIODE ≅ IF(AVG) • VD(IOUT,TJ) The Schottky diode forward voltage is a function of both IF(AVG) and TJ, so several iterations may be required to satisfy both equations. The Schottky forward voltage, VD, should be taken from the Schottky diode data sheet curve showing instantaneous forward voltage. The forward voltage will change as a function of both TJ and IF(AVG). The nominal forward voltage will also tend to increase as the reverse breakdown voltage increases. It is therefore advantageous to select a Schottky diode appropriate to the input voltage requirements. The diode reverse breakdown voltage must meet the following condition: VR > VIN(MAX) + |VOUT| CIN and COUT Selection The worst-case condition for diode conduction is a shortcircuit condition where the Schottky must handle the maximum current as its duty factor approaches 100% (and the P-channel MOSFET’s duty factor approaches 0%). The diode therefore must be chosen carefully to meet worstcase voltage and current requirements. A good practice is to choose a diode that has a forward current rating two times higher than IOUT(MAX). The input and output capacitance, CIN/COUT, are required to filter the square wave current through the P-channel MOSFET and diode respectively. Use a low ESR capacitor sized to handle the maximum RMS current: ICIN(RMS) =ICOUT(RMS) =IOUT • | VOUT | +VD VIN 3863f For more information www.linear.com/3863 17 LTC3863 APPLICATIONS INFORMATION The formula shows that the RMS current is greater than the maximum IOUT when VOUT is greater than VIN. Choose capacitors with higher RMS rating with sufficient margin. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life, which makes it advisable to derate the capacitor. The selection of COUT is primarily determined by the ESR required to minimize voltage ripple and load step transients. The ∆VOUT is approximately bounded by: ∆VOUT ≤IL(PEAK) •ESR + IOUT •D f •COUT where IL(PEAK) is the peak inductor current and it’s given as: IL(PEAK) = IOUT ( VIN +| VOUT | +VD ) VIN + characteristics but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an overvoltage hazard to the power switch and controller. To dampen input voltage transients, add a small 5μF to 40μF aluminum electrolytic capacitor with an ESR in the range of 0.5Ω to 2Ω. High performance through-hole capacitors may also be used, but an additional ceramic capacitor in parallel is recommended to reduce the effect of lead inductance. Discontinuous and Continuous Operation The LTC3863 operates in discontinuous conduction (DCM) until the load current is high enough for the inductor current to be positive at the end of the switching cycle. The output load current at the continuous/discontinuous boundary, IOUT(CDB), is given by the following equation: VIN • (| VOUT | +VD ) 2 •L • f • ( VIN +| VOUT | +VD ) Since IL(PEAK) and D reach their maximum values at minimum VIN, the output voltage ripple is highest at minimum VIN and maximum IOUT. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, specialty polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Specialty polymer capacitors offer very low ESR but have lower specific capacitance than other types. Tantalum capacitors have the highest specific capacitance, but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications provided that consideration is given to ripple current ratings and longterm reliability. Ceramic capacitors have excellent low ESR IOUT(CDB) = VIN(MAX)2 • (| VOUT | +VD ) ( 2 •L • f • VIN(MAX) +| VOUT | +VD )2 The continuous/discontinuous boundary is inversely proportional to the inductor value. Therefore, if required, IOUT(CDB) can be reduced by increasing the inductor value. External Soft-Start and Output Tracking Start-up characteristics are controlled by the voltage on the SS pin. When the voltage on the SS pin is less than the internal 0.8V reference, the LTC3863 regulates the VFB pin voltage to the voltage on the SS pin. When the SS pin is greater than the internal 0.8V reference, the VFB pin voltage regulates to the 0.8V internal reference. The SS pin can be used to program an external soft-start function or to allow VOUT to track another supply during start-up. 3863f 18 For more information www.linear.com/3863 LTC3863 APPLICATIONS INFORMATION Soft-start is enabled by connecting a capacitor from the SS pin to ground. An internal 10µA current source charges the capacitor, providing a linear ramping voltage at the SS pin that causes VOUT to rise smoothly from 0V to its final regulated value. The total soft-start time will be approximately: 0.8V 10µA When the LTC3863 is configured to track another supply, a voltage divider can be used from the tracking supply to the SS pin to scale the ramp rate appropriately. Two common implementations of tracking as shown in Figure 5a are coincident and ratiometric. For coincident tracking, choose the divider ratio for the external supply as shown in Figure 5b. Ratiometric tracking could be achieved by using a different ratio than the feedback (Figure 5b). Note that the soft-start capacitor charging current is always flowing, producing a small offset error. To minimize this error, select the tracking resistive divider values to be small enough to make this offset error negligible. The inductor current limit is inherently set in a current mode controller by the maximum sense voltage and RSENSE. In the LTC3863, the maximum sense voltage is 95mV, measured across the inductor sense resistor, RSENSE, placed across the VIN and SENSE pins. The output current limit is approximately: VIN(MIN) 95mV ∆IL ILIMIT(MIN) = – • 2 VIN(MIN) + | VOUT | + VD R SENSE ( Short-circuit fault protection is assured by the combination of current limit and frequency foldback. When the output feedback voltage, VFB, drops below 0.4V, the operating VOLTAGE EXTERNAL SUPPLY –VOUT (VOUT < 0V) ) The current limit must be chosen to ensure that ILIMIT(MIN) > IOUT(MAX) under all operating conditions. The inductor current limit should be greater than the inductor current required to produce maximum output power at worst-case efficiency. For the LTC3863, both minimum and maximum VIN cases should be checked to determine the worst-case efficiency. EXTERNAL SUPPLY VOLTAGE t SS = CSS • Short-Circuit Faults: Current Limit and Foldback –VOUT (VOUT < 0V) TIME TIME Coincident Tracking Ratiometric Tracking 3863 F05a Figure 5a. Two Different Modes of Output Tracking EXT. V |VOUT| > 0.8V R1 = RFB1 – RFB2 VOUT TO VFBN TO SS R2 = RFB2 RFB2 VOUT EXT. V RFB1 R1 TO SS R2 0.8V ≥ R1+ R2 EXT. V R2 TO FB RFB1 TO VFBN RFB2 TO FB 3863 F05b Coincident Tracking Setup Ratiometric Tracking Setup Figure 5b. Setup for Ratiometric and Coincident Tracking For more information www.linear.com/3863 3863f 19 LTC3863 APPLICATIONS INFORMATION VIN ––VOUT VOLTAGE frequency, f, will fold back to a minimum value of 0.18 • f when VFB reaches 0V. Both current limit and frequency foldback are active in all modes of operation. In a shortcircuit fault condition, the output current is first limited by current limit and then further reduced by folding back the operating frequency as the short becomes more severe. The worst-case fault condition occurs when VOUT is shorted to ground. INTERNAL SOFT-START INDUCED START-UP (NO EXTERNAL SOFT-START CAPACITOR) ~650µs TIME (6a) Short-Circuit Recovery and Internal Soft-Start The internal soft-start voltage and the external SS pin operate independently. The output will track the lower of the two voltages. The slew rate of the internal soft-start voltage is roughly 1.2V/ms, which translates to a total soft-start time of 650µs. If the slew rate of the SS pin is greater than 1.2V/ms the output will track the internal softstart ramp. To assure robust fault recovery, the internal soft-start feature is active in all operational cases. If a short-circuit condition occurs which causes the output to drop significantly, the internal soft-start will assure a soft recovery when the fault condition is removed. The internal soft-start assures a clean soft ramp-up from any fault condition that causes the output to droop, guaranteeing a maximum ramp rate in soft-start, short-circuit fault release. Figure 6 illustrates how internal soft-start controls the output ramp-up rate under varying scenarios. VIN Undervoltage Lockout (UVLO) The LTC3863 is designed to accommodate applications requiring widely varying power input voltages from 3.5V to 60V. To accommodate the cases where VIN drops significantly once in regulation, the LTC3863 is guaranteed to operate down to a VIN of 3.5V over the full temperature range. –VOUT VOLTAGE An internal soft-start feature guarantees a maximum positive output voltage slew rate in all operational cases. In a short-circuit recovery condition for example, the output recovery rate is limited by the internal soft-start so that output voltage overshoot and excessive inductor current buildup is prevented. SHORT-CIRCUIT INTERNAL SOFT-START INDUCED RECOVERY TIME 3863 F06 (6b) Figure 6. Internal Soft-Start (6a) Allows Soft-Start without an External Soft-Start Capacitor and Allows Soft Recovery from (6b) a Short-Circuit The implications of both the UVLO rising and UVLO falling specifications must be carefully considered for low VIN operation. The UVLO threshold with VIN rising is typically 3.5V (with a maximum of 3.8V) and UVLO falling is typically 3.25V (with a maximum of 3.5V). The operating input voltage range of the LTC3863 is guaranteed to be 3.5V to 60V over temperature, but the initial VIN ramp must exceed 3.8V to guarantee start-up. Minimum On-Time Considerations The minimum on-time, tON(MIN), is the smallest time duration that the LTC3863 is capable of turning on the power MOSFET, and is typically 220ns. It is determined by internal timing delays and the gate charge required to turn on the MOSFET. Low duty cycle applications may approach this minimum on-time limit, so care should be taken to ensure that: t ON(MIN) < ( (| VOUT | + VD ) f • VIN(MAX) + | VOUT | + VD ) 3863f 20 For more information www.linear.com/3863 LTC3863 APPLICATIONS INFORMATION If the duty cycle falls below what can be accommodated by the minimum on-time, the controller will skip cycles. However, the output voltage will continue to regulate. Efficiency Considerations 2. Transition Loss: Transition loss of the P-channel MOSFET becomes significant only when operating at high input voltages (typically 20V or greater.) The P‑channel transition losses (PMOSTRL) can be determined from the following equation: The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine the dominant contributors and therefore where efficiency improvements can be made. Percent efficiency can be expressed as: f •CMILLER • ( VIN +| VOUT | +VD ) PPMOSTRL = 2 I RDN RUP • OUT • + 1–D ( VIN – VCAP ) – VMILLER VMILLER % Efficiency = 100% - (L1+L2+L3+…) where L1, L2, L3, etc., are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses in LTC3863 application circuits. 1. Conduction Loss: Conduction losses result from the P-channel MOSFET RDS(ON), inductor resistance DCR, the current sense resistor RSENSE, and input and output capacitor ESR. The current through DCR is continuous. The currents through both the P-channel MOSFET and Schottky diode are discontinuous. The following equation may be used to determine the total conduction loss (PCOND) in continuous conduction mode: I 2 ∆I 2 PCOND ≈ OUT 2 + L 12 (1–D) ( 2 3. Gate Charging Loss: Charging and discharging the gate of the MOSFET will result in an effective gate charging current. Each time the P-channel MOSFET gate is switched from low to high and low again, a packet of charge, dQ, moves from the capacitor across VIN – VCAP and is then replenished from ground by the internal VCAP regulator. The resulting dQ/dt current is a current out of VIN flowing to ground. The total power loss in the controller including gate charging loss is determined by the following equation: PCNTRL = VIN • (IQ + f • QG(PMOSFET)) 4. Schottky Loss: The Schottky loss is independent of duty factors. The critical component is the Schottky forward voltage as a function of junction temperature and current. The Schottky power loss is given by the equation: PDIODE = IOUT • VD(IOUT,TJ) RDCR +D • RDS(ON) +RSENSE +RESR(CIN) • + (1–D) •RESR(COUT) ) When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If changes cause the input current to decrease, then the efficiency has increased. If there is no change in input current, there is no change in efficiency. 3863f For more information www.linear.com/3863 21 LTC3863 APPLICATIONS INFORMATION OPTI-LOOP® Compensation OPTI-LOOP compensation, through the availability of the ITH pin, allows the transient response to be optimized for a wide range of loads and output capacitors. The ITH pin not only allows optimization of the control loop behavior but also provides a test point for the regulator ’s DC-coupled and AC-filtered closed-loop response. The DC step, rise time and settling at this test point truly reflects the closedloop response. Assuming a predominantly second order system, phase margin and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also be estimated by examining the rise time at this pin. The ITH series RITH-CITH1 filter sets the dominant pole-zero loop compensation. Additionally, a small capacitor placed from the ITH pin to signal ground, CITH2, may be required to attenuate high frequency noise. The values can be modified to optimize transient response once the final PCB layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because their various types and values determine the loop feedback factor gain and phase. An output current pulse of 20% to 100% of full load current having a rise time of 1μs to 10μs will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. The general goal of OPTI-LOOP compensation is to realize a fast but stable ITH response with minimal output droop due to the load step. For a detailed explanation of OPTI-LOOP compensation, refer to Application Note 76. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ∆ILOAD • ESR, where ESR is the effective series resistance of COUT . ∆ILOAD also begins to charge or discharge COUT , generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. Connecting a resistive load in series with a power MOSFET, then placing the two directly across the output capacitor and driving the gate with an appropriate signal generator is a practical way to produce a realistic load-step condition. The initial output voltage step resulting from the step change in output current may not be within the bandwidth of the feedback loop, so this signal cannot be used to determine phase margin. This is why it is better to look at the ITH pin signal which is in the feedback loop and is the filtered and compensated feedback loop response. The gain of the loop increases with RITH and the bandwidth of the loop increases with decreasing CITH1. If RITH is increased by the same factor that CITH1 is decreased, the zero frequency will be kept the same, thereby keeping the phase the same in the most critical frequency range of the feedback loop. In addition, a feedforward capacitor, CFF , can be added to improve the high frequency response, as shown in Figure 1. Capacitor CFF provides phase lead by creating a high frequency zero with RB1 which improves the phase margin. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate overall performance of the regulator. In some applications, a more severe transient can be caused by switching in loads with large (>10μF) input capacitors. If the switch connecting the load has low resistance and is driven quickly, then the discharged input capacitors are effectively put in parallel with COUT , causing a rapid drop in VOUT . No regulator can deliver enough current to prevent this problem. The solution is to limit the turn-on speed of the load switch driver. A Hot Swap™ controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection and soft-start. Large-Signal Effects on ITH Inverting controllers have a wide range of applications and operating conditions which affect compensation. Low switching frequencies and the inverting buck-boost right-half-plane zero can result in very low gain crossover frequency requirements. Low crossover frequencies often require a compensation RITH and CITH that are too small for 3863f 22 For more information www.linear.com/3863 LTC3863 APPLICATIONS INFORMATION the error amplifier output drive current on ITH of 100µA. The effect causes ITH to appear clamped in response to a transient load current step which causes excessive output droop. An RITH greater than 20k allows ITH to swing 1.5V with margin for temperature and part to part variation and should never have this issue. In applications with less severe transient load step requirements, RITH can safely be set as low as 10k. We do not recommend less than 10k in any application. If RITH is too small then either the operating frequency will need to be increased or the output capacitor increased to increase the RITH required to stabilize the system. We strongly recommend that any system with an RITH less than 20k be experimentally verified with worst-case load steps. Design Example Consider an inverting converter with the following specifications: VIN = 4.5V to 55V, VOUT = –5V, IOUT(MAX) = 1.8A, and f = 320kHz (Figure 7). The output voltage is programmed according to: The FREQ pin is tied to signal ground in order to program the switching frequency to 350kHz. The on-time required to generate –5V output from 55V VIN in continuous mode can be calculated as: tON(CCM) = 5V + 0.5V = 260ns 320kHz • ( 55V + 5V + 0.5V ) This on-time, tON, is larger than LTC3863’s minimum ontime with sufficient margin to prevent cycle skipping. Use a lower frequency if a larger on-time margin is needed to account for variations from minimum on-time and switching frequency. As load current decreases, the converter will eventually start cycle skipping. Next, set the inductor value such that the inductor ripple current is 60% of the average inductor current at maximum VIN = 55V and full load = 1.8A: L= 55V 2 • ( 5V + 0.5V ) 0.6 •1.8A • 320kHz • ( 55V + 5V + 0.5V ) 2 ≈ 13.1µH Select a standard value of 12μH. RFB2 RFB1 The resulting ripple current at minimum VIN of 4.5V is: ∆IL = 0.47µF 320kHz CAP RUN VIN 16mΩ PLLIN/MODE 0.1µF 15nF 220pF 52.3k CIN2 4.7µF 100V ×2 + VIN CIN1 4.5V TO 55V 68µF 63V SENSE SS GATE 10k 5V • ( 5V + 0.5V ) = 0.644A 12µH • 320kHz • ( 5V + 5V + 0.5V ) LTC3863 Q1 D1 L1 12µH ITH 187k FREQ SGND VFBN 12pF PGND VFB 3863 F07 COUT1 33µF 16V ×2 + VOUT = –0.8V • If RFB2 is chosen to be 188k, then RFB1 needs to be 30.1k. VOUT –5V COUT3 1.8A 100µF 20V CIN1: CDE AFK686M63G24T-F C : TDK CGA6M3X7S2A475K 30.1k CIN2 : TDK C4532X7R1C336M OUT1 COUT3: PANASONIC 20SVP100M D1: VISHAY SS8PH9-M3/87A L1: MSS1278-123ML Q1: VISHAY Si7469DP Figure 7. Design Example (4.5V to 55V Input, –5V, 1.8A at 320kHz) 3863f For more information www.linear.com/3863 23 LTC3863 APPLICATIONS INFORMATION The boundary output current for continuous/discontinuous mode is calculated: IOUT(CDB) = 55V 2 • 5V 2 •12µH• 320kHz • ( 55V + 5V ) 2 = 0.55A The maximum inductor peak current occurs at minimum VIN of 4.5V and full load of 1.8A where LTC3863 operates in continuous mode is: 1.8A • ( 4.5V + 5V + 0.5V ) ∆IL + 5V 2 0.644A = 3.6A + ≈ 4.25A 2 IL(PEAK _ MAX) = Next, set the RSENSE resistor value to ensure that the converter can deliver the maximum peak inductor current of 4.25A with sufficient margin to account for component variations and worst-case operating conditions. Using a 30% margin factor: RSENSE = 95mV = 17.2mΩ 1.3 • 4.25A D= 5V + 0.5V ≈ 0.55 4.5V + 5V + 0.5V 2 1.8A PPMOS = 0.55 • •1.8 • 30mΩ 1– 0.55 320kHz • 235pF • ( 4.5V+| –5V| +0.5V ) + 2 1.8A 2Ω 0.9Ω • • + (1– 0.55) 4.5V – 3.2V 3.2V 2 ≈ 0.475W + 0.020W ≈ 0.495W Use a more readily available 16mΩ sense resistor. This results in peak inductor current limit: IL(PEAK) = Next choose a P-channel MOSFET with the appropriate BVDSS and ID rating. The BVDSS rating should be greater than (55V + 5V + VD) with sufficient margin. In this example, a good choice is the Vishay Si7469DP (BVDSS = 80V, ID = 10A, RDS(ON) = 30mΩ, ρ100°C = 1.8, VMILLER = 3.2V, CMILLER = 235pF, θJA = 24°C/W). The highest power dissipation and the resulting junction temperature for the P-channel MOSFET occurs at the minimum VIN of 5V and maximum output current of 1.8A. They can be calculated at TA = 70°C as follows: 95mV = 5.94A 16mΩ TJ = 70°C + 0.495W • 24°C/W = 82°C The same calculations can be repeated for VIN(MAX) = 55V: D= 5V + 0.5V ≈ 0.091 55V + 5V + 0.5V 2 Choose an inductor that has rated saturation current higher than 5.94A with sufficient margin. 1.8A PPMOS ≈ 0.091• •1.8 • 30mΩ 1– 0.091 The output current limit can be calculated from the peak inductor current limit and its minimum occurs at minimum VIN of 5V: 320kHz • 235pF • ( 55V+| –5V| +0.5V ) + 2 1.8A 2Ω 0.9Ω • • + (1– 0.091) 5V – 3.2V 3.2V 5V 95mV 0.644A ILIMIT(MIN) = – • 16mΩ 2 ( 4.5V + 5V + 0.5V ) = 2.8A In this example, 2.8A is the maximum output current the switching regulator can support at VIN = 4.5V. It is larger than the full load of 1.8A by a margin of 1A. If a larger margin is needed, use a smaller RSENSE. 2 ≈ 0.019W + 0.39W ≈ 0.411W TJ = 70°C + 0.411W • 24°C/W = 80°C Next choose an appropriate Schottky diode that will handle the power requirements. The reverse voltage of the diode, VR, should be greater than (55V + 5V). The Fairchild S38 Schottky diode is selected (VF (3A,125°C) = 0.45V, 3863f 24 For more information www.linear.com/3863 LTC3863 APPLICATIONS INFORMATION VR = 80V, θJA = 55°C/W) for this application. The power dissipation and junction temperature at TA = 70° and full load = 1.8A can be calculated as: PDIODE = 1.8A • 0.45V = 0.81W TJ = 70°C + 0.81W • 55°C/W = 114°C These power dissipation calculations show that careful attention to heat sinking will be necessary. For the input bypass capacitors, choose low ESR ceramic capacitors that can handle the maximum RMS current at the minimum VIN of 4.5V: ICIN(RMS) ≈ 1.8A • | –5V| = 1.9A 4.5V COUT will be selected based on the ESR that is required to satisfy the output voltage ripple requirement. For this design, two 47μF ceramic capacitors are chosen to offer low ripple in both normal operation and in Burst Mode operation. The selected COUT must support the maximum RMS operating current at a minimum VIN of 4.5V: ICIN(RMS) ≈ 1.8A • | –5V| = 1.9A 4.5V A soft-start time of 8ms can be programmed through a 0.1μF capacitor on the SS pin: CSS = 8ms •10µA = 0.1µF 0.8V Loop compensation components on the ITH pin are chosen based on load step transient behavior (as described under OPTI-LOOP Compensation) and is optimized for stability. A pull-up resistor is used on the RUN pin for FMEA compliance (see Failure Modes and Effects Analysis). An application with complementary dual outputs of ±5V can be designed by using two LTC3863 parts with one configured into an inverting regulator and the other into a step-down buck regulator as shown in Figure 11. Refer to LTC3864 data sheet for the actual design of a buck output of 5V. Gate Driver Component Placement, Layout and Routing It is important to follow recommended power supply PC board layout practices such as placing external power elements to minimize loop area and inductance in switching paths. Be careful to pay particular attention to gate driver component placement, layout and routing. The effective CCAP capacitance should be greater than 0.1µF minimum in all operating conditions. Operating voltage and temperature both decrease the rated capacitance to varying degrees depending on dielectric type. The LTC3863 is a PMOS controller with an internal gate driver and bootstrapped LDO that regulates the differential CAP voltage (VIN – VCAP) to 8V nominal. The CCAP capacitance needs to be large enough to assure stability and provide cycleto-cycle current to the PMOS switch with minimum series inductance. We recommend a ceramic 0.47µF 16V capacitor with a high quality dielectric such as X5R or X7R. Some high current applications with large Qg PMOS switches may benefit from an even larger CCAP capacitance. Figure 8 shows the LTC3863 Generic Application Schematic which includes an optional current sense filter and series gate resistor. Figure 9 illustrates the recommended gate driver component placement, layout and routing of the GATE, VIN, SENSE and CAP pins and key gate driver components. It is recommended that the gate driver layout follow the example shown in Figure 9 to assure proper operation and long term reliability. The LTC3863 gate driver should connect to the external power elements in the following manner. First route the VIN pin using a single low impedance isolated trace to the positive RSENSE resistor PAD without connection to the VIN plane. The reason for this precaution is that the VIN pin is internally Kelvin connected to the current sense comparator, internal VIN power and the PMOS gate driver. Connecting the VIN pin to the VIN power plane adds noise and can result in jitter or instability. Figure 9 shows a single VIN trace from the positive RSENSE pad connected to CSF, CCAP, VIN pad and CINB. The total trace length to RSENSE should be minimized and the capacitors CCF, CCAP and CINB should be placed near the VIN pin of the LTC3863. 3863f For more information www.linear.com/3863 25 LTC3863 APPLICATIONS INFORMATION CINB VIN CCAP RUN CSS CAP CIN VIN CPITH SENSE SS GATE CITH CSF PLLIN/MODE RITH RFREQ RSF RGATE + RSENSE – Q1 D1 VOUT COUT LTC3863 L1 ITH RFB1 FREQ SGND VFBN RFB2 CFB2 GROUND PLANE TO PGND PGND VFB 3863 F08 Figure 8: LTC3863 Generic Application Schematic with Optional Current Sense Filter and Series Gate Resistor CINB TO Q1 GATE RGATE GATE CSF TO RSENSE+ VIN SENSE CAP CCAP RSF TO RSENSE– 3863 F09 Figure 9: LTC3863 Recommended Gate Driver PC Board Placement, Layout and Routing CCAP should be placed near the VIN and CAP pins. Figure 9 shows CCAP placed adjacent to the VIN and CAP pins with SENSE routed between the pads. This is the recommended layout and results in the minimum parasitic inductance. The gate driver is capable of providing high peak current. Parasitic inductance in the gate drive and the series inductance between VIN to CAP can cause a voltage spike between VIN and CAP on each switching cycle. The voltage spike can result in electrical over-stress to the gate driver and can result in gate driver failures in extreme cases. It is recommended to follow the example shown in Figure 9 for the placement of CCAP as close as is practical. RGATE resistor pads can be added with a 0Ω resistor to allow the damping resistor to be added later. The total length of the gate drive trace to the PMOS gate should be minimized and ideally be less than 1cm. In most cases with a good layout the RGATE resistor is not needed. The RGATE resistor should be located near the gate pin to reduce peak current through GATE and minimize reflected noise on the gate pin. The RSF and CSF pads can be added with a zero ohm resistor for RSF and CSF not populated. In most applications, external filtering is not needed. The current sense filter RSF and CSF can be added later if noise if demonstrated to be a problem. The bypass capacitor CINB is used to locally filter the VIN supply. CINB should be tied to the VIN pin trace and to the PGND exposed pad. The CINB positive pad should connect to RSENSE positive though the VIN pin trace. The CINB ground trace should connect to the PGND exposed pad connection. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3863. 1. Multilayer boards with dedicated ground layers are preferable for reduced noise and for heat sinking purposes. Use wide rails and/or entire planes for VIN, VOUT and GND for good filtering and minimal copper loss. If a ground layer is used, then it should be immediately below (and/or above) the routing layer for the power train components which consist of CIN, sense resistor, P-channel MOSFET, Schottky diode, inductor, and COUT. Flood unused areas of all layers with copper for better heat sinking. 2. Keep signal and power grounds separate except at the point where they are shorted together. Short the signal and power ground together only at a single point with a narrow PCB trace (or single via in a multilayer board). All power train components should be referenced to power ground and all small-signal components (e.g., CITH1, RFREQ, CSS etc.) should be referenced to the signal ground. 3863f 26 For more information www.linear.com/3863 LTC3863 APPLICATIONS INFORMATION 3. Place CIN, sense resistor, P-channel MOSFET, inductor, and primary COUT capacitors close together in one compact area. The junction connecting the drain of the P‑channel MOSFET, cathode of the Schottky, and (+) terminal of the inductor (this junction is commonly referred to as switch or phase node) should be compact but be large enough to handle the inductor currents without large copper losses. Place the sense resistor and source of P-channel MOSFET as close as possible to the (+) plate of the CIN capacitor(s) that provides the bulk of the AC current (these are normally the ceramic capacitors), and connect the (–) terminal of the inductor as close as possible to the (–) terminal of the same CIN capacitor(s). The high dI/dt loop formed by CIN, the MOSFET, and the Schottky diode should have short leads and PCB trace lengths to minimize high frequency EMI and voltage stress from inductive ringing. The (+) terminal of the primary COUT capacitor(s) which filter the bulk of the inductor ripple current (these are normally the ceramic capacitors) should also be connected close to the (–) terminal of CIN. 4. Place Pins 7 to 12 facing the power train components. Keep high dV/dt signals on GATE and switch away from sensitive small-signal traces and components. 5. Place the sense resistor close to the (+) terminal of CIN and source of P-channel MOSFET. Use a Kelvin (4-wire) connection across the sense resistor and route the traces together as a differential pair into the VIN and SENSE pins. An optional RC filter could be placed near the VIN and SENSE pins to filter the current sense signal. 6. Place the resistive feedback divider RFB1/2 as close as possible to the VFB and VFBN pins. The (–) terminal of the feedback divider should connect to the output regulation point and the (+) terminal of the feedback divider should connect to VFB. 7. Place the ceramic CCAP capacitor as close as possible to the VIN and CAP pins. This capacitor provides the gate discharging current for the power P-channel MOSFET. and RFREQ pins. Use sufficient isolation when routing a clock signal into the PLLIN /MODE pin so that the clock does not couple into sensitive small-signal pins. Failure Mode and Effects Analysis (FMEA) A FMEA study on the LTC3863 has been conducted through adjacent pin opens and shorts. The device was tested in a step-down application (Figure 15) from VIN = 12V to VOUT = –5V with a current load of 2A on the output. One group of tests involved the application being monitored while each pin was disconnected from the PC board and left open while all other pins remained intact. The other group of tests involved each pin being shorted to its adjacent pins while all other pins were connected as it would be normally in the application. The results are shown in Table 2. For FMEA compliance, the following design implementations are recommended: • If the RUN pin is being pulled up to a voltage greater than 6V, then it is done so through a pull-up resistor (100k to 1M) so that the VFBN pin is not damaged in case of a RUN to VFBN short. • The gate of the external P-channel MOSFET should be pulled through a resistor (20k to 100k) to the input supply, VIN so that the P-channel MOSFET is guaranteed to turn off in case of a GATE open. • To protect against the VFBN open condition it is necessary to add an output shutdown clamp. The output shutdown clamp is comprised of a Zener, VZ, NPN and Zener bias resistor, RZ, to ground as found in Figure 10. The clamp voltage will be the Zener forward voltage VZ plus a VBE. The clamp needs to be designed such that the worst-case minimum Zener voltage is less than the maximum operating voltage. The worst-case Zener leakage current times the Zener bias resistor should not exceed 200mV. 8. Place small signal components as close to their respective pins as possible. This minimizes the possibility of PCB noise coupling into these pins. Give priority to VFB, ITH, CZ VZ RZ SS 2N3904 –VOUT 3863 F10 Figure 10 3863f For more information www.linear.com/3863 27 LTC3863 APPLICATIONS INFORMATION Table 2 FAILURE MODE VOUT IOUT IVIN f None –5V 1A 453mA 350kHz RECOVERY WHEN FAULT IS REMOVED? BEHAVIOR N/A Normal Operation. Pin Open Open Pin 1 (PLLIN/MODE) –5V 1A 453mA 350kHz OK Pin already left open in normal application, so no difference. Open Pin 2 (FREQ) –5V 1A 453mA 535kHz OK Frequency jumps to default open value. Open Pin 3 (GND) –5V 1A 453mA 350kHz OK Exposed pad still provides GND connection to device. Open Pin 4 (SS) –5V 1A 453mA 350kHz OK External soft-start removed, but internal soft-start still available. Open Pin 5 (VFB) 0V 0A 0.7mA 0kHz OK Controller stops switching. VFB internally self biases HI to prevent switching. Open Pin 6 (ITH) –5V 1A 507mA 40kHz OK Output still regulating, but the switching is erratic. Loop not stable. Open Pin 7 (VFBN) –6V pk 1A 502mA 350kHz OK Use a 5.1V Zener VZ, 10k RZ and 0.01µF CZ. Output Voltage is –6V peak and averages –4.9V. Open Pin 8 (RUN) –5V 1A 453mA 350kHz OK Controller does not start-up. Open Pin 9 (CAP) –5V 1A 453mA 350kHz OK More jitter during switching, but regulates normally. Open Pin 10 (SENSE) 0V 0A 0.7mA 0kHz OK SENSE internally prebiases to 0.6V below VIN. This prevents controller from switching. –5.4V 1A 597mA 20kHz OK VIN able to bias internally through SENSE. Regulates with high VOUT ripple. Open Pin 12 (GATE) 0V 0A 0.7mA 0kHz OK Gate does not drive external power FET, preventing output regulation. Open Pin 13 (PGND) –5V 453mA 350kHz OK Pin 3 (GND) still provides GND connection to device. Open Pin 11 (VIN) Pins Shorted Short Pins 1, 2 (PLLIN/MODE and FREQ) –5V 1A 453mA 350kHz OK Burst Mode operation disabled, but runs normally as in pulseskipping mode. Short Pins 2, 3 (FREQ and GND) –5V 1A 453mA 0kHz OK FREQ already shorted to GND, so regulates normally. Short Pins 3, 4 (GND and SS) 0V 0A 0.7mA 0kHz OK SS short to GND prevents device from starting up. Short Pins 4, 5 (SS and VFB) –1V(DC) –3VP-P 50mA 9mA Erratic OK VOUT oscillates from 0V to 3V. Short Pins 5, 6 (VFB and ITH) –3.15V 625mA 181mA 350kHz OK Controller loop does not regulate to proper output voltage. Short Pins 7, 8 (VFBN and RUN) 5V 0A 860µA 350kHz OK Controller does not start-up. Short Pins 8, 9 (RUN and CAP) –5V 1A 453mA 350kHz OK Able to start-up and regulate normally. Short Pins 9, 10 (CAP and SENSE) 0V 0A 181mA 0kHz OK CAP ~ VIN, which prevents turning on external P-MOSFET. Short Pins 10, 11 (SENSE and VIN) –5V 1A 453mA 50kHz OK Regulates with high VOUT ripple. Short Pins 11, 12 (VIN and GATE) 0V 0A 29mA 0kHz OK Power MOSFET is always kept OFF, preventing regulation. 3863f 28 For more information www.linear.com/3863 LTC3863 TYPICAL APPLICATIONS 0.47µF CAP VIN 25mΩ RUN + COUT4 100µF 6.3V 0.47µF MOD 320kHz SS CAP RUN 220pF 16mΩ PLLIN/MODE 0.1µF SENSE SS 220pF D2 ITH PGOOD *VOUT FOLLOWS VIN WHEN 3.5V < VIN < 5.2V Q1 D1 GATE FREQ 15k 52.3k 10nF 15nF 10k 52.3k LTC3863 L1 12µH ITH SGND SGND PGND NOTE: LTC3863 CAN BE USED IN PLACE OF LTC3864 IF VFBN IS TIED > 2V D2: DIODES SBR3U100LP-7 L2: TOKO B1134AS-100M Q2: FAIRCHILD FDMC5614P COUT4: TDK C4532X5R0J07M COUT5: PANASONIC EEE-FK1V221P PGND 10 80 8 6 50 5 PULSE-SKIPPING MODE Burst Mode OPERATION 30 4 3 20 10 0 0.002 POWER LOSS 0.02 0.2 LOAD CURRENT (A) 2 90 9 70 7 60 8 6 EFFICIENCY 50 5 4 40 PULSE-SKIPPING MODE Burst Mode OPERATION 30 2 20 1 10 0 2 POWER LOSS 0 0.002 1 0.02 0.2 LOAD CURRENT (A) 3863 F11b 2 0 3863 F11d Negative 5V Gain/Phase Positive 5V Gain/Phase 70 105 60 90 60 90 50 75 50 75 40 60 45 30 20 10 15 GAIN 0 30 10 15 GAIN 0 0 –10 –15 –20 –30 –20 –30 –45 100 –30 10 FREQUENCY (kHz) 45 20 –10 1 60 PHASE 30 1 10 FREQUENCY (kHz) 3863 F11c 0 –15 –30 PHASE (DEG) 30 PHASE (DEG) PHASE GAIN (dB) 40 GAIN (dB) 3 POWER LOSS (W) 7 EFFICIENCY 40 3863 F11a CIN1: CDE AFK686M63G24T-F CIN2: TDK CGA6M3X7S2A475K COUT1 TDK C4532X7R1C336M COUT3 PANASONIC 20SVP100M D1: VISHAY SS8PH9-M3/87A L1: MSS1278-123ML Q1: VISHAY Si7469DP VIN = 12V 80 VOUT = –5V EFFICIENCY (%) 9 POWER LOSS (W) EFFICIENCY (%) VIN = 12V 90 VOUT = 5V 60 VFB 30.1k VOUT –5V COUT3 1.8A 100µF 20V Negative 5V Efficiency Positive 5V Efficiency 100 70 187k VFBN 12pF VFB COUT1 33µF 16V ×2 FREQ 422k 80.6k VIN* CIN1 4.5V TO 55V 68µF 63V + VIN GATE LTC3864 COUT5 220µF 35V SET 0.1µF Q2 L2 10µH CIN2 4.7µF 100V ×2 GND SYNC2 PLLIN/MODE SENSE VOUT* 5V 1.8A 316k LTC6908-1 SYNC1 V+ + 0.1µF –45 100 3863 F11e Figure 11. Design Example, 4.5V to 55V Input, ±5V, 1.8A at 320kHz Gain/Phase Measurements Taken with OMICRON Lab Bode 100 Vector Network Analyzer. For more information www.linear.com/3863 3863f 29 LTC3863 TYPICAL APPLICATIONS CAP RUN VIN 25mΩ PLLIN/MODE 0.1µF 4.7nF 50pF SENSE SS GATE 36.5k 100k LTC3863 Q1 D1 L1 15µH ITH COUT1 10µF 25V ×2 191k FREQ SGND VFBN 47pF PGND VFB 3863 F12a PULSE-SKIPPING MODE Burst Mode OPERATION VIN = 12V VOUT = –18V 50 60 8 50 7 40 6 5 EFFICIENCY 4 40 90 VIN = 12V VOUT = –18V 75 60 PHASE 30 45 20 30 10 15 GAIN 30 3 20 2 –10 –15 1 –20 –30 0 –30 10 0 0.001 POWER LOSS 0.01 0.1 LOAD CURRENT (A) 1 0 1 10 FREQUENCY (kHz) 0 PHASE (DEG) 60 9 POWER LOSS (W) EFFICIENCY (%) Gain/Phase GAIN (dB) 80 70 VOUT –18V COUT3 700mA 100µF 20V CIN1: NICHICON UCJ1H101MCL1GS C : TDK C3225X7R1E108M 8.45k CIN2 : MURATA GRM32DR61E106KA12L OUT1 COUT3: PANASONIC 20SVO100M D1: DIODES SBR8U60P5 L1: WURTH 744770115 Q1: VISHAY SI7469DP Efficiency 90 + 750kHz VIN CIN2 5V TO 23V 10µF 25V ×2 CIN1 + 100µF 50V 0.47µF –45 100 3863 F12c 3863 F12b Figure 12. 5V to 23V Input, –18V/700mA Output, 750kHz Inverting Converter Gain/Phase Measurements Taken with OMICRON Lab Bode 100 Vector Network Analyzer. 30 For more information www.linear.com/3863 3863f LTC3863 TYPICAL APPLICATIONS 0.47µF CAP RUN VIN 82mΩ PLLIN/MODE 0.1µF 3.3nF 82pF SENSE SS GATE 5k 61.9k LTC3863 Q1 D1 L1 15µH ITH COUT1 10µF 25V ×2 95.3k FREQ SGND VFBN 10pF VFB PGND 3863 F13a 191k Gain/Phase 0.4 0.2 10 0.1 POWER LOSS 0 0.2 60 PHASE 30 GAIN (dB) EFFICIENCY 20 40 45 20 30 10 15 GAIN 0 0 –10 –15 –20 –30 –30 1 10 FREQUENCY (kHz) PHASE (DEG) 0.3 0.02 LOAD CURRENT (A) 90 VIN = 12V VOUT = –0.4V 75 50 30 0 0.002 60 0.5 PULSE-SKIPPING MODE Burst Mode OPERATION VIN = 12V VOUT = –0.4V POWER LOSS (W) EFFICIENCY (%) 40 VOUT –0.4V COUT3 200mA 100µF 20V CIN1: NICHICON UCJ1H101MCL1GS CIN2: MURATA GRM32ER71H108H COUT1: TDK C4532X7R1C336M COUT3: PANASONIC 16TQC150MYF D1: DIODES B540C-13-F L1: WURTH 7447779115 Q1: FAIRCHILD FDMC5614P Efficiency 50 + 80kHz VIN CIN1 3.5V TO 28V 100µF 50V + CIN2 10µF 50V ×2 –45 100 3863 F13c 3863 F13b Figure 13. 3.5V to 28V Input, –0.4V/200mA Output, 80kHz Inverting Converter Gain/Phase Measurements Taken with OMICRON Lab Bode 100 Vector Network Analyzer. For more information www.linear.com/3863 3863f 31 LTC3863 TYPICAL APPLICATIONS 0.47µF 440kHz RUN CAP VIN 27mΩ PLLIN/MODE + VIN CIN1 12V TO 42V 100µF 50V SENSE SS 1M 64.9k GATE LTC3863 Q1 D1 L1 10µH ITH COUT1 4.7µF 100V ×2 196k FREQ SGND VFBN 100pF PGND VFB 3863 F14a PULSE-SKIPPING MODE Burst Mode OPERATION VIN = 24V VOUT = –48V 50 70 8 60 7 50 75 40 60 6 EFFICIENCY 5 105 VIN = 24V VOUT = –48V 90 PHASE 30 45 30 20 40 4 30 3 20 2 –10 –15 1 –20 –30 0 –30 10 0 0.001 POWER LOSS 0.01 LOAD CURRENT (A) 0.1 10 15 GAIN PHASE (DEG) 60 9 POWER LOSS (W) EFFICIENCY (%) Gain/Phase GAIN (dB) 80 70 VOUT –48V COUT3 300mA 100µF 63V CIN1: NICHICON UCJ1H101MCL1GS C : MURATA GRM32ER71H106H 3.32k CIN2 : TDK CGA6M3X7S2A475K OUT1 COUT3: UCC EMVH630ARA101MKE D1: DIODES PDS5100H L1: WURTH 744314101 Q1: VISHAY SI7113DN Efficiency 90 + 0.1µF 18nF CIN2 10µF 50V ×2 0 0 2 20 FREQUENCY (kHz) –45 200 3863 F14c 3863 F14b Figure 14. 12V to 42V Input, –48V/300mA Output, 440kHz Inverting Converter Gain/Phase Measurements Taken with OMICRON Lab Bode 100 Vector Network Analyzer. 32 For more information www.linear.com/3863 3863f LTC3863 TYPICAL APPLICATIONS 0.47µF RUN CAP VIN 16mΩ PLLIN/MODE 0.1µF 27nF 390pF 61.9k VIN CIN1 4.5V TO 16V 100µF 20V SENSE SS Q1 D1 GATE 14.7k + LTC3863 L1 10µH ITH RFBO 698k 511k FREQ SGND VFBN 68pF VFB PGND 3863 F15a COUT1 33µF 16V ×2 + 350kHz CIN2 10µF 25V ×2 CIN1: PANASONIC 20SVP100M C : TDK C3225X7R1E106M 80.6k CIN2 : TDK C4532X7R1C336M OUT1 COUT3: PANASONIC 16TQC150MYF D1: DIODES B540C L1: TOKO 919AS-100M Q1: VISHAY SI7129DN-T1-GE3 –5V Efficiency –12V Efficiency 9 70 90 9 8 VIN = 12V 80 VOUT = –12V 7 70 7 50 5 40 4 30 PULSE-SKIPPING MODE Burst Mode OPERATION 3 60 8 EFFICIENCY 6 50 40 30 5 4 PULSE-SKIPPING MODE Burst Mode OPERATION 3 20 2 20 2 10 1 10 1 0 0.002 POWER LOSS 0.02 0.2 LOAD CURRENT (A) 2 0 0 0.002 POWER LOSS 0.02 0.2 LOAD CURRENT (A) 3863 F15b 2 POWER LOSS (W) 6 EFFICIENCY POWER LOSS (W) EFFICIENCY (%) VIN = 12V 80 VOUT = –5V EFFICIENCY (%) 90 60 VOUT –5V 1.7A (SHORT RFBO) COUT3 –12V 1A 150µF 16V ×2 0 3863 F15b Figure 15. 4.5V to 16V Input, –5V/1.7A, –12V/1A Output, 350kHz Inverting Converter 3863f For more information www.linear.com/3863 33 LTC3863 PACKAGE DESCRIPTION Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. DE/UE Package 12-Lead Plastic DFN (4mm × 3mm) (Reference LTC DWG # 05-08-1695 Rev D) 0.70 ±0.05 3.30 ±0.05 3.60 ±0.05 2.20 ±0.05 1.70 ±0.05 PACKAGE OUTLINE 0.25 ±0.05 0.50 BSC 2.50 REF RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 4.00 ±0.10 (2 SIDES) 7 R = 0.115 TYP 0.40 ±0.10 12 R = 0.05 TYP PIN 1 TOP MARK (NOTE 6) 0.200 REF 3.30 ±0.10 3.00 ±0.10 (2 SIDES) 1.70 ±0.10 0.75 ±0.05 6 0.25 ±0.05 1 PIN 1 NOTCH R = 0.20 OR 0.35 × 45° CHAMFER (UE12/DE12) DFN 0806 REV D 0.50 BSC 2.50 REF 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING PROPOSED TO BE A VARIATION OF VERSION (WGED) IN JEDEC PACKAGE OUTLINE M0-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3863f 34 For more information www.linear.com/3863 LTC3863 PACKAGE DESCRIPTION Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. MSE Package 12-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1666 Rev F) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ±0.102 (.112 ±.004) 5.23 (.206) MIN 2.845 ±0.102 (.112 ±.004) 0.889 ±0.127 (.035 ±.005) 6 1 1.651 ±0.102 (.065 ±.004) 1.651 ±0.102 3.20 – 3.45 (.065 ±.004) (.126 – .136) 12 0.65 0.42 ±0.038 (.0256) (.0165 ±.0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 0.35 REF 4.039 ±0.102 (.159 ±.004) (NOTE 3) 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 7 NO MEASUREMENT PURPOSE 0.406 ±0.076 (.016 ±.003) REF 12 11 10 9 8 7 DETAIL “A” 0° – 6° TYP 3.00 ±0.102 (.118 ±.004) (NOTE 4) 4.90 ±0.152 (.193 ±.006) GAUGE PLANE 0.53 ±0.152 (.021 ±.006) DETAIL “A” 1.10 (.043) MAX 0.18 (.007) SEATING PLANE 0.22 – 0.38 (.009 – .015) TYP 1 2 3 4 5 6 0.650 (.0256) BSC NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.86 (.034) REF 0.1016 ±0.0508 (.004 ±.002) MSOP (MSE12) 0911 REV F 3863f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection its circuits as described herein will not infringe on existing patent rights. Forofmore information www.linear.com/3863 35 LTC3863 TYPICAL APPLICATION Efficiency 90 9 70 7 RUN CAP VIN 39mΩ PLLIN/MODE 180pF SS 52.3k LTC3863 Q1 D1 1M 1M FREQ SGND VFBN 3.3pF 40 VFB 3863 F16a COUT1 1µF 250V ×2 30 3 2 POWER LOSS 0.01 LOAD CURRENT (A) 1 0.1 0 3863 F16b Gain/Phase 40 CIN1: CDE AFK686M63G24T-F C : MURATA GRM32ER71H106H 10.7k CIN2 : TDK CGA8P3X7T2E105K/SOFT OUT1 COUT3: LELON VEJ-470M2DTR-1616 D1: ON SEMI MBRS3201T3G L1: TOKO 1217AS-H-150M Q1: VISHAY SI7119 4 PULSE-SKIPPING MODE Burst Mode OPERATION 135 VIN = 24V 35 V OUT = –150V 30 120 105 PHASE 25 90 20 75 15 60 10 45 5 30 0 15 GAIN –5 0 –10 –15 –15 –30 –20 0.1 PHASE (DEG) PGND 5 0 0.001 VOUT –150V COUT3 40mA 47µF 200V 6 50 10 L1 15µH ITH EFFICIENCY 20 SENSE GATE 845k VIN CIN3 12V TO 40V 68µF 63V EFFICIENCY (%) 0.1µF 1.8pF + + 320kHz CIN2 10µF 50V ×2 60 8 POWER LOSS (W) 0.47µF EFFICIENCY (%) VIN = 24V 80 VOUT = –150V 1 10 FREQUENCY (Hz) –45 100 3963 F16c Figure 16. 12V to 40V Input, –150V/40mA Output, 320kHz Inverting Converter Gain/Phase Measurements Taken with OMICRON Lab Bode 100 Vector Network Analyzer. RELATED PARTS PART NUMBER LTC3864 DESCRIPTION Low IQ, High Voltage Step-Down DC/DC Controller with 100% Duty Cycle LTC3891 60V, Low IQ, Synchronous Step-Down DC/DC Controller LTC3890/LTC3890-1 60V, Low IQ, Dual 2-Phase Synchronous LTC3890-2/LTC3890-3 Step-Down DC/DC Controller LTC3630 High Efficiency, 65V, 500mA Synchronous Step-Down Regulator LTC3834/LTC3834-1 LTC3835/LTC3835-1 LT3758A LTC3826/LTC3826-1 LTC3859AL Low IQ, Single Output Synchronous Step-Down DC/DC Controllers with 99% Duty Cycle High Input Voltage, Boost, Flyback, SEPIC and Inverting Controller COMMENTS Fixed Frequency 50kHz to 850kHz, 3.5V≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ VIN, IQ = 40µA, MSOP-12E, 3mm × 4mm DFN-12 PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ 24V, IQ = 50µA PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ 24V, IQ = 50µA 4V ≤ VIN ≤ 65V, 0.8V ≤ VOUT ≤ VIN, IQ = 12µA, 3mm × 5mm DFN-16 and MSOP-16E PLL Fixed Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, IQ = 30µA/80µA 5.5V ≤ VIN ≤ 100V, Positive or Negative VOUT, 3mm × 3mm DFN-10 and MSOP-10E PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, Low IQ, Dual Output 2-Phase Synchronous Step-Down DC/DC Controllers with 99% Duty Cycle IQ = 30µA All Outputs Remain in Regulation Through Cold Crank 2.5V ≤ VIN ≤ 38V, Low IQ, Triple Output Buck/Buck/Boost Synchronous DC/DC Controller VOUT(BUCKS) Up to 24V, VOUT(BOOST) Up to 60V, IQ = 28µA 3863f 36 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/3863 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/3863 LT 0313 • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2013