AD ADXL50 Monolithic accelerometer with signal conditioning Datasheet

a
Monolithic Accelerometer
With Signal Conditioning
ADXL50*
For convenience, the ADXL50 has an internal buffer amplifier
with a full 0.25 V to 4.75 V output range. This may be used to
set the zero-g level and change the output sensitivity by using
external resistors. External capacitors may be added to the resistor network to provide 1 or 2 poles of filtering. No external
active components are required to interface directly to most
analog-to-digital converters (ADCs) or microcontrollers.
FEATURES
Complete Acceleration Measurement System
on a Single Monolithic IC
Full-Scale Measurement Range: 650 g
Self-Test on Digital Command
+5 V Single Supply Operation
Sensitivity Precalibrated to 19 mV/g
Internal Buffer Amplifier for User Adjustable Sensitivity
and Zero-g Level
Frequency Response: DC to 10 kHz
Post Filtering with External Passive Components
High Shock Survival: >2000 g Unpowered
Other Versions Available: ADXL05 (65 g)
The ADXL50 uses a capacitive measurement method. The analog output voltage is directly proportional to acceleration, and is
fully scaled, referenced and temperature compensated, resulting
in high accuracy and linearity over a wide temperature range.
Internal circuitry implements a forced-balance control loop that
improves accuracy by compensating for any mechanical sensor
variations.
OBS
OLE
GENERAL DESCRIPTION
The ADXL50 is powered from a standard +5 V supply and is
robust for use in harsh industrial and automotive environments
and will survive shocks of more than 2000 g unpowered.
The ADXL50 is a complete acceleration measurement system on
a single monolithic IC. Three external capacitors and a +5 volt
power supply are all that is required to measure accelerations up
to ± 50 g. Device sensitivity is factory trimmed to 19 mV/g,
resulting in a full-scale output swing of ± 0.95 volts for a ± 50 g
applied acceleration. Its zero g output level is +1.8 volts.
TE
The ADXL50 is available in a hermetic 10-pin TO-100 metal
can, specified over the 0°C to +70°C commercial, and –40°C to
+85°C industrial temperature ranges. Contact factory for availability of devices specified for operation over the –40°C to
+105°C automotive temperature range.
A TTL compatible self-test function can electrostatically deflect
the sensor beam at any time to verify device functionality.
FUNCTIONAL BLOCK DIAGRAM
ADXL50
+3.4V
6
REFERENCE
VREF
OUTPUT
+1.8V
OSCILLATOR
DECOUPLING
CAPACITOR
4
OSCILLATOR
SENSOR
DEMODULATOR
C2
BUFFER
AMP
PREAMP
SELF TEST 7
(ST)
1
5
COM
2
C1
C3
+5V
3
8
C1
DEMODULATOR
CAPACITOR
VPR
10
R1
9
VIN–
R3
VOUT
R2
*Patents pending.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
© Analog Devices, Inc., 1996
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
(TA = TMIN to TMAX, TA = +258C for J Grade Only, VS = +5 V, @ Acceleration = 0 g,
ADXL50–SPECIFICATIONS unless otherwise noted)
Parameter
SENSOR INPUT
Measurement Range
Nonlinearity
Alignment Error1
Transverse Sensitivity2
Conditions
Min
Guaranteed Full Scale
Best Fit Straight Line, 50 g FS
–50
SENSITIVITY
Initial Sensitivity at VPR
Temperature Drift3
+25°C
16.1
1.55/1.60
ZERO g BIAS LEVEL
Initial Offset
vs. Temperature3
vs. Supply
at VPR
NOISE PERFORMANCE
Voltage Noise Density
Noise in 100 Hz Bandwidth
Noise in 10 Hz Bandwidth
at VPR
BW = 10 Hz to 1 kHz
SELF TEST INPUT
Output Change at VPR5
Logic “1” Voltage
Logic “0” Voltage
Input Resistance
+3.4 V REFERENCE
Output Voltage
Output Temperature Drift3
Power Supply Rejection
Output Current
PREAMPLIFIER OUTPUT
Voltage Swing
Current Output
Capacitive Load Drive
BUFFER AMPLIFIER
Input Offset Voltage6
Input Bias Current
Open-Loop Gain
Unity Gain Bandwidth
Output Voltage Swing
Capacitive Load Drive
Power Supply Rejection
Max
Units
+50
g
% of FS
Degrees
%
19.0
0.75/1.0
21.9
mV/g
% of Reading
1.80
± 15/35
10
2.05/2.00
V
mV
mV/V
6.6
66
20
12
0.2
±1
±2
VS = 4.75 V to 5.25 V
OBS
FREQUENCY RESPONSE
3 dB Bandwidth4
3 dB Bandwidth4
Sensor Resonant Frequency
ADXL50J/A
Typ
C1 = 0.022 µF (See Figure 22)
C1 = 0.0068 µF
800
ST Pin from Logic “0” to “1”
–0.85
2.0
32
1300
10
24
Hz
kHz
kHz
OLE
To Common
–1.00
50
3.350
DC, VS = +4.75 V to +5.25 V
Sourcing
500
Source or Sink
0.25
30
80
100
± 10
5
80
200
Delta from Nominal 1.800 V
DC
IOUT = ± 100 µA
3.400
± 10
1
DC, VS = +4.75 V to +5.25 V
0
–40
–40
0.8
V
V
V
kΩ
TE
3.450
10
V
mV
mV/V
µA
VS – 1.4
V
µA
pF
± 25
20
1
10
10
5.25
13
V
mA
+70
+85
+125
°C
°C
°C
VS – 0.25
4.75
TEMPERATURE RANGE
Operating Range J
Specified Performance A
Automotive Grade*
–1.15
mV
nA
dB
kHz
V
pF
mV/V
0.25
1000
POWER SUPPLY
Operating Voltage Range
Quiescent Supply Current
mg/√Hz
mg rms
mg rms
NOTES
1
Alignment error is specified as the angle between the true and indicated axis of sensitivity, (see Figure 2).
Transverse sensitivity is measured with an applied acceleration that is 90° from the indicated axis of sensitivity. Transverse sensitivity is specified as the percent of
transverse acceleration that appears at the V PR output. This is the algebraic sum of the alignment and the inherent sensor sensitivity errors, (see Figure 2).
3
Specification refers to the maximum change in parameter from its initial at +25°C to its worst case value at T MIN to TMAX.
4
Frequency at which response is 3 dB down from dc response assuming an exact C1 value is used. Maximum recommended BW is 10 kHz using a 0.007 µF capacitor, refer to
Figure 22.
5
Applying logic high to the self-test input has the effect of applying an acceleration of –52.6 g to the ADXL50.
6
Input offset voltage is defined as the output voltage differential from 1.800 V when the amplifier is connected as a follower (i.e., Pins 9 and 10 tied together). The voltage at
Pin 9 has a temperature drift proportional to that of the 3.4 V reference.
*Contact factory for availability of automotive grade devices.
2
All min and max specifications are guaranteed. Typical specifications are not tested or guaranteed.
Specifications subject to change without notice.
–2–
REV. B
ADXL50
ABSOLUTE MAXIMUM RATINGS*
Package Characteristics
Acceleration (Any Axis, Unpowered for 0.5 ms) . . . . . . 2000 g
Acceleration (Any Axis, Powered for 0.5 ms) . . . . . . . . . . 500 g
+VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7.0 V
Output Short Circuit Duration
(VPR, VOUT, VREF Terminals to Common) . . . . . . . Indefinite
Operating Temperature . . . . . . . . . . . . . . . . . –55°C to +125°C
Storage Temperature . . . . . . . . . . . . . . . . . . . –65°C to +150°C
Package
uJA
uJC
Device Weight
10-Pin TO-100
130°C/W
30°C/W
5 Grams
ORDERING GUIDE
*Stresses above those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. This is a stress rating only; the functional
operation of the device at these or any other conditions above those indicated in the
operational sections of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
Model
Temperature
Range
ADXL50JH
ADXL50AH
0°C to +70°C
–40°C to +85°C
OBS
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADXL50 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
OLE
The power supply input pin.
C2
Connection for an external bypass capacitor (nominally
0.022 µF) used to prevent oscillator switching noise
from interfering with other ADXL50 circuitry. Please
see the section on component selection.
C1
TOP VIEW
COM
VREF
ST
7
88
COM The power supply common (or “ground”) connection.
Output of the internal 3.4 V voltage reference.
ST
The digital self-test input. It is both CMOS and TTL
compatible.
VPR
The ADXL50 preamplifier output providing an output
voltage of 19 mV per g of acceleration.
VPR
REV. B
2
THE CASE OF THE METAL CAN
PACKAGE IS CONNECTED TO
PIN 5 (COMMON).
+5V
VIN–
–3–
NOTES:
AXIS OF SENSITIVITY IS ALONG
A LINE BETWEEN PIN 5 AND
THE TAB.
C1
10
VOUT
The inverting input of the uncommitted buffer amplifier.
TE
C1
3
1
9
VOUT Output of the uncommitted buffer amplifier.
VIN–
44
66
Connections for the demodulator capacitor, nominally
0.022 µF. See the section on component selection for
application information.
VREF
C2
5
SENSITIVITY
+5 V
ESD SENSITIVE DEVICE
CONNECTION DIAGRAM
10-Header (TO-100)
AXIS OF
PIN DESCRIPTION
WARNING!
ARROW INDICATES DIRECTION
OF POSITIVE ACCELERATION
ALONG AXIS OF SENSITIVITY.
ADXL50
Polarity of the Acceleration Output
The polarity of the ADXL50 output is shown in the Figure 1.
When oriented to the earth’s gravity (and held in place), the
ADXL50 will experience an acceleration of +1 g. This corresponds to a change of approximately +19 mV at the VPR output
pin. Note that the polarity will be reversed to a negative going
signal at the buffer amplifier output VOUT, due to its inverting
configuration.
TAB
+1g
PIN 5
Mounting Considerations
There are three main causes of measurement error when using
accelerometers. The first two are alignment and transverse sensitivity errors. The third source of error is due to resonances or
vibrations of the sensor in its mounting fixture.
OBS
Errors Due to Misalignment
Figure 1. Output Polarity at VPR
The ADXL50 is a sensor designed to measure accelerations that
result from an applied force. Because these forces act on the
sensor in a vector manner, the alignment of the sensor to the
force to be measured may be critical.
Z
TRANSVERSE Z AXIS
SIDE VIEW
X
X
PIN 5
TAB
OLE
The ADXL50 responds to the component of acceleration on its
sensitive X axis. Figures 2a and 2b show the relationship between the sensitive “X” axis and the transverse “Z” and “Y”
axes as they relate to the TO-100 package.
SENSITIVE (X) AXIS
Z
TE
Figure 2c describes a three dimensional acceleration vector
(AXYZ) which might act on the sensor, where AX is the component of interest. To determine AX, first, the component of acceleration in the XY plane (AXY) is found using the cosine law:
Figure 2a. Sensitive X and Transverse Z Axis
Y
TRANSVERSE Y AXIS
AXY = AXYZ (cosθXY) then
AX = AXY (cosθX)
Therefore: Typical VPR = 19 mV/g (AXYZ) (cosθXY) cosθX
TOP VIEW
X
Note that an ideal sensor will react to forces along or at angles
to its sensitive axis but will reject signals from its various transverse axes, i.e., those exactly 90° from the sensitive “X” axis.
But even an ideal sensor will produce output signals if the transverse signals are not exactly 90° to the sensitive axis. An acceleration that is acting on the sensor from a direction different
from the sensitive axis will show up at the ADXL50 output at a
reduced amplitude.
X
PIN 5
TAB
SENSITIVE (X) AXIS
Y
Figure 2b. Sensitive X and Transverse Y Axis
Table I. Ideal Output Signals for Off Axis Applied
Accelerations Disregarding Device Alignment and
Transverse Sensitivity Errors
–Z Axis
Axyz
θxy
Ax
X Axis
θx
Axy
Y Axis
Figure 2c. A Vector Analysis of an Acceleration Acting
Upon the ADXL50 in Three Dimensions
–4–
θX
% of Signal Appearing
at Output
Output in gs for a 50 g
Applied Acceleration
0
1°
2°
3°
5°
10°
30°
45°
60°
80°
85°
87°
88°
89°
90°
100%
99 98%
99.94%
99.86%
99.62%
98.48%
86.60%
70.71%
50.00%
17.36%
8.72%
5.25%
3.49%
1.7%
0%
50 (On Axis)
49.99
49.97
49.93
49.81
49.24
43.30
35.36
25.00
8.68
4.36
2.63
1.75
0.85
0.00 (Transverse Axis)
REV. B
ADXL50
Table I shows the percentage signals resulting from various θX
angles. Note that small errors in alignment have a negligible
effect on the output signal. A 1° error will only cause a 0.02%
error in the signal. Note, however, that a signal coming 1° off of
the transverse axis (i.e., 89° off the sensitive axis) will still contribute 1.7% of its signal to the output. Thus large transverse
signals could cause output signals as large as the signals of
interest.
Sensitivity: The output voltage change per g unit of acceleration applied, specified at the VPR pin in mV/g.
Sensitive Axis (X): The most sensitive axis of the accelerometer sensor. Defined by a line drawn between the package tab
and Pin 5 in the plane of the pin circle. See Figures 2a and 2b.
Sensor Alignment Error: Misalignment between the
ADXL50’s on-chip sensor and the package axis, defined by
Pin 5 and the package tab.
Table I may also be used to approximate the effect of the
ADXL50’s internal errors due to misalignment of the die to the
package. For example: a 1 degree sensor alignment error will
allow 1.7% of a transverse signal to appear at the output. In a
nonideal sensor, transverse sensitivity may also occur due to inherent sensor properties. That is, if the sensor physically moves
due to a force applied exactly 90° to its sensitive axis, then this
might be detected as an output signal, whereas an ideal sensor
would reject such signals. In every day use, alignment errors
may cause a small output peak with accelerations applied close
to the sensitive axis but the largest errors are normally due to
large accelerations applied close to the transverse axis.
Total Alignment Error: Net misalignment of the ADXL50’s
on-chip sensor and the measurement axis of the application.
This error includes errors due to sensor die alignment to the
package, and any misalignment due to installation of the sensor
package in a circuit board or module.
OBS
Transverse Acceleration: Any acceleration applied 90° to the
axis of sensitivity.
Transverse Sensitivity Error: The percent of a transverse acceleration that appears at the VPR output. For example, if the
transverse sensitivity is 1%, then a +10 g transverse acceleration
will cause a 0.1 g signal to appear at VPR (1% of 10 g). Transverse sensitivity can result from a sensitivity of the sensor to
transverse forces or from misalignment of the internal sensor to
its package.
OLE
Errors Due to Mounting Fixture Resonances
A common source of error in acceleration sensing is resonance
of the mounting fixture. For example, the circuit board that the
ADXL50 mounts to may have resonant frequencies in the same
range as the signals of interest. This could cause the signals
measured to be larger than they really are. A common solution
to this problem is to dampen these resonances by mounting the
ADXL50 near a mounting post or by adding extra screws to
hold the board more securely in place.
TE
Transverse Y Axis: The axis perpendicular (90°) to the package axis of sensitivity in the plane of the package pin circle. See
Figure 2.
Transverse Z Axis: The axis perpendicular (90°) to both the
package axis of sensitivity and the plane of the package pin
circle. See Figure 2.
When testing the accelerometer in your end application, it is
recommended that you test the application at a variety of frequencies in order to ensure that no major resonance problems
exist.
100
GLOSSARY OF TERMS
90
Acceleration: Change in velocity per unit time.
Acceleration Vector: Vector describing the net acceleration
acting upon the ADXL50 (AXYZ).
g: A unit of acceleration equal to the average force of gravity
occurring at the earth’s surface. A g is approximately equal to
32.17 feet/s2, or 9.807 meters/s2.
10
0%
Nonlinearity: The maximum deviation of the ADXL50 output
voltage from a best fit straight line fitted to a plot of acceleration
vs. output voltage, calculated as a % of the full-scale output
voltage (@ 50 g).
1V
0.5ms
Figure 3. 500 g Shock Overload Recovery. Top Trace:
ADXL50 Output. Bottom Trace: Reference Accelerometer
Output
Resonant Frequency: The natural frequency of vibration of
the ADXL50 sensor’s central plate (or “beam”). At its resonant
frequency of 24 kHz, the ADXL50’s moving center plate has a
peak in its frequency response with a Q of 3 or 4.
REV. B
0.5V
–5–
ADXL50–Typical Characteristics
7
9
5 g p-p SIGNAL T A = +25°C
6
6
0
NOISE – mV, RMS
NORMALIZED SENSITIVITY – dB
C1 = C2 = 0.022µF
3
–3
–6
–9
5
4
–12
3
–15
–18
2
0.010
–21
OBS
1
10
100
FREQUENCY – Hz
1k
10k
Figure 4. Normalized Sensitivity vs. Frequency
Figure 7. RMS Noise vs. Value of Demodulator
Capacitor, C1
0.15%
0.10%
0.05%
0.00%
–0.05%
–0.10%
100
TA = +25°C, ACL = 2
80
OUTPUT IMPEDANCE – Ω
0.20%
TE
60
40
20
–0.15%
–0.20%
–0.25%
0.1
OLE
0.25%
LINEARITY IN % OF FULL SCALE
DEMODULATOR CAPACITANCE – µF
0
20
10
30
40
50
0
60
10
100
1000
FREQUENCY – Hz
g LEVEL APPLIED
Figure 5. Linearity in Percent of Full Scale
10000
Figure 8. Buffer Amplifier Output Impedance vs. Frequency
1350
30
TA = +25°C
25
G = 10
20
1300
15
1250
GAIN – dB
–3dB BW – Hz
C1 = C2 = 0.022µF
1200
10
G=2
5
0
–5
–10
1150
–15
1100
–60
–20
–40
–20
0
20
40
60
80
TEMPERATURE – °C
100
120
10
140
Figure 6. –3 dB Bandwidth vs. Temperature at VPR
100
1k
10k
FREQUENCY – Hz
100k
1M
Figure 9. Buffer Amplifier Closed-Loop Gain vs. Frequency
–6–
REV. B
ADXL50
0.5
TA = +25 °C
C1 = C2 = 0.022µF
CHANGE IN SENSITIVITY – %
CHANGE IN SENSITIVITY AT VPR – %
+0.50
+0.25
0
–0.25
0.0
–0.5
–1.0
–1.5
OBS
–0.50
4.8
4.9
5.0
5.1
SUPPLY VOLTAGE – V
5.2
5.3
–60
Figure 10. Change in Sensitivity vs. Supply Voltage
45
35
25
10
100
1k
FREQUENCY – Hz
120
140
OLE
10k
TE
60
40
100k
1
10
100
1k
FREQUENCY – Hz
10k
100k
Figure 14. +3.4 V REF PSRR vs. Frequency
3.404
40
VREF
20
10
0
–60
–1004
3.400
30
VREF OUTPUT – Volts
CHANGE IN 0g OUTPUT LEVEL – mV
100
TA = +25 °C
VS = +5V + (0.5Vp-p)
20
1
20
40
60
80
TEMPERATURE – °C
80
Figure 11. VPR 0 g PSRR vs. Frequency
–40
–20
0
20
40
60
80
TEMPERATURE – °C
100
120
–1000
3.396
SELF–TEST
3.392
–0.996
3.388
–0.992
3.384
–60
140
Figure 12. 0 g Bias Level vs. Temperature
REV. B
0
CHANGE IN SELF–TEST OUTPUT SWING – mV
VPR 0g PSRR – dB
TA = +25 °C
VS = +5V + (0.5Vp-p)
C1 = C2 = 0.022µF
–20
Figure 13. Percent Change in Sensitivity at VPR vs.
Temperature
+3.4V REF PSRR – dB
55
–40
–0.988
–40
–20
0
20
40
60
80
TEMPERATURE – °C
100
120
140
Figure 15. VREF Output and Change in Self-Test Output
Swing vs. Temperature
–7–
ADXL50
demodulator will rectify any voltage which is in sync with its
clock signal. If the applied voltage is in sync and in phase with
the clock, a positive output will result. If the applied voltage is in
sync but 180° out of phase with the clock, then the demodulator’s output will be negative. All other signals will be rejected.
An external capacitor, C1, sets the bandwidth of the demodulator.
THEORY OF OPERATION
The ADXL50 is a complete acceleration measurement system
on a single monolithic IC. It contains a polysilicon surface-micro machined sensor and signal conditioning circuitry. The
ADXL50 is capable of measuring both positive and negative acceleration to a maximum level of ± 50 g.
The output of the synchronous demodulator drives the preamp
—an instrumentation amplifier buffer which is referenced to
+1.8 volts. The output of the preamp is fed back to the sensor
through a 3 MΩ isolation resistor. The correction voltage required to hold the sensor’s center plate in the 0 g position is a
direct measure of the applied acceleration and appears at the
VPR pin.
Figure 16 is a simplified view of the ADXL50’s acceleration
sensor at rest. The actual structure of the sensor consists of 42
unit cells and a common beam. The differential capacitor sensor
consists of independent fixed plates and a movable “floating”
central plate which deflects in response to changes in relative
motion. The two capacitors are series connected, forming a capacitive divider with a common movable central plate. A force
balance technique counters any impeding deflection due to acceleration and servos the sensor back to its 0 g position.
OBS
TOP VIEW
TOP VIEW
CS1
TETHER
CS2
CENTER
PLATE
BEAM
CENTER
PLATE
FIXED
OUTER
PLATES
APPLIED
ACCELERATION
OLE
BEAM
CS1
CS2
CS1
CS2
UNIT CELL
CS1 = CS2
TE
UNIT CELL
CS1 < CS2
DENOTES ANCHOR
Figure 17. The ADXL50 Sensor Momentarily Responding
to an Externally Applied Acceleration
DENOTES ANCHOR
When the ADXL50 is subjected to an acceleration, its capacitive
sensor begins to move creating a momentary output signal. This
is signal conditioned and amplified by the demodulator and
preamp circuits. The dc voltage appearing at the preamp output
is then fed back to the sensor and electrostatically forces the
center plate back to its original center position.
Figure 16. A Simplified Diagram of the ADXL50 Sensor at
Rest
Figure 17 shows the sensor responding to an applied acceleration. When this occurs, the common central plate or “beam”
moves closer to one of the fixed plates while moving further
from the other. The sensor’s fixed capacitor plates are driven
deferentially by a 1 MHz square wave: the two square wave amplitudes are equal but are 180° out of phase from one another.
When at rest, the values of the two capacitors are the same and
therefore, the voltage output at their electrical center (i.e., at the
center plate) is zero.
At 0 g the ADXL50 is calibrated to provide +1.8 volts at the
VPR pin. With an applied acceleration, the VPR voltage changes
to the voltage required to hold the sensor stationary for the duration of the acceleration and provides an output which varies
directly with applied acceleration.
The loop bandwidth corresponds to the time required to apply
feedback to the sensor and is set by external capacitor C1. The
loop response is fast enough to follow changes in g level up to
and exceeding 1 kHz. The ADXL50’s ability to maintain a flat
response over this bandwidth keeps the sensor virtually motionless. This essentially eliminates any nonlinearity or aging effects
due to the sensor beam’s mechanical spring constant, as compared to an open-loop sensor.
When the sensor begins to move, a mismatch in the value of
their capacitance is created producing an output signal at the
central plate. The output amplitude will increase with the
amount of acceleration experienced by the sensor. Information
concerning the direction of beam motion is contained in the
phase of the signal with synchronous demodulation being used
to extract this information. Note that the sensor needs to be positioned so that the measured acceleration is along its sensitive
axis.
An uncommitted buffer amplifier provides the capability to adjust the scale factor and 0 g offset level over a wide range. An internal reference supplies the necessary regulated voltages for
powering the chip and +3.4 volts for external use.
Figure 18 shows a block diagram of the ADXL50. The voltage
output from the central plate of the sensor is buffered and then
applied to a synchronous demodulator. The demodulator is also
supplied with a (nominal) 1 MHz clock signal from the same
oscillator which drives the fixed plates of the sensor. The
–8–
REV. B
ADXL50
+3.4V
DENOTES EXTERNAL
PIN CONNECTION
+3.4V
+5V
33kΩ
75Ω
33kΩ
PREAMP
C2
C1
EXTERNAL
OSCILLATOR
DECOUPLING
CAPACITOR
0°
CS1
180°
CS2
1MHz
OSCILLATOR
EXTERNAL
DEMODULATION
CAPACITOR
+1.8V
+1.8V
BEAM
SYNCHRONOUS
DEMODULATOR
INTERNAL
FEEDBACK
LOOP
LOOP GAIN = 10
+5V
+5V
SYNC
3MΩ
+5V
COMMON
RST
VREF
OBS
+3.4V
+3.4V
+0.2V
INTERNAL
REFERENCE
+3.4V
BUFFER
AMPLIFIER
+1.8V
50kΩ
+1.8V +0.2V
VOUT
SELF–TEST
(ST)
OLE
COM
VIN–
Figure 18. Functional Block Diagram
the inverting input and the output of this amplifier via pins
VOUT and VIN–, while the noninverting input is connected internally to a +1.8 V reference. The +1.8 V is derived from a
resistor divider connected to the 3.4 V reference.
The sensor’s tight mechanical spacing allows it to be electrostatically deflected to full scale while operating on a 5 volt supply. A self-test is initiated by applying a TTL “high” level
voltage (>+2.0 V) to the ADXL50’s self-test pin which causes
the chip to apply a deflection voltage to the beam which moves
it an amount equal to –50 g (the negative full-scale output of the
device). Note that the ± 10% tolerance of the self-test circuit is
not proportional to the sensitivity error, see Self-Test section.
TE
BASIC CONNECTIONS FOR THE ADXL50
Figure 19 shows the basic connections needed for the ADXL50
to measure accelerations in the ± 50 g range with an output scale
factor 40 mV/g corresponding to a 2.5 V 0 g level, a ± 2.0 V fullscale swing around 0 g and a 3 dB bandwidth of approximately
1 kHz.
The output of the ADXL50’s preamplifier is 1.8 V at 0 g acceleration with an output range of ± 0.95 V for a ± 50 g input, i.e.,
19 mV/g. An uncommitted buffer amplifier has been included
on-chip to enhance the user’s ability to offset the 0 g signal level
and to amplify and filter the signal. Access is provided to both
In general, the designer will need to take into account the initial
zero g bias when designing circuits. For the ADXL50J this offset is 1.8 V ± 250 mV. When microprocessors and software
ADXL50
+3.4V
REFERENCE
4
OSCILLATOR
SENSOR
VREF
OUTPUT
DEMODULATOR
C2
0.022µF
SELF TEST
(ST)
6
+1.8V
OSCILLATOR
DECOUPLING
CAPACITOR
BUFFER
AMP
PREAMP
7
1
5
COM
2
C1
C3
+5V
3
8
C1
0.022µF
DEMODULATOR
CAPACITOR
10
VPR
9
VIN–
R3
105kΩ
VOUT
R1
49.9kΩ
R2
274kΩ
Figure 19. ADXL50 Application Providing an Output Sensitivity of
40 mV/g, a +2.5 V 0 g Level and a Bandwidth of 1 kHz
REV. B
VPR
C1
–9–
ADXL50
calibration are used and there is a desire to eliminate trim potentiometers, the design should leave room at either supply rail
to account for signal swing and or variations in initial zero g bias.
For example, in the circuit in Figure 19, the initial zero g bias of
± 250 mV will be reflected to the output by the gain of the R3/R1
network, resulting in an output offset of ± 526 mV worst case.
The offset, combined with a full-scale signal of 50 g, (+2.0 V)
will cause the output buffer amplifier to saturate at the supply
rail.
ranges listed by keeping R1 > 49.9 kΩ, with the subsequent
tradeoff that the required values for R3 will become very large.
The user always has the option of adding external gain and filtering stages after the ADXL50 to make lower full-scale ranges.
Measuring Full-Scale Accelerations Less than 65 g
Applications, such as motion detection, and tilt sensing, have
signal amplitudes in the 1 g to 2 g range. Although designed for
higher full-scale ranges, the ADXL50 may be adapted for use in
C2
The full ± 2.25 V output swing of the buffer amplifier can be
utilized if the user is able to trim the zero-g bias to exactly
2.5 V. In applications where the full-scale range will be ± 25 g or
less, a bias trim such as that shown in Figure 20 will almost always be required.
4
C1
OBS
+5V
0.1µF
PRE-AMP
1.8V
2
BUFFER
AMP
0.022µF
3
9
C1
VARYING THE OUTPUT SENSITIVITY AND 0 g LEVEL
USING THE INTERNAL BUFFER AMPLIFIER
5
OLE
Table II. Recommended Resistor Values for Setting the
Circuit of Figure 20 to Several Common Full-Scale Ranges
FS (g)
Buffer
Gain
SF in
mV/g
R1
R3
R2
± 50.0
± 40.0
± 30.8
± 26.7
± 20.0
± 10.0
2.11
2.63
3.42
3.95
5.26
10.53
40
50
65
75
100
200
49.9 k
39.2 k
40.2 k
28.7 k
26.1 k
23.7 k
105 k
103 k
137 k
113 k
137 k
249 k
100 k
100 k
100 k
100 k
100 k
100 k
VOUT
COM
The uncommitted buffer amplifier may be used to change the
output sensitivity to provide useful full-scale ranges of ± 50 g
and below. Table II provides recommended resistor values for
several standard ranges down to ± 10 g. As the full-scale range is
decreased, buffer amplifier gain is increased, and the noise contribution as a percentage of full scale will also increase. For all
ranges, the signal-to-noise ratio can be improved by reducing
the circuit bandwidth, either by increasing the demodulator capacitor, C1, or by adding a post filter using the buffer amplifier.
6
VREF
8
VPR
+3.4V
0g
LEVEL 50kΩ
TRIM
R1
10
VIN–
+1.8V
R3
R2
VX
TE
Figure 20. ADXL50 Circuit Using the Buffer Amplifier to
Set the Output Scaling and 0 g Offset Level
low g applications; the two main design considerations are noise
and 0 g offset drift (BH, KH grades recommended).
At its full 1 kHz bandwidth, the ADXL50 will typically exhibit
1 g p-p of noise. With ± 50 g accelerations this is generally not a
problem, but at a ± 2 g full-scale level the signal-to-noise ratio
will be very poor. However, reducing the bandwidth to 100 Hz
or less considerably improves the S/N ratio. Figure 25 shows the
relationship between ADXL50 bandwidth and noise.
The ADXL50 exhibits offset drifts that are typically 0.02 g per
°C but which may be as large as 0.1 g per °C. With the buffer
amplifier configured for a 2 g full scale, the ADXL50 will typically drift 1/2 of its full-scale range with a 50°C increase in
temperature.
Note that the value of resistor R1 should be selected to limit the
output current flowing into VPR to less than 25 µA (to provide a
safety margin). For a “J” grade device, this current is equal to:
I PR =
1
ADXL50
0.022µF
There are several cures for offset drift. If a dc response is not
required, for example in motion sensing or vibration measurement
applications, consider ac coupling the acceleration signal to remove the effects of offset drift. See the section on ac coupling.
(2.05 V – The peak full -scale output voltage at VPR ) – 1.8 V
R1 in ohms
For a ± 50 g full-scale range, R1 needs to be 49.9 kΩ or larger in
value; but at the lower full-scale g ranges, if the VPR swing is
much less, then it is possible to use much lower resistance values. For this table, the circuit of Figure 20 is used, as a 0 g offset trim will be required for most applications. In all cases, it is
assumed that the zero-g bias level is 2.5 V with an output span
of ± 2 V.
Note that for full scales below ± 20 g the self-test is unlikely to
operate correctly because the VPR pull-down current is not guaranteed to be large enough to drive R1 to the required –1.0 V
swing. In these cases, the self-test command will cause VOUT to
saturate at the rail, and it will be necessary to monitor the selftest at VPR. Self-test can remain operational at VPR for all g
Periodically recalibrating the accelerometer’s 0 g level is another
option. Autozero or long term averaging can be used to remove
long term drift using a microprocessor or the autozero circuit of
Figure 29. Be sure to keep the buffer amplifier’s full-scale output range much larger than the measurement range to allow for
the 0 g level drift.
CALCULATING COMPONENT VALUES FOR SCALE
FACTOR AND 0 g SIGNAL LEVEL
The ADXL50 buffer’s scale factor is set by –R3/R1 (since the
amplifier is in the inverter mode).
–10–
REV. B
ADXL50
As an example, if the desired span is ± 2.0 V for a = ± 50 g input,
then R3/R1 should be chosen such that
R3/R1 = VOUT Span/VPR Span = 2.00/0.95 = 2.105
(1)
where VPR span is the output from the preamplifier and VOUT
span is the buffer amplifier’s output, giving
R3 = 2.105 × R1
(2)
In noncritical applications, a resistor, R2, may simply be connected between VIN– and common to provide an approximate
0 g offset level (see Figure 19). In this simplified configuration
R2 is found using:
SELF-TEST FUNCTION
R2 = (1.8 V × R3)/(VOUT @ 0 g – 1.8 V)
When used with a trim potentiometer, as in Figure 20, resistor
R2 sets the 0 g offset range and also sets the resolution of the
offset trim. A value of 100 kΩ is typical. Increasing R2 above
this value makes trimming the offset easier, but may not provide
enough trim range to set VOUT equal to +2.5 V for all devices.
OBS
To provide an output span of ± 2.00 V, with a 0 g output of
+2.5 V, R1 could be set to the standard value of 49.9 kΩ and
from Equation 2, R3 = 105 kΩ.
For Figure 20, the circuit transfer function is:
The digital self-test input is compatible with both CMOS and
TTL signals. A Logic “l” applied to the self-test (ST) input will
cause an electrostatic force to be applied to the sensor which
will cause it to deflect to the approximate negative full-scale output of the device. Accordingly, a correctly functioning accelerometer will respond by initiating an approximate –1 volt output
change at VPR. If the ADXL50 is experiencing an acceleration
when the self-test is initiated, the VPR output will equal the algebraic sum of the two inputs. The output will stay at the self-test
level as long as the ST input remains high and will return to the
0 g level when the ST voltage is removed.
OLE
A self-test output that varies more than ± 10% from the nominal
–1.0 V change indicates a defective beam or a circuit problem
such as an open or shorted pin or component.
TE
 R3
  R3

V OUT = 
(1.8 V – VPR ) + 
(1.8 V – VX )  + 1.8 V
 R1
  R2

Operating the ADXL50’s buffer amplifier at Gains > 2, to provide full-scale outputs of less than ± 50 g, may cause the self-test
output to overdrive the buffer into saturation. The self-test may
still be used in the case, but the change in the output must then
be monitored at the VPR pin instead of the buffer output.
The summing amplifier configuration allows noninteractive
trimming of offset and span. Since VPR is not always exactly
1.8 V at 0 g, it will contribute to output offset. Therefore, span
must be trimmed first, followed by 0 g offset adjustment.
Note that the value of the self-test delta is not an exact indication of the sensitivity (mV/g) of the ADXL50 and, therefore,
may not be used to calibrate the device for sensitivity error.
LOAD DRIVE CAPABILITIES OF THE V PR AND BUFFER
OUTPUTS
In critical applications, it may be desirable to monitor shifts in
the zero-g bias voltage from its initial value. A shift in the 0 g
bias level may indicate that the 0 g level has shifted which may
warrant an alarm.
The VPR and the buffer amplifier outputs are both capable of
driving a load to voltage levels approaching that of the supply
rail. However, both outputs are limited in how much current
they can supply, affecting component selection.
POWER SUPPLY DECOUPLING
VPR Output
The VPR pin has the ability to source current up to 500 µA but
only has a sinking capability of 30 µA which limits its ability to
drive loads. It is recommended that the buffer amplifier be used
in most applications, to avoid loading down VPR. In standard
± 50 g applications, the resistor R1 from VPR to VIN– is recommended to have a value greater than 50 kΩ to reduce loading
effects.
Capacitive loading of the VPR pin should be minimized. A load
capacitance between the VPR pin and common will introduce an
offset of approximately 1 mV for every 10 pF of load. The VPR
pin may be used to directly drive an A/D input or other source
as long as these sensitivities are taken into account. It is always
preferable to drive A/D converters or other sources using the
buffer amplifier (or an external op amp) instead of the VPR pin.
Buffer Amplifier Output
The buffer output can drive a load to within 0.25 V of either
power supply rail and is capable of driving 1000 pF capacitive
REV. B
loads. Note that a capacitance connected across the buffer feedback resistor for low-pass filtering does not appear as a capacitive load to the buffer. The buffer amplifier is limited to
sourcing or sinking a maximum of 100 µA. Component values
for the resistor network should be selected to ensure that the
buffer amplifier can drive the filter under worst case transient
conditions.
The ADXL50 power supply should be decoupled with a 0.1 µF
ceramic capacitor from +5 V pin of the ADXL50 to common
using very short component leads. For other decoupling considerations, see EMI/RFI section.
OSCILLATOR DECOUPLING CAPACITOR, C2
An oscillator decoupling capacitor, C2, is used to remove
1 MHz switching transients in the sensor excitation signal, and
is required for proper operation of the ADXL50. A ceramic capacitor with a minimum value of 0.022 µF is recommended
from the oscillator decoupling capacitor pin to common. Small
amounts of capacitor leakage due to a dc resistance greater than
l MΩ will not affect operation (i.e., a high quality capacitor is
not needed here). As with the power supply bypass capacitor,
very short component leads are recommended. Although
0.022 µF is a good typical value, it may be increased for reasons
of convenience, but doing this will not improve the noise performance of the ADXL50.
–11–
ADXL50
DEMODULATOR CAPACITOR, C1
For example, to reduce the average power to 5 mW from its
typical 50 mW, the power should be on 10% of the time. With
the power on for 1 ms and off for 9 ms, a maximum sample rate
of 100 Hz is achievable. Further reduction in average power can
be realized with lower sample rates.
The demodulator capacitor is connected across Pins 2 and 3 to
filter the demodulated signal from the sensor beam and to set
the bandwidth of the force balance control loop. This capacitor
may be used to approximately set the bandwidth of the accelerometer. A capacitor is always required for proper operation.
The frequency response of the ADXL50 exhibits a single pole
roll-off response whose nominal 3 dB frequency is set by the
following equation:
+5V
POWER
SUPPLY
(V)
0V
f3 dB = (28.60/C1 in µF) ± 40%
VFINAL
A nominal value of 0.022 µF is recommended for C1. In general, the design bandwidth should be set 40% higher than the
minimum desired system bandwidth due to the ± 40% tolerance.
OBS
A minimum value of 0.015 µF is required, (over temperature
and system life), to prevent device instability or oscillation. The
demodulation capacitor should be a low leakage, low drift ceramic type with an NPO (best) or X7R (good) dielectric.
VOUT
(V)
OLE
0.4
0.6
TIME – ms
0.8
1.0
Figure 21. Power-On Settling Time when Power Cycling
TE
SYSTEM BANDWIDTH CONTROL AND POST
FILTERING
Unlike piezoresistive sensors, the resonant frequency of the
ADXL50’s capacitive sensor element is typically greater than
20 kHz and does not limit the useful bandwidth of the device.
Usually, the resonant frequency of the beam appears as a peak
in the bandwidth response at approximately 24 kHz with a Q of
3 to 4, as shown in Figure 22.
When using the recommended 0.022 µF demodulator capacitor,
be advised that the nominal 1300 Hz pole it establishes within
the device can vary ± 40%. Therefore, if additional low-pass
filtering is used—at frequencies much above 600 Hz—the two
poles may interact and result in a net circuit bandwidth that is
lower than expected.
190
REDUCING THE AVERAGE POWER CONSUMPTION OF
THE ADXL50
AMPLITUDE
19
0.022µF
1.9
0.015µF
0.010µF
0.005µF
0.19
The ADXL50 is a versatile accelerometer that can be used in a
wide variety of applications. In some battery powered applications, such as shipping recorders, power consumption is a critical parameter. The ADXL50 typically draws 10 mA current
from a 5 V power supply which may exceed the power budgeted
for the accelerometer.
For such applications, the ADXL50 can be successfully power
cycled, where the power is turned on only during the period
when data is sampled. Figure 21 illustrates the power-on settling
of the ADXL50 during cycling where the output amplifier has
a gain of one with no filtering. The settling time-constant is
approximately 0.12 ms, waiting l ms before sampling ensures
maximally accurate readings.
0.2
0
–90
PHASE
–180
100
1k
FREQUENCY – Hz
PHASE SHIFT – Degrees
Care should be taken to reduce or eliminate any leakage paths
from the demodulator capacitor pins to common or to the +5 V
pin. Even a small imbalance in the leakage paths from these pins
will result in offset shifts in the zero-g bias level. As an example,
an unbalanced parasitic resistance of 30 MΩ from either
demodulator pin to ground will result in an offset shift at VPR of
approximately 50 mV. Conformal coating of PC boards with a
high impedance material is recommended to avoid leakage problems due to aging or moisture.
T0
SENSITIVITY – mV/g
In general, it’s best to use the recommended 0.022 µF capacitor
across the demodulator pins and perform any additional lowpass filtering using the buffer amplifier. Using a large denominator capacitor for low-pass filtering has the disadvantage that the
capacitive sensor will be slow to respond to rapid changes in
acceleration and, therefore, the full shock survivability of the
device could be compromised. The use of the buffer for lowpass filtering generally results in smaller capacitance values and
better overall performance. It is also a convenient and more precise way to set the system bandwidth. Post filtering allows bandwidth to be controlled accurately by component selection and
avoids the ± 40% demodulation tolerance. Note that signal noise
is proportional to the square root of the bandwidth of the
ADXL50 and may be a consideration in component selection—
see section on noise.
10k
Figure 22. Frequency Response of the ADXL50 for Various
Demodulator Capacitors
–12–
REV. B
ADXL50
C2
4
RECOMMENDED COMPONENT VALUES FOR VARIOUS
FULL SCALE RANGES AND A 300Hz BANDWIDTH
ADXL50
0.022µF
C1
FULL
SCALE
mV
per g
3dB
BW (Hz)
R1a
kΩ
R1b
kΩ
R3
kΩ
R2
kΩ
C4
µF
±10 g
200
300
5
21.5
249
100
0.0022
±20 g
100
300
5
23.7
137
100
0.0039
±40 g
50
300
10
34
105
100
0.0056
±50 g
40
300
10
45.3
105
100
0.0056
1
0.1µF
BUFFER
AMP
PRE-AMP
2
+5V
1.8V
0.022µF
9
VOUT
3
C1
5
COM
6
8
1
3dB BW =
2π R3 C4
VIN–
C4
R1a R1b
OPTIONAL SCALE
FACTOR TRIM*
0g
LEVEL
TRIM
OBS
10
VPR
+3.4V
REF
R3
50kΩ
R2
*TO OMIT THE OPTIONAL SCALE FACTOR TRIM,
REPLACE R1a AND R1b WITH A FIXED VALUE 1%
METAL FILM RESISTOR. SEE VALUES SPECIFIED
IN TABLE II.
OLE
Figure 23. Using the Buffer Amplifier to Provide One Pole Post Filtering Plus Scale Factor and 0 g Level Trimming
ONE POLE POST FILTERING
need to be twice as large as its 100 Hz value or 0.012 µF × 2 =
0.024 µF. The closest standard value of 0.022 µF should then
be used.
Figure 23 shows the ADXL50 buffer amplifier connected to
provide one pole post filtering, 0 g offset trimming, and output
scaling. The table included with the figure lists practical component values for various full-scale g levels and approximate circuit
bandwidths. For bandwidths other than those listed, use the
formula:
TE
TWO POLE POST FILTERING
Figure 24 shows a circuit which uses the ADXL50’s buffer amplifier to provide two pole post filtering. An AD820 external op
amp allows noninteractive adjustment of 0 g offset and scale
factor. Component values for the two pole filter were selected to
operate the buffer at unity gain with a Q of one.
Capacitor C4 in Farads = 1/(2 π × R3 in Ohms
× 3 dB BW in Hertz)
or simply scale the value of capacitor C4 accordingly; i.e., for a
± 20 g application with a 50 Hz bandwidth, the value of C4 will
0.022µF
ADXL50
4
C2
C1
0.022µF
2 POLE FILTER
COMPONENT VALUES
3dB
BW(Hz) C3µF
0.1µF
BUFFER
AMP
PRE-AMP
2
+5V
1
1.8V
OPTIONAL CAPACITOR
FOR 3 POLE FILTERING
9
VOUT
3
R5
C1
C4µF
5
COM
+5V
8
6
10
300
0.027
0.0033
100
0.082
0.01
30
0.27
0.033
R1
10
0.82
0.1
82.5kΩ
VREF
C4
VIN–
VPR
42.2kΩ
R5
C3
0.01µF
R4a
R3
82.5kΩ
SCALE
FACTOR
TRIM
2 POLE FILTER
+3.4V
OFFSET & SCALING AMPLIFIER
COMPONENT VALUES
FULL
SCALE
mV
per g
GAIN
R4a
kΩ
R4b
kΩ
R5
kΩ
40.2kΩ
±10 g
200
10.53
5
21.5
249
20kΩ
±20 g
100
5.26
5
23.7
137
±40 g
50
2.63
10
34
105
±50 g
40
2.11
10
45.3
105
71.5kΩ
R4b
2
7
3 AD820
4
6
OFFSET AND
SCALING
AMPLIFIER
R6
R7
0g
LEVEL
TRIM
Figure 24. Circuit Providing Two Pole Post Filtering and 0 g Offset and Scale Factor Trimming
REV. B
–13–
OUTPUT
ADXL50
Capacitors C3 and C4 are chosen to provide the desired 3 dB
bandwidth. Component values are specified for bandwidths of
10 Hz, 30 Hz, 100 Hz, and 300 Hz. For other 3 dB bandwidths
simply scale the capacitor values; i.e., for a 3 dB bandwidth of
20 Hz, divide the 10 Hz bandwidth numbers by 2.0. The nominal buffer amplifier output will be +1.8 V ± 19 mV/g. Note that
the ADXL50’s self-test will be fully functional since the buffer
amplifier is operated at unity gain and resistor R1 is large. The
external op amp offsets and scales the output to provide a +2.5 V
± 2 V output over a wide range of full-scale g levels. The external op amp may be omitted in high g, low gain applications.
NOISE CONSIDERATIONS
The output noise of the ADXL50 scales with the square root of
its bandwidth. The noise floor may be reduced by lowering the
bandwidth of the ADXL50 either by increasing the value of the
demodulator capacitor or by adding an external filter.
OBS
Nominal Peak-toPeak Value
% of Time that Noise will Exceed
Nominal Peak-to-Peak Value
2.0 × rms
3.0 × rms
4.0 × rms
5.0 × rms
6.0 × rms
6.6 × rms
7.0 × rms
8.0 × rms
32%
13%
4.6%
1.2%
0.27%
0.1%
0.046%
0.006%
AC COUPLING VPR TO BUFFER INPUT
If a dc response is not required, as in applications such as motion detection or vibration measurement, then ac coupling
should be considered. In low g applications, the output voltage
change due to acceleration is small compared to the 0 g offset
voltage drift. Because ac coupling removes the dc component of
the output, the preamp output signal may be amplified considerably without increasing the 0 g level drift. The most effective
way to ac couple the ADXL50 is between the preamp output at
VPR and the buffer input, VIN–, as shown in Figure 26.
The typical rms noise of the ADXL50J with a bandwidth of
100 Hz and a noise density of 125 µV/√Hz is estimated as
follows:
OLE
Noise (rms) = (125 µV/√Hz) √100 = 1.25 mV rms
Peak-to-peak noise may be estimated with the following
equation:
Noise p-p = (6.6) Noise rms
Table III.
1.7
0.26
1.4
0.21
1.0
0.16
0.73
0.11
0.33
ADXL50
PRE-AMP
VPR
100
3dB BANDWIDTH – Hz
VOUT
10
C4
VIN–
R1
R3
R2
Figure 26. AC Coupling the VPR Output to the Buffer Input
Using this configuration, the system’s ac response is now rolled
off—at the low frequency end at FL, and at the high frequency
end at FH. The normalized frequency response of the system
can be seen in Figure 27.
0
10
BUFFER
AMP
8
VPR
0.05
0
TE
1.8V
9
NOISE LEVEL – g rms
NOISE LEVEL – g p-p
Peak-to-peak noise is thus estimated at 8.25 mV or approximately 0.4 g p-p. The ADXL50 noise is characteristic of white
noise. Typical rms and p-p noise for various 3 dB bandwidths is
estimated in Figure 25.
1k
The low frequency roll-off, FL, due to the ac coupling network
is:
FL = 1/(2 π R1 C4)
Figure 25. ADXL50 Noise Level and Resolution vs. –3 dB
Bandwidth
Because the ADXL50’s noise is for all practical purposes Gaussian in amplitude distribution, the highest noise amplitudes have
the smallest (yet nonzero) probability. Peak-to-peak noise is,
therefore, difficult to measure and can only be estimated due to
its statistical nature. Table III is useful for estimating the probabilities of exceeding various peak values, given the rms value.
The high frequency roll-off FH is determined by the dominant
pole of the system which is controlled by either the demodulator
capacitor and its associated time-constant or by a dominant post
filter.
As a consequence of ac coupling, any constant acceleration
component gravity will not be detected (because this too is a dc
voltage present at the VPR output). The self-test feature, if used,
must be monitored at VPR, rather than at the buffer output.
–14–
REV. B
ADXL50
and package orientation affect the ADXL50’s output (TO-100
package shown). Note that the output polarity is that which appears at VPR; the output at VOUT will have the opposite sign.
With its axis of sensitivity in the vertical plane, the ADXL50
should register a 1 g acceleration, either positive or negative, depending on orientation. With the axis of sensitivity in the horizontal plane, no acceleration (0 g) should be indicated.
NORMALIZED OUTPUT LEVEL – dB
+20
+10
LOW FREQUENCY ROLL-OFF
(F L ) DUE TO AC COUPLING
0
–10
HIGH FREQUENCY ROLL-OFF (FH )
DUE TO DEMODULATOR BANDWIDTH
–20
0g
(a)
0g
(b)
–1g
(c)
+1g
(d)
–30
OBS
1
10
100
FREQUENCY – Hz
1k
10k
INDICATED POLARITY IS THAT
OCCURING AT V PR .
Figure 27. Normalized Output Level vs. Frequency for a
Typical Application Using AC Coupling Between VPR and
Buffer Amplifier
MINIMIZING EMI/RFI
OLE
The architecture of the ADXL50 and its use of synchronous demodulation make the device immune to most electromagnetic
(EMI) and radio frequency (RFI) interference. The use of synchronous demodulation allows the circuit to reject all signals except those at the frequency of the oscillator driving the sensor
element. However, the ADXL50 does have a sensitivity to RFI
that is within ± 5 kHz of the internal oscillator’s nominal frequency of 1 MHz. The internal oscillator frequency will exhibit
part to part variation in the range of 0.6 MHz to 1.4 MHz.
In general the effect is difficult to notice as the interference
must match the internal oscillator within ± 5 kHz and must be
large in amplitude. For example: a 1 MHz interference signal of
20 mV p-p applied to the +5 V power supply pin will produce a
200 mV p-p signal at the VPR pin if the internal oscillator and
interference signals are matched exactly. If the same 20 mV interference is applied but 5 kHz above or below the internal
oscillator’s frequency, the signal level at VPR will only be 20 mV
p-p in amplitude.
Power supply decoupling, short component leads (especially for
capacitors C1 and C2), physically small (surface mount, etc.)
components and attention to good grounding practices all help
to prevent RFI and EMI problems. Please consult the factory
for applications assistance in instances where this may be of
concern.
SELF-CALIBRATING THE ADXL50
If a calibrated shaker is not available, both the 0 g level and
scale factor of the ADXL50 may be easily set to fair accuracy by
using a self-calibration technique based on the 1 g (average) acceleration of the earth’s gravity. Figure 28 shows how gravity
REV. B
Figure 28. Using the Earth’s Gravity to Self-Calibrate the
ADXL50
To self-calibrate the ADXL50, place the accelerometer on its
side with its axis of sensitivity oriented as shown in “a.” The 0 g
offset potentiometer, Rt, is then roughly adjusted for midscale:
+2.5 V at the buffer output. If the optional scale factor trimming is to be used, it should be adjusted next.
TE
Next, the package axis should be oriented as in “c” (pointing
down) and the output reading noted. The package axis should
then be rotated 180° to position “d” and the scale factor potentiometer, R1a, adjusted so that the output voltage indicates a
change of 2 g’s in acceleration. For example, if the circuit scale
factor at the buffer output is 100 mV per g, then the scale factor
trim should be adjusted so that an output change of 200 mV is
indicated.
Adjusting the circuit’s scale factor will have some effect on its 0
g level so this should be readjusted, as before, but this time
checked in both positions “a” and “b.” If there is a difference in
the 0 g reading, a compromise setting should be selected so that
the reading in each direction is equidistant from +2.5 V. Scale
factor and 0 g offset adjustments should be repeated until both
are correct. Temporarily placing a capacitor across the buffer
amplifier’s feedback resistor will reduce output noise and so aid
in trimming the device. Note that, for high full-scale g ranges,
± 2 g may be a very small fraction of the full-scale range and
device nonlinearity will, therefore, affect the circuit’s high g level
accuracy.
–15–
ADXL50
Compensating for the 0 g Drift of the ADXL50 Accelerometer
+15
The circuit of Figure 29 provides a linear temperature compensation for the ADXL50. Figure 30 shows the 0 g drift over temperature for a typical ADXL50 with and without this circuit. As
shown by Figure 30, the linear portion of the drift curve has
been subtracted out. In effect, the curve has been rotated counterclockwise until it is horizontal, leaving just the bow of the
curve: that portion which is not linear. As shown by Figure 30,
over a +25°C to +70°C range, a 10× reduction in drift is achieved.
+10
OBS
0.022µF
ADXL50
4
BUFFER
AMP
2
PRE-AMP
0.022µF
6
49.9kΩ
VIN–
–40
–45
–55
–35
3101kΩ
R5
TC
COMP
SET
RB
3
30k
R6
30k
R8
+45
+65
TE
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
+5V
0.1µF
C3
REFERENCE PLANE
0.185 (4.70)
0.165 (4.19)
TEMPERATURE
COMPENSATED
ACCELERATION
OUTPUT
0.750 (19.05)
0.500 (12.70)
0.160 (4.06)
0.110 (2.79)
7
0.370 (9.40)
0.335 (8.51)
20kΩ
8
0.115
(2.92)
BSC
4
0.045 (1.14)
0.027 (0.69)
9
3
2
0.040 (1.02) MAX
0.045 (1.14)
0.010 (0.25)
0.230 (5.84)
BSC
10
1
0.034 (0.86)
0.027 (0.69)
SEATING PLANE
0g OUTPUT
LEVEL
R9
6
TEMPCO
AMPLIFIER
+105
Contact the Analog Devices Literature Center for a copy of
publication number G2112, the Accelerometer Application
Guide. This includes all current application notes and data
sheets for Analog Devices’ accelerometers.
499kΩ
7
+85
ADXL50 Application Literature Available
6
TEST
0.01µF POINT
"A"
4
BRIDGE
BALANCE
RA
+25
Figure 30. ADXL50 0 g Drift With and Without the Compensation Circuit of Figure 29
49.9kΩ
R2
2 AD820
+5
5
R3
310kΩ
R7
+5V
–15
TEMPERATURE – °C
0.335 (8.51)
0.305 (7.75)
1µA/°K
10kΩ
1kΩ
–35
10
8
VPR
R1
R4
UNCOMPENSATED XL50
–30
5
+5V
500Ω
–25
VOUT
+3.4V
REF
AD590
–20
9
3
C1
COM
COMPENSATED XL50
–15
R10
RC
20kΩ
25kΩ
PRINTED IN U.S.A.
C1
–5
–10
OLE
1
1.8V
C2
LOW TEMP TRIM
0
C1808b–5–3/96
MEASURED 0g DRIFT – mV
The circuit of Figure 29 is essentially a temperature sensor
coupled to a Whetstone bridge. The AD590 provides a 1 µA/°K
current output whose voltage scale factor is set by resistor RA.
The bridge circuit subtracts out the nominal 298 mV output of
the AD590 at +25°C and leaves only the change in temperature,
which is what is needed. Without the bridge, the 298 mV room
temperature “offset” would “swamp” the much smaller change
in output with temperature.
Resistors R5 and R6 form a resistor divider (one half of the
bridge) which divides down the +3.4 V reference output of the
ADXL50 to 0.3 V which appears at the noninverting input of
the AD820 op amp. Resistors R7 and R8 form the other half of
the bridge, and because they have the same ratio as R5 and R6,
the op amp will have a +3.4 V output at room temperature.
HIGH TEMP TRIM
+5
CALIBRATION PROCEEDURE:
AT T MIN OR LOWER TEMP CAL POINT...
1.
2.
3.
4.
5.
6.
7.
8.
9.
SET RB ALL THE WAY TO ONE SIDE.
ADJUST RA FOR +3.4V AT TEST POINT "A."
SET RC FOR +2.5V V OUT (AT PIN 9 OF ADXL50).
TEMPORARILY CONNECT A 1.5kΩ RESISTOR BETWEEN
THE CENTER OF RB AND GROUND.
ADJUST RB FOR +2.5V AT VOUT .
REMOVE THE 1.5kΩ RESISTOR. VOUT SHOULD NOT CHANGE.
GO TO T MAX OR HIGH TEMP CAL POINT.
READJUST RB FOR +2.5V.
CALIBRATION COMPLETE.
Figure 29. ADXL50 0 g Drift Compensation Circuit
–16–
REV. B