MPS MP3213 700khz/1.3mhz boost converter with a 3.5a switch Datasheet

MP3213
700KHz/1.3MHz Boost Converter
with a 3.5A Switch
The Future of Analog IC Technology
DESCRIPTION
FEATURES
The MP3213 is a current mode step-up
converter with a 3.5A, 0.18Ω internal switch to
provide a highly efficient regulator with fast
response. The MP3213 operates at 700KHz or
1.3MHz allowing for easy filtering and low
noise. An external compensation pin gives the
user flexibility in setting loop dynamics, which
allows the use of small, low-ESR ceramic
output capacitors. Soft-start results in small
inrush current and can be programmed with an
external capacitor. The MP3213 operates from
an input voltage as low as 2.5V and can
generate 12V at up to 500mA from a 5V supply.
•
•
•
The MP3213 includes under-voltage lockout,
current limiting and thermal overload protection
to prevent damage in the event of an output
overload. The MP3213 is available in a low
profile 8-pin MSOP package with exposed pad.
EVALUATION BOARD REFERENCE
Board Number
Dimensions
EV3213DH-00A
2.1”X x 2.1”Y x 0.5”Z
•
•
•
•
•
•
3.5A, 0.18Ω, 25V Power MOSFET
Uses Tiny Capacitors and Inductors
Pin Selectable 700KHz or 1.3MHz Fixed
Switching Frequency
Programmable Soft-Start
Operates with Input Voltage as Low as 2.5V
and Output Voltage as High as 22V
12V at 500mA from 5V Input
UVLO, Thermal Shutdown
Internal Current Limit
Available in an 8-Pin MSOP Package with
Exposed Pad
APPLICATIONS
•
•
•
•
LCD Displays
Portable Applications
Handheld Computers and PDAs
Digital Still and Video Cameras
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
TYPICAL APPLICATION
D1
VIN
5V
Efficiency vs
Load Current
VOUT
12V
100
95
7
OFF ON
3
8
5
IN
FSEL
EN
SS
SW
MP3213
GND
4
C4
10nF
MP3213 Rev. 1.1
5/12/2006
FB
2
COMP
1
C3
2.2nF
EFFICIENCY (%)
90
6
85
80
75
70
65
VIN = 5V
VOUT = 12V
60
55
50
1
10
100
LOAD CURRENT (mA)
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1
MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
ABSOLUTE MAXIMUM RATINGS (1)
PACKAGE REFERENCE
TOP VIEW
COMP
1
8
SS
FB
2
7
FSEL
EN
3
6
IN
GND
4
5
SW
Part Number*
Package
Temperature
MP3213DH
MSOP8
(Exposed Pad)
–40°C to +85°C
*
For Tape & Reel, add suffix –Z (eg. MP3213DH–Z)
For RoHS Compliant Packaging, add suffix –LF (eg.
MP3213DH–LF–Z)
SW ............................................... –0.5V to +25V
IN ............................................... –0.5V to +25V
All Other Pins.............................. –0.3V to +6.5V
Junction Temperature...............................150°C
Lead Temperature ....................................260°C
Storage Temperature ..............–65°C to +150°C
Recommended Operating Conditions
(2)
Supply Voltage VIN ........................... 2.5V to 22V
Output Voltage VOUT ........................... 3V to 22V
Operating Temperature .............–40°C to +85°C
Thermal Resistance
θJA
θJC
MSOP8 ................................... 80 ...... 12... °C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The device is not guaranteed to function outside of its
operating conditions.
ELECTRICAL CHARACTERISTICS
VIN = VEN = 5V, TA = +25°C, unless otherwise noted.
Parameter
Operating Input Voltage
Undervoltage Lockout
Undervoltage Lockout
Hysteresis
Supply Current (Shutdown)
Supply Current (Quiescent)
Switching Frequency
Symbol Condition
VIN
VIN Rising
Typ
Max
22
2.45
100
fSW
FSEL High Threshold
FSEL Low Threshold
VEN = 0V
VFB = 1.35V
VFSEL = VIN
VFSEL = GND
VFSEL Rising
VFB = 0V, VFSEL = VIN
VFB = 0V, VFSEL = GND
VEN Rising
Maximum Duty Cycle
EN High Threshold
EN Low Threshold
EN Input Bias Current
Soft-Start Current
FB Voltage
FB Input Bias Current
Error Amp Voltage Gain
Error Amp
Transconductance
Error Amp Output Current
SW On-Resistance (3)
SW Current Limit (3)
SW Current Limit (3)
SW Leakage
Thermal Shutdown (3)
Min
2.5
2.15
1.1
560
0.1
700
1.3
700
0.5
85
92
90
95
mV
1
900
1.5
840
1.5
µA
µA
MHz
KHz
V
V
%
1.5
V
V
µA
µA
V
nA
V/V
0.5
VEN = 0V, 5V
Units
V
V
1
AVEA
6
1.25
–100
1000
GEA
350
µmho
35
0.18
3.5
2.7
µA
Ω
A
A
µA
°C
1.225
–200
RON
Duty Cycle = 0%
Duty Cycle = 50%
VSW = 20V
1.275
1
160
Note:
3) Guaranteed by design.
MP3213 Rev. 1.1
5/12/2006
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MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
TYPICAL PERFORMANCE CHARACTERISTICS
100
100
90
95
95
85
90
90
80
75
70
65
VIN = 3.3V
VOUT = 12V
55
50
1
10
100
LOAD CURRENT (mA)
85
80
75
70
65
60
VIN = 3.3V
VOUT = 8V
55
50
1000
1
100
95
50
1000
1.258
80
660
75
650
70
65
640
VIN = 5V
VOUT = 12V
10
100
LOAD CURRENT (mA)
630
620
-45 -25 0 25 45 65 85 105125145
TEMPERATURE (°C)
1000
1.254
1.253
1.252
1.251
1.250
-45 -25 0 25 45 65 85 105125145
TEMPERATURE (°C)
Current Limit vs
Duty Cycle
680
1.26
3.0
FREQUENCY (MHz)
3.5
650
640
1.22
1.20
1.18
630
1.16
620
-45 -25 0 25 45 65 85 105125145
TEMPERATURE (°C)
1.14
-45 -25 0 25 45 65 85 105125145
TEMPERATURE (°C)
MP3213 Rev. 1.1
5/12/2006
1000
1.255
1.28
660
10
100
LOAD CURRENT (mA)
1.256
Frequency (1.3MHz) vs
Temperature
1.24
1
1.257
690
670
VIN = 5V
VOUT = 18V
Feedback Voltage vs
Temperature
680
Frequency (700KHz) vs
Temperature
FREQUENCY (KHz)
65
1.259
670
1
70
690
85
55
75
55
CURRENT LIMIT (A)
EFFICIENCY (%)
90
60
80
Quiescent Current vs
Temperature
Efficiency vs
Load Current
50
10
100
LOAD CURRENT (mA)
85
60
FEEDBACK VOLTAGE (V)
60
EFFICIENCY (%)
95
EFFICIENCY (%)
EFFICIENCY (%)
Circuit on front page, VIN = 5V, VOUT = 12V, TA = +25°C, C2 = 4.7µF, C4 = 10nF, unless otherwise
noted.
Efficiency vs
Efficiency vs
Efficiency vs
Load Current
Load Current
Load Current
2.5
2.0
1.5
1.0
0.5
0
0 10 20 30 40 50 60 70 80 90
DUTY CYCLE (%)
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MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Circuit on front page, VIN = 5V, VOUT = 12V, TA = +25°C, C2 = 4.7µF, C4 = 10nF, unless otherwise
noted.
VSW
5V/div.
VSW
5V/div.
IINDUCTOR
0.5A/div.
IINDUCTOR
0.5A/div.
400ns/div.
VOUT
AC Coupled
0.2V/div.
IOUT
0.2A/div.
VOUT
AC Coupled
0.2V/div.
IOUT
0.2A/div.
VEN
2V/div.
VEN
2V/div.
VOUT
5V/div.
VOUT
5V/div.
IINDUCTOR
0.5A/div.
IINDUCTOR
0.5A/div.
MP3213 Rev. 1.1
5/12/2006
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MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
PIN FUNCTIONS
Pin #
1
2
3
4
5
6
7
8
Name Description
COMP Compensation Pin. Connect a capacitor and resistor in series to ground for loop stability.
FB
Feedback Input. Reference voltage is 1.25V. Connect a resistor divider to this pin.
Regulator On/Off Control Input. A high input at EN turns on the converter, and a low input turns
EN
it off. When not used, connect EN to the input source (through a 100kΩ pull-up resistor if VIN >
6V) for automatic startup. EN cannot be left floating.
GND Ground. The exposed pad is connected to GND.
Power Switch Output. SW is the drain of the internal MOSFET switch. Connect the power
SW
inductor and output rectifier to SW. SW can swing between GND and 25V.
IN
Input Supply Pin. IN must be locally bypassed.
Frequency Select Pin. Tie to IN (through a 100kΩ resistor if VIN > 6V) for 1.3MHz operation or to
FSEL
GND for 700KHz operation.
Soft-Start Control Pin. Connect a soft-start capacitor to this pin. The soft-start capacitor is
SS
charged with a constant current of 6µA. Leave SS disconnected if the soft-start is not used.
OPERATION
The MP3213 uses a constant frequency, peak
current mode boost regulation architecture to
regulate the feedback voltage.
The operation of the MP3213 can be
understood by referring to the block diagram of
Figure 1.
IN
EN
FSEL
INTERNAL REGULATOR
AND ENABLE CIRCUITRY
OSCILLATOR
SW
+
--
PWM
CONTROL
LOGIC
CURRENT
SENSE
AMP
+
---
GND
FB
GM
SS
+
1.25V
COMP
Figure 1—Functional Block Diagram
MP3213 Rev. 1.1
5/12/2006
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MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
At the beginning of each cycle, the N-Channel
MOSFET switch is turned on, forcing the
inductor current to rise. The current at the
source of the switch is internally measured and
converted to a voltage by the current sense
amplifier. That voltage is compared to the error
voltage at COMP. The voltage at the output of
the error amplifier is an amplified version of the
difference between the 1.25V reference voltage
and the feedback voltage. When these two
voltages are equal, the PWM comparator turns
off the switch forcing the inductor current to the
output capacitor through the external rectifier.
This causes the inductor current to decrease.
The peak inductor current is controlled by the
voltage at COMP, which in turn is controlled by
the output voltage. Thus the output voltage is
regulated by the inductor current to satisfy the
load. The use of current mode regulation
improves transient response and control loop
stability.
APPLICATION INFORMATION
Components referenced below apply to Typical Application Circuit on page 1.
Selecting the Soft-Start Capacitor
The MP3213 includes a soft-start timer that
limits the voltage at COMP during startup to
prevent excessive current at the input. This
prevents fault tripping of the input voltage at
startup due to input current overshoot. When
power is applied to the MP3213, and enable is
asserted, a 6µA internal current source charges
the external capacitor at SS. As the SS
capacitor is charged, the voltage at SS rises.
The MP3213 internally clamps the voltage at
COMP to 700mV above the voltage at SS. The
soft-start ends when the voltage at SS reaches
0.45V. This limits the inductor current at startup,
forcing the input current to rise slowly to the
current required to regulate the output voltage.
The soft-start period is determined by the
equation:
t SS = 75 × C SS
Where CSS (in nF) is the soft-start capacitor
from SS to GND, and tSS (in µs) is the soft-start
period.
Determine the capacitor required for a given
soft-start period by the equation:
C SS = 0.0133 × t SS
Setting the Output Voltage
Set the output voltage by selecting the resistive
voltage divider ratio. Use 10kΩ for the low-side
resistor R2 of the voltage divider. Determine the
high-side resistor R1 by the equation:
R1 =
MP3213 Rev. 1.1
5/12/2006
where VOUT is the output voltage.
For R2 = 10kΩ and VFB = 1.25V, then
R1 (kΩ) = 8kΩ (VOUT – 1.25V).
Selecting the Input Capacitor
An input capacitor (C1) is required to supply the
AC ripple current to the inductor, while limiting
noise at the input source. A low ESR capacitor
is required to keep the noise at the IC to a
minimum. Ceramic capacitors are preferred, but
tantalum or low-ESR electrolytic capacitors may
also suffice.
Use an input capacitor value greater than
4.7µF. The capacitor can be electrolytic,
tantalum or ceramic. However since it absorbs
the input switching current it requires an
adequate ripple current rating. Use a capacitor
with RMS current rating greater than the
inductor ripple current (see Selecting The
Inductor to determine the inductor ripple
current).
To ensure stable operation, place the input
capacitor as close to the IC as possible.
Alternately a smaller high quality ceramic 0.1µF
capacitor may be placed closer to the IC with
the larger capacitor placed further away. If
using this technique, the larger capacitor can be
a tantalum or electrolytic type. All ceramic
capacitors should be placed close to the
MP3213.
R2( VOUT − VFB )
VFB
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MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
Selecting the Output Capacitor
The output capacitor is required to maintain the
DC output voltage. Low ESR capacitors are
preferred to keep the output voltage ripple to a
minimum. The characteristic of the output
capacitor also affects the stability of the
regulation control system. Ceramic, tantalum, or
low
ESR
electrolytic
capacitors
are
recommended. In the case of ceramic
capacitors, the impedance of the capacitor at
the switching frequency is dominated by the
capacitance, and so the output voltage ripple is
mostly independent of the ESR. The output
voltage ripple is estimated to be:
VRIPPLE
⎛
V ⎞
⎜1 - IN ⎟ × ILOAD
⎜ V
⎟
OUT ⎠
⎝
≈
C2 × f SW
Where VRIPPLE is the output ripple voltage, VIN
and VOUT are the DC input and output voltages
respectively, ILOAD is the load current, fSW is the
switching frequency, and C2 is the capacitance
of the output capacitor.
In the case of tantalum or low-ESR electrolytic
capacitors, the ESR dominates the impedance
at the switching frequency, and so the output
ripple is calculated as:
(1 −
VRIPPLE ≈
VIN
) × ILOAD
VOUT
I
× R ESR × VOUT
+ LOAD
C2 × f SW
VIN
Where RESR is the equivalent series resistance
of the output capacitors.
Choose an output capacitor to satisfy the output
ripple and load transient requirements of the
design. A 4.7µF-22µF ceramic capacitor is
suitable for most applications.
A 4.7µH inductor is recommended for most
1.3MHz applications and a 10µH inductor is
recommended for most 700KHz applications.
However, a more exact inductance value can
be calculated. A good rule of thumb is to allow
the peak-to-peak ripple current to be
approximately 30-50% of the maximum input
current. Make sure that the peak inductor
current is below 75% of the current limit at the
operating duty cycle to prevent loss of
regulation due to the current limit. Also make
sure that the inductor does not saturate under
the worst-case load transient and startup
conditions. Calculate the required inductance
value by the equation:
L=
VIN × (VOUT - VIN )
VOUT × f SW × ∆I
IIN(MAX ) =
VOUT × ILOAD (MAX )
VIN × η
∆I = (30% − 50%)IIN(MAX )
Where ILOAD(MAX) is the maximum load current, ∆I
is the peak-to-peak inductor ripple current, and η
is efficiency.
Selecting the Diode
The output rectifier diode supplies current to the
inductor when the internal MOSFET is off. To
reduce losses due to diode forward voltage and
recovery time, use a Schottky diode with the
MP3213. The diode should be rated for a
reverse voltage equal to or greater than the
output voltage used. The average current rating
must be greater than the maximum load current
expected, and the peak current rating must be
greater than the peak inductor current.
Selecting the Inductor
The inductor is required to force the higher
output voltage while being driven by the input
voltage. A larger value inductor results in less
ripple current that results in lower peak inductor
current, reducing stress on the internal
N-Channel.switch. However, the larger value
inductor has a larger physical size, higher
series resistance, and/or lower saturation
current.
MP3213 Rev. 1.1
5/12/2006
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7
MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
Compensation
The output of the transconductance error
amplifier (COMP) is used to compensate the
regulation control system. The system uses two
poles and one zero to stabilize the control loop.
The poles are fP1 set by the output capacitor C2
and load resistance and fP2 set by the
compensation capacitor C3. The zero fZ1 is set
by the compensation capacitor C3 and the
compensation resistor R3. These are
determined by the equations:
fP1 =
fP2 =
1
π × C2 × RLOAD
G EA
2 × π × C3 × A VEA
f Z1 =
1
2 × π × C3 × R3
Where RLOAD is the load resistance, GEA is the
error amplifier transconductance, and AVEA is
the error amplifier voltage gain.
The DC loop gain is:
A VDC =
1.5 × A VEA × VIN × R LOAD × VFB
VOUT
2
Where VFB is the feedback regulation threshold.
There is also a right-half-plane zero (fRHPZ) that
exists in continuous conduction mode (inductor
current does not drop to zero on each cycle)
step-up converters. The frequency of the right
half plane zero is:
2
fRHPZ =
VIN × R LOAD
2 × π × L × VOUT
2
Table
1
lists
generally
recommended
compensation components for different input
voltage, output voltage and capacitance of most
frequently used output ceramic capacitors.
Ceramic capacitors have extremely low ESR,
therefore the second compensation capacitor
(from COMP to GND) is not required.
MP3213 Rev. 1.1
5/12/2006
Table 1—Component Selection
VIN
(V)
VOUT
(V)
C2
(µF)
R3
(kΩ)
C3
(nF)
3.3
3.3
3.3
3.3
3.3
3.3
3.3
3.3
3.3
5
5
5
5
5
5
5
5
5
12
12
12
12
12
12
8
8
8
12
12
12
18
18
18
8
8
8
12
12
12
18
18
18
15
15
15
18
18
18
4.7
10
22
4.7
10
22
4.7
10
22
4.7
10
22
4.7
10
22
4.7
10
22
4.7
10
22
4.7
10
22
10
10
10
15
15
15
20
20
30
10
10
15
15
15
20
20
20
30
10
10
15
5.1
5.1
15
2.2
2.2
2.2
1
1
2.2
1
1
2.2
4.7
4.7
1
2.2
2.2
1
1
1
1
2.2
2.2
1
2.2
2.2
1
For faster control loop and better transient
response, set the capacitor C3 to the
recommended value in Table 1. Then slowly
increase the resistor R3 and check the load
step response on a bench to make sure the
ringing and overshoot on the output voltage at
the edge of the load steps is minimal. Finally,
the compensation needs to be checked by
calculating the DC loop gain and the crossover
frequency. The crossover frequency where the
loop gain drops to 0dB or a gain of 1 can be
obtained visually by placing a –20dB/decade
slope at each pole, and a +20dB/decade slope
at each zero. The crossover frequency should
be at least one decade below the frequency of
the right-half-plane zero at maximum output
load current to obtain high enough phase
margin for stability.
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MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
Layout Consideration
High frequency switching regulators require very
careful layout for stable operation and low noise.
All components must be placed as close to the IC
as possible. Keep the path between the SW pin,
output diode, output capacitor and GND pin
extremely short for minimal noise and ringing. The
input capacitor must be placed close to the IN pin
for best decoupling. All feedback components
must be kept close to the FB pin to prevent noise
injection on the FB pin trace. The ground return of
the input and output capacitors should be tied
close to the GND pin. See the MP3213 demo
board layout for reference.
TYPICAL APPLICATION CIRCUIT
D1
VIN
5V
6
7
OFF ON
3
8
5
IN
FSEL
EN
SS
SW
FB
2
MP3213
GND
4
C4
10nF
VOUT
12V
COMP
1
C3
2.2nF
Figure 2—Typical Application Circuit
MP3213 Rev. 1.1
5/12/2006
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MP3213 – 700KHZ/1.3MHZ BOOST CONVERTER WITH A 3.5A SWITCH
PACKAGE INFORMATION
MSOP8
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS
products into any application. MPS will not assume any legal responsibility for any said applications.
MP3213 Rev. 1.1
5/12/2006
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