AD AD9236BCPRL7-80 12-bit, 80 msps, 3 v a/d converter Datasheet

12-Bit, 80 MSPS, 3 V A/D Converter
AD9236
FEATURES
Single 3 V supply operation (2.7 V to 3.6 V)
SNR = 70.4 dBc to Nyquist
SFDR = 87.8 dBc to Nyquist
Low power: 366 mW
Differential input with 500 MHz bandwidth
On-chip reference and sample-and-hold
DNL = ±0.4 LSB
Flexible analog input: 1 V p-p to 2 V p-p range
Offset binary or twos complement data format
Clock duty cycle stabilizer
FUNCTIONAL BLOCK DIAGRAM
AVDD
DRVDD
AD9236
VIN+
8-STAGE
1 1/2-BIT PIPELINE
MDAC1
SHA
VIN–
4
A/D
16
3
A/D
REFT
REFB
CORRECTION LOGIC
OTR
12
OUTPUT BUFFERS
D11 (MSB)
VREF
APPLICATIONS
High end medical imaging equipment
IF sampling in communications receivers
WCDMA, CDMA-One, CDMA-2000
Battery-powered instruments
Hand-held scopemeters
Low cost digital oscilloscopes
DTV subsystems
GENERAL DESCRIPTION
The AD9236 is a monolithic, single 3 V supply, 12-bit, 80 MSPS
analog-to-digital converter featuring a high performance sampleand-hold amplifier (SHA) and voltage reference. The AD9236
uses a multistage differential pipelined architecture with output
error correction logic to provide 12-bit accuracy at 80 MSPS
and guarantee no missing codes over the full operating
temperature range.
The wide bandwidth, truly differential SHA allows a variety of
user-selectable input ranges and common modes, including
single-ended applications. It is suitable for multiplexed systems
that switch full-scale voltage levels in successive channels and
for sampling single-channel inputs at frequencies well beyond
the Nyquist rate. Combined with power and cost savings over
previously available analog-to-digital converters, the AD9236 is
suitable for applications in communications, imaging, and
medical ultrasound.
A single-ended clock input is used to control all internal
conversion cycles. A duty cycle stabilizer (DCS) compensates
for wide variations in the clock duty cycle while maintaining
excellent overall ADC performance. The digital output data is
D0 (LSB)
SENSE
0.5V
REF
SELECT
CLOCK
DUTY CYCLE
STABILIZER
AGND
CLK
MODE
SELECT
PDWN
MODE DGND
03066-0-001
Figure 1.
presented in straight binary or twos complement formats. An
out-of-range (OTR) signal indicates an overflow condition that
can be used with the most significant bit to determine low or
high overflow. Fabricated on an advanced CMOS process, the
AD9236 is available in a 28-lead TSSOP and a 32-lead LFCSP
and is specified over the industrial temperature range
(−40°C to +85°C).
PRODUCT HIGHLIGHTS
1. The AD9236 operates from a single 3 V power supply and
features a separate digital output driver supply to
accommodate 2.5 V and 3.3 V logic families.
2. Operating at 80 MSPS, the AD9236 consumes a low 366 mW.
3. The patented SHA input maintains excellent performance for
input frequencies up to 100 MHz, and can be configured for
single-ended or differential operation.
4. The AD9236 is pin compatible with the AD9215, AD9235,
and AD9245. This allows a simplified migration from 10 bits
to 14 bits and 20 MSPS to 80 MSPS.
5. The DCS maintains overall ADC performance over a wide
range of clock pulse widths.
6. The OTR output bit indicates when the signal is beyond the
selected input range.
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
© 2006 Analog Devices, Inc. All rights reserved.
AD9236
TABLE OF CONTENTS
Features .............................................................................................. 1
Equivalent Circuits......................................................................... 10
Applications....................................................................................... 1
Typical Performance Characteristics ........................................... 11
General Description ......................................................................... 1
Theory of Operation ...................................................................... 14
Functional Block Diagram .............................................................. 1
Analog Input and Reference Overview ................................... 14
Product Highlights ........................................................................... 1
Clock Input Considerations...................................................... 15
Revision History ............................................................................... 2
Power Dissipation and Standby Mode .................................... 16
DC Specifications ............................................................................. 3
Digital Outputs ........................................................................... 16
AC Specifications.............................................................................. 4
Timing ......................................................................................... 17
Digital Specifications........................................................................ 5
Voltage Reference ....................................................................... 17
Switching Specifications .................................................................. 6
Operational Mode Selection ..................................................... 18
Absolute Maximum Ratings............................................................ 7
Evaluation Board ........................................................................ 18
Thermal Resistance ...................................................................... 7
Outline Dimensions ....................................................................... 33
ESD Caution.................................................................................. 7
Ordering Guide .......................................................................... 34
Terminology ...................................................................................... 8
Pin Configurations and Function Descriptions ........................... 9
REVISION HISTORY
1/06—Rev. A to Rev. B
Changes to Figure 29...................................................................... 15
Changes to Equation in Jitter Considerations Section .............. 16
Changes to Internal Reference Connection Section, Figure 34,
and Table 10..................................................................................... 17
Changes to Figure 35...................................................................... 18
Changes to Figure 38...................................................................... 20
Changes to Figure 39...................................................................... 21
Changes to Figure 48...................................................................... 27
Changes to Figure 49...................................................................... 28
Changes to Figure 50...................................................................... 29
Changes to Table 12........................................................................ 32
Updated Outline Dimensions ....................................................... 33
Changes to Ordering Guide .......................................................... 34
10/03—Rev. 0 to Rev. A
Changes to Figure 30...................................................................... 15
Changes to Figure 33 ..................................................................... 17
Changes to Figure 40...................................................................... 22
Changes to Figure 49...................................................................... 28
Changes to Figure 50...................................................................... 29
Changes to Table 11 ....................................................................... 32
Changes to Ordering Guide ......................................................... 33
Rev. B | Page 2 of 36
AD9236
DC SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, sample rate = 80 MSPS, 2 V p-p differential input, 1.0 V external reference, unless otherwise noted.
Table 1.
Parameter
RESOLUTION
ACCURACY
No Missing Codes
Offset Error 1
Gain Error
Gain Error1
Differential Nonlinearity (DNL) 2
Integral Nonlinearity (INL)2
TEMPERATURE DRIFT
Offset Error1
Gain Error
Gain Error1
INTERNAL VOLTAGE REFERENCE
Output Voltage Error (1 V)
Load Regulation @ 1.0 mA
Output Voltage Error (0.5 V)
Load Regulation @ 0.5 mA
INPUT REFERRED NOISE
VREF = 0.5 V
VREF = 1.0 V
ANALOG INPUT
Input Span, VREF = 0.5 V
Input Span, VREF = 1.0 V
Input Capacitance 3
REFERENCE INPUT RESISTANCE
POWER SUPPLIES
Supply Voltage
AVDD
DRVDD
Supply Current
IAVDD 4
IDRVDD4
PSRR
POWER CONSUMPTION
Low Frequency Input4
Standby Power 5
AD9236BRU/AD9236BCP
Typ
Max
Temp
Full
Test Level
VI
Full
Full
25°C
Full
Full
Full
VI
VI
V
VI
VI
VI
Guaranteed
±0.30
±0.10
±0.30
±0.40
±0.35
Full
Full
Full
V
V
V
±6
±12
±18
Full
25°C
25°C
25°C
VI
V
V
V
±2
0.8
±1
0.1
25°C
25°C
V
V
0.55
0.28
LSB rms
LSB rms
Full
Full
Full
Full
IV
IV
V
V
1
2
7
7
V p-p
V p-p
pF
kΩ
Full
Full
IV
IV
Full
25°C
25°C
25°C
25°C
Min
12
±4.34
±0.65
±1.20
% FSR
% FSR
% FSR
LSB
LSB
ppm/°C
ppm/°C
ppm/°C
±35
mV
mV
mV
mV
3.0
2.5
3.6
3.6
V
V
VI
V
V
122
8
±0.01
137
mA
mA
% FSR
V
V
366
1.0
1
2.7
2.25
±1.30
Unit
Bits
mW
mW
With a 1.0 V internal reference.
Measured at low input frequency, full-scale sine wave, with approximately 5 pF loading on each output bit.
3
Input capacitance refers to the effective capacitance between one differential input pin and AGND. Refer to Figure 5 for the equivalent analog input structure.
4
Measured at AC Specifications conditions without output drivers.
5
Measured with a dc input, CLK pin inactive (that is, set to AVDD or AGND).
2
Rev. B | Page 3 of 36
AD9236
AC SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, sample rate = 80 MSPS, 2 V p-p differential input, 1.0 V external reference, AIN = –0.5 dBFS, DCS off,
unless otherwise noted.
Table 2.
Parameter
SIGNAL-TO-NOISE-RATIO (SNR)
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
SIGNAL-TO-NOISE AND DISTORTION (SINAD)
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
EFFECTIVE NUMBER OF BITS (ENOB)
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
WORST SECOND OR THIRD
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
SPURIOUS FREE DYNAMIC RANGE (SFDR)
fIN = 2.4 MHz
fIN = 40 MHz
fIN = 70 MHz
fIN = 100 MHz
Temp
Test Level
Min
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
68.6
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
VI
V
V
Full
25°C
25°C
Full
25°C
25°C
VI
V
V
IV
V
V
Rev. B | Page 4 of 36
AD9236BRU/AD9236BCP
Typ
Max
dB
dB
dB
dB
dB
dB
70.9
70.4
67.8
70.1
69.0
68.4
dB
dB
dB
dB
dB
dB
70.8
70.2
67.4
69.8
68.0
11.1
Bits
Bits
Bits
Bits
Bits
Bits
11.5
11.4
10.9
11.3
11.0
–75.6
–91.3
–87.8
–73.2
–81.4
–76.4
75.6
91.3
87.8
73.2
81.4
76.4
Unit
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
AD9236
DIGITAL SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, 1.0 V external reference, unless otherwise noted.
Table 3.
Parameter
LOGIC INPUTS (CLK, PDWN)
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Capacitance
DIGITAL OUTPUTS (D0–D11, OTR) 1
DRVDD = 3.3 V
High Level Output Voltage (IOH = 50 μA)
High Level Output Voltage (IOH = 0.5 mA)
Low Level Output Voltage (IOH = 1.6 mA)
Low Level Output Voltage (IOH = 50 μA)
DRVDD = 2.5 V
High Level Output Voltage (IOH = 50 μA)
High Level Output Voltage (IOH = 0.5 mA)
Low Level Output Voltage (IOH = 1.6 mA)
Low Level Output Voltage (IOH = 50 μA)
1
AD9236BRU/AD9236BCP
Min
Typ
Max
Temp
Test Level
Full
Full
Full
Full
Full
IV
IV
IV
IV
V
Full
Full
Full
Full
IV
IV
IV
IV
3.29
3.25
Full
Full
Full
Full
IV
IV
IV
IV
2.49
2.45
Output voltage levels measured with 5 pF load on each output.
Rev. B | Page 5 of 36
2.0
0.8
+10
+10
–10
–10
2
Unit
V
V
μA
μA
pF
0.2
0.05
V
V
V
V
0.2
0.05
V
V
V
V
AD9236
SWITCHING SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, unless otherwise noted.
Table 4.
Parameter
CLOCK INPUT PARAMETERS
Maximum Conversion Rate
Minimum Conversion Rate
CLK Period
CLK Pulse Width High 1
CLK Pulse Width Low1
DATA OUTPUT PARAMETERS
Output Propagation Delay (tPD) 2
Pipeline Delay (Latency)
Aperture Delay (tA)
Aperture Uncertainty (Jitter, tJ)
Wake-Up Time 3
OUT OF RANGE RECOVERY TIME
1
2
3
Temp
Test Level
Min
Full
Full
Full
Full
Full
VI
V
V
V
V
80
Full
Full
Full
Full
Full
Full
V
V
V
V
V
V
AD9236BRU/AD9236BCP
Typ
Max
Unit
MSPS
MSPS
ns
ns
ns
1
12.5
4.0
4.0
3.5
7
1.0
0.3
7
2
ns
Cycles
ns
ps rms
ms
Cycles
With duty cycle stabilizer (DCS) enabled.
Output propagation delay is measured from CLK 50% transition to DATA 50% transition, with 5 pF load.
Wake-up time is dependant on the value of the decoupling capacitors; typical values shown with 0.1 μF and 10 μF capacitors on REFT and REFB.
N
N+1
N+2
N–1
tA
ANALOG
INPUT
N+8
N+3
N+7
N+4
N+5
N+6
CLK
DATA
OUT
N–9
N–8
N–7
N–6
N–5
N–4
N–3
N–2
N–1
N
tPD = 6.0ns MAX
2.0ns MIN
03066-0-002
Figure 2. Timing Diagram
Table 5. Explanation of Test Levels
Test Level
I
II
III
IV
V
VI
Definitions
100% production tested.
100% production tested at 25°C and guaranteed by design and characterization at specified temperatures.
Sample tested only.
Parameter is guaranteed by design and characterization testing.
Parameter is a typical value only.
100% production tested at 25°C and guaranteed by design and characterization for industrial temperature range.
Rev. B | Page 6 of 36
AD9236
ABSOLUTE MAXIMUM RATINGS
Table 6.
THERMAL RESISTANCE
With
Respect to
Parameter
ELECTRICAL
AVDD
AGND
DRVDD
DGND
AGND
DGND
AVDD
DRVDD
D0 to D11
DGND
CLK, MODE
AGND
VIN+, VIN–
AGND
VREF
AGND
SENSE
AGND
REFT, REFB
AGND
PDWN
AGND
ENVIRONMENTAL
Storage Temperature
Operating Temperature Range
Lead Temperature
(Soldering 10 sec)
Junction Temperature
Min
Max
Unit
θJA is specified for the worst-case conditions on a 4-layer board
in still air, in accordance with EIA/JESD51-1.
–0.3
–0.3
–0.3
–3.9
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
+3.9
+3.9
+0.3
+3.9
DRVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
V
V
V
V
V
V
V
V
V
V
V
Table 7.
–65
–40
+125
+85
300
°C
°C
°C
150
°C
Package Type
RU-28
CP-32-2
θJA
67.7
32.5
θJC
Unit
32.71
°C/W
°C/W
Airflow increases heat dissipation effectively, reducing θJA. In
addition, more metal directly in contact with the package leads
from metal traces, through holes, ground, and power planes
reduces the θJA. It is recommended that the exposed paddle be
soldered to the ground plane for the LFCSP package. There is
an increased reliability of the solder joints, and maximum
thermal capability of the package is achieved with the exposed
paddle soldered to the customer board.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only and functional operation of the device at these or
any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. B | Page 7 of 36
AD9236
TERMINOLOGY
Analog Bandwidth (Full Power Bandwidth)
The analog input frequency at which the spectral power of the
fundamental frequency (as determined by the FFT analysis) is
reduced by 3 dB.
Signal-to-Noise and Distortion (SINAD)1
The ratio of the rms input signal amplitude to the rms value of
the sum of all other spectral components below the Nyquist
frequency, including harmonics but excluding dc.
Aperture Delay (tA)
The delay between the 50% point of the rising edge of the clock
and the instant at which the analog input is sampled.
Effective Number of Bits (ENOB)
The effective number of bits for a sine wave input at a given
input frequency can be calculated directly from its measured
SINAD using the following formula
Aperture Uncertainty (Jitter, tJ)
The sample-to-sample variation in aperture delay.
ENOB =
Integral Nonlinearity (INL)
The deviation of each individual code from a line drawn from
negative full scale through positive full scale. The point used as
negative full scale occurs ½ LSB before the first code transition.
Positive full scale is defined as a level 1½ LSB beyond the last
code transition. The deviation is measured from the middle of
each particular code to the true straight line.
Differential Nonlinearity (DNL, No Missing Codes)
An ideal ADC exhibits code transitions that are exactly 1 LSB
apart. DNL is the deviation from this ideal value. Guaranteed
no missing codes to 12-bit resolution indicates that all 4096
codes must be present over all operating ranges.
Offset Error
The major carry transition should occur for an analog value
½ LSB below VIN+ = VIN–. Offset error is defined as the deviation
of the actual transition from that point.
Gain Error
The first code transition should occur at an analog value
½ LSB above negative full scale. The last transition should occur
at an analog value 1½ LSB below positive full scale. Gain error
is the deviation of the actual difference between first and last
code transitions and the ideal difference between first and last
code transitions.
Temperature Drift
The temperature drift for offset error and gain error specifies
the maximum change from the initial (25°C) value to the value
at TMIN or TMAX.
Power Supply Rejection Ratio
The change in full scale from the value with the supply at the
minimum limit to the value with the supply at its maximum limit.
Total Harmonic Distortion (THD) 1
The ratio of the rms input signal amplitude to the rms value of
the sum of the first six harmonic components.
(SINAD − 1.76 )
6.02
Signal-to-Noise Ratio (SNR)1
The ratio of the rms input signal amplitude to the rms value of
the sum of all other spectral components below the Nyquist
frequency, excluding the first six harmonics and dc.
Spurious Free Dynamic Range (SFDR)1
The difference in dB between the rms input signal amplitude
and the peak spurious signal. The peak spurious component
may or may not be a harmonic.
Two-Tone SFDR1
The ratio of the rms value of either input tone to the rms value
of the peak spurious component. The peak spurious component
may or may not be an IMD product.
Clock Pulse Width and Duty Cycle
Pulse width high is the minimum amount of time that the clock
pulse should be left in the Logic 1 state to achieve rated
performance. Pulse width low is the minimum time the clock
pulse should be left in the low state. At a given clock rate, these
specifications define an acceptable clock duty cycle.
Minimum Conversion Rate
The clock rate at which the SNR of the lowest analog signal
frequency drops by no more than 3 dB below the guaranteed limit.
Maximum Conversion Rate
The clock rate at which parametric testing is performed.
Output Propagation Delay (tPD)
The delay between the clock rising edge and the time when all
bits are within valid logic levels.
Out-of-Range Recovery Time
The time it takes for the ADC to reacquire the analog input
after a transition from 10% above positive full scale to 10%
above negative full scale, or from 10% below negative full scale
to 10% below positive full scale.
1
AC specifications may be reported in dBc (degrades as signal levels are
lowered) or in dBFS (always related back to converter full scale).
Rev. B | Page 8 of 36
AD9236
REFB
5
24 DRVDD
DNC 1
24 VREF
CLK 2
23 SENSE
DNC 3
REFT
6
AVDD
7
AGND
8
VIN+
AD9236
23 DGND
25 REFB
25 D8
26 REFT
26 D9
4
27 AVDD
3
VREF
28 AGND
SENSE
29 VIN+
28 D11 (MSB)
27 D10
30 VIN–
1
2
31 AGND
OTR
MODE
32 AVDD
PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
22 MODE
AD9236
PDWN 4
21 OTR
CSP
22 D7
DNC 5
21 D6
DNC 6
9
20 D5
(LSB) D0 7
18 D9
VIN– 10
19 D4
D1 8
17 D8
AGND 11
18 D3
AVDD 12
17 D2
CLK 13
16 D1
PDWN 14
20 D11 (MSB)
DRVDD 16
19 D10
DGND 15
D7 14
D6 13
D5 12
D4 11
D2 9
TOP VIEW
(Not to Scale)
D3 10
TOP VIEW
(Not to Scale)
15 D0 (LSB)
03066-0-022
03066-0-021
Figure 4. 32-Lead LFCSP
Figure 3. 28-Lead TSSOP
Table 8. Pin Function Descriptions—28-Lead TSSOP
Table 9. Pin Function Descriptions—32-Lead LFCSP
Pin No.
1
2
Mnemonic
OTR
MODE
3
4
5
6
7, 12
8, 11
9
10
13
14
15 to 22,
25 to 28
23
24
SENSE
VREF
REFB
REFT
AVDD
AGND
VIN+
VIN–
CLK
PDWN
D0 (LSB) to
D11 (MSB)
DGND
DRVDD
Pin No.
1, 3, 5, 6
2
4
7 to 14,
17 to 20
15
16
21
22
Mnemonic
DNC
CLK
PDWN
D0 (LSB) to
D11 (MSB)
DGND
DRVDD
OTR
MODE
23
24
25
26
27, 32
28, 31
29
30
SENSE
VREF
REFB
REFT
AVDD
AGND
VIN+
VIN–
Description
Out-of-Range Indicator
Data Format Select and DCS
Mode Selection
Reference Mode Selection
Voltage Reference Input/Output
Differential Reference (–)
Differential Reference (+)
Analog Power Supply
Analog Ground
Analog Input Pin (+)
Analog Input Pin (–)
Clock Input Pin
Power-Down Function Select
Data Output Bits
Digital Output Ground
Digital Output Driver Supply
Rev. B | Page 9 of 36
Description
Do Not Connect
Clock Input Pin
Power-Down Function Select
Data Output Bits
Digital Output Ground
Digital Output Driver Supply
Out-of-Range Indicator
Data Format Select and DCS
Mode Selection
Reference Mode Selection
Voltage Reference Input/Output
Differential Reference (–)
Differential Reference (+)
Analog Power Supply
Analog Ground
Analog Input Pin (+)
Analog Input Pin (–)
AD9236
EQUIVALENT CIRCUITS
AVDD
DRVDD
D11-D0,
OTR
VIN+, VIN–
03600-0-005
03600-0-003
Figure 5. Equivalent Analog Input Circuit
Figure 7. Equivalent Digital Output Circuit
AVDD
AVDD
CLK,
PDWN
MODE
20kΩ
03600-0-004
03600-0-006
Figure 6. Equivalent MODE Input Circuit
Figure 8. Equivalent Digital Input Circuit
Rev. B | Page 10 of 36
AD9236
TYPICAL PERFORMANCE CHARACTERISTICS
AVDD = 3.0 V, DRVDD = 2.5 V, sample rate = 80 MSPS, DCS disabled, TA = 25°C, 2 V p-p differential input, AIN = –0.5 dBFS,
VREF = 1.0 V external, unless otherwise noted.
0
100
AIN = –0.5dBFS
SNR = 71.0dBc
ENOB = 11.5 BITS
SFDR = 93.6dBc
–10
90
SNR/SFDR (dBc AND dBFS)
–20
SFDR (dBFS)
AMPLITUDE (dBFS)
–30
–40
–50
–60
–70
–80
–90
–100
SFDR (dBc)
80
SFDR = 90dB
REFERENCE LINE
70
SNR (dBFS)
60
SNR (dBc)
50
–110
–120
0
5
10
15
20
25
30
35
40
–30
40
–25
FREQUENCY (MHz)
–20
–15
–10
–5
03066-0-031
Figure 9. Single Tone 8K FFT @ 2.5 MHz
03066-0-048
Figure 12. Single Tone SNR/SFDR vs. Input Amplitude (AIN) @ 2.5 MHz
0
100
AIN = –0.5dBFS
SNR = 70.6dBc
ENOB = 11.4 BITS
SFDR = 87.8dBc
–10
SFDR (dBFS)
90
SNR/SFDR (dBc AND dBFS)
–20
–30
AMPLITUDE (dBFS)
0
INPUT AMPLITUDE (dBFS)
–40
–50
–60
–70
–80
–90
–100
SFDR (dBc)
80
SFDR = 90dB
REFERENCE LINE
70
SNR (dBFS)
SNR (dBc)
60
50
–110
–120
0
5
10
15
20
25
30
35
40
–30
40
–25
FREQUENCY (MHz)
–20
–15
–10
–5
0
INPUT AMPLITUDE (dBFS)
03066-0-032
Figure 10. Single Tone 8K FFT @ 39 MHz
03066-0-049
Figure 13. Single Tone SNR/SFDR vs. Input Amplitude (AIN) @ 39 MHz
0
100
AIN = –0.5dBFS
SNR = 70.1dBc
ENOB = 11.3 BITS
SFDR = 81.9dBc
–10
–20
SFDR (DIFF)
90
–40
SNR/SFDR (dBc)
AMPLITUDE (dBFS)
–30
–50
–60
–70
–80
SFDR (SE)
80
SNR (DIFF)
70
SNR (SE)
–90
60
–100
–110
–120
50
0
5
10
15
20
25
30
35
40
0
FREQUENCY (MHz)
20
40
60
80
100
SAMPLE RATE (MSPS)
03066-0-033
03066-0-042
Figure 11. Single Tone 8K FFT @ 70 MHz
Figure 14. SNR/SFDR vs. Sample Rate @ 10 MHz
Rev. B | Page 11 of 36
AD9236
0
SFDR (dBFS)
AIN = –6.5dBFS
SNR = 71.3dBFS
SFDR = 92.5dBc
–10
100
SNR/SFDR (dBc AND dBFS)
–20
AMPLITUDE (dBFS)
–30
–40
–50
–60
–70
–80
–90
90
SFDR (dBc)
80
70
SNR (dBFS)
SFDR = 90dB
REFERENCE LINE
60
SNR (dBc)
50
–100
–110
40
–30
–120
0
5
10
15
20
25
30
35
40
–27
–24
–21
–18
–15
–12
–9
–6
INPUT AMPLITUDE (dBFS)
FREQUENCY (MHz)
03066-0-039
03066-0-036
Figure 15. Two-Tone 8K FFT @ 30 MHz and 31 MHz
Figure 18. Two-Tone SNR/SFDR vs. Input Amplitude @ 30 MHz and 31 MHz
0
100
AIN = –6.5dBFS
SNR = 71.0dBFS
SFDR = 79.3dBc
–10
–20
SNR/SFDR (dBc AND dBFS)
–30
AMPLITUDE (dBFS)
SFDR (dBFS)
90
–40
–50
–60
–70
–80
–90
–100
SFDR (dBc)
80
70
SNR (dBFS)
60
SFDR = 90dB
REFERENCE LINE
SNR(dBc)
50
–110
–120
0
5
10
15
20
25
30
35
40
–30
40
–27
–24
FREQUENCY (MHz)
–21
–18
–15
–12
03066-0-037
Figure 16. Two-Tone 8K FFT @ 69 MHz and 70 MHz
–6
03066-0-040
Figure 19. Two-Tone SNR/SFDR vs. Input Amplitude @ 69 MHz and 70 MHz
1.0
1.0
0.8
0.8
0.6
0.6
0.4
0.4
0.2
0.2
DNL (LSB)
INL (LSB)
–9
INPUT AMPLITUDE (dBFS)
0
–0.2
0
–0.2
–0.4
–0.4
–0.6
–0.6
–0.8
–0.8
–1.0
–1.0
0
1024
2048
3072
4096
0
CODE
1024
2048
3072
4096
CODE
03066-0-038
03066-0-041
Figure 17. Typical INL
Figure 20. Typical DNL
Rev. B | Page 12 of 36
AD9236
72.0
100
71.5
95
71.0
–40°C
–40°C
90
SFDR (dBc)
SNR (dBc)
70.5
+25°C
70.0
+85°C
69.5
+85°C
85
+25°C
80
69.0
75
68.5
68.0
70
0
25
50
75
100
0
125
25
50
75
100
125
INPUT FREQUENCY (MHz)
INPUT FREQUENCY (MHz)
03066-0-045
03066-0-047
Figure 21. SNR vs. Input Frequency
Figure 24. SFDR vs. Input Frequency
95
0
SFDR (DCS ON)
–10
90
–20
–30
SFDR (DCS OFF)
AMPLITUDE (dBFS)
SNR/SFDR (dBc)
85
80
SNR (DCS OFF)
75
70
65
–40
–50
–60
–70
–80
–90
SNR (DCS ON)
–100
60
–110
55
30
–120
35
40
45
50
55
60
65
70
0
DUTY CYCLE (%)
0
–20
AMPLITUDE (dBFS)
–30
–40
–50
–60
–70
–80
–90
–100
–110
–120
15.36
23.04
23.04
30.72
Figure 25. 32K FFT WCDMA Carrier @ FIN =76.8 MHz,
Sample Rate = 61.44 MSPS
–10
7.68
15.36
03066-0-061
Figure 22. SNR/SFDR vs. Clock Duty Cycle
0
7.68
FREQUENCY (MHz)
03066-0-046
30.72
FREQUENCY (MHz)
03066-0-060
Figure 23. 32K FFT CDMA-2000 Carrier @ FIN = 46.08 MHz, Sample Rate =
61.44 MSPS
Rev. B | Page 13 of 36
AD9236
THEORY OF OPERATION
The AD9236 architecture consists of a front-end sample-andhold amplifier (SHA) followed by a pipelined switched capacitor
ADC. The pipelined ADC is divided into three sections,
consisting of a 4-bit first stage followed by eight 1.5-bit stages
and a final 3-bit flash. Each stage provides sufficient overlap to
correct for flash errors in the preceding stages. The quantized
outputs from each stage are combined into a final 12-bit result
in the digital correction logic. The pipelined architecture
permits the first stage to operate on a new input sample, while
the remaining stages operate on preceding samples. Sampling
occurs on the rising edge of the clock.
Each stage of the pipeline, excluding the last, consists of a low
resolution flash ADC connected to a switched capacitor DAC
and interstage residue amplifier (MDAC). The residue amplifier
magnifies the difference between the reconstructed DAC output
and the flash input for the next stage in the pipeline. One bit of
redundancy is used in each stage to facilitate digital correction
of flash errors. The last stage simply consists of a flash ADC.
Referring to Figure 27, the clock signal alternately switches the
SHA between sample mode and hold mode. When the SHA is
switched into sample mode, the signal source must be capable
of charging the sample capacitors and settling within one-half
of a clock cycle. A small resistor in series with each input can
help reduce the peak transient current required from the output
stage of the driving source. In addition, a small shunt capacitor
can be placed across the inputs to provide dynamic charging
currents. This passive network creates a low-pass filter at the
ADC’s input; therefore, the precise values are dependant upon
the application. In IF undersampling applications, any shunt
capacitors should be reduced or removed. In combination with the
driving source impedance, they would limit the input bandwidth.
H
T
CPAR
T
The input stage contains a differential SHA that can be ac- or
dc-coupled in differential or single-ended modes. The outputstaging block aligns the data, carries out the error correction,
and passes the data to the output buffers. The output buffers are
powered from a separate supply, allowing adjustment of the
output voltage swing. During power-down, the output buffers
go into a high impedance state.
ANALOG INPUT AND REFERENCE OVERVIEW
The analog input to the AD9236 is a differential switched
capacitor SHA that has been designed for optimum
performance while processing a differential input signal. The
SHA input can support a wide common-mode range (VCM)
and maintain excellent performance, as shown in Figure 26. An
input common-mode voltage of midsupply minimizes signaldependant errors and provides optimum performance.
T
5pF
VIN+
5pF
VIN–
CPAR
T
H
03066-0-012
Figure 27. Switched-Capacitor SHA Input
For best dynamic performance, the source impedances driving
VIN+ and VIN– should be matched such that common-mode
settling errors are symmetrical. These errors are reduced by the
common-mode rejection of the ADC.
An internal differential reference buffer creates positive and
negative reference voltages, REFT and REFB, that define the
span of the ADC core. The output common mode of the
reference buffer is set to midsupply, and the REFT and REFB
voltages and span are defined as follows:
100
REFT = ½(AVDD + VREF)
95
SFDR (2.5MHz)
REFB = ½(AVDD + VREF)
90
Span = 2 × (REFT − REFB) = 2 × VREF
SNR/SFDR (dBc)
85
SFDR (39MHz)
80
It can be seen from the previous equations that the REFT and
REFB voltages are symmetrical about the midsupply voltage and,
by definition, the input span is twice the value of the VREF voltage.
75
SNR (2.5MHz)
70
SNR (39MHz)
65
60
55
50
0.5
1.0
1.5
2.0
2.5
3.0
COMMON-MODE LEVEL (V)
03066-0-016
The internal voltage reference can be pin strapped to fixed
values of 0.5 V or 1.0 V, or adjusted within the same range as
discussed in the Internal Reference Connection section.
Maximum SNR performance is achieved with the AD9236 set
to the largest input span of 2 V p-p. The relative SNR degradation is
3 dB when changing from 2 V p-p mode to 1 V p-p mode.
Figure 26. SNR, SFDR vs. Common-Mode Level
Rev. B | Page 14 of 36
AD9236
The SHA can be driven from a source that keeps the signal
peaks within the allowable range for the selected reference
voltage. The minimum and maximum common-mode input
levels are defined as:
VCM MIN =
33Ω
2V p-p
49.9Ω
20pF
33Ω
VREF
2
VCM MAX =
AVDD
VIN+
AD9236
VIN–
AGND
1kΩ
( AVDD + VREF )
0.1μF
2
1kΩ
03600-0-014
The minimum common-mode input level allows the AD9236 to
accommodate ground referenced inputs.
Although optimum performance is achieved with a differential
input, a single-ended source can be applied to VIN+ or VIN–.
In this configuration, one input accepts the signal, while the
opposite input should be set to midscale by connecting it to an
appropriate reference. For example, a 2 V p-p signal can be
applied to VIN+ while a 1 V reference is applied to VIN–. The
AD9236 then accepts an input signal varying between 2 V and
0 V. In the single-ended configuration, distortion performance
can degrade significantly as compared to the differential case.
However, the effect is less noticeable at lower input frequencies.
Differential Input Configurations
Figure 29. Differential Transformer-Coupled Configuration
The signal characteristics must be considered when selecting
a transformer. Most RF transformers saturate at frequencies
below a few MHz, and excessive signal power can also cause
core saturation, which leads to distortion.
Single-Ended Input Configuration
A single-ended input can provide adequate performance in
cost-sensitive applications. In this configuration, there is a
degradation in SFDR and distortion performance due to the
large input common-mode swing (see Figure 14). However, if
the source impedances on each input are matched, there should
be little effect on SNR performance. Figure 30 details a typical
single-ended input configuration.
As previously detailed, optimum performance is achieved while
driving the AD9236 in a differential input configuration. For
baseband applications, the AD8138 differential driver provides
excellent performance and a flexible interface to the ADC. The
output common-mode voltage of the AD8138 is easily set to
AVDD/2, and the driver can be configured in a Sallen-Key filter
topology to provide band limiting of the input signal.
1kΩ
2V p-p
33Ω
0.33μF 1kΩ
49.9Ω
20pF
1kΩ
+
10μF
0.1μF
33Ω
AVDD
VIN+
AD9236
VIN–
1kΩ
AGND
03600-A-015
1V p-p
49.9Ω
Figure 30. Single-Ended Input Configuration
499Ω
33Ω
499Ω
AD8138
1kΩ
VIN+
CLOCK INPUT CONSIDERATIONS
AD9236
33Ω
523Ω
0.1μF
20pF
AVDD
1kΩ
499Ω
VIN–
AGND
03066-0-013
Figure 28. Differential Input Configuration Using the AD8138
At input frequencies in the second Nyquist zone and above, the
performance of most amplifiers is not adequate to achieve the
true performance of the AD9236. This is especially true in IF
undersampling applications where frequencies in the 70 MHz
to 100 MHz range are being sampled. For these applications,
differential transformer coupling is the recommended input
configuration. The value of the shunt capacitor is dependent
on the input frequency and source impedance and should be
reduced or removed. An example is shown in Figure 29.
Typical high speed ADCs use both clock edges to generate a
variety of internal timing signals, and as a result can be sensitive
to clock duty cycle. Commonly a 5% tolerance is required on
the clock duty cycle to maintain dynamic performance
characteristics. The AD9236 contains a clock
duty cycle stabilizer (DCS) that retimes the nonsampling edge,
providing an internal clock signal with a nominal 50% duty
cycle. This allows a wide range of clock input duty cycles
without affecting the performance of the AD9236. As shown in
Figure 22, noise and distortion performance is nearly flat for a
30% to 70% duty cycle with the DCS on.
The duty cycle stabilizer uses a delay-locked loop (DLL) to
create the nonsampling edge. As a result, any changes to the
sampling frequency require approximately 100 clock cycles to
allow the DLL to acquire and lock to the new rate.
Rev. B | Page 15 of 36
AD9236
Jitter Considerations
High speed, high resolution ADCs are sensitive to the quality of
the clock input. The degradation in SNR at a given input frequency
(fINPUT) due only to aperture jitter (tJ) can be calculated with the
following equation:
425
120
400
⎤
⎥
⎥⎦
The clock input should be treated as an analog signal in cases
where aperture jitter can affect the dynamic range of the
AD9236. Power supplies for clock drivers should be separated
from the ADC output driver supplies to avoid modulating the
clock signal with digital noise. Low jitter, crystal controlled
oscillators make the best clock sources. If the clock is generated
from another type of source (by gating, dividing, or other
methods), it should be retimed by the original clock at the last step.
75
SNR (dBc)
0.2ps
MEASURED
SNR
60
0.5ps
55
1.0ps
1.5ps
50
2.0ps
2.5ps
3.0ps
45
40
10
100
80
TOTAL POWER
60
350
40
325
20
DIGITAL CURRENT
300
10
20
30
40
50
60
70
80
90
0
100
SAMPLE RATE (MSPS)
03066-0-044
Figure 32. Power and Current vs. Sample Rate @ 2.5 MHz
Reducing the capacitive load presented to the output drivers
can minimize digital power consumption. The data in Figure 32
was taken with the same operating conditions as the Typical
Performance Characteristics, and with a 5 pF load on each
output driver.
By asserting the PDWN pin high, the AD9236 is placed in
standby mode. In this state, the ADC typically dissipates
1 mW if the CLK and analog inputs are static. During
standby, the output drivers are placed in a high impedance
state. Reasserting the PDWN pin low returns the AD9236
to its normal operational mode.
70
65
375
CURRENT (mA)
100
In the equation, the rms aperture jitter represents the rootmean square of all jitter sources, which include the clock input,
analog input signal, and ADC aperture jitter specification. IF
undersampling applications are particularly sensitive to jitter
(see Figure 31).
1
140
ANALOG CURRENT
POWER (mW)
⎡
1
SNR = 20 log 10 ⎢
⎢⎣ 2πf INPUT × t J
which is determined by the sample rate and the characteristics
of the analog input signal.
1000
INPUT FREQUENCY (MHz)
03066-0-043
Figure 31. SNR vs. Input Frequency and Jitter
POWER DISSIPATION AND STANDBY MODE
As shown in Figure 32, the power dissipated by the AD9236 is
proportional to its sample rate. The digital power dissipation is
determined primarily by the strength of the digital drivers and
the load on each output bit. The maximum DRVDD current
(IDRVDD) can be calculated as
IDRVDD = VDRVDD × CLOAD × fCLK × N
I DRVDD = VDRVDD × C LOAD × f CLK × N
where N is the number of output bits, 12 in the case of the
AD9236. This maximum current occurs when every output bit
switches on every clock cycle, that is, a full-scale square wave at
the Nyquist frequency, fCLK/2. In practice, the DRVDD current is
established by the average number of output bits switching,
Low power dissipation in standby mode is achieved by shutting
down the reference, reference buffer, and biasing networks. The
decoupling capacitors on REFT and REFB are discharged when
entering standby mode and then must be recharged when
returning to normal operation. As a result, the wake-up time is
related to the time spent in standby mode, and shorter standby
cycles result in proportionally shorter wake-up times. With the
recommended 0.1 μF and 10 μF decoupling capacitors on REFT
and REFB, it takes approximately 1 second to fully discharge the
reference buffer decoupling capacitors and 7 ms to restore full
operation.
DIGITAL OUTPUTS
The AD9236 output drivers can be configured to interface with
2.5 V or 3.3 V logic families by matching DRVDD to the digital
supply of the interfaced logic. The output drivers are sized to
provide sufficient output current to drive a wide variety of logic
families. However, large drive currents tend to cause current
glitches on the supplies, which can affect converter performance.
Applications requiring the ADC to drive large capacitive loads
or large fanouts can require external buffers or latches.
As detailed in Table 11, the data format can be selected for
either offset binary or twos complement.
Rev. B | Page 16 of 36
AD9236
TIMING
The AD9236 provides latched data outputs with a pipeline delay
of seven clock cycles. Data outputs are available one propagation
delay (tPD) after the rising edge of the clock signal. Refer to
Figure 2 for a detailed timing diagram.
In all reference configurations, REFT and REFB drive the A/D
conversion core and establish its input span. The input range of
the ADC always equals twice the voltage at the reference pin for
either an internal or an external reference.
VIN+
The length of the output data lines and the loads placed on
them should be minimized to reduce transients within the
AD9236. These transients can degrade the converter’s dynamic
performance.
VIN–
0.1μF
ADC
CORE
+
0.1μF
10μF
REFB
The lowest typical conversion rate of the AD9236 is 1 MSPS. At
clock rates below 1 MSPS, dynamic performance can degrade.
0.1μF
VREF
10μF
VOLTAGE REFERENCE
REFT
+
0.1μF
SELECT
LOGIC
A stable and accurate 0.5 V voltage reference is built into the
AD9236. The input range can be adjusted by varying the
reference voltage applied to the AD9236 using either the
internal reference or an externally applied reference voltage.
The input span of the ADC tracks reference voltage changes
linearly. The various reference modes are summarized in Table 10
and described in the following sections.
SENSE
0.5V
AD9236
03066-A-017
Figure 33. Internal Reference Configuration
If the ADC is being driven differentially through a transformer,
the reference voltage can be used to bias the center tap
(common-mode voltage).
VIN+
VIN–
REFT
Internal Reference Connection
A comparator within the AD9236 detects the potential at the
SENSE pin and configures the reference into four possible
states, which are summarized in Table 10. If SENSE is
grounded, the reference amplifier switch is connected to the
internal resistor divider (see Figure 33), setting VREF to 1 V.
Connecting the SENSE pin to VREF switches the reference
amplifier output to the SENSE pin, completing the loop and
providing a 0.5 V reference output. If a resistor divider is
connected as shown in Figure 34, the switch is again set to the
SENSE pin. This puts the reference amplifier in a noninverting
mode with the VREF output defined as follows:
ADC
CORE
0.1μF
0.1μF
REFB
0.1μF
VREF
+
10μF
0.1μF
R2 ⎞
VREF = 0.5 × ⎛⎜1 +
⎟
R1 ⎠
⎝
SELECT
LOGIC
R2
SENSE
R1
0.5V
AD9236
03066-0-018
Figure 34. Programmable Reference Configuration
Table 10. Reference Configuration Summary
Selected Mode
External Reference
Internal Fixed Reference
Programmable Reference
SENSE Voltage
AVDD
VREF
0.2 V to VREF
Resulting VREF (V)
N/A
0.5
Internal Fixed Reference
AGND to 0.2 V
1.0
R2 ⎞ (See Figure 34)
0 . 5 × ⎛⎜ 1 +
⎟
R1 ⎠
⎝
Resulting Differential Span (V p-p)
2 × External Reference
1.0
2 × VREF
2.0
Rev. B | Page 17 of 36
+
10μF
AD9236
If the internal reference of the AD9236 is used to drive multiple
converters to improve gain matching, the loading of the reference
by the other converters must be considered. Figure 35 depicts
how the internal reference voltage is affected by loading. A
2 mA load is the maximum recommended load.
0.05
As discussed in the Digital Outputs section, the AD9236 can
output data in either offset binary or twos complement format.
There is also a provision for enabling or disabling the clock duty
cycle stabilizer (DCS). The MODE pin is a multilevel input that
controls the data format and DCS state. The input threshold
values and corresponding mode selections are outlined in Table 11.
Table 11. Mode Selection
0
–0.05
ERROR (%)
OPERATIONAL MODE SELECTION
MODE Voltage
AVDD
2/3 AVDD
1/3 AVDD
AGND (Default)
0.5V ERROR (%)
–0.10
1.0V ERROR (%)
–0.15
Data Format
Twos Complement
Twos Complement
Offset Binary
Offset Binary
Duty Cycle
Stabilizer
Disabled
Enabled
Enabled
Disabled
–0.20
EVALUATION BOARD
–0.25
0
0.5
1.0
1.5
LOAD (mA)
2.0
2.5
3.0
03066-0-019
Figure 35. VREF Accuracy vs. Load
External Reference Operation
The use of an external reference can be necessary to enhance
the gain accuracy of the ADC or to improve thermal drift
characteristics. When multiple ADCs track one another, a
single reference (internal or external) can be necessary to
reduce gain matching errors to an acceptable level. Figure 36
shows the typical drift characteristics of the internal reference
in both 1.0 V and 0.5 V modes.
When the SENSE pin is tied to AVDD, the internal reference is
disabled, allowing the use of an external reference. An internal
reference buffer loads the external reference with an equivalent
7 kΩ load. The internal buffer still generates the positive and
negative full-scale references, REFT and REFB, for the ADC
core. The input span is always twice the value of the reference
voltage; therefore, the external reference must be limited to a
maximum of 1.0 V.
1.0
0.9
0.8
VREF ERROR (%)
0.7
0.6
VREF = 1.0V
0.5
0.4
VREF = 0.5V
0.3
0.2
0.1
0
–40 –30 –20 –10
0
10
20
30
40
50
60
70
80
The AD9236 evaluation board provides all of the support
circuitry required to operate the ADC in its various modes and
configurations. Complete schematics and layout plots follow
and demonstrate the proper routing and grounding techniques
that should be applied at the system level.
It is critical that signal sources with very low phase noise (< 1 ps
rms jitter) be used to realize the ultimate performance of the
converter. Proper filtering of the input signal, to remove
harmonics and lower the integrated noise at the input, is also
necessary to achieve the specified noise performance.
TSSOP Evaluation Board
Figure 37 shows the typical bench setup used to evaluate the ac
performance of the AD9236. The AD9236 can be driven singleended or differentially through an AD8138 driver or a
transformer. Separate power pins are provided to isolate the
DUT from the support circuitry. Each input configuration can
be selected by proper connection of various jumpers (refer to
the schematics).
The AUXCLK input should be selected in applications requiring
the lowest jitter and SNR performance (that is, IF undersampling
characterization). It allows the user to apply a clock input signal
that is 4× the target sample rate of the AD9236. A low jitter,
differential divide-by-4 counter, the MC100LVEL33D, provides
a 1× clock output that is subsequently returned back to the CLK
input via JP9. For example, a 260 MHz signal (sinusoid) is
divided down to a 65 MHz signal for clocking the ADC. Note
that R1 must be removed with the AUXCLK interface. Lower
jitter is often achieved with this interface since many RF signal
generators display improved phase noise at higher output
frequencies and the slew rate of the sinusoidal output signal is
4× that of a 1× signal of equal amplitude.
TEMPERATURE (°C)
03066-0-011
Figure 36. Typical VREF Drift
Rev. B | Page 18 of 36
AD9236
LFCSP Evaluation Board
An alternative differential analog input path using an AD8351
op amp is included in the layout but is not populated in
production. Designers interested in evaluating the op amp with
the ADC should remove C15, R12, and R3 and populate the op
amp circuit. The passive network between the AD8351 outputs
and the AD9236 allows the user to optimize the frequency
response of the op amp for their application.
The typical bench setup used to evaluate the ac performance of
the AD9236 is similar to the TSSOP evaluation board
connections. The AD9236 can be driven single-ended or
differentially through a transformer. Separate power pins are
provided to isolate the DUT from the support circuitry. Each
input configuration can be selected by proper connection of
various jumpers (see Figure 48).
3V
–
REFIN
HP8644, 2V p-p
SIGNAL SYNTHESIZER
BAND-PASS
FILTER
3V
+
–
3V
+
–
3V
+
AVDD GND DUT GND DUT
S4
AVDD
DRVDD
XFMR
INPUT
AD9236
EVALUATION BOARD
10MHz
HP8644, 2V p-p
REFOUT CLOCK SYNTHESIZER
CLOCK
DIVIDER
S1
CLOCK
–
+
DVDD
J1
DATA
CAPTURE
AND
PROCESSING
03066-0-024
Figure 37. TSSOP Evaluation Board Connections
Rev. B | Page 19 of 36
TB1
TB1
TB1
TB1
TB1
AVDDIN
DRVDDIN
AGND
DVDDIN
TB1
Figure 38. TSSOP Evaluation Board Schematic, DUT
Rev. B | Page 20 of 36
6
4
5
1
3
2
C6
22μF
25V
25V
C48
22μF
25V
C47
22μF
25V
C58
22μF
4 RP4 22Ω 5
3 RP4 22Ω 6
2 RP4 22Ω 7
1 RP4 22Ω 8
4 RP3 22Ω 5
3 RP3 22Ω 6
2 RP3 22Ω 7
1 RP3 22Ω 8
AGND
DUTAVDDIN
D7O
D6O
D5O
D4O
D3O
D2O
D1O
D0O
L1
OTRO
D11O
D10O
D9O
D8O
C14
0.1μF
FBEAD
21
C53
0.1μF
FBEAD
2
C52
0.1μF
FBEAD
2
L4
L3
1
L2
1
C59
0.1μF
F
FBEAD
21
D7
D6
D5
D4
D3
D2
D1
D0
JP11
AVDD
DUTDRVDD
5kΩ
R27
JP13
DUTAVDD
OTR
D11
D10
D9
D8
AVDD
TP11 TP12 TP13 TP14
BLK
BLK BLK
BLK
TP4
RED
DVDD
TP9
TP10 TP15 TP16
BLK
BLK
BLK BLK
TP3
RED
JP12
AVDD
RED
TP1
RED
TP2
4 RP6 22Ω 5
3 RP6 22Ω 6
2 RP6 22Ω 7
1 RP6 22Ω 8
4 RP5 22Ω 5
3 RP5 22Ω 6
2 RP5 22Ω 7
22Ω
1 RP5
8
1kΩ
R42
1kΩ
R17
1kΩ
R20
R4
10kΩ
R3
10kΩ
JP2
JP1
JP6
JP7
C21
10μF
10V
C57
0.1μF
0.1μF
C35
C33
0.1μF
JP24
JP25
JP23
C23
10μF
10V
DUTAVDD
C32
0.1μF
C34
0.1μF
10V
10μF
C20
WHT
TP5
C38
0.1μF
C22
10μF
10V
JP22
C41
0.001μF
C50
0.1μF
VIN+
VIN–
WHT
TP17
0.001μF
C39
C36
0.1μF
DUTAVDD
AD9236
C1
10μF
10V
C37
0.1μF
AVDD
OTR
71
AGND
D11
8
28
SENSE
D10
3
27
VREF
D9
4
26
PDWN
D8
14
25
REFB
D7
22
5
REFT
D6
21
6
MODE U1 D5
2
20
VIN+
D4
19
9
VIN–
D3
10
18
AGND
D2
17
11
D1
AVDD
16
12
D0
DGND
15
23
DRVDD
CLK
24
13
TP6
C40
0.001μF
03066-0-007
DUTDRVDD
DUTCLK
WHT
OTRO
D0O
D1O
D2O
D3O
D4O
D5O
D6O
D7O
D8O
D9O
D10O
D11O
AD9236
1
Rev. B | Page 21 of 36
2
1
CLOCK
S1
R14
90Ω
R1
49.9Ω
JP9
C27
0.1μF
R11
49.9Ω
AUXCLK
R12
113Ω
AVDD
2
S5
0.1μF
C13
R19
500Ω
AVDD
CW
R2
10kΩ
C24
0.1μF
1
2
3
4
R18
500Ω
74VHC04
2
C26
0.1μF
5
Figure 39. TSSOP Evaluation Board Schematic, Clock Inputs and Output Buffering
9
11
13
3
6
1N5712
8
74VHC04
U8
10
74VHC04
U8
12
74VHC04
U8
4
74VHC04
U8
JP3
JP4
R7
22Ω
R26
10kΩ
D2
D1
C28
10μF
10V
74VHC04
U8
AVDD;14
AGND;7
AVDD
U3 DECOUPLING
AVDD
TP7 WHT
U8
1
R15
90Ω
R13
113Ω
AVDD
MC100LVEL33D
8
VCC
NC
7
OUT
U3 INA
6
REF
INB
5
VEE
INCOM
3
4
T2
2
1
5
6
T1-1T
R25
10kΩ
D0
D1
D2
D3
D4
D5
D6
D7
C10
0.1μF
OTR
D9
D10
D11
D8
G1
G2
2 A1
3 A2
4 A3
5 A4
6 A5
7 A6
8 A7
9 A8
1
19
U8 DECOUPLING
AVDD
R9
22Ω
DUTCLK
AVDD
C3
10μF
10V
16
15
14
13
12
11
18
17
G2
Y1
Y2
Y3
Y4
Y5
Y6
Y7
Y8
GND 10
U7
74VHC541
VCC
20
C5
10μF
10V
2
1
C11
0.1μF
15
14
13
12
11
18
17
16
DVDD
C4
10μF
1
G1
2 A1
3 A2
4 A3
5 A4
6 A5
7 A6
8 A7
9 A8
1
19
10V
2
VCC 20
10
GND
U6
74VHC541
Y1
Y2
Y3
Y4
Y5
Y6
Y7
Y8
C12
0.1μF
8
7
6
5
4
3
2
1
8
7
6
5
4
3
2
1
22Ω
22Ω
22Ω 12
RP1
RP1
RP1
22Ω 11
22Ω 10
22Ω 9
RP2
RP2
RP2
12
22Ω
RP2
14
15
22Ω 13
RP2
9
16
RP2
22Ω
RP2
22Ω
22Ω
RP2
22Ω
22Ω 10
RP1
RP1
22Ω 11
RP1
13
14
15
22Ω
RP1
16
22Ω
RP1
DACLK
DOTR
DD0
DD1
DD2
DD3
DD4
DD5
DD6
DD7
DD8
DD9
DD10
DD11
13
15
17
19
21
23
25
27
29
31
33
35
37
39
11
1
3
5
7
9
HEADER RIGHT ANGLE MALE NO EJECTORS
J1
03066-0-008
14
16
18
20
22
24
26
28
30
32
34
36
38
40
10
12
2
4
6
8
AD9236
1N5712
Rev. B | Page 22 of 36
2
AMP INPUT
1
S2
Figure 40. TSSOP Evaluation Board Schematic, Analog Inputs
12
R31
49.9Ω
C19
10μF
1
10V
C18
0.1μF
2
B
JP8
A
R35
499Ω
R34
523Ω
3
C2
R37
499Ω
VAL
C69
0.33μF
C15
10μF
10V
1 +
2
ALT VEE
TP8
RED
3
6
VAL
C17
R36
499Ω
VCC 4
VO+
–IN
2
1
VOC
U2
8 +IN VO–
AD8138
VEE 5
AVDD
C8
0.1μF
1
R10
40Ω
R6
40Ω
2
XFMR INPUT
1
S4
2
R33
1kΩ
R32
1kΩ
AVDD
S3
SINGLE INPUT
R24
49.9Ω
VAL
VAL
C9
0.33μF
R5
49.9Ω
JP5
4
5
6
C45
C42
T1
T1-1T
R41
1kΩ
R23
1kΩ
3
2
1
AVDD
C7
0.1μF
R8
1kΩ
R16
1kΩ
AVDD
C25
0.33μF
C16
0.1μF
JP43
JP41
JP46
JP45
JP40
JP42
03066-A-009
R22
33Ω
R21
33Ω
C43
DNP
C44B
20pF
C44
DNP
VIN–
VIN+
AD9236
AD9236
DACLK
DD0
DD1
DD2
DD3
DD4
DD5
DD6
DD7
DD8
DD9
DD10
DD11
1
2
3
4
5
6
7
8
9
10
11
12
13
14
MSB–DB11
CLOCK
DB10
DVDD
DB19
DCOM
DB8 AD9762 NC3
DB7
AVDD
DB6
COMP2
DB5
IOUTA
U4
DB4
IOUTB
DB3
ACOM
DB2
COMP1
DB1
FSADJ
DB0
REFIO
NC1
REFLO
SLEEP
NC2
28
27
26
25
24
23
22
21
20
19
18
17
16
15
DVDD
C30
0.1μF
C31
0.01μF
C29
0.1μF
C46
0.01μF
TP18
WHT
C56
0.01μF
S6
R29
49.9Ω
C51
0.01μF
R30
2kΩ
C49
0.01μF
C55
22pF
R28
49.9Ω
C54
22pF
03066-0-010
Figure 41. TSSOP Evaluation Board Schematic, Optional D/A Converter
03066-0-025
Figure 42. TSSOP Evaluation Board Layout, Primary Side
Rev. B | Page 23 of 36
AD9236
03066-0-026
Figure 43. TSSOP Evaluation Board Layout, Secondary Side
03066-0-027
Figure 44. TSSOP Evaluation Board Layout, Ground Plane
Rev. B | Page 24 of 36
AD9236
03066-0-028
Figure 45. TSSOP Evaluation Board Layout, Power Plane
03066-0-029
Figure 46. TSSOP Evaluation Board Layout, Primary Silkscreen
Rev. B | Page 25 of 36
AD9236
03066-0-030
Figure 47. TSSOP Evaluation Board Layout, Secondary Silkscreen
Rev. B | Page 26 of 36
Rev. B | Page 27 of 36
GND
J1
GND
0.1μF
C12
Figure 48. LFCSP Evaluation Board Schematic, Analog Inputs and DUT
PRI SEC
GND
C9
0.10μF
GND
R11
36Ω
R12
0.1μF
C11
C18
0.10μF
GND
R2
XX
AVDD
GND
D
R36
1kΩ
R15
33Ω
R13
1kΩ
C23
10pF
GND
OR L1
FOR FILTER
AVDD
C19
20pF
P4
P3
R25
1kΩ
GND
AVDD
3
2
4
P1
AVDD
GND
VIN+
VIN–
R6
1kΩ
R7
1kΩ
R5
1kΩ
GND
GND
GND
C21
10pF
R26
1kΩ
C22
10μF
GND GND
R4
33Ω
R SINGLE ENDED
R3
0Ω
C5
0.1μF
C26
10pF
GND
R18
25Ω
AMPINB
R3, R16, C18
ONLY ONE SHOULD BE
ON BOARD AT A TIME
XOUTB
GND
C16
0.1μF
R10
36Ω
E 45
XOUT
AMPIN
p10
C
C29
10μF
E
C7
0.1μF
R12, R42, C17
ONLY ONE SHOULD BE
ON BOARD AT A TIME.
6
2 CT
4
PRI SEC
OPTIONAL XFR
T2
FT C1–1–13
5
1
XOUT
X FRIN
2
CT
3
4
GND
XOUTB
C15
AMP 0.1μF
R42
0Ω
T1
ADT1–1WT
XFRIN1 1
5
NC
3
GND
L1 10nH
C6
0.1μF
FOR SINGLE ENDED INPUT
PLACE R18, R19, R42, C6, AND C18.
REMOVE R3, R12, C15, C17, AND C27.
GND
R9
10kΩ
P7 A B
AVDD
C13
GND
0.10μF
P11
P9 P8
REFB
GND
26
REFT
27
AVDD
28
AGND
29
VIN+
30
VIN–
31
AGND
32
AVDD
25
MODE
2
P5
24
R1
10kΩ
23
22
21
P6
GND
C8
0.1μF
U4
AD9236
AVDD
P14
CLK
P13
R8
1kΩ
12
11
10
9
6
7
8
GND
MODE PIN SOLDERABLE JUMPER:
5 TO 1: TWOS COMPLEMENT/DCS OFF
5 TO 2: TWOS COMPLEMENT/DCS ON
5 TO 3: OFFSET BINARY/DCS ON
5 TO 4: OFFSET BINARY/DCS OFF
03066-A-050
D0X
10
9
7
8
SENSE PIN SOLDERABLE JUMPER:
E TO A: EXTERNAL VOLTAGE DIVIDER
E TO B: INTERNAL 1V REFERENCE (DEFAULT)
E TO C: EXTERNAL REFERENCE
E TO D: INTERNAL 0.5V REFERENCE
D2X
D1X
11
12
5
D4X
D3X
D6X
D5X
D8X
D7X
D9X
D10X
D12X
D11X
DRX
D13X
6
13
15
14
4
RP1 220Ω
13
4
5
16
15
14
2
3
1
16
1
RP2 220Ω
H4
MTHOLE6
H3
MTHOLE6
H2
MTHOLE6
H1
MTHOLE6
10
D2 9
(LSB)
DRVDD
GND
(MSB)
6
5
GND
2
3
12
13
15
14
16
3
GND
OVERRANGE BIT
4
GND
2.5V DRVDD
2
P2
11
DRVDD
DGND
D7
D6
D5
D4
D3
1
AVDD
3.0V
1
20
19
VDL
2.5V
AVDD
18
17
VREF
SENSE
MODE
OTR
D11
D10
D9
D8
1
DNC
2
CLK
3
DNC
4
PDWN
5
DNC
6
DNC
7 D0
8
D1
VAMP
5.0V
EXTREF
1V MAX E1
AD9236
LSB
MSB
Rev. B | Page 28 of 36
IN
2DB
26
2D7
27
GND
28
2D6
29
2D5
30
V
31 CC
2D4
32
2D3
33
GND
34
2D2
35
2D1
36
1D8
37
1D7
38
GND
39
1D6
40
1D5
41
VCC
42
1D4
43
1D3
44
GND
45
1D2
46
1D1
47
1CLK
48
25
2CLK
1
U1
Figure 49. LFCSP Evaluation Board Schematic, Digital Path
GND
R19
50Ω
AMP
AMP IN
R41
10kΩ
VAMP
CC
24
7
6
GND
GND
DRVDD
GND
GND
DRVDD
GND
GND
C35
0.10μF
R33 RPG2 5
25Ω
6 COMM
7 OPLO
INLO 4
10 VOCM
9 VPOS
8 OPHI
R34
1.2kΩ
U3
AD8351
C44
0.1μF
R38
1kΩ
INHI 3
PWDN 1
RGP1 2
DRY
VAMP
GND
TO USE AMPLIFIER
PLACE ALL COMPONENTS
SHOWN HERE (RIGHT)
EXCEPT R40 OR R41.
REMOVE R12, R3, R18, R42, C6,
C15, AND C18.
OUT
1Q3
5
GND
4
1Q2
3
1Q1
2
1OE
1
1Q4
C28
0.1μF
R35
25Ω
GND
2OE
2QB
23
2Q7
22
GND
21
2Q6
20
2Q5
19
VCC
18
2Q4
17
2Q3
16
GND
15
2Q2
14
2Q1
13
1Q8
12
1Q7
11
GND
10
1Q6
9
1Q5
8
V
R40
10kΩ
GND
POWER DOWN
USE R40 OR R41
CLKLAT/DAC
GND
D0X
D2X
D1X
DRVDD
D4X
D3X
D5X
GND
D7X
D6X
GND
D8X
D10X
D9X
D11X
DRVDD
GND
D12X
DRX
D13X
CLKAT/DAC
74LVTH162374
GND
R14
25Ω
VAMP
R39
1kΩ
C45
0.1μF
C24
10μF
R17
0Ω
R16
0Ω
GND
GND
GND
MSB
C17
0.1μF
C27
0.1μF
GND
DRY
GND
DR
GND
AMPIN
AMPINB
7
3
5
1
9
9
11
11
13
13
15
15
17
17
19
19
21
21
23
23
25
25
27
27
29
29
31
31
33
33
35
35
37
37
39
39
7
8
10
1
3
5
P12
HEADER 40
4
6
2
12
14
14
16
16
18
18
20
20
22
22
24
24
26
26
28
28
30
30
32
32
34
34
36
36
38
38
40
40
12
10
4
6
8
2
03066-A-051
GND
AD9236
C4
10μF
GND
Figure 50. LFCSP Evaluation Board Schematic, Clock Input
Rev. B | Page 29 of 36
J2
GND
R29
50Ω
C43
0.1μF
ENC
ENCX
GND
R30
1kΩ
R31
1kΩ
VDL
R27
0Ω
R28
0Ω
VDL
VDL
E43
E44
E35
E51
E52
VDL
E31
VDL
E50
CLK
ENC
C33
C14
0.1μF 0.001μF
ANALOG BYPASSING
C32
0.001μF
CLOCK TIMING ADJUSTMENTS
GND
ENCODE
C25
10μF
GND
AVDD
FOR A BUFFERED ENCODE USE R28
FOR A DIRECT ENCODE USE R27
AVDD
C3
10μF
DUT BYPASSING
C10
10μF
VDL DRVDD
R20
1kΩ
GND
GND
R24
1kΩ
GND
R21
1kΩ
GND
E53
GND
R32
1kΩ
C41
0.1μF
DRVDD
C30
0.001μF
5
9
10
12
13
3A
3B
4A
4B
2B
1 1A
2 1B
4 2A
C31
0.1μF
U5
4Y
3Y
2Y
1Y
74VCX86
DIGITAL BYPASSING
C2
10μF
PWR
GND
14
8
11
6
7
3
C34
0.1μF
VDL
GND
ENCX
C36
0.1μF
R23
0Ω
C1
C39
0.001μF 0.1μF
CLKLAT/DAC
R37
25Ω
Rx
DNP
DR
VDL
R22
0Ω
GND
C49
0.001μF
LATCH BYPASSING
C47
0.1μF
SCHEMATIC SHOWS TWO GATE DELAY SETUP.
FOR ONE DELAY, REMOVE R22 AND R37 AND
ATTACH Rx (Rx = 0Ω).
C38
0.001μF
C48
0.001μF
GND
VAMP
C20
10μF
C40
0.001μF
03066-A-052
C46
10μF
C37
0.1μF
AD9236
AD9236
03066-0-055
03066-0-053
Figure 53. LFCSP Evaluation Board Layout, Ground Plane
Figure 51. LFCSP Evaluation Board Layout, Primary Side
03066-0-056
03066-0-054
Figure 54. LFCSP Evaluation Board Layout, Power Plane
Figure 52. LFCSP Evaluation Board Layout, Secondary Side
Rev. B | Page 30 of 36
AD9236
03066-0-057
03066-0-058
Figure 55. LFCSP Evaluation Board Layout, Primary Silkscreen
Figure 56. LFCSP Evaluation Board Layout, Secondary Silkscreen
Rev. B | Page 31 of 36
AD9236
Table 12. LFCSP Evaluation Board Bill of Materials
Item Qty. Omit 1 Reference Designator
1
18
C1, C5, C7, C8, C9, C11, C12,
C13, C15, C16, C31, C33, C34,
C36, C37, C41, C43, C47
8
C6, C18, C27, C17, C28,
C35, C45, C44
2
8
C2, C3, C4, C10, C20,
C22, C25, C29
2
C46, C24
Device
Chip Capacitors
Package
0603
Value
0.1 μF
Tantalum Capacitors
TAJC
10 μF
3
8
4
1
5
1
2
6
9
C14, C30, C32, C38,
C39, C40, C48, C49
C19
Chip Capacitors
0603
0.001 μF
Chip Capacitor
0603
20 pF
C26
Chip Capacitors
0603
10 pF
Headers
EHOLE
Jumper Blocks
E1, E45
7
2
J1, J2
SMA Connectors/50 Ω
SMA
8
1
L1
Inductor
0603
9
1
P2
Terminal Block
TB6
10
1
P12
Header Dual 20-Pin RT Angle
HEADER40
11
5
R3, R12, R23, R28, RX
Chip Resistors
0603
0Ω
12
2
R4, R15
Chip Resistors
0603
33 Ω
13
14
Chip Resistors
0603
1 kΩ
14
2
R5, R6, R7, R8, R13, R20,
R21, R24, R25, R26, R30,
R31, R32, R36
R10, R11
Chip Resistors
0603
36 Ω
15
1
R29
Chip Resistor
0603
50 Ω
220 Ω
6
1
Supplied
by ADI
C21, C23
E31, E35, E43, E44,
E50, E51, E52, E53
2
Recommended
Vendor/Part No.
10 nH
Coilcraft/0603CS10NXGBU
Wieland/25.602.2653.0,
z5-530-0625-0
Digi-Key S2131-20-ND
R37, R22, R42, R16, R17, R27
R19
16
2
RP1, RP2
Resistor Pack
R_742
17
1
T1
ADT1-1WT
AWT1-1T
18
1
U1
74LVTH162374 CMOS Register TSSOP-48
19
1
U4
AD9236BCP ADC (DUT)
CSP-32
Digi-Key
CTS/742C163221JTR
Mini-Circuits
Analog Devices, Inc.
X
20
1
U5
74VCX86M
SOIC-14
Fairchild
21
1
PCB
AD92XXBCP/PCB
PCB
Analog Devices, Inc.
X
X
22
1
U3
AD8351 Op Amp
MSOP-8
Analog Devices, Inc.
23
1
T2
M/A-COM Transformer
ETC1-1-13 1-1 TX
M/A-COM/ETC1-1-13
24
5
R9, R1, R2, R38, R39
Chip Resistors
0603
SELECT
25
4
R18, R14, R33, R35
Chip Resistors
0603
25 Ω
26
2
R40, R41
Chip Resistors
0603
10 kΩ
27
1
R34
Chip Resistor
Total 81
35
1
These items are included in the PCB design, but are omitted at assembly.
Rev. B | Page 32 of 36
1.2 kΩ
AD9236
OUTLINE DIMENSIONS
9.80
9.70
9.60
28
15
4.50
4.40
4.30
6.40 BSC
1
14
PIN 1
0.65
BSC
1.20 MAX
0.15
0.05
0.30
0.19
COPLANARITY
0.10
SEATING
PLANE
8°
0°
0.20
0.09
0.75
0.60
0.45
COMPLIANT TO JEDEC STANDARDS MO-153-AE
Figure 57. 28-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-28)
Dimensions shown in millimeters
0.60 MAX
5.00
BSC SQ
0.60 MAX
PIN 1
INDICATOR
TOP
VIEW
0.50
BSC
4.75
BSC SQ
0.50
0.40
0.30
12° MAX
1.00
0.85
0.80
PIN 1
INDICATOR
25
24
32
17
16
9
8
0.25 MIN
3.50 REF
0.80 MAX
0.65 TYP
0.30
0.23
0.18
3.25
3.10 SQ
2.95
EXPOSED
PAD
(BOTTOM VIEW)
0.05 MAX
0.02 NOM
SEATING
PLANE
1
0.20 REF
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VHHD-2
Figure 58. 32-Lead Frame Chip Scale Package [LFCSP_VQ]
5 mm × 5 mm Body, Very Thin Quad
(CP-32-2)
Dimensions shown in millimeters
Rev. B | Page 33 of 36
AD9236
ORDERING GUIDE
Model
AD9236BRU-80
AD9236BRURL7-80
AD9236BRUZ-80 1
AD9236BRUZRL7-801
AD9236BCP-80 2
AD9236BCPRL7-802
AD9236BCPZ-801, 2
AD9236BCPZRL7-801, 2
AD9236BRU-80EB
AD9236BCP-80EB2
1
2
Temperature Range
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Package Description
28-Lead Thin Shrink Small Outline Package (TSSOP)
28-Lead Thin Shrink Small Outline Package (TSSOP)
28-Lead Thin Shrink Small Outline Package (TSSOP)
28-Lead Thin Shrink Small Outline Package (TSSOP)
32-Lead Lead Frame Chip Scale (LFCSP_VQ)
32-Lead Lead Frame Chip Scale (LFCSP_VQ)
32-Lead Lead Frame Chip Scale (LFCSP_VQ)
32-Lead Lead Frame Chip Scale (LFCSP_VQ)
TSSOP Evaluation Board
LFCSP Evaluation Board
Package Option
RU-28
RU-28
RU-28
RU-28
CP-32-2
CP-32-2
CP-32-2
CP-32-2
Z = Pb-free part.
It is recommended that the exposed paddle be soldered to the ground plane for the LFCSP. There is an increased reliability of the solder joints, and the maximum
thermal capability of the package is achieved with the exposed paddle soldered to the customer board.
Rev. B | Page 34 of 36
AD9236
NOTES
Rev. B | Page 35 of 36
AD9236
NOTES
© 2006 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C03066-0-1/06(B)
Rev. B | Page 36 of 36