TI LM25117QPSQ/NOPB Wide input range synchronous buck controller with analog current monitor Datasheet

LM25117
LM25117-Q1
www.ti.com
SNVS714D – APRIL 2011 – REVISED APRIL 2012
Wide Input Range Synchronous Buck Controller with Analog Current Monitor
Check for Samples: LM25117, LM25117-Q1
FEATURES
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LM25117Q is an Automotive Grade product
that is AEC-Q100 grade 1 qualified (-40°C to
+125°C operating junction temperature)
Emulated peak current mode control
Wide operating range from 4.5V to 42V
Robust 3.3A peak gate drives
Adaptive dead-time output driver control
Free-run or synchronizable clock up to 750kHz
Optional diode emulation mode
Programmable output from 0.8V
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Precision 1.5% voltage reference
Analog current monitor
Programmable current limit
Hiccup mode over current protection
Programmable soft-start and tracking
Programmable line under-voltage lockout
Programmable switch-over to external bias
supply
Thermal shutdown
DESCRIPTION
The LM25117 is a synchronous buck controller intended for step-down regulator applications from a high voltage
or widely varying input supply. The control method is based upon current mode control utilizing an emulated
current ramp. Current mode control provides inherent line feed-forward, cycle-by-cycle current limiting and ease
of loop compensation. The use of an emulated control ramp reduces noise sensitivity of the pulse-width
modulation circuit, allowing reliable control of very small duty cycles necessary in high input voltage applications.
The operating frequency is programmable from 50 kHz to 750 kHz. The LM25117 drives external high-side and
low-side NMOS power switches with adaptive dead-time control. A user-selectable diode emulation mode
enables discontinuous mode operation for improved efficiency at light load conditions. A high voltage bias
regulator that allows external bias supply further improves efficiency. The LM25117’s unique analog telemetry
feature provides average output current information. Additional features include thermal shutdown, frequency
synchronization, hiccup mode current limit and adjustable line under-voltage lockout.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2011–2012, Texas Instruments Incorporated
LM25117
LM25117-Q1
SNVS714D – APRIL 2011 – REVISED APRIL 2012
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Typical Application
VIN
UVLO
SW
VIN
DEMB
VCC
HB
RAMP
LM25117
VOUT
HO
VOUT
VCCDIS
SW
COMP
LO
FB
CS
CM
RT
CSG
RES
SS AGND PGND
2
19
HB
RES
3
18
HO
SS
4
17
SW
RT
5
16
VCC
15
LO
HO
DEMB
HB
VIN
NC
20
VIN
1
NC
UVLO
UVLO
Connection Diagram
24
23
22
21
20
19
DEMB
1
18
SW
RES
2
17
NC
SS
3
16
VCC
EP
PGND
FB
8
13
CSG
COMP
9
12
CS
CM
10
11
RAMP
RT
4
15
LO
AGND
5
14
PGND
NC
6
13
CSG
VCCDIS
7
Figure 1. Top View
20-Lead HTSSOP EP
2
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8
9
10
11
12
CS
14
RAMP
7
CM
VCCDIS
EP
COMP
6
FB
AGND
Figure 2. Top View
WQFN-24 (4mmx4mm)
Copyright © 2011–2012, Texas Instruments Incorporated
Product Folder Links: LM25117 LM25117-Q1
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LM25117-Q1
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SNVS714D – APRIL 2011 – REVISED APRIL 2012
PIN DESCRIPTIONS
HTSSOP
Pin
WQFN
Pin
Name
Description
1
24
UVLO
Under-voltage lockout programming pin. If the UVLO pin voltage is below 0.4V, the regulator is in the
shutdown mode with all functions disabled. If the UVLO pin voltage is greater than 0.4V and less than
1.25V, the regulator is in standby mode with the VCC regulator operational, the SS pin grounded, and no
switching at the HO and LO outputs. If the UVLO pin voltage is above 1.25V, the SS pin is allowed to
ramp and pulse width modulated gate drive signals are delivered to the HO and LO pins. A 20μA current
source is enabled when UVLO exceeds 1.25V and flows through the external UVLO resistors to provide
hysteresis.
2
1
DEMB
Optional logic input that enables diode emulation when in the low state. In diode emulation mode, the
low-side NMOS is latched off for the remainder of the PWM cycle after detecting reverse current flow
(current flow from output to ground through the low-side NMOS). When DEMB is high, diode emulation
is disabled allowing current to flow in either direction through the low-side NMOS. A 50kΩ pull-down
resistor internal to the LM25117 holds DEMB pin low and enables diode emulation if the pin is left
floating.
3
2
RES
The restart timer pin that configures the hiccup mode current limiting. A capacitor on the RES pin
determines the time the controller remains off before automatically restarting. The hiccup mode
commences when the controller experiences 256 consecutive PWM cycles of cycle-by-cycle current
limiting. After this occurs, an 10μA current source charges the RES pin capacitor to the 1.25V threshold
and restarts LM25117.
4
3
SS
An external capacitor and an internal 10μA current source set the ramp rate of the error amplifier
reference during soft-start. The SS pin is held low when VCC< 4V, UVLO < 1.25V or during thermal
shutdown.
5
4
RT
The internal oscillator is programmed with a single resistor between RT and the AGND. The
recommended maximum oscillator frequency is 750kHz. The internal oscillator can be synchronized to
an external clock by coupling a positive pulse into the RT pin through a small coupling capacitor.
6
5
AGND
7
7
VCCDIS
Optional input that disables the internal VCC regulator. If VCCDIS>1.25V, the internal VCC regulator is
disabled. VCCDIS has an internal 500kΩ pull-down resistor to enable the VCC regulator when the pin is
left floating. The internal 500kΩ pull-down resistor can be overridden by pulling VCCDIS above 1.25V
with a resistor divider connected to an external bias supply.
8
8
FB
Feedback. Inverting input of the internal error amplifier. A resistor divider from the output to this pin sets
the output voltage level. The regulation threshold at the FB pin is 0.8V.
9
9
COMP
Output of the internal error amplifier. The loop compensation network should be connected between this
pin and the FB pin.
10
10
CM
Current monitor output. Average of the sensed inductor current is provided. Monitor directly between CM
and AGND. CM should be left floating when the pin is not used.
11
11
RAMP
PWM ramp signal. An external resistor and capacitor connected between the SW pin, the RAMP pin and
the AGND pin sets the PWM ramp slope. Proper selection of component values produces a RAMP
signal that emulates the AC component of the inductor with a slope proportional to input supply voltage.
12
12
CS
13
13
CSG
Kelvin ground connection to the current sense resistor. Connect directly to the low-side of the current
sense resistor.
14
14
PGND
Power ground return pin for low-side NMOS gate driver. Connect directly to the low-side of the current
sense resistor.
15
15
LO
Low-side NMOS gate drive output. Connect to the gate of the low-side synchronous NMOS transistor
through a short, low inductance path.
16
16
VCC
Bias supply pin. Locally decouple to PGND using a low ESR/ESL capacitor located as close to controller
as possible.
17
18
SW
Switching node of the buck regulator. Connect to the bootstrap capacitor, the source terminal of the
high-side NMOS transistor and the drain terminal of the low-side NMOS through a short, low inductance
path.
18
19
HO
High-side NMOS gate drive output. Connect to the gate of the high-side NMOS transistor through a
short, low inductance path.
19
20
HB
High-side driver supply for the bootstrap gate drive. Connect to the cathode of the external bootstrap
diode and to the bootstrap capacitor. The bootstrap capacitor supplies current to charge the high-side
NMOS gate and should be placed as close to controller as possible.
20
22
VIN
Supply voltage input source for the VCC regulator.
EP
EP
EP
Exposed pad of the package. Electrically isolated. Should be soldered to the ground plane to reduce
thermal resistance.
6
NC
No electrical contact.
Analog ground. Return for the internal 0.8V voltage reference and analog circuits.
Current sense amplifier input. Connect to the high-side of the current sense resistor.
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PIN DESCRIPTIONS (continued)
HTSSOP
Pin
WQFN
Pin
Name
Description
17
NC
No electrical contact.
21
NC
No electrical contact.
23
NC
No electrical contact.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1)
VIN to AGND
-0.3 to 45V
SW to AGND
-3.0 to 45V
HB to SW
-0.3 to 15V
VCC to AGND (2)
-0.3 to 15V
HO to SW
-0.3 to HB+0.3V
LO to AGND
-0.3 to VCC+0.3V
FB, DEMB, RES, VCCDIS, UVLO to AGND
-0.3 to 15V
CM, COMP to AGND (3)
-0.3 to 7V
SS, RAMP, RT to AGND
-0.3 to 7V
CS, CSG, PGND, to AGND
-0.3 to 0.3V
ESD Rating HBM (4)
2kV
Storage Temperature
-55°C to +150°C
Junction Temperature
+150°C
(1)
(2)
(3)
(4)
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is intended to be functional, but does not ensure specific performance limits. For specifications and test
conditions see the Electrical Characteristics Table.
See Application Information when input supply voltage is less than the VCC voltage.
These pins are output pins. As such they are not specified to have an external voltage applied.
The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin.
Operating Ratings (1)
VIN (2)
4.5V to 42V
VCC
4.5V to 14V
HB to SW
4.5V to 14V
Junction Temperature
(1)
(2)
4
-40°C to +125°C
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is intended to be functional, but does not ensure specific performance limits. For specifications and test
conditions see the Electrical Characteristics Table.
Minimum VIN operating voltage is defined with VCC supplied by the internal HV startup regulator and no external load on VCC.
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Electrical Characteristics
Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature range of -40°C to
+125°C. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent
the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise specified, the
following conditions apply: VVIN = 24V, VVCCDIS = 0V, RT = 25kΩ, no load on LO & HO.
Symbol
Parameter
Test Conditions
Min
Typ
Max
Unit
VIN Supply
IBIAS
ISHUTDOWN
VIN Operating Current
(1)
VIN Shutdown Current
VSS = 0V
4.8
6.2
mA
VSS = 0V, VVCCDIS = 2V
0.4
0.55
mA
VSS = 0V, VUVLO = 0V
16
40
µA
7.6
8.2
V
V
VCC Regulator
VCC(REG)
VCC Regulation
No Load
VCC Dropout (VIN to VCC)
VVIN = 4.5V, No external load
0.05
0.14
VVIN = 5.0V, ICC = 20mA
0.4
0.5
VCC Sourcing Current Limit
IVCC
VCC Operating Current
VVCC = 0V
(1)
6.85
30
VSS = 0V, VVCCDIS = 2V
4.0
VSS = 0V, VVCCDIS = 2V, VVCC = 14V
VCC Under-voltage Threshold
VCC Rising
42
3.75
VCC Under-voltage Hysteresis
V
mA
5.0
mA
5.8
7.3
mA
4.0
4.15
V
0.2
V
VCC Disable
VCCDIS Threshold
VCCDIS Rising
1.22
VCCDIS Hysteresis
1.25
1.29
0.06
VCCDIS Input Current
VVCCDIS = 0V
VCCDIS Pull-down Resistance
V
V
-20
nA
500
kΩ
UVLO
UVLO Threshold
UVLO Rising
UVLO Hysteresis Current
VUVLO = 1.4V
UVLO Shutdown Threshold
UVLO Falling
1.22
1.25
1.29
V
15
20
25
µA
0.23
0.3
V
0.1
V
UVLO Shutdown Hysteresis
Soft Start
ISS
SS Current Source
VSS = 0V
7
SS Pull-down Resistance
10
12
µA
13
24
Ω
800
812
mV
Error Amplifier
VREF
FB Reference Voltage
Measured at FB, FB = COMP
FB Input Bias Current
VFB = 0.8V
788
VOH
COMP Output High Voltage
ISOURCE = 3mA
VOL
COMP Output Low Voltage
ISINK = 3mA
AOL
DC Gain
80
dB
fBW
Unity Gain Bandwidth
3
MHz
1
nA
2.8
V
0.26
V
PWM Comparator
tHO(OFF)
Forced HO Off-time
tON(MIN)
Minimum HO On-time
260
VVIN = 42V
COMP to PWM comparator offset
320
440
ns
100
ns
1.2
V
Oscillator
fSW1
Frequency 1
RT = 25kΩ
180
200
220
kHz
fSW2
Frequency 2
RT = 10kΩ
430
480
530
kHz
RT Output Voltage
(1)
1.25
RT Sync Positive Threshold
2.6
Sync Pulse Width
100
3.2
V
3.95
V
ns
Operating current does not include the current into the RT resistor.
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Electrical Characteristics (continued)
Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature range of -40°C to
+125°C. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent
the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise specified, the
following conditions apply: VVIN = 24V, VVCCDIS = 0V, RT = 25kΩ, no load on LO & HO.
Symbol
Parameter
Test Conditions
Min
Typ
Max
Unit
135
mV
Current Limit
VCS(TH)
Cycle-by-cycle Sense Voltage
Threshold
VRAMP = 0, CSG to CS
106
120
CS Input Bias Current
VCS = 0V
-100
-66
CSG Input Bias Current
VCSG = 0V
-100
-66
µA
Current Sense Amplifier Gain
10
V/V
Hiccup Mode Fault Timer
256
Cycles
µA
RES
IRES
RES Current Source
VRES
RES Threshold
10
RES Rising
1.22
µA
1.25
1.285
V
2.0
1.65
V
Diode Emulation
VIL
DEMB Input Low Threshold
VIH
DEMB Input High Threshold
2.95
2.5
V
SW Zero Cross Threshold
-5
mV
DEMB Input Pull-down Resistance
50
kΩ
Current Monitor
Current Monitor Amplifier Gain
CS to CM
Current Monitor Amplifier Gain
Drift over Temperature
17.5
20.5
23.5
-2
0
+2
%
25
120
mV
V
Zero Input Offset
V/V
HO Gate Driver
VOHH
HO High-state Voltage Drop
IHO = –100mA, VOHH = VHB - VHO
0.17
0.3
VOLH
HO Low-state Voltage Drop
IHO = 100mA, VOLH = VHO - VSW
0.1
0.2
HO Rise Time
C-load = 1000pF
(2)
HO Fall Time
C-load = 1000pF
(2)
IOHH
Peak HO Source Current
VHO = 0V, SW = 0V, HB = 7.6V
IOLH
Peak HO Sink Current
VHO = VHB = 7.6V
3.3
A
HB to SW Under-voltage
HB DC Bias Current
2.56
HB - SW = 7.6V
V
6
ns
5
ns
2.2
A
2.9
3.32
V
65
100
µA
V
LO Gate Driver
VOHL
LO High-state Voltage Drop
ILO = –100mA, VOHL = VCC-VLO
0.17
0.27
VOLL
LO Low-state Voltage Drop
ILO = 100mA, VOLL = VLO
0.1
0.2
LO Rise Time
C-load = 1000pF
(2)
6
ns
LO Fall Time
C-load = 1000pF
(2)
5
ns
IOHL
Peak LO Source Current
VLO = 0V
2.5
A
IOLL
Peak LO Sink Current
VLO = 7.6V
3.3
A
LO Fall to HO Rise Delay
No load
72
ns
HO Fall to LO Rise Delay
No load
71
ns
Thermal Shutdown
Temperature Rising
165
°C
V
Switching Characteristics
TDLH
Thermal
TSD
Thermal Shutdown Hysteresis
(2)
6
25
°C
θJA
Junction to Ambient
HTSSOP-20EP
40
°C/W
θJC
Junction to Case
HTSSOP-20EP
4
°C/W
High and low reference are 80% and 20% of the pulse amplitude respectively.
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Electrical Characteristics (continued)
Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature range of -40°C to
+125°C. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent
the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise specified, the
following conditions apply: VVIN = 24V, VVCCDIS = 0V, RT = 25kΩ, no load on LO & HO.
Symbol
Parameter
Test Conditions
Min
Typ
Max
Unit
θJA
Junction to Ambient
WQFN-24 (4mmx4mm)
40
°C/W
θJC
Junction to Case
WQFN-24 (4mmx4mm)
6
°C/W
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Typical Performance Characteristics
8
HO Peak Driver Current vsOutput Voltage
LO Peak Driver Current vs Output Voltage
Figure 3.
Figure 4.
Driver Dead Time vs VVCC
Driver Dead Time vs Temperature
Figure 5.
Figure 6.
Forced HO Off-time vs Temperature
Switching Frequency vs RT
Figure 7.
Figure 8.
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Typical Performance Characteristics (continued)
VVCC vs IVCC
VVCC vs VVIN
Figure 9.
Figure 10.
VCS(TH) vs Temperature
VREF vs Temperature
Figure 11.
Figure 12.
VVCC vs Temperature
Error Amp Gain and Phase vs Frequency
Figure 13.
Figure 14.
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Typical Performance Characteristics (continued)
10
VCM vs IOUT
VCM vs VCSG-CS
Figure 15.
Figure 16.
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SYNC
CSYNC
CSS
REF
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RES
RES
Current
COMP
FB
SS
RT
+
+
ERR
+ AMP
-
+
+
-
RESTART
TIMER
-
1.2V
+
STANDBY
REF
STANDBY
SS Current
OSCILLATOR/
SYNC DETECTOR
UVLO
Shutdown
Threshold
UVLO
UVLO
Threshold
UVLO Hysteresis
Current
LM25117
+
AGND
CLK
Q
Q
R
S
STANDBY
VCC OFF
HO_ENABLE
+
PGND
HICCUP MODE
FAULT TIMER
256 CYCLES
C/L
Comparator
RES RESET
HICCUP
10uVCS(TH)
+
500 k:
VCCDIS
Threshold
THERMAL
SHUTDOWN
PWM
Comparator
DE_ENABLE
CLK
STANDBY
VCC OFF
RES RESET
STANDBY
VCCDIS
+
50 k:
DE_ENABLE
DIODE
EMULATION
CONTROL
HB
UVLO
-5 mV
CONDITIONER
VCC
CSG
A=2
RAMP
40 k:
CM
CCM
RCM
RCS2
CCS
RCS1
CS
RGH
RGL
CHB
DHB
RS
CVCC
LO
SW
HO
HB
DEMB
VCC
UVLO
VCC
Current Monitor
Amplifier
Current Sense
Amplifier
A=10
LO Driver
+
-
HO Driver
DISABLE
STANDBY
ZCD
Comparator
2.0/2.5V
VCC Regulator
CVIN
LEVEL SHIFT/
ADAPTIVE
TIMER
VCC OFF
LO_ENABLE
HO_ENABLE
DE_ENABLE
VCC OFF
VIN
RVIN
QL
QH
CSNB
RSNB
CRAMP
RRAMP
COUT1
LO
RFB1
RFB2
COUT2
VOUT
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CRES
CHF
RT
RUV1
RUV2
RCOMP CCOMP
CFT
CIN
+
-
VIN
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BLOCK DIAGRAM AND TYPICAL APPLICATION CIRCUIT
Figure 17. Block Diagram and Typical Application Circuit
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FUNCTIONAL DESCRIPTION
The LM25117 high voltage switching controller features all of the functions necessary to implement an efficient
high voltage buck regulator that operates over a very wide input voltage range. This easy to use controller
integrates high-side and low-side NMOS drivers. The regulator control method is based upon peak current mode
control utilizing an emulated current ramp. Peak current mode control provides inherent line feed-forward, cycleby-cycle current limiting and ease of loop compensation. The use of an emulated control ramp reduces noise
sensitivity of the PWM circuit, allowing reliable processing of the very small duty cycles necessary in high input
voltage applications.
The switching frequency is user programmable up to 750 kHz. The RT pin allows the switching frequency to be
programmed by a single resistor or synchronized to an external clock. Fault protection features include cycle-bycycle and hiccup mode current limiting, thermal shutdown and remote shutdown capability by pulling down UVLO
pin. The UVLO input enables the regulator when the input voltage reaches a user selected threshold and
provides a very low quiescent shutdown current when pulled low. A unique analog telemetry feature provides
averaged output current information, allowing various applications that need either a current monitor or current
control. The functional block diagram and typical application circuit of the LM25117 are shown in Figure 17
The device is available in HTSSOP-20EP and WQFN-24 package featuring an exposed pad to aid in thermal
dissipation.
High Voltage Startup Regulator and VCC Disable
The LM25117 contains an internal high voltage bias regulator that provides the VCC bias supply for the PWM
controller and NMOS gate drivers. The VIN pin can be connected to an input voltage source as high as 42V. The
output of the VCC regulator is set to 7.6V. When the input voltage is below the VCC set-point level, the VCC
output tracks the VIN with a small dropout voltage. The output of the VCC regulator is current limited at 30mA
minimum.
Upon power-up, the regulator sources current into the capacitor connected to the VCC pin. The recommended
capacitance range for the pin VCC is 0.47µF to 10µF. When the VCC pin voltage exceeds the VCC UV threshold
and the UVLO pin is greater than UVLO threshold, the HO and LO drivers are enabled and a soft-start sequence
begins. The HO and LO drivers remain enabled until either the VCC pin voltage falls below VCC UV threshold,
the UVLO pin voltage falls below UVLO threshold, hiccup mode is activated or the die temperature exceeds the
thermal shutdown threshold. Enabling/Disabling the IC by controlling UVLO is recommended in most of cases.
An output voltage derived bias supply can be applied to the VCC pin to reduce the controller power dissipation at
higher input voltage. The VCCDIS input can be used to disable the internal VCC regulator when external biasing
is supplied. The externally supplied bias should be coupled to the VCC pin through a diode, preferably a
Schottky diode. If the VCCDIS pin voltage exceeds the VCCDIS threshold, the internal VCC regulator is disabled.
VCCDIS has a 500kΩ internal pull-down resistor to ground for normal operation with no external bias.
The VCC regulator series pass transistor includes a diode between VCC (Anode) and VIN (Cathode) that should
not be forward biased in normal operation. If the voltage of the external bias supply is greater than the VIN pin
voltage, an external blocking diode is required from the input power supply to the VIN pin to prevent the external
bias supply from passing current to the input supply through VCC.
VIN
VIN
LM25117
External
VCC Supply
VCC
Figure 18. VIN Configuration for VVIN<VVCC
For VOUT between 5V and 14.5V, the output can be connected directly to VCC through a diode.
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VOUT
VCC
LM25117
VCCDIS resistor divider is required
when external VCC supplying voltage is
smaller than 8.5V
VCCDIS
Figure 19. External VCC Supply for 5V< VOUT<14.5V
For VOUT < 5V, a bias winding on the output inductor can be added to generate the external VCC supply voltage.
VCC
LM25117
VCCDIS
VOUT
SW
VCCDIS resistor divider is required
when external VCC supplying voltage is
smaller than 8.5V
Figure 20. External VCC Supply for VOUT <5V
For 14.5V <VOUT, the external supply voltage can be regulated by using a series Zener diode from the output to
VCC.
VOUT
R1
VCC
Zener
30 k:
LM25117
R1 is required to limit maximum reverse zener current
30 k: minimum resistive loss at VCC
guarantees minimum reverse zener current
Figure 21. External VCC Supply for 14.5V <VOUT
In high input voltage applications, extra care should be taken to ensure the VIN pin does not exceed the absolute
maximum voltage rating of 45V. During line or load transients, voltage ringing on the VIN that exceeds the
Absolute Maximum Rating can damage the IC. Both careful PC board layout and the use of quality bypass
capacitors located close to the VIN and AGND pin are essential. Adding an R-C filter (RVIN, CVIN) on VIN is
optional and helps to prevent faulty operation caused by poor PC board layout and high frequency switching
noise injection. The recommended capacitance and resistance range are 0.1µF to 10µF and 1Ω to 10Ω.
UVLO
The LM25117 contains a dual level UVLO (under-voltage lockout) circuit. When the UVLO is less than 0.4V, the
LM25117 is in shutdown mode. The shutdown comparator provides 100mV of hysteresis to avoid chatter during
transitions. When the UVLO pin voltage is greater than 0.4V but less than 1.25V, the controller is in standby
mode. In the standby mode, the VCC bias regulator is active but the HO and LO drivers are disabled and the SS
pin is held low. This feature allows the UVLO pin to be used as a remote shutdown function by pulling the UVLO
pin down below 0.4V with an external open collector or open drain device. When the VCC pin exceeds its undervoltage lockout threshold and the UVLO pin voltage is greater than 1.25V, the HO and LO drivers are enabled
and normal operation begins.
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UVLO Hysteresis
Current
VIN
UVLO
Threshold
RUV2
UVLO
CFT
RUV1
LM25117
UVLO
Shutdown
Threshold
+
+
STANDBY
SHUTDOWN
Figure 22. UVLO Configuration
The UVLO pin should not be left floating. An external UVLO set-point voltage divider from the VIN to AGND is
used to set the minimum input operating voltage of the regulator. The divider must be designed such that the
voltage at the UVLO pin is greater than 1.25V and never exceeds 15V when the input voltage is in the desired
operating range. If necessary, the UVLO pin can be clamped with a Zener diode.
UVLO hysteresis is accomplished with an internal 20μA current source that is switched on or off into the
impedance of the UVLO set-point divider. When the UVLO pin voltage exceeds the 1.25V threshold, the current
source is enabled to quickly raise the voltage at the UVLO pin. When the UVLO pin voltage falls below the 1.25V
threshold, the current source is disabled causing the voltage at the UVLO pin to quickly fall. The use of a CFT
capacitor in parallel with RUV1 helps to minimize switching noise injection into UVLO pin, but it may slow down
the falling speed of the UVLO pin when the 20μA current source is disabled. The recommended range for CFT is
10pF to 220pF.
The values of RUV1 and RUV2 can be determined from the following equations:
VHYS
[5]
RUV2 =
20 µA
RUV1 =
1.25V x RUV2
[5]
VIN(STARTUP) - 1.25V
(1)
(2)
Where VHYS is the desired UVLO hysteresis and VIN(STARTUP) is the desired startup voltage of the regulator during
turn-on.
Oscillator and Sync Capability
The LM25117 switching frequency is programmed by a single external resistor connected between the RT pin
and the AGND pin. The resistor should be located very close to the device and connected directly to the RT and
AGND pins. To set a desired switching frequency (fSW), the resistor value can be calculated from the following
equation:
9
RT =
5.2 x 10
- 948 [5]
fSW
(3)
The RT pin can be used to synchronize the internal oscillator to an external clock. The internal oscillator can be
synchronized by AC coupling a positive edge into the RT pin. The voltage at the RT pin is nominally 1.25V and
the voltage at the RT pin must exceed the RT Sync Positive Threshold to trip the internal synchronization pulse
detector. A 5V amplitude pulse signal coupled through 100pF capacitor is a good starting point. The frequency of
the external synchronization pulse is recommended to be within +/-10% of the frequency programmed by the RT
resistor but will operate to +100/-40% of the programmed frequency. Care should be taken to make sure that the
RT pin voltage does not go below -0.3V at the falling edge of the external pulse. This may limit the duty cycle of
external synchronization pulse.
The RT resistor is always required, whether the oscillator is free running or externally synchronized.
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Ramp Generator and Emulated Current Sense
The ramp signal used in the pulse width modulator for traditional current mode control is typically derived directly
from the high-side switch current. This switch current corresponds to the positive slope portion of the inductor
current. Using this signal for the PWM ramp simplifies the control loop transfer function to a single pole response
and provides inherent input voltage feed-forward compensation.
The disadvantage of using the high-side switch current signal for PWM control is the large leading edge spike
due to circuit parasitics that must be filtered or blanked. Minimum achievable pulse width is limited by the
filtering, blanking time and propagation delay with a high-side current sensing scheme.
In the applications where the input voltage may be relatively large in comparison to the output voltage, controlling
small pulse widths and duty cycles are necessary for regulation. The LM25117 utilizes a unique ramp generator
which does not actually measure the high-side switch current but rather reconstructs the signal. Representing or
emulating the inductor current provides a ramp signal to the PWM comparator that is free of leading edge spikes
and measurement or filtering delays, while maintaining the advantages of traditional peak current mode control.
The current reconstruction is comprised of two elements: a sample-and-hold DC level and the emulated inductor
current ramp as shown in Figure 23. The sample-and-hold DC level is derived from a measurement of the
recirculating current flowing through the current sense resistor. The voltage across the sense resistor is sampled
and held just prior to the onset of the next conduction interval of the high-side switch. The current sense amplifier
with a gain of 10 and sample-and-hold circuit provide the DC level of the reconstructed current signal as shown
in Figure 24.
Additional Slope
RAMPPK =
VIN x tON
RRAMP x CRAMP
ILO x RS x 10
Sample and Hold
DC Level
tON
Figure 23. Composition of Emulated Current Sense Signal
CS
10 x VCS(TH)
Current Sense
Amplifier
+
-
AS=10
+
Current Limit
Comparator
RS
IL
CSG
HO_ENABLE
LM25117
RAMP
SW
RRAMP
CRAMP
Figure 24. RAMP Generator and Current Limit Circuit
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The positive slope inductor current ramp is emulated by CRAMP connected between RAMP and AGND and RRAMP
connected between SW and RAMP. RRAMP should not be connected to VIN directly because the RAMP pin
absolute maximum voltage rating could be exceeded under high input voltage conditions. CRAMP is discharged by
an internal switch during the off-time and must be fully discharged during the minimum off-time. This limits the
ramp capacitor to be less than 2nF. A good quality, thermally stable ceramic capacitor is recommended for
CRAMP.
The selection of RRAMP and CRAMP can be simplified by adopting a K factor, which is defined as:
LO
K=
RRAMP x CRAMP x RS x AS
(4)
Where AS is the current sense amplifier gain which is normally 10. By choosing 1 as the K factor, the regulator
removes any error after one switching cycle and the design procedure is simplified. See Application Information
for detailed information.
Error Amplifier and PWM Comparator
The internal high-gain error amplifier generates an error signal proportional to the difference between the FB pin
voltage and the internal precision 0.8V reference. The output of error amplifier is connected to the COMP pin
allowing the user to provide Type 2 loop compensation components, RCOMP, CCOMP and optional CHF.
VOUT
LM25117
PWM
Comparator
REF
RFB2
FB
+
+
Error
Amplifier
RCOMP
CCOMP
-
+
-
COMP
RAMP Generator
Output
RFB1
CHF (optional)
Type 2 Compensation
Components
Figure 25. Feedback Configuration and PWM Comparator
RCOMP, CCOMP and CHF configure the error amplifier gain and phase characteristics to achieve a stable voltage
loop gain. This network creates a pole at DC (FP1), a mid-band zero (FZ) for phase boost, and a high frequency
pole (FP2). The recommended range of RCOMP is 2kΩ to 40kΩ. See Application Information for detailed
information.
FP1 = 0
FZ =
[Hz]
1
2S x RCOMP x CCOMP
(5)
[Hz]
(6)
1
[Hz]
FP2 =
CCOMP x CHF·
2S x RCOMP x §
©CCOMP + CHF¹
(7)
The PWM comparator compares the emulated current sense signal from Ramp Generator to the voltage at the
COMP pin through a 1.2V internal voltage drop and terminates the present cycle when the emulated current
sense signal is greater than VCOMP - 1.2V.
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Diode Emulation
A fully synchronous buck regulator implemented with a freewheeling NMOS rather than a diode has the
capability to sink current from the output in certain conditions such as light load, over-voltage or pre-bias startup.
The LM25117 provides a diode emulation feature that can be enabled to prevent reverse current flow in the lowside NMOS device. When configured for diode emulation, the low-side NMOS driver is disabled when SW pin
voltage is greater than -5mV during the off-time of the high-side NMOS driver, preventing reverse current flow.
A benefit of the diode emulation is lower power loss at no load or light load conditions. The negative effect of
diode emulation is degraded light load transient response.
The diode emulation feature is configured with the DEMB pin. To enable diode emulation, connect the DEMB pin
to GND or leave the pin floating. If continuous conduction operation is desired, the DEMB pin should be tied to a
voltage greater than 3V and may be connected to VCC. The LM25117 forces the regulator to operate in diode
emulation mode when SS pin voltage is less than the internal 0.8V reference, allowing for startup into a prebiased load with the continuous conduction configuration.
Soft-Start
The soft-start feature allows the regulator to gradually reach the steady state operating point, thus reducing
startup stresses and surges. The LM25117 regulates the FB pin to the SS pin voltage or the internal 0.8V
reference, whichever is lower. The internal 10µA soft-start current source gradually increases the voltage on an
external soft-start capacitor connected to the SS pin. This results in a gradual rise of the output voltage. Soft-start
time (tss) can be calculated from the following equation:
tSS =
CSS x 0.8V
10 µA
[sec]
(8)
The LM25117 can track the output of a master power supply during soft-start by connecting a voltage divider
from the output of master power supply to the SS pin. At the beginning of the soft-start sequence, VSS should be
allowed to go below 25mV by the internal SS pull-down switch. During soft-start period, when SS pin voltage is
less than 0.8V, the LM25117 forces diode emulation for startup into a pre-biased load. If the tracking feature is
desired, connect the DEMB pin to GND or leave the pin floating.
Cycle-by-Cycle Current Limit
The LM25117 contains a current limit monitoring scheme to protect the regulator from possible over-current
conditions as shown in Figure 24. If the emulated ramp signal exceeds 1.2V, the present cycle is terminated. For
the case where the switch current overshoots when the inductor is saturated or the output is shorted to ground,
the sample-and-hold circuit detects the excess recirculating current before the high-side NMOS driver is turned
on again. The high-side NMOS driver is disabled and will skip pulses until the current has decayed below the
current limit threshold. This approach prevents current runaway conditions since the inductor current is forced to
decay to a controlled level following any current overshoot. Maximum peak inductor current can be calculated as:
VCS(TH)
VOUT
[A]
IL(MAX)_PK =
+ IPP RS
fSW x AS x RS x RRAMP x CRAMP
(9)
IL(MAX)_AVE = IL(MAX)_PK -
IPP
2
[A]
(10)
Where IPP represents inductor peak to peak ripple current in Figure 26, and is defined as:
IPP =
VOUT
LO x fSW
§
©
x ¨1 -
VOUT
VIN
· [A]
¸
¹
(11)
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IPP
IOUT
T=
1
fSW
0
Figure 26. Inductor Current
During an output short condition, the worst case peak inductor current is limited to:
VIN(MAX) x tON(MIN)
VCS(TH)
ILIM_PK =
RS
+
LO
[A]
(12)
Where tON(MIN) is the minimum HO on-time.
In most of cases, especially if the output voltage is relatively high, it is recommended that a soft-saturating
inductor such as a powder core device is used. If a sharp-saturating inductor is used, the inductor saturation
level must be above ILIM_PK. The temperatures of the NMOS devices, RS and inductor should be checked under
this output short condition.
Hiccup Mode Current Limiting
To further protect the regulator during prolonged current limit conditions, LM25117 provides a hiccup mode
current limit. An internal hiccup mode fault timer counts the PWM clock cycles during which cycle-by-cycle
current limiting occurs. When the hiccup mode fault timer detects 256 consecutive cycles of current limiting, an
internal restart timer forces the controller to enter a low power dissipation standby mode and starts sourcing
10μA current into the RES pin capacitor CRES. In this standby mode, HO and LO outputs are disabled and the
soft-start capacitor CSS is discharged.
CRES is connected from RES pin to AGND and determines the time (tRES) in which the LM25117 remains in the
standby before automatically restarting. When the RES pin voltage exceeds the 1.25V RES threshold, RES
capacitor is discharged and a soft-start sequence begins. tRES can be calculated from the following equation:
CRES x 1.25V
[sec]
tRES =
10 PA
(13)
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Current Limit
Detected
1.25V RES Threshold
IRES = 10 µA
RES
0V
ISS = 10 µA
SS
0.8V REF
HO
LO
Current Limit
persists during 256
consecutive cycles
tRES
tSS
Figure 27. Hiccup Mode Current Limit Timing Diagram
LM25117
STANDBY
RES
Current
RESTART
TIMER
HICCUP
RES
CRES
HICCUP MODE
FAULT TIMER
256 CYCLES
Current Limit
Comparator
+
Figure 28. Hiccup Mode Current Limit Circuit
The RES pin can also be configured for latch-off mode current limiting or non-hiccup mode cycle-by-cycle current
limiting. If the RES pin is tied to VCC or a voltage greater than the RES threshold at initial power-on, the restart
timer is disabled and the regulator operates with non-hiccup mode cycle-by-cycle current limit. If the RES pin is
tied to GND, the regulator enters into the standby mode after 256 consecutive cycles of current limiting and then
never restarts until UVLO shutdown is cycled. The restart timer is configured during initial power-on when UVLO
is above the UVLO threshold and VCC is above the VCC UV threshold.
RES
LM25117
RES
CRES
LM25117
VCC
RES
LM25117
VCC
(a) Hiccup Mode
Current Limit
(b) Latch-off Mode
Current Limit
(c) Cycle-by-cycle
Current Limit
Figure 29. RES Configurations
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HO and LO Drivers
The LM25117 contains high current NMOS drivers and an associated high-side level shifter to drive the external
high-side NMOS device. This high-side gate driver works in conjunction with an external diode DHB, and
bootstrap capacitor CHB. A 0.1μF or larger ceramic capacitor, connected with short traces between the HB and
SW pin, is recommended. During the off-time of the high-side NMOS driver, the SW pin voltage is approximately
0V and the CHB is charged from VCC through the DHB. When operating with a high PWM duty cycle, the highside NMOS device is forced off each cycle for 320ns to ensure that CHB is recharged.
The LO and HO outputs are controlled with an adaptive dead-time methodology which insures that both outputs
are never enabled at the same time. When the controller commands HO to be enabled, the adaptive dead-time
logic first disables LO and waits for the LO voltage to drop. HO is then enabled after a small delay (LO Fall to HO
Rise Delay). Similarly, the LO turn-on is delayed until the HO voltage has discharged. LO is then enabled after a
small delay (HO Fall to LO Rise Delay). This technique insures adequate dead-time for any size NMOS device,
especially when VCC is supplied by a higher external voltage source. The adaptive dead-time circuitry monitors
the voltages of HO and LO outputs and insures the dead-time between the HO and LO outputs. Adding a gate
resister, RGH or RGL, may decrease the effective dead-time.
Care should be exercised in selecting an output NMOS device with the appropriate threshold voltage, especially
if VCC is supplied by an external bias supply voltage below the VCC regulation level. During startup at low input
voltages, the low-side NMOS device gate plateau voltage should be lower than the VCC under-voltage lockout
threshold. Otherwise, there may be insufficient VCC voltage to completely enhance the NMOS device as the
VCC under-voltage lockout is released during startup. If the high-side NMOS drive voltage is lower than the highside NMOS device gate plateau voltage during startup, the regulator may not start or it may hang up momentarily
in a high power dissipation state. This condition can be addressed by selecting an NMOS device with a lower
threshold voltage. This situation can be avoided if the minimum input voltage programmed by the UVLO resistor
is above the VCC regulation level.
Current Monitor
The LM25117 provides average output current information, enabling various applications requiring monitoring or
control of the output current.
Current Sense
Amplifier Output
LM25117
Current Monitor
Amplifier
CM
CONDITIONER
RCM
AM = 2
40 k:
CCM
Figure 30. Current Monitor
The average of CM output can be calculated by:
VCM_AVE = (IPEAK + IVALLEY) x RS x AS
[V
(14)
The current monitor output is only valid in continuous conduction operation. The current monitor has a limited
bandwidth of approximately one tenth of fSW. Adding an R-C filter, RCM and CCM, on the output of current monitor
with the cut off frequency below one tenth of fSW is recommended to attenuate sampling noise.
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Maximum Duty Cycle
When operating with a high PWM duty cycle, the high-side NMOS device is forced off each cycle for 320ns to
ensure that CHB is recharged and to allow time to sample and hold the current in the low-side NMOS FET. This
forced off-time limits the maximum duty cycle of the controller. When designing a regulator with high switching
frequency and high duty cycle requirements, a check should be made of the required maximum duty cycle
against the graph shown in Figure 31. The actual maximum duty cycle varies with the switching frequency as
follows:
Figure 31. Maximum Duty Cycle vs Switching Frequency
Thermal Protection
Internal thermal shutdown circuitry is provided to protect the controller in the event the maximum junction
temperature is exceeded. When activated, typically at 165°C, the controller is forced into a low power shutdown
mode, disabling the output drivers and the VCC regulator. This feature is designed to prevent overheating and
destroying the device.
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APPLICATION INFORMATION
FEEDBACK COMPENSATION
Open loop response of the regulator is defined as the product of modulator transfer function and feedback
transfer function. When plotted on a dB scale, the open loop gain is shown as the sum of modulator gain and
feedback gain.
The modulator transfer function includes a power stage transfer function with an embedded current loop and can
be simplified as one pole and one zero system as shown in the following equations.
s
1+
^
ZZ_ESR
VOUT
= AM x
^
V
§1+ s ·
COMP
©
ZP_LF ¹
(15)
RLOAD
Where AM (Modulator DC gain) =
,
R S x AS
(16)
1
,
ZZ_ESR (ESR zero) =
RESR x COUT
(17)
1
ZP_LF (Load pole) =
RLOAD x COUT
(18)
If the ESR of COUT (RESR) is very small, the modulator transfer function can be further simplified to a one pole
system and the voltage loop can be closed with only two loop compensation components, RCOMP and CCOMP,
leaving a single pole response at the crossover frequency. A single pole response at the crossover frequency
yields a very stable loop with 90 degrees of phase margin.
The feedback transfer function includes the feedback resistor divider and loop compensation of the error
amplifier. RCOMP, CCOMP and optional CHF configure the error amplifier gain and phase characteristics and create
a pole at origin, a low frequency zero and a high frequency pole. This is shown mathematically in the following
equations.
s
1+
^
ZZ_EA
VCOMP
= AFB x
s
^
s x (1+
V
OUT
Z )
P_EA
(19)
Where AFB (Feedback DC gain) =
1
,
RFB2 x (CCOMP + CHF)
wZ_EA (Low frequency zero) =
1
,
RCOMP x CCOMP
wP_EA (High frequency pole) =
1
RCOMP x CHF
(20)
The pole at the origin minimizes output steady state error. The low frequency zero should be placed to cancel the
load pole of the modulator. The high frequency pole can be used to cancel the zero created by the output
capacitor ESR or to decrease noise susceptibility of the error amplifier. By placing the low frequency zero an
order of magnitude less than the crossover frequency, the maximum amount of phase boost can be achieved at
the crossover frequency. The high frequency pole should be placed well beyond the crossover frequency since
the addition of CHF adds a pole in the feedback transfer function.
The crossover frequency (loop bandwidth) is usually selected between one twentieth and one fifth of the fSW. In a
simplified formula, the crossover frequency can be defined as:
fCROSS =
22
RCOMP
[Hz]
2 x ' x RS x RFB2 x AS x COUT
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(21)
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For higher crossover frequency, RCOMP can be increased, while proportionally decreasing CCOMP. Conversely,
decreasing RCOMP while proportionally increasing CCOMP, results in lower bandwidth while keeping the same zero
frequency in the feedback transfer function.
The sampled gain inductor pole is inversely proportional to the K factor, which is defined as:
fSW
Zp_HF = K - 0.5
(22)
The maximum achievable loop bandwidth, in fact, is limited by this sampled gain inductor pole. In traditional
current mode control, the maximum achievable loop bandwidth varies with input voltage. With the LM25117’s
unique slope compensation scheme, the sampled gain inductor pole is independent of changes to the input
voltage. This frees the user from additional concerns in wide varying input range applications and is an
advantage of the LM25117.
If the sampled gain inductor pole or the ESR zero is close to the crossover frequency, it is recommended that the
comprehensive formulas in Table 1 be used and the stability should be checked by a network analyzer. The
modulator transfer function can be measured and the feedback transfer function can be configured for the
desired open loop transfer function. If a network analyzer is not available, step load transient tests can be
performed to verify acceptable performance. The step load goal is minimum overshoot/undershoot with a
damped response.
SUB-HARMONIC OSCILLATION
Peak current mode regulators can exhibit unstable behavior when operating above 50% duty cycle. This
behavior is known as sub-harmonic oscillation and is characterized by alternating wide and narrow pulses at the
SW pin. Sub-harmonic oscillation can be prevented by adding an additional voltage ramp (slope compensation)
on top of the sensed inductor current shown in Figure 23. By choosing K≥1, the regulator will not be subject to
sub-harmonic oscillation caused by a varying input voltage.
In time-domain analysis, the steady-state inductor current starts and ends at the same value during one clock
cycle. If the magnitude of the end-of-cycle current error, dI1, caused by an initial perturbation, dI0, is less than the
magnitude of dI0 or dI1/dI0 > -1, the perturbation naturally disappears after a few cycles. When dI1/dI0 < -1, the
initial perturbation does not disappear, resulting in sub-harmonic oscillation in steady-state operation.
Steady-State
Inductor
Current
dI0
tON
dI1
Inductor Current with
Initial Perturbation
Figure 32. Effect of Initial Perturbation when dl1/dl0 < -1
dI1/dI0 can be calculated by:
dl1
1
=1dl0
K
(23)
The relationship between dI1/dI0 and K factor is illustrated graphically in Figure 33.
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Figure 33. dl1/dl0 vs K Factor
The minimum value of K is 0.5. When K<0.5, the amplitude of dI1 is greater than the amplitude of dI0 and any
initial perturbation results in sub-harmonic oscillation. If K=1, any initial perturbation will be removed in one
switching cycle. This is known as one-cycle damping. When -1<dl1/dl0<0, any initial perturbation will be underdamped. Any perturbation will be over-damped when 0<dl1/dl0<1.
In the frequency-domain, Q, the quality factor of the sampling gain term in the modulator transfer function, is
used to predict the tendency for sub-harmonic oscillation, which is defined as:
1
Q=
S(K-0.5)
(24)
The relationship between Q and K factor is illustrated graphically in Figure 34.
Figure 34. Sampling gain Q vs K Factor
The minimum value of K is 0.5 again. This is the same as time domain analysis result. When K<0.5, the regulator
is unstable. High gain peaking at 0.5 results in sub-harmonic oscillation at FSW/2. When K=1, one-cycle damping
is realized. Q is equal to 0.673 at this point. A higher K factor may introduce additional phase shift by moving the
sampled gain inductor pole closer to the crossover frequency, but will help reduce noise sensitivity in the current
loop. The maximum allowable value of K factor can be calculated by the Maximum Crossover Frequency
equation in Table 1.
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PC BOARD LAYOUT RECOMMENDATION
In a buck regulator the primary switching loop consists of the input capacitor, NMOS power switches and current
sense resistor. Minimizing the area of this loop reduces the stray inductance and minimizes noise and possible
erratic operation. High quality input capacitors should be placed as close as possible to the NMOS power
switches, with the VIN side of the capacitor connected directly to the high-side NMOS drain and the ground side
of the capacitor connected as close as possible to the current sense resistor ground connection.
Connect all of the low power ground connections (RUV1, RT, RFB1, CSS, CRES, CCM, CVIN, CRAMP) directly to the
regulator AGND pin. Connect CVCC directly to the regulator PGND pin. Note that CVIN and CVCC must be as
physically close as possible to the IC. AGND and PGND must be directly connected together through a top-side
copper pattern connected to the exposed pad. Ensure no high current flows beneath the underside exposed pad.
The LM25117 has an exposed thermal pad to aid power dissipation. Adding several vias under the exposed pad
helps conduct heat away from the IC. The junction to ambient thermal resistance varies with application. The
most significant variables are the area of copper in the PC board, the number of vias under the exposed pad and
the amount of forced air cooling. The integrity of the solder connection from the IC exposed pad to the PC board
is critical. Excessive voids greatly decrease the thermal dissipation capacity.
The highest power dissipating components are the two power switches. Selecting NMOS switches with exposed
pads aids the power dissipation of these devices.
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Frequency Analysis Formulas
Table 1. LM25117 Frequency Analysis Formulas
SIMPLE FORMULA
s
1+
^
ZZ_ESR
VOUT
= AM x
^
VCOMP
(1 + s )
Modulator
Transfer
Function
ZP_LF
Modulator DC
Gain
ZZ_ESR =
ESR Pole
Dominant Load
Pole
ZP_LF = R
1
RESR x COUT
x COUT
Not considered
Quality Factor
Not considered
Sub-harmonic
Double Pole
Not considered
-
^
VCOMP
^
V
OUT
High Frequency
Pole
26
1+
= AFB x
AFB_MID =
2
s
ZZ_EA
s
ZP_EA )
1
x COUT1
1
x (COUT1 // COUT2 )
-
or
ZP_HF = Q x Zn
1
S(K - 0.5)
= S x fSW or fn =
fSW
2
LO
RRAMP x CRAMP x RS x AS
^
VCOMP
^
V
OUT
1+
= AFB x
s
ZZ_EA
s x (1+
s
ZP_EA )
1
RFB2 x (CCOMP + CHF)
AFB =
RCOMP
RFB2
1
RCOMP x CHF
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ZSW
Zn =
AFB_MID =
1
=
fSW
0.5
Q=
RCOMP
RFB2
1
ZZ_EA = RCOMP x CCOMP
ZP_EA
K
K=
s x (1+
ESR1
ESR1
1
1
+
+ RESR1) x (COUT1 + COUT2) LO x (COUT1 + COUT2) x ZP_HF
ZP_HF =
1
RFB2 x (CCOMP + CHF)
AFB =
Mid-band Gain
Low Frequency
Zero
LOAD
K=1
K Factor
Feedback DC
Gain
ZZ_ESR = R
ZP_LF = (R
1
RLOAD
ZP_HF x LO
RLOAD
x
R S x AS 1 +
ZP_ESR = R
1
LOAD
Sampled Gain
Inductor Pole
Feedback
Transfer
Function
ZZ_ESR
s
s
s
s2
x
ZP_LF) (1 + ZP_ESR ) x (1 + ZP_HF + Zn2 )
AM =
Not considered
s
1+
^
VOUT
= AM x
^
VCOMP
(1 +
RLOAD
R S x AS
AM =
ESR Zero
COMPREHENSIVE FORMULA*
ZZ_EA = RCOMP x CCOMP
ZP_EA
=
1
RCOMP x (CHF // CCOMP)
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Table 1. LM25117 Frequency Analysis Formulas (continued)
SIMPLE FORMULA
s
s
1+
1+
ZZ_ESR
ZZ_EA
T(s) = AM x AFB x
x
(1 + s ) s x (1 + s
1+
T(s) = AM x AFB x
ZP_EA)
ZP_LF
Open-Loop
Response
COMPREHENSIVE FORMULA*
(1 +
AM x AFB
x
T(s) =
s
AM x AFB
T(s) =
s
when ZZ_EA = ZP_LF & ZP_EA = ZZ_ESR
Crossover
Frequency
(Open Loop
Bandwidth)
when
&
Maximum
Crossover
Frequency
fSW
fCROSS_MAX =
5
s
ZZ_ESR
2
s
s
(1 + Z ) x (1 + Z
) x (1 + Z s + s 2 )
P_EA
P_ESR
P_HF
Zn
fCROSS =
when ZZ_EA = ZP_LF & ZP_EA = ZZ_ESR
s
1+
when
RCOMP
fCROSS =
2 x ' x RS x RFB2 x AS x COUT
s
1+
ZZ_ESR
ZZ_EA
x
2
s
s
s
s
s
x
x
(
1
+
s
x
ZP_EA)
ZP_LF) (1 + ZP_ESR ) (1 + ZP_HF + Zn2 )
ZZ_EA = ZP_LF
RCOMP
2 x ' x RS x RFB2 x AS x (COUT1 + COUT2)
ZZ_EA = ZP_LF
ZP_HF
fCROSS <
2 x S x 10
fCROSS_MAX =
ZP_EA = ZZ_ESR
&
&
fCROSS <
ZP_ESR
2 x S x 10
fSW
x ( 1 + 4 x Q2 -1)
4xQ
The frequency at which 45° phase shift occurs in modulator phase
characteristics.
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DESIGN EXAMPLE
OPERATING CONDITIONS
•
•
•
•
•
•
•
Output Voltage VOUT = 3.3V
Full Load Current IOUT = 9A
Minimum Input Voltage VIN(MIN) = 6V
Maximum Input Voltage VIN(MAX) = 36V
Switching Frequency fSW = 230kHz
Diode Emulation Yes
External VCC Supply No
TIMING RESISTOR RT
Generally, higher frequency applications are smaller but have higher losses. Operation at 230 kHz was selected
for this example as a reasonable compromise between small size and high efficiency. The value of RT for 230
kHz switching frequency can be calculated from the equation as follows:
9
RT =
5.2 x 10
- 948 = 21.7 k:
230 x 103
(25)
A standard value of 22.1kΩ was chosen for RT.
OUTPUT INDUCTOR LO
The maximum inductor ripple current occurs at the maximum input voltage. Typically, 20% to 40% of the full load
current is a good compromise between core loss and copper loss of the inductor. Higher ripple current allows for
a smaller inductor size, but places more of a burden on the output capacitor to smooth the ripple voltage on the
output. For this example, a ripple current of 20% of 9A was chosen. Knowing the switching frequency, maximum
ripple current, maximum input voltage and the nominal output voltage, the inductor value can be calculated as
follows:
VOUT
§ VOUT · [H]
x ¨1 LO =
IPP(MAX) x fSW
VIN(MAX)¸
¹
©
LO =
(26)
§ 3.3V· = 7.2 PH
3.3V
x 19A x 0.2 x 230 kHz ¨ 36V ¸
©
¹
(27)
The closest standard value of 6.8μH was chosen for LO. Using the value of 6.8μH for LO, calculate IPP again. This
step is necessary if the chosen value of LO differs significantly from the calculated value.
From Equation 11,
IPP(MAX) =
§ 3.3V· = 1.9A
3.3V
x ¨1 36V¸
6.8 PH x 230 kHz
©
¹
(28)
At the minimum input voltage, this value is 0.95A.
DIODE EMULATION FUNCTION
The DEMB pin is left floating since this example uses diode emulation to reduce the power loss under no load or
light load conditions.
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CURRENT SENSE RESISTOR RS
The performance of the converter will vary depending on the K value. For this example, K=1 was chosen to
control sub-harmonic oscillation and achieve one-cycle damping. The maximum output current capability
(IOUT(MAX)) should be 20~50% higher than the required output current, to account for tolerances and ripple
current. For this example, 150% of 9A was chosen. The current sense resistor value can be calculated from
Equation 9 and Equation 10 as follows:
VCS(TH)
RS =
RS =
VOUT x K _ IPP
IOUT(MAX) +
2
fSW x LO
>:@
(29)
0.12V
= 7.9 m:
3.3 x 1
_ 0.95A
9A x 1.5 +
230 kHz x 6.8 µH
2
(30)
A value of 8mΩ was chosen for RS. A larger value resistor can be placed in parallel with RS to adjust the
maximum output current capability. The sense resistor must be rated to handle the power dissipation at
maximum input voltage when current flows through the low-side NMOS for the majority of the PWM cycle. The
maximum power dissipation of RS can be calculated as:
§
©
VIN(MAX)¸
§
©
3.3V·
2
x 9A x 8 m: = 0.59W
36V ¸
PRS = ¨1 -
PRS = ¨1 -
VOUT
· x I 2x R
OUT
S
¹
[W]
(31)
¹
(32)
The worst case peak inductor current under the output short condition can be calculated from Equation 12 as
follows:
0.12V 36V x 100 ns
+
= 15.5A
ILIM_PK =
8 m:
6.8 PH
(33)
Where tON(MIN) is normally 100ns.
CURRENT SENSE FILTER RCS and CCS
The LM25117 itself is not affected by the large leading edge spike because it samples valley current just prior to
the onset of the high-side switch. A current sense filter is used to minimize a noise injection from any external
noise sources. In general, a current sense filter is not necessary. In this example, a current sense filter is not
used
Adding RCS resistor changes the current sense amplifier gain which is defined as AS=10k / (1k+RCS). A small
value of RCS resistor below 100Ω is recommended to minimize the gain change which is caused by the
temperature coefficient difference between internal and external resistors.
RAMP RESISTOR RRAMP and RAMP CAPACITOR CRAMP
The positive slope of the inductor current ramp signal is emulated by RRAMP and CRAMP. For this example, the
value of CRAMP was set at the standard capacitor value of 820pF. With the inductor, sense resistor and the K
factor selected, the value of RRAMP can be calculated from Equation 4 as follows:
LO
[:@
RRAMP =
K x CRAMP x RS x AS
(34)
RRAMP =
6.8 PH
= 104 k:
1 x 820 pF x 8 m: x 10
(35)
The standard value of 105 kΩ was selected for RRAMP.
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UVLO DIVIDER RUV2, RUV1 AND CFT
The desired startup voltage and the hysteresis are set by the voltage divider RUV1 and RUV2. Capacitor CFT
provides filtering for the divider. For this design, the startup voltage was set to 5.7V, 0.3V below VIN(MIN). VHYS
was set to 1V. The value of RUV1, RUV2 can be calculated from Equation 1 and Equation 2 as follows:
1V
= 50 k:
RUV2 =
20 µA
(36)
RUV1 =
1.25V x 50 k:
= 14.0 k:
5.7V - 1.25V
(37)
The standard value of 50kΩ was selected for RUV2. RUV1 was selected to be 14kΩ. A value of 100pF was chosen
for CFT.
VCC DISABLE AND EXTERNAL VCC SUPPLY
In this example, VCCDIS is left floating to enable the internal VCC regulator.
POWER SWITCHES QH and QL
Selection of the power NMOS devices is governed by the same trade-offs as switching frequency. Breaking
down the losses in the high-side and low-side NMOS devices is one way to compare the relative efficiencies of
different devices. Losses in the power NMOS devices can be broken down into conduction loss, gate charging
loss, and switching loss.
Conduction loss PDC is approximately:
PDC (High-Side) = D x (IOUT2 x RDS(ON) x 1.3)
PDC (Low-Side) = (1 ± D) x
(IOUT2
[W]
x RDS(ON) x 1.3)
(38)
[W]
(39)
Where D is the duty cycle and the factor of 1.3 accounts for the increase in the NMOS device on-resistance due
to heating. Alternatively, the factor of 1.3 can be eliminated and the high temperature on-resistance of the NMOS
device can be estimated using the RDS(ON) vs Temperature curves in the MOSFET datasheet.
Gate charging loss (PGC) results from the current driving the gate capacitance of the power NMOS devices and is
approximated as:
PGC = n x VVCC x Qg x fSW [W]
(40)
Qg refers to the total gate charge of an individual NMOS device, and ‘n’ is the number of NMOS devices. Gate
charge loss differs from conduction and switching losses in that the actual dissipation occurs in the controller IC.
Switching loss (PSW) occurs during the brief transition period as the high-side NMOS device turns on and off.
During the transition period both current and voltage are present in the channel of the NMOS device. The
switching loss can be approximated as:
PSW = 0.5 x VIN x IOUT x (tR + tF) x fSW [W]
(41)
tR and tF are the rise and fall times of the high-side NMOS device. The rise and fall times are usually mentioned
in the MOSFET datasheet or can be empirically observed with an oscilloscope. Switching loss is calculated for
the high-side NMOS device only. Switching loss in the low-side NMOS device is negligible because the body
diode of the low-side NMOS device turns on before and after the low-side NMOS device switches. For this
example, the maximum drain-to-source voltage applied to either NMOS device is 36V. The selected NMOS
devices must be able to withstand 36V plus any ringing from drain to source and must be able to handle at least
the VCC voltage plus any ringing from gate to source.
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SNUBBER COMPONENTS RSNB AND CSNB
A resistor-capacitor snubber network across the low-side NMOS device reduces ringing and spikes at the
switching node. Excessive ringing and spikes can cause erratic operation and can couple noise to the output
voltage. Selecting the values for the snubber is best accomplished through empirical methods. First, make sure
the lead lengths for the snubber connections are very short. Start with a resistor value between 5 and 50Ω.
Increasing the value of the snubber capacitor results in more damping, but higher snubber losses. Select a
minimum value for the snubber capacitor that provides adequate damping of the spikes on the switch waveform
at heavy load. A snubber may not be necessary with an optimized layout.
BOOTSTRAP CAPACITOR CHB AND BOOTSTRAP DIODE DHB
The bootstrap capacitor between the HB and SW pin supplies the gate current to charge the high-side NMOS
device gate during each cycle’s turn-on and also supplies recovery charge for the bootstrap diode. These current
peaks can be several amperes. The recommended value of the bootstrap capacitor is at least 0.1μF. CHB should
be a good quality, low ESR, ceramic capacitor located at the pins of the IC to minimize potentially damaging
voltage transients caused by trace inductance. The absolute minimum value for the bootstrap capacitor is
calculated as:
Qg
CHB t
[F]
'VHB
(42)
Where Qg is the high-side NMOS gate charge and ΔVHB is the tolerable voltage droop on CHB, which is typically
less than 5% of VCC or 0.15V conservatively. A value of 0.47μF was selected for this design.
VCC CAPACITOR CVCC
The primary purpose of the VCC capacitor (CVCC) is to supply the peak transient currents of the LO driver and
bootstrap diode as well as provide stability for the VCC regulator. These peak currents can be several amperes.
The recommended value of CVCC should be no smaller than 0.47μF, and should be a good quality, low ESR,
ceramic capacitor. CVCC should be placed at the pins of the IC to minimize potentially damaging voltage
transients caused by trace inductance. A value of 1μF was selected for this design.
OUTPUT CAPACITOR CO
The output capacitors smooth the output voltage ripple caused by inductor ripple current and provide a source of
charge during transient loading conditions. For this design example, a 680μF electrolytic capacitor with maximum
10mΩ ESR was selected as the main output capacitor. The fundamental component of the output ripple voltage
with maximum ESR is approximated as:
'VOUT = IPP x
RESR
2
§
+¨
©8 x
1
·
fSW x COUT¸
¹
2
[V]
(43)
2
'VOUT = 1.9 x
1
·
2 §
= 19 mV
0.01: + ¨
8 x 230 kHz x 680 PF¸
©
¹
(44)
Additional low ERS / ESL ceramic capacitors can be placed in parallel with the main output capacitor to further
reduce the output voltage ripple and spikes. In this example, two 22μF capacitors were added.
INPUT CAPACITOR CIN
The regulator input supply voltage typically has high source impedance at the switching frequency. Good quality
input capacitors are necessary to limit the ripple voltage at the VIN pin while supplying most of the switch current
during the on-time. When the high-side NMOS device turns on, the current into the device steps to the valley of
the inductor current waveform, ramps up to the peak value, and then drops to the zero at turnoff. The input
capacitor should be selected for RMS current rating and minimum ripple voltage. A good approximation for the
required ripple current rating necessary is IRMS > IOUT / 2.
In this example, seven 2.2μF ceramic capacitors were used. With ceramic capacitors, the input ripple voltage will
be triangular. The input ripple voltage can be approximated as:
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'VIN =
IOUT
4 x fSW x CIN
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[V]
(45)
'VIN =
9A
= 0.63V
4 x 230 kHz x 2.2 PH x 7
(46)
Capacitors connected in parallel should be evaluated for RMS current rating. The current will split between the
input capacitors based on the relative impedance of the capacitors at the switching frequency.
VIN FILTER RVIN, CVIN
An R-C filter (RVIN, CVIN) on VIN is optional. The filter helps to prevent faults caused by high frequency switching
noise injection into the VIN pin. 0.47μF ceramic capacitor is used for CVIN in the example.
SOFT-START CAPACITOR CSS
The capacitor at the SS pin (CSS) determines the soft-start time (tSS), which is the time for the output voltage to
reach the final regulated value. The tSS for a given CSS can be calculated from Equation 8 as follows:
0.047 µF x 0.8V
= 3.8 ms
tSS =
10 µA
(47)
For this example, a value of 0.047μF was chosen for a soft-start time of 3.8ms.
RESTART CAPACITOR CRES
The capacitor at the RES pin (CRES) determines tRES, which is the time the LM25117 remains off before a restart
attempt is made in hiccup mode current limiting. tRES for a given CRES can be calculated from Equation 13 as
follows:
0.47 µF x 1.25V
= 59 ms
tRES =
10 µA
(48)
For this example, a value of 0.47μF was chosen for a restart time of 59ms.
OUTPUT VOLTAGE DIVIDER RFB2 and RFB1
RFB1 and RFB2 set the output voltage level. The ratio of these resistors is calculated as:
RFB2 VOUT
-1
=
RFB1 0.8V
(49)
The ratio between RCOMP and RFB2 determines the mid-band gain, AFB_MID. A larger value for RFB2 may require a
corresponding larger value for RCOMP. RFB2 should be large enough to keep the total divider power dissipation
small. 3.24kΩ was chosen for RFB2 in this example, which results in a RFB1 value of 1.05kΩ for 3.3V output.
LOOP COMPENSATION COMPONENTS CCOMP, RCOMP and CHF
RCOMP, CCOMP and CHF configure the error amplifier gain and phase characteristics to produce a stable voltage
loop. For a quick start, follow the 4 steps listed below. For detailed information, see Application Information.
STEP1: Select fCROSS
By selecting one tenth of the switching frequency, fCROSS is calculated as follows:
fSW
= 23 kHz
fCROSS =
10
(50)
STEP2: Determine required RCOMP
Knowing fCROSS, RCOMP is calculated as follows:
RCOMP = 2S x RS x AS x COUT x RFB2 x fCROSS
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[:@
(51)
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SNVS714D – APRIL 2011 – REVISED APRIL 2012
RCOMP = 2S x 8 m: x 10 x 724 µF x 3.24 k: x 23 kHz = 27.1 k:
(52)
The standard value of 27.4kΩ was selected for RCOMP
STEP3: Determine CCOMP to cancel load pole
Knowing RCOMP, CCOMP is calculated as follows:
RLOAD x COUT
[F]
CCOMP =
RCOMP
CCOMP =
3.3V
9A
(53)
x 724 µF
27.4 k:
= 10 nF
(54)
The standard value of 10nF was selected for CCOMP
STEP4: Determine CHF to cancel ESR zero
Knowing RCOMP and CCOMP, CHF is calculated as follows:
RESR x COUT x CCOMP
CHF =
[F]
RCOMP x CCOMP - RESR x COUT
CHF =
(55)
5 m: x 724 µF x 10 nF
= 134 pF
27.4 k: x 10 nF - 5 m: x 724 µF
(56)
Half of the maximum ESR is assumed as a typical ESR. The standard value of 150pF was selected for CHF.
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Application Circuit
Figure 35. 3.3V, 9A Typical Application Schematic
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Example of Constant Current Regulator
The LM25117 can be configured as a constant current regulator by using the current monitor feature (CM) as the
feedback input. A voltage divider at the VCCDIS pin from VOUT to AGND can be used to protect against output
over-voltage. When the VCCDIS pin voltage is greater than the VCCDIS threshold, the controller disables the
VCC regulator and the VCC pin voltage decays. When the VCC pin voltage is less than the VCC UV threshold,
both HO and LO outputs stop switching. Due to the time delay required for VCC to decay below the VCC UV
threshold, the over-voltage protection operates in hiccup mode. See Figure 36.
VIN
100k:
CIN
UVLO VIN
SW
RES DEMB VCC
15 k:
HB
RAMP
VOUT
DHB
LM25117
Hiccup Mode OVP 100 k:
Triggered at 13.4V
CVCC
CHB
1500 pF
HO
3.24 k:
QH
CC Mode: 2A
SW
VCCDIS
LO
332:
VOUT
68 µH
QL
80 µF
CS
47 m:
0.022 µF
3.24 k:
CSG
CM
COMP
FB
RT
SS
AGND PGND
2.37 k:
22.1 k:
0.47 µF
Current Control (CC)
Figure 36. Constant Current Regulator with Hiccup Mode Output OVP
The LM25117 also can be configured as a constant voltage and constant current regulator, known as CV+CC
regulator. In this configuration, there is much less variation in the current limiting as compared to peak cycle-bycycle current limiting of the inductor current. The LMV431 and the PNP transistor create a voltage-to-current
amplifier in the current loop. This amplifier circuitry does not affect the normal operation when the output current
is less than the current limit set-point. When the output current is greater than the set-point, the PNP transistor
sources a current into CRAMP and increases the positive slope of emulated inductor current ramp until the output
current is less than or equal to the current limit set-point. See Figure 37 and Figure 38.
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VIN
Current Control (CC)
100 k:
VCC
UVLO VIN
SW
10 k:
100:
CIN
15 k:
RES DEMB VCC
LM25117
100 k:
PNP
CVCC
DHB
HB
RAMP
CHB
1 nF
1500 pF
QH
HO
100 k:
CM
68 PH
VOUT
CV Mode : 5V
CC Mode: 2A
SW
LMV431
200 k:
QL
LO
VCCDIS
80 PF
CS
47 m:
VOUT
3.24 k:
CSG
34.8 k:
0.1 PF
COMP
FB
RT
SS
AGND PGND
619:
22.1 k:
0.33 PF x2
Voltage Control (CV)
Figure 37. Constant Voltage Regulator with Accurate Current Limit
Figure 38. Current Limit Comparison
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PACKAGE OPTION ADDENDUM
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24-Jan-2013
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package Qty
Drawing
Eco Plan
Lead/Ball Finish
(2)
MSL Peak Temp
Op Temp (°C)
Top-Side Markings
(3)
(4)
LM25117PMH/NOPB
ACTIVE
HTSSOP
PWP
20
73
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM25117
PMH
LM25117PMHE/NOPB
ACTIVE
HTSSOP
PWP
20
250
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM25117
PMH
LM25117PMHX/NOPB
ACTIVE
HTSSOP
PWP
20
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM25117
PMH
LM25117PSQ/NOPB
ACTIVE
WQFN
RTW
24
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
L25117P
LM25117PSQE/NOPB
ACTIVE
WQFN
RTW
24
250
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
L25117P
LM25117PSQX/NOPB
ACTIVE
WQFN
RTW
24
4500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
L25117P
LM25117QPMH/NOPB
ACTIVE
HTSSOP
PWP
20
73
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM25117
QPMH
LM25117QPMHE/NOPB
ACTIVE
HTSSOP
PWP
20
250
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM25117
QPMH
LM25117QPMHX/NOPB
ACTIVE
HTSSOP
PWP
20
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
-40 to 125
LM25117
QPMH
LM25117QPSQ/NOPB
ACTIVE
WQFN
RTW
24
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
L25117Q
LM25117QPSQE/NOPB
ACTIVE
WQFN
RTW
24
250
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
L25117Q
LM25117QPSQX/NOPB
ACTIVE
WQFN
RTW
24
4500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
L25117Q
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
24-Jan-2013
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
Only one of markings shown within the brackets will appear on the physical device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM25117, LM25117-Q1 :
• Catalog: LM25117
• Automotive: LM25117-Q1
NOTE: Qualified Version Definitions:
• Catalog - TI's standard catalog product
• Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
17-Nov-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
LM25117PMHE/NOPB
HTSSOP
PWP
20
250
178.0
16.4
LM25117PMHX/NOPB
HTSSOP
PWP
20
2500
330.0
LM25117PSQ/NOPB
WQFN
RTW
24
1000
178.0
LM25117PSQE/NOPB
WQFN
RTW
24
250
LM25117PSQX/NOPB
WQFN
RTW
24
LM25117QPMHE/NOPB HTSSOP
PWP
LM25117QPMHX/NOPB HTSSOP
PWP
W
Pin1
(mm) Quadrant
6.95
7.1
1.6
8.0
16.0
Q1
16.4
6.95
7.1
1.6
8.0
16.0
Q1
12.4
4.3
4.3
1.3
8.0
12.0
Q1
178.0
12.4
4.3
4.3
1.3
8.0
12.0
Q1
4500
330.0
12.4
4.3
4.3
1.3
8.0
12.0
Q1
20
250
178.0
16.4
6.95
7.1
1.6
8.0
16.0
Q1
20
2500
330.0
16.4
6.95
7.1
1.6
8.0
16.0
Q1
1000
178.0
12.4
4.3
4.3
1.3
8.0
12.0
Q1
LM25117QPSQ/NOPB
WQFN
RTW
24
LM25117QPSQE/NOPB
WQFN
RTW
24
250
178.0
12.4
4.3
4.3
1.3
8.0
12.0
Q1
LM25117QPSQX/NOPB
WQFN
RTW
24
4500
330.0
12.4
4.3
4.3
1.3
8.0
12.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
17-Nov-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM25117PMHE/NOPB
HTSSOP
PWP
LM25117PMHX/NOPB
HTSSOP
PWP
20
250
203.0
190.0
41.0
20
2500
358.0
343.0
63.0
LM25117PSQ/NOPB
WQFN
RTW
24
1000
203.0
190.0
41.0
LM25117PSQE/NOPB
WQFN
RTW
24
250
203.0
190.0
41.0
LM25117PSQX/NOPB
WQFN
RTW
24
4500
349.0
337.0
45.0
LM25117QPMHE/NOPB
HTSSOP
PWP
20
250
203.0
190.0
41.0
LM25117QPMHX/NOPB
HTSSOP
PWP
20
2500
358.0
343.0
63.0
LM25117QPSQ/NOPB
WQFN
RTW
24
1000
203.0
190.0
41.0
LM25117QPSQE/NOPB
WQFN
RTW
24
250
203.0
190.0
41.0
LM25117QPSQX/NOPB
WQFN
RTW
24
4500
349.0
337.0
45.0
Pack Materials-Page 2
MECHANICAL DATA
PWP0020A
MXA20A (Rev C)
www.ti.com
MECHANICAL DATA
RTW0024A
SQA24A (Rev B)
www.ti.com
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