AD ADL5387-EVALZ1 50 mhz to 2 ghz quadrature demodulator Datasheet

50 MHz to 2 GHz
Quadrature Demodulator
ADL5387
Operating RF frequency
50 MHz to 2 GHz
LO input at 2 × fLO
100 MHz to 4 GHz
Input IP3: 31 dBm @ 900 MHz
Input IP2: 62 dBm @ 900 MHz
Input P1dB: 13 dBm @ 900 MHz
Noise figure (NF)
12.0 dB @ 140 MHz
14.7 dB @ 900 MHz
Voltage conversion gain > 4 dB
Quadrature demodulation accuracy
Phase accuracy ~0.4°
Amplitude balance ~0.05 dB
Demodulation bandwidth ~240 MHz
Baseband I/Q drive 2 V p-p into 200 Ω
Single 5 V supply
FUNCTIONAL BLOCK DIAGRAM
24
23
22
21
20
19
1
CMRF CMRF RFIP RFIN CMRF VPX
VPA
VPB 18
2
COM
3
BIAS
4
VPL
5
VPL
6
VPL
CML
7
VPB 17
QHI 16
DIVIDE-BY-2
PHASE SPLITTER
QLO 15
IHI 14
ILO 13
LOIP LOIN CML
8
9
10
CML
COM
11
12
06764-001
FEATURES
Figure 1.
APPLICATIONS
QAM/QPSK RF/IF demodulators
W-CDMA/CDMA/CDMA2000/GSM
Microwave point-to-(multi)point radios
Broadband wireless and WiMAX
Broadband CATVs
GENERAL DESCRIPTION
The ADL5387 is a broadband quadrature I/Q demodulator that
covers an RF/IF input frequency range from 50 MHz to 2 GHz.
With a NF = 13.2 dB, IP1dB = 12.7 dBm, and IIP3 = 32 dBm @
450 MHz, the ADL5387 demodulator offers outstanding dynamic
range suitable for the demanding infrastructure direct-conversion
requirements. The differential RF/IF inputs provide a wellbehaved broadband input impedance of 50 Ω and are best
driven from a 1:1 balun for optimum performance.
The fully balanced design minimizes effects from second-order
distortion. The leakage from the LO port to the RF port is
<−70 dBc. Differential dc-offsets at the I and Q outputs are
<10 mV. Both of these factors contribute to the excellent IIP2
specifications > 60 dBm.
Ultrabroadband operation is achieved with a divide-by-2 method
for local oscillator (LO) quadrature generation. Over a wide
range of LO levels, excellent demodulation accuracy is
achieved with amplitude and phase balances ~0.05 dB and
~0.4°, respectively. The demodulated in-phase (I) and
quadrature (Q) differential outputs are fully buffered and
provide a voltage conversion gain of >4 dB. The buffered
baseband outputs are capable of driving a 2 V p-p differential
signal into 200 Ω.
The ADL5387 is fabricated using the Analog Devices, Inc.
advanced silicon-germanium bipolar process and is available in
a 24-lead exposed paddle LFCSP.
The ADL5387 operates off a single 4.75 V to 5.25 V supply. The
supply current is adjustable with an external resistor from the
BIAS pin to ground.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2007 Analog Devices, Inc. All rights reserved.
ADL5387
TABLE OF CONTENTS
Features .............................................................................................. 1
Mixers .......................................................................................... 14
Applications....................................................................................... 1
Emitter Follower Buffers ........................................................... 14
Functional Block Diagram .............................................................. 1
Bias Circuit.................................................................................. 14
General Description ......................................................................... 1
Applications..................................................................................... 15
Revision History ............................................................................... 2
Basic Connections...................................................................... 15
Specifications..................................................................................... 3
Power Supply............................................................................... 15
Absolute Maximum Ratings............................................................ 5
Local Oscillator (LO) Input ...................................................... 15
ESD Caution.................................................................................. 5
RF Input....................................................................................... 16
Pin Configuration and Function Descriptions............................. 6
Baseband Outputs ...................................................................... 16
Typical Performance Characteristics ............................................. 7
Error Vector Magnitude (EVM) Performance ....................... 17
Distributions for fRF = 140 MHz ............................................... 10
Low IF Image Rejection............................................................. 18
Distributions for fRF = 450 MHz ............................................... 11
Example Baseband Interface..................................................... 18
Distributions for fRF = 900 MHz ............................................... 12
Characterization Setups................................................................. 21
Distributions for fRF = 1900 MHz............................................. 13
Evaluation Board ............................................................................ 23
Circuit Description......................................................................... 14
Outline Dimensions ....................................................................... 26
LO Interface................................................................................. 14
Ordering Guide .......................................................................... 26
V-to-I Converter......................................................................... 14
REVISION HISTORY
10/07—Revision 0: Initial Version
Rev. 0 | Page 2 of 28
ADL5387
SPECIFICATIONS
VS = 5 V, TA = 25°C, fRF = 900 MHz, fIF = 4.5 MHz, PLO = 0 dBm, BIAS pin open, ZO = 50 Ω, unless otherwise noted, baseband outputs
differentially loaded with 450 Ω.
Table 1.
Parameter
OPERATING CONDITIONS
LO Frequency Range
RF Frequency Range
LO INPUT
Input Return Loss
LO Input Level
I/Q BASEBAND OUTPUTS
Voltage Conversion Gain
Demodulation Bandwidth
Quadrature Phase Error
I/Q Amplitude Imbalance
Output DC Offset (Differential)
Output Common-Mode
0.1 dB Gain Flatness
Output Swing
Peak Output Current
POWER SUPPLIES
Voltage
Current
DYNAMIC PERFORMANCE @ RF = 140 MHz
Conversion Gain
Input P1dB (IP1dB)
Second-Order Input Intercept (IIP2)
Third-Order Input Intercept (IIP3)
LO to RF
RF to LO
I/Q Magnitude Imbalance
I/Q Phase Imbalance
LO to I/Q
Noise Figure
Noise Figure under Blocking Conditions
Condition
Min
External input = 2xLO frequency
0.1
0.05
Typ
Max
Unit
4
2
GHz
GHz
LOIP, LOIN
−10
AC-coupled into LOIP with LOIN bypassed,
measured at 2 GHz
−6
QHI, QLO, IHI, ILO
450 Ω differential load on I and Q outputs
(@ 900 MHz)
200 Ω differential load on I and Q outputs
(@ 900 MHz)
1 V p-p signal 3 dB bandwidth
@ 900 MHz
0 dBm LO input
Differential 200 Ω load
Each pin
VPA, VPL, VPB, VPX
0
−5 dBm each input tone
−5 dBm each input tone
RFIN, RFIP terminated in 50 Ω, 1xLO
appearing at the RF port
LOIN, LOIP terminated in 50 Ω
RFIN, RFIP terminated in 50 Ω, 1xLO
appearing at the BB port
With a −5 dBm interferer 5 MHz away
Rev. 0 | Page 3 of 28
+6
dBm
4.3
dB
3.2
dB
240
0.4
0.1
±5
VPOS − 2.8
40
2
12
MHz
Degrees
dB
mV
V
MHz
V p-p
mA
4.75
BIAS pin open
RBIAS = 4 kΩ
RFIP, RFIN
dB
180
157
5.25
V
mA
mA
4.7
13
67
31
−100
dB
dBm
dBm
dBm
dBm
−95
0.05
0.2
−39
dBc
dB
Degrees
dBm
12.0
14.4
dB
dB
ADL5387
Parameter
DYNAMIC PERFORMANCE @ RF = 450 MHz
Conversion Gain
Input P1dB (IP1dB)
Second-Order Input Intercept (IIP2)
Third-Order Input Intercept (IIP3)
LO to RF
RF to LO
I/Q Magnitude Imbalance
I/Q Phase Imbalance
LO to I/Q
Noise Figure
DYNAMIC PERFORMANCE @ RF = 900 MHz
Conversion Gain
Input P1dB (IP1dB)
Second-Order Input Intercept (IIP2)
Third-Order Input Intercept (IIP3)
LO to RF
RF to LO
I/Q Magnitude Imbalance
I/Q Phase Imbalance
LO to I/Q
Noise Figure
Noise Figure under Blocking Conditions
DYNAMIC PERFORMANCE @ RF = 1900 MHz
Conversion Gain
Input P1dB (IP1dB)
Second-Order Input Intercept (IIP2)
Third-Order Input Intercept (IIP3)
LO to RF
RF to LO
I/Q Magnitude Imbalance
I/Q Phase Imbalance
LO to I/Q
Noise Figure
Noise Figure under Blocking Conditions
Condition
Min
−5 dBm each input tone
−5 dBm each input tone
RFIN, RFIP terminated in 50 Ω, 1xLO
appearing at the RF port
LOIN, LOIP terminated in 50 Ω
RFIN, RFIP terminated in 50 Ω, 1xLO
appearing at the BB port
−5 dBm each input tone
−5 dBm each input tone
RFIN, RFIP terminated in 50 Ω, 1xLO
appearing at the RF port
LOIN, LOIP terminated in 50 Ω
RFIN, RFIP terminated in 50 Ω,
1XLO appearing at the BB port
With a −5 dBm interferer 5 MHz away
−5 dBm each input tone
−5 dBm each input tone
RFIN, RFIP terminated in 50 Ω, 1xLO
appearing at the RF port
LOIN, LOIP terminated in 50 Ω
RFIN, RFIP terminated in 50 Ω, 1xLO
appearing at the BB port
With a −5 dBm interferer 5 MHz away
Rev. 0 | Page 4 of 28
Typ
Max
Unit
4.4
12.7
69.2
32.8
−87
dB
dBm
dBm
dBm
dBm
−90
0.05
0.6
−38
dBc
dB
Degrees
dBm
13.2
dB
4.3
12.8
61.7
31.2
−79
dB
dBm
dBm
dBm
dBm
−88
0.05
0.2
−41
dBc
dB
Degrees
dBm
14.7
15.8
dB
dB
3.8
12.8
59.8
27.4
−75
dB
dBm
dBm
dBm
dBm
−70
0.05
0.3
−43
dBc
dB
Degrees
dBm
16.5
18.7
dB
dB
ADL5387
ABSOLUTE MAXIMUM RATINGS
ESD CAUTION
Table 2.
Parameter
Supply Voltage VPOS1, VPOS2, VPOS3
LO Input Power
RF/IF Input Power
Internal Maximum Power Dissipation
θJA
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Rating
5.5 V
13 dBm (re: 50 Ω)
15 dBm (re: 50 Ω)
1100 mW
54°C/W
150°C
−40°C to +85°C
−65°C to +125°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Rev. 0 | Page 5 of 28
ADL5387
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
23
22
21
20
19
1
2
COM
3
BIAS
4
VPL
5
VPL
6
VPL
CML
7
VPB 17
QHI 16
ADL5387
TOP VIEW
(Not to Scale)
QLO 15
IHI 14
ILO 13
LOIP LOIN CML
8
9
10
CML
COM
11
12
06764-002
24
CMRF CMRF RFIP RFIN CMRF VPX
VPA
VPB 18
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1, 4 to 6,
17 to 19
2, 7, 10 to 12,
20, 23, 24
3
Mnemonic
VPA, VPL, VPB, VPX
BIAS
8, 9
LOIP, LOIN
13 to 16
ILO, IHI, QLO, QHI
21, 22
RFIN, RFIP
COM, CML, CMRF
EP
Description
Supply. Positive supply for LO, IF, biasing and baseband sections, respectively. These pins should
be decoupled to board ground using appropriate sized capacitors.
Ground. Connect to a low impedance ground plane.
Bias Control. A resistor can be connected between BIAS and COM to reduce the mixer core current.
The default setting for this pin is open.
Local Oscillator. External LO input is at 2xLO frequency. A single-ended LO at 0 dBm can be applied
through a 1000 pF capacitor to LOIP. LOIN should be ac-grounded, also using a 1000 pF. These inputs
can also be driven differentially through a balun (recommended balun is M/A-COM ETC1-1-13).
I-Channel and Q-Channel Mixer Baseband Outputs. These outputs have a 50 Ω differential output
impedance (25 Ω per pin). The bias level on these pins is equal to VPOS − 2.8 V. Each output pair can
swing 2 V p-p (differential) into a load of 200 Ω. Output 3 dB bandwidth is 240 MHz.
RF Input. A single-ended 50 Ω signal can be applied to the RF inputs through a 1:1 balun (recommended
balun is M/A-COM ETC1-1-13). Ground-referenced inductors must also be connected to RFIP and
RFIN (recommended values = 120 nH).
Exposed Paddle. Connect to a low impedance ground plane
Rev. 0 | Page 6 of 28
ADL5387
TYPICAL PERFORMANCE CHARACTERISTICS
VS = 5 V, TA = 25°C, LO drive level = 0 dBm, RBIAS = open, unless otherwise noted.
5
TA = –40°C
TA = +25°C
TA = +85°C
NORMALIZED TO 1MHz
0
INPUT P1dB
15
BB RESPONSE (dB)
GAIN (dB), IP1dB (dBm)
20
10
GAIN
5
–5
–10
–15
–20
400
600
800
1000 1200 1400 1600 1800 2000
RF FREQUENCY (MHz)
–30
I CHANNEL
Q CHANNEL
70
19
NOISE FIGURE (dB)
IIP2, IIP3 (dBm)
50
40
30
0
200
400
600
800
1000 1200 1400 1600 1800 2000
RF FREQUENCY (MHz)
13
11
7
06764-004
10
15
9
INPUT IP3
(I AND Q CHANNELS)
20
0
200
QUADRATURE PHASE ERROR (Degrees)
0.5
0
–0.5
–1.0
–1.5
0
200
400
600
800
1000 1200 1400 1600 1800 2000
RF FREQUENCY (MHz)
06764-005
MAGNITUDE ERROR (dB)
4
1.0
–2.0
600
800
1000 1200 1400 1600 1800 2000
Figure 7. Noise Figure vs. RF Frequency
TA = –40°C
TA = +25°C
TA = +85°C
1.5
400
RF FREQUENCY (MHz)
Figure 4. Input Third-Order Intercept (IIP3) and
Input Second-Order Intercept Point (IIP2) vs. RF Frequency
2.0
1000
TA = –40°C
TA = +25°C
TA = +85°C
17
INPUT IP2
100
Figure 6. Normalized I/Q Baseband Frequency Response
TA = +85°C
TA = +25°C
TA = –40°C
60
10
BB FREQUENCY (MHz)
Figure 3. Conversion Gain and Input 1 dB Compression Point (IP1dB) vs.
RF Frequency
80
1
06764-007
200
Figure 5. I/Q Gain Mismatch vs. RF Frequency
TA = –40°C
TA = +25°C
TA = +85°C
3
2
1
0
–1
–2
–3
–4
0
200
400
600
800
1000 1200 1400 1600 1800 2000
RF FREQUENCY (MHz)
Figure 8. I/Q Quadrature Phase Error vs. RF Frequency
Rev. 0 | Page 7 of 28
06764-008
0
06764-003
0
06764-006
–25
NOISE FIGURE
GAIN
5
35
INPUT IP3
0
–6
–5
–4
–3
–2
–1
0
1
2
3
4
5
6
20
LO LEVEL (dBm)
15
GAIN
INPUT IP3
0
–6
16
165
155
NOISE FIGURE
12
8
145
1
135
100
10
RBIAS (kΩ)
RBIAS = 10kΩ
15
RBIAS = 4kΩ
RBIAS = 1.4kΩ
5
–25
–20
–15
–10
–5
0
5
RF BLOCKER INPUT POWER (dBm)
Figure 11. Noise Figure vs. Input Blocker Level, fRF = 900 MHz
(RF Blocker 5 MHz Offset)
06764-011
NOISE FIGURE (dB)
20
0
–30
–2
–1
0
1
2
3
4
5
6
20
INPUT IP3
24
20
NOISE FIGURE
16
12
8
1
10
100
100
RBIAS (kΩ)
Figure 13. IIP3 and Noise Figure vs. RBIAS, fRF = 900 MHz
GAIN (dB), IP1dB, IIP2, I AND Q CHANNELS (dBm)
25
10
–3
TA = –40°C
TA = +25°C
TA = +85°C
28
Figure 10. Noise Figure, IIP3, and Supply Current vs. RBIAS, fRF = 140 MHz
RBIAS = 100kΩ
–4
LO LEVEL (dBm)
IIP3 (dBm) AND NOISE FIGURE (dB)
SUPPLY
CURRENT
20
–5
32
SUPPLY CURRENT (mA)
175
35
Figure 12. Conversion Gain, Noise Figure, IIP3, IIP2, and IP1dB vs.
LO Level, fRF = 900 MHz
185
INPUT IP3
24
50
5
06764-010
IIP3 (dBm) AND NOISE FIGURE (dB)
28
INPUT P1dB
INPUT IP2, Q CHANNEL
195
TA = –40°C
TA = +25°C
TA = +85°C
65
10
Figure 9. Conversion Gain, Noise Figure, IIP3, IIP2, and IP1dB vs.
LO Level, fRF = 140 MHz
32
NOISE FIGURE
INPUT IP2, INPUT IP3 (dBm)
50
INPUT IP2, I CHANNEL
06764-013
INPUT P1dB
10
80
06764-014
65
INPUT IP2, I CHANNEL
INPUT IP2, INPUT IP3 (dBm)
15
20
06764-012
80
INPUT IP2, Q CHANNEL
GAIN (dB), INPUT P1dB (dBm), NOISE FIGURE (dB)
20
06764-009
GAIN (dB), INPUT P1dB (dBm), NOISE FIGURE (dB)
ADL5387
80
70
60
140MHz: GAIN
140MHz: IP1dB
50
140MHz: IIP2, I CHANNEL
140MHz: IIP2, Q CHANNEL
40
450MHz: GAIN
450MHz: IP1dB
30
450MHz: IIP2, I CHANNEL
450MHz: IIP2, Q CHANNEL
20
10
0
1
10
RBIAS (kΩ)
Figure 14. Conversion Gain, IP1dB, IIP2 I Channel, and IIP2 Q Channel vs. RBIAS
Rev. 0 | Page 8 of 28
ADL5387
80
75
INPUT IP2,
I CHANNEL
25
70
20
65
INPUT IP2,
Q CHANNEL
15
60
10
55
0
5
10
IP1dB
–40
–50
–60
1xLO
–70
–80
–90
2xLO
15
20
25
30
35
40
45
50
50
–100
BB FREQUENCY (MHz)
0
200
400
600
800
1000 1200 1400 1600 1800 2000
INTERNAL 1xLO FREQUENCY (MHz)
Figure 15. IIIP3, IIP2, IP1dB vs. Baseband Frequency
06764-018
5
TA = –40°C
TA = +25°C
TA = +85°C
–30
LO LEAKAGE (dBm)
IIP3
06764-015
IP1dB, IIP3 (dBm)
30
–20
INPUT IP2, I AND Q CHANNELS (dBm)
35
Figure 18. LO-to-RF Leakage vs. Internal 1xLO Frequency
0
–20
–10
LEAKAGE (dBc)
FEEDTHROUGH (dBm)
–40
–20
–30
1xLO (INTERNAL)
–40
–50
–60
–80
2xLO (EXTERNAL)
–60
–100
0
200
400
600
800
1000 1200 1400 1600 1800 2000
INTERNAL 1xLO FREQUENCY (MHz)
–120
06764-016
0
600
800
1000 1200 1400 1600 1800 2000
Figure 19. RF-to-LO Leakage vs. RF Frequency
0
0
–5
RETURN LOSS (dB)
–5
RETURN LOSS (dB)
400
RF FREQUENCY (MHz)
Figure 16. LO-to-BB Feedthrough vs. 1xLO Frequency (Internal LO Frequency)
–10
–15
–20
–10
–15
–20
–25
0
200
400
600
800
1000 1200 1400 1600 1800 2000
RF FREQUENCY (MHz)
06764-017
–25
200
Figure 17. RF Port Return Loss vs. RF Frequency, Measured on
Characterization Board through ETC1-1-13 Balun with 120 nH Bias Inductors
Rev. 0 | Page 9 of 28
–30
0
500
1000
1500
2000
2500
3000
3500
FREQUENCY (MHz)
Figure 20. Single-Ended LO Port Return Loss vs.
LO Frequency, LOIN AC-Coupled to Ground
4000
06764-020
–80
06764-019
–70
ADL5387
DISTRIBUTIONS FOR fRF = 140 MHz
80
60
40
60
40
20
29
30
31
32
33
INPUT IP3 (dBm)
0
60
06764-021
100
TA = –40°C
TA = +25°C
TA = +85°C
80
60
40
20
11
12
13
14
15
INPUT P1dB (dBm)
60
40
0
10.5
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
80
60
40
12.0
12.5
13.0
13.5
TA = –40°C
TA = +25°C
TA = +85°C
60
40
20
20
0
–0.2
11.5
Figure 25. Noise Figure Distributions
–0.1
0
0.1
I/Q GAIN MISMATCH (dB)
0.2
06764-023
PERCENTAGE (%)
80
11.0
NOISE FIGURE (dB)
Figure 22. IP1dB Distributions
100
TA = –40°C
TA = +25°C
TA = +85°C
20
06764-022
0
10
75
Figure 24. IIP2 Distributions for I Channel and Q Channel
PERCENTAGE (%)
PERCENTAGE (%)
80
70
INPUT IP2 (dBm)
Figure 21. IIP3 Distributions
100
65
06764-025
0
28
I CHANNEL
Q CHANNEL
06764-024
20
TA = –40°C
TA = +25°C
TA = +85°C
0
–1.0
–0.5
0
0.5
QUADRATURE PHASE ERROR (Degrees)
Figure 26. I/Q Quadrature Error Distributions
Figure 23. I/Q Gain Mismatch Distributions
Rev. 0 | Page 10 of 28
1.0
06764-026
PERCENTAGE (%)
80
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
100
ADL5387
DISTRIBUTIONS FOR fRF = 450 MHz
80
60
40
60
40
20
31
32
33
34
35
INPUT IP3 (dBm)
0
60
06764-027
100
TA = –40°C
TA = +25°C
TA = +85°C
40
20
60
40
20
12
13
14
15
0
12.0
06764-028
11
INPUT P1dB (dBm)
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
80
60
40
13.5
14.0
14.5
15.0
TA = –40°C
TA = +25°C
TA = +85°C
60
40
20
20
0
–0.2
13.0
Figure 31. Noise Figure Distributions
–0.1
0
0.1
I/Q GAIN MISMATCH (dB)
0.2
06764-029
PERCENTAGE (%)
80
12.5
NOISE FIGURE (dB)
Figure 28. IP1dB Distributions
100
TA = –40°C
TA = +25°C
TA = +85°C
80
60
0
10
75
Figure 30. IIP2 Distributions for I Channel and Q Channel
PERCENTAGE (%)
PERCENTAGE (%)
80
70
INPUT IP2 (dBm)
Figure 27. IIP3 Distributions
100
65
06764-031
0
30
I CHANNEL
Q CHANNEL
06764-030
20
TA = –40°C
TA = +25°C
TA = +85°C
0
–1.0
–0.5
0
0.5
QUADRATURE PHASE ERROR (Degrees)
Figure 32. I/Q Quadrature Error Distributions
Figure 29. I/Q Gain Mismatch Distributions
Rev. 0 | Page 11 of 28
1.0
06764-032
PERCENTAGE (%)
80
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
100
ADL5387
DISTRIBUTIONS FOR fRF = 900 MHz
80
60
40
60
40
20
31
32
33
34
35
INPUT IP3 (dBm)
0
55
06764-033
100
TA = –40°C
TA = +25°C
TA = +85°C
80
60
40
20
11
12
13
14
15
INPUT P1dB (dBm)
40
0
13.0
14.0
14.5
15.0
15.5
16.0
Figure 37. Noise Figure Distributions
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
80
60
40
TA = –40°C
TA = +25°C
TA = +85°C
60
40
20
20
0
–0.2
13.5
NOISE FIGURE (dB)
–0.1
0
0.1
I/Q GAIN MISMATCH (dB)
0.2
06764-035
PERCENTAGE (%)
80
TA = –40°C
TA = +25°C
TA = +85°C
60
Figure 34. IP1dB Distributions
100
75
20
06764-034
0
10
70
Figure 36. IIP2 Distributions for I Channel and Q Channel
PERCENTAGE (%)
PERCENTAGE (%)
80
65
INPUT IP2 (dBm)
Figure 33. IIP3 Distributions
100
60
06764-037
0
30
I CHANNEL
Q CHANNEL
06764-036
20
TA = –40°C
TA = +25°C
TA = +85°C
0
–1.0
–0.5
0
0.5
QUADRATURE PHASE ERROR (Degrees)
Figure 38. I/Q Quadrature Error Distributions
Figure 35. I/Q Gain Mismatch Distributions
Rev. 0 | Page 12 of 28
1.0
06764-038
PERCENTAGE (%)
80
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
100
ADL5387
DISTRIBUTIONS FOR fRF = 1900 MHz
PERCENTAGE (%)
80
100
TA = –40°C
TA = +25°C
TA = +85°C
80
PERCENTAGE (%)
100
60
40
20
60
40
TA = –40°C
TA = +25°C
TA = +85°C
20
27
28
29
30
31
INPUT IP3 (dBm)
0
52
06764-039
100
TA = –40°C
TA = +25°C
TA = +85°C
40
12
13
14
15
68
40
0
15.0
06764-040
11
15.5
16.0
16.5
17.0
17.5
18.0
NOISE FIGURE (dB)
Figure 43. Noise Figure Distributions
100
TA = –40°C
TA = +25°C
TA = +85°C
PERCENTAGE (%)
80
60
40
TA = –40°C
TA = +25°C
TA = +85°C
60
40
20
20
–0.1
0
0.1
I/Q GAIN MISMATCH (dB)
0.2
06764-041
PERCENTAGE (%)
66
60
Figure 40. IP1dB Distributions
0
–0.2
64
20
INPUT P1dB (dBm)
80
62
06764-043
20
100
60
TA = –40°C
TA = +25°C
TA = +85°C
80
60
0
10
58
Figure 42. IIP2 Distributions for I Channel and Q Channel
PERCENTAGE (%)
PERCENTAGE (%)
80
56
INPUT IP2 (dBm)
Figure 39. IIP3 Distributions
100
54
0
–1.0
–0.5
0
0.5
QUADRATURE PHASE ERROR (Degrees)
Figure 44. I/Q Quadrature Error Distributions
Figure 41. I/Q Gain Mismatch Distributions
Rev. 0 | Page 13 of 28
1.0
06764-044
0
26
06764-042
I CHANNEL
Q CHANNEL
ADL5387
CIRCUIT DESCRIPTION
The ADL5387 can be divided into five sections: the local
oscillator (LO) interface, the RF voltage-to-current (V-to-I)
converter, the mixers, the differential emitter follower outputs,
and the bias circuit. A detailed block diagram of the device is
shown in Figure 45.
BIAS
The ADL5387 has two double-balanced mixers: one for the
in-phase channel (I channel) and one for the quadrature channel
(Q channel). These mixers are based on the Gilbert cell design
of four cross-connected transistors. The output currents from
the two mixers are summed together in the resistive loads that
then feed into the subsequent emitter follower buffers.
ILO
LOIP
RFIN
The differential RF input signal is applied to a resistively
degenerated common base stage, which converts the differential
input voltage to output currents. The output currents then
modulate the two half-frequency LO carriers in the mixer stage.
MIXERS
IHI
RFIP
V-TO-I CONVERTER
DIVIDE-BY-TWO
QUADRATURE
PHASE SPLITTER
EMITTER FOLLOWER BUFFERS
The output emitter followers drive the differential I and Q
signals off-chip. The output impedance is set by on-chip 25 Ω
series resistors that yield a 50 Ω differential output impedance
for each baseband port. The fixed output impedance forms a
voltage divider with the load impedance that reduces the effective
gain. For example, a 500 Ω differential load has 1 dB lower
effective gain than a high (10 kΩ) differential load impedance.
LOIN
QLO
06764-045
QHI
Figure 45. Block Diagram
The LO interface generates two LO signals at 90° of phase
difference to drive two mixers in quadrature. RF signals are
converted into currents by the V-to-I converters that feed into
the two mixers. The differential I and Q outputs of the mixers
are buffered via emitter followers. Reference currents to each
section are generated by the bias circuit. A detailed description
of each section follows.
LO INTERFACE
The LO interface consists of a buffer amplifier followed by a
frequency divider that generate two carriers at half the input
frequency and in quadrature with each other. Each carrier is
then amplified and amplitude-limited to drive the doublebalanced mixers.
BIAS CIRCUIT
A band gap reference circuit generates the proportional-toabsolute temperature (PTAT) as well as temperature-independent
reference currents used by different sections. The mixer current
can be reduced via an external resistor between the BIAS pin
and ground. When the BIAS pin is open, the mixer runs at
maximum current and hence the greatest dynamic range. The
mixer current can be reduced by placing a resistance to ground;
therefore, reducing overall power consumption, noise figure,
and IIP3. The effect on each of these parameters is shown in
Figure 10, Figure 13, and Figure 14.
Rev. 0 | Page 14 of 28
ADL5387
APPLICATIONS INFORMATION
BASIC CONNECTIONS
LO INPUT
8
LOIP
9
LOIN
POWER SUPPLY
06764-047
1000pF
Figure 47 shows the basic connections schematic for the ADL5387.
1000pF
The nominal voltage supply for the ADL5387 is 5 V and is
applied to the VPA, VPB, VPL, and VPX pins. Ground should
be connected to the COM, CML, and CMRF pins. Each of
the supply pins should be decoupled using two capacitors;
recommended capacitor values are 100 pF and 0.1 μF.
Figure 46. Single-Ended LO Drive
The recommended LO drive level is between −6 dBm and
+6 dBm. The LO frequency at the input to the device should be
twice that of the desired LO frequency at the mixer core. The
applied LO frequency range is between 100 MHz and 4 GHz.
LOCAL OSCILLATOR (LO) INPUT
The LO port is driven in a single-ended manner. The LO signal
must be ac-coupled via a 1000 pF capacitor directly into LOIP,
and LOIN is ac-coupled to ground also using a 1000 pF capacitor.
The LO port is designed for a broadband 50 Ω match and
therefore exhibits excellent return loss from 100 MHz to 4 GHz.
The LO return loss can be seen in Figure 20. Figure 46 shows
the LO input configuration.
ETC1-1-13
RFC
120nH
24
23
22
21
20
19
CMRF
CMRF
RFIP
RFIN
CMRF
VPX
VPOS
1 VPA
VPB 17
3 BIAS
QHI 16
ADL5387
4 VPL
QHI
QLO 15
5 VPL
IHI 14
6 VPL
ILO 13
LOIP
LOIN
CML
CML
COM
100pF
CML
0.1µF
0.1µF
100pF
2 COM
VPOS
VPOS
VPB 18
100pF
7
8
9
10
11
12
QLO
IHI
ILO
1000pF
1000pF
LO
Figure 47. Basic Connections Schematic for ADL5387
Rev. 0 | Page 15 of 28
06764-046
0.1µF
120nH
1000pF 1000pF
ADL5387
120nH
21 RFIN
1000pF
ETC1-1-13
1000pF
22 RFIP
RF INPUT
06764-048
120nH
–10
–12
–14
–16
–18
–20
–22
–24
–28
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
FREQUENCY (GHz)
1.8
2.0
06764-049
–26
Figure 49. Differential RF Port Return Loss
BASEBAND OUTPUTS
The baseband outputs QHI, QLO, IHI, and ILO are fixed
impedance ports. Each baseband pair has a 50 Ω differential
output impedance. The outputs can be presented with differential
loads as low as 200 Ω (with some degradation in linearity and
gain) or high impedance differential loads (500 Ω or greater
impedance yields the same excellent linearity) that is typical of
an ADC. The TCM9-1 9:1 balun converts the differential IF
output to single-ended. When loaded with 50 Ω, this balun
presents a 450 Ω load to the device. The typical maximum
linear voltage swing for these outputs is 2 V p-p differential.
The bias level on these pins is equal to VPOS − 2.8 V. The
output 3 dB bandwidth is 240 MHz. Figure 50 shows the
baseband output configuration.
Figure 48. RF Input
QHI 16
QHI
QLO 15
QLO
IHI 14
IHI
ILO 13
ILO
06764-050
The RF inputs have a differential input impedance of
approximately 50 Ω. For optimum performance, the RF port
should be driven differentially through a balun. The recommended
balun is M/A-COM ETC1-1-13. The RF inputs to the device
should be ac-coupled with 1000 pF capacitors. Ground-referenced
choke inductors must also be connected to RFIP and RFIN
(recommended value = 120 nH, Coilcraft 0402CS-R12XJL) for
appropriate biasing. Several important aspects must be taken
into account when selecting an appropriate choke inductor for
this application. First, the inductor must be able to handle the
approximately 40 mA of standing dc current being delivered
from each of the RF input pins (RFIP, RFIN). (The suggested
0402 inductor has a 50 mA current rating). The purpose of the
choke inductors is to provide a very low resistance dc path to
ground and high ac impedance at the RF frequency so as not to
affect the RF input impedance. A choke inductor that has a selfresonant frequency greater than the RF input frequency ensures
that the choke is still looking inductive and therefore has a more
predictable ac impedance (jωL) at the RF frequency. Figure 48
shows the RF input configuration.
The differential RF port return loss has been characterized as
shown in Figure 49.
S(1, 1) (dB)
RF INPUT
Figure 50. Baseband Output Configuration
Rev. 0 | Page 16 of 28
ADL5387
Figure 52 shows the EVM performance of the ADL5387 when
ac-coupled, with an IEEE 802.16e WiMAX signal.
The ADL5387 shows excellent EVM performance for various
modulation schemes. Figure 51 shows typical EVM performance
over input power range for a point-to-point application with
16 QAM modulation schemes and zero-IF baseband. The
differential dc offsets on the ADL5387 are in the order of a
few mV. However, ac coupling the baseband outputs with 10 μF
capacitors helps to eliminate dc offsets and enhances EVM
performance. With a 10 MHz BW signal, 10 μF ac coupling
capacitors with the 500 Ω differential load results in a high-pass
corner frequency of ~64 Hz which absorbs an insignificant
amount of modulated signal energy from the baseband signal.
By using ac coupling capacitors at the baseband outputs, the dc
offset effects, which can limit dynamic range at low input power
levels, can be eliminated.
0
–5
–10
–15
–25
–30
–35
–40
–45
–50
–50
–30
–20
–10
0
10
20
Figure 52. RF = 750MHz MHz, IF = 0 Hz, EVM vs. Input Power for a 16 QAM
10 MHz Bandwidth Mobile WiMAX Signal (AC-Coupled Baseband Outputs)
Figure 53 exhibits the zero IF EVM performance of a WCDMA
signal over a wide RF input power range.
0
–5
0
–10
–15
EVM (dB)
–10
–15
–20
–20
–25
–30
–25
–35
–40
–40
–45
–70
–60
–50
–40
–30
–20
INPUT POWER (dBm)
–60
–50
–40
–30
–20
INPUT POWER (dBm)
–10
0
10
06764-051
–45
–10
0
10
06764-053
–35
–30
–50
–70
–40
INPUT POWER (dBm)
–5
EVM (dB)
–20
06764-052
EVM is a measure used to quantify the performance of a digital
radio transmitter or receiver. A signal received by a receiver
would have all constellation points at the ideal locations; however,
various imperfections in the implementation (such as carrier
leakage, phase noise, and quadrature error) cause the actual
constellation points to deviate from the ideal locations.
EVM (dB)
ERROR VECTOR MAGNITUDE (EVM)
PERFORMANCE
Figure 53. RF = 1950 MHz, IF = 0 Hz, EVM vs. Input Power for a WCDMA
(AC-Coupled Baseband Outputs)
Figure 51. RF = 140 MHz, IF = 0 Hz, EVM vs. Input Power for a 16 QAM
10 Msym/s Signal (AC-Coupled Baseband Outputs)
Rev. 0 | Page 17 of 28
ADL5387
COSωLOt
0°
ωIF
ωIF
–ωIF
0
+ωIF
–90°
0
+ωIF
0
+ωIF
+90°
ωLSB
ωLO
ωUSB
0°
0
+ωIF
06764-054
–ωIF
SINωLOt
Figure 54. Illustration of the Image Problem
LOW IF IMAGE REJECTION
EXAMPLE BASEBAND INTERFACE
The image rejection ratio is the ratio of the intermediate
frequency (IF) signal level produced by the desired input
frequency to that produced by the image frequency. The image
rejection ratio is expressed in decibels. Appropriate image
rejection is critical because the image power can be much
higher than that of the desired signal, thereby plaguing the
down conversion process. Figure 54 illustrates the image
problem. If the upper sideband (lower sideband) is the desired
band, a 90° shift to the Q channel (I channel) cancels the image
at the lower sideband (upper sideband).
In most direct conversion receiver designs, it is desirable to
select a wanted carrier within a specified band. The desired
channel can be demodulated by tuning the LO to the appropriate
carrier frequency. If the desired RF band contains multiple
carriers of interest, the adjacent carriers would also be down
converted to a lower IF frequency. These adjacent carriers can
be problematic if they are large relative to the wanted carrier as
they can overdrive the baseband signal detection circuitry. As a
result, it is often necessary to insert a filter to provide sufficient
rejection of the adjacent carriers.
Figure 55 shows the excellent image rejection capabilities of the
ADL5387 for low IF applications, such as CDMA2000. The
ADL5387 exhibits image rejection greater than 45 dB over the
broad frequency range for an IF = 1.23 MHz.
It is necessary to consider the overall source and load impedance
presented by the ADL5387 and ADC input to design the filter
network. The differential baseband output impedance of the
ADL5387 is 50 Ω. The ADL5387 is designed to drive a high
impedance ADC input. It may be desirable to terminate the
ADC input down to lower impedance by using a terminating
resistor, such as 500 Ω. The terminating resistor helps to better
define the input impedance at the ADC input. The order and
type of filter network depends on the desired high frequency
rejection required, pass-band ripple, and group delay. Filter
design tables provide outlines for various filter types and orders,
illustrating the normalized inductor and capacitor values for a
1 Hz cutoff frequency and 1 Ω load. After scaling the normalized
prototype element values by the actual desired cut-off frequency
and load impedance, the series reactance elements are halved to
realize the final balanced filter network component values.
–10
–20
–30
–40
–50
–60
–70
50
250
450
650
850
1050 1250 1450 1650 1850
RF INPUT FREQUENCY (MHz)
06764-055
IMAGE REJECTION AT 1.23MHz (dB)
0
Figure 55. Image Rejection vs.
RF Input Frequency for a CDMA2000 Signal, IF = 1.23 MHz
Rev. 0 | Page 18 of 28
ADL5387
The balanced configuration is realized as the 0.54 μH inductor
is split in half to realize the network shown in Figure 56.
LN = 0.074H
NORMALIZED
SINGLE-ENDED
CONFIGURATION
VS
CN
14.814F
RS
= 0.1
RL
RS = 50Ω
5
0
–5
–10
–15
RL= 500Ω
–20
fC = 1Hz
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
FREQUENCY (MHz)
0.54µH
DENORMALIZED
SINGLE-ENDED
EQUIVALENT
VS
10
06764-157
RS = 50Ω
Figure 57 and Figure 58 show the measured frequency response
and group delay of the filter.
MAGNITUDE RESPONSE (dB)
As an example, a second-order, Butterworth, low-pass filter
design is shown in Figure 56 where the differential load impedance
is 500 Ω, and the source impedance of the ADL5387 is 50 Ω.
The normalized series inductor value for the 10-to-1, load-tosource impedance ratio is 0.074 H, and the normalized shunt
capacitor is 14.814 F. For a 10.9 MHz cutoff frequency, the
single-ended equivalent circuit consists of a 0.54 μH series
inductor followed by a 433 pF shunt capacitor.
Figure 57. Baseband Filter Response
433pF
900
RL= 500Ω
800
0.27µH
RL
2 = 250Ω
RL
= 250Ω
2
600
500
400
Figure 56. Second-Order, Butterworth, Low-Pass Filter Design Example
300
A complete design example is shown in Figure 59. A sixth-order
Butterworth differential filter having a 1.9 MHz corner frequency
interfaces the output of the ADL5387 to that of an ADC input.
The 500 Ω load resistor defines the input impedance of the
ADC. The filter adheres to typical direct conversion WCDMA
applications, where 1.92 MHz away from the carrier IF frequency,
1 dB of rejection is desired and 2.7 MHz away 10 dB of rejection
is desired.
200
Rev. 0 | Page 19 of 28
100
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
FREQUENCY (MHz)
Figure 58. Baseband Filter Group Delay
1.8
06764-158
RS
= 25Ω
2
433pF
DELAY (ns)
BALANCED
CONFIGURATION
VS
700
0.27µH
06764-056
RS
= 25Ω
2
fC = 10.9MHz
ADL5387
ETC1-1-13
120nH
20
19
RFIP
RFIN
CMRF
VPX
1 VPA
100pF
100pF
2 COM
27µH
27µH
10µH
27µH
27µH
10µH
QHI 16
ADL5387
4 VPL
QLO 15
5 VPL
IHI 14
6 VPL
ILO 13
CML
CML
COM
8
9
10
11
12
1000pF
CAC
10µF
1000pF
27µH
27µH
10µH
06764-159
LO
68pF
LOIN
7
CAC
10µF
91pF
LOIP
100pF
CML
0.1µF
0.1µF
VPB 17
3 BIAS
VPOS
CAC
10µF
270pF
0.1µF
VPOS
VPB 18
ADC INPUT
21
10µH
ADC INPUT
22
27µH
500Ω
23
27µH
500Ω
24
CMRF
VPOS
CMRF
CAC
10µF
68pF
1000pF 1000pF
91pF
120nH
270pF
RFC
Figure 59. Sixth Order Low-Pass Butterworth Baseband Filter Schematic
Rev. 0 | Page 20 of 28
ADL5387
CHARACTERIZATION SETUPS
10 MHz. For the case where a blocker was applied, the output
blocker was at 15 MHz baseband frequency. Note that great care
must be taken when measuring NF in the presence of a blocker.
The RF blocker generator must be filtered to prevent its noise
(which increases with increasing generator output power) from
swamping the noise contribution of the ADL5387. At least
30 dB of attention at the RF and image frequencies is desired.
For example, with a 2xLO of 1848 MHz applied to the ADL5387,
the internal 1xLO is 924 MHz. To obtain a 15 MHz output
blocker signal, the RF blocker generator is set to 939 MHz and
the filters tuned such that there is at least 30 dB of attenuation
from the generator at both the desired RF frequency (934 MHz)
and the image RF frequency (914 MHz). Finally, the blocker
must be removed from the output (by the 10 MHz low-pass
filter) to prevent the blocker from swamping the analyzer.
Figure 60 to Figure 62 show the general characterization bench
setups used extensively for the ADL5387. The setup shown in
Figure 62 was used to do the bulk of the testing and used sinusoidal
signals on both the LO and RF inputs. An automated AgilentVEE program was used to control the equipment over the IEEE
bus. This setup was used to measure gain, IP1dB, IIP2, IIP3, I/Q
gain match, and quadrature error. The ADL5387 characterization
board had a 9-to-1 impedance transformer on each of the
differential baseband ports to do the differential-to-singleended conversion.
The two setups shown in Figure 60 and Figure 61 were used
for making NF measurements. Figure 60 shows the setup for
measuring NF with no blocker signal applied while Figure 61
was used to measure NF in the presence of a blocker. For both
setups, the noise was measured at a baseband frequency of
SNS
RF
ADL5387
VPOS CHAR BOARD
I
LO
INPUT
6dB PAD
HP 6235A
POWER SUPPLY
R1
50Ω
AGILENT N8974A
NOISE FIGURE ANALYZER
LOW-PASS
FILTER
IEEE
GND
Q
FROM SNS PORT
CONTROL
OUTPUT
AGILENT 8665B
SIGNAL GENERATOR
PC CONTROLLER
Figure 60. General Noise Figure Measurement Setup
Rev. 0 | Page 21 of 28
06764-057
IEEE
ADL5387
BAND-PASS
TUNABLE FILTER
BAND-REJECT
TUNABLE FILTER
6dB PAD
R&S SMT03
SIGNAL GENERATOR
RF
GND
ADL5387
6dB PAD
VPOS CHAR BOARD
LOW-PASS
FILTER
I
LO
6dB PAD
HP 6235A
POWER SUPPLY
R&S FSEA30
SPECTRUM ANALYZER
R1
50Ω
Q
HP87405
LOW NOISE
PREAMP
06764-058
BAND-PASS
CAVITY FILTER
AGILENT 8665B
SIGNAL GENERATOR
Figure 61. Measurement Setup for Noise Figure in the Presence of a Blocker
3dB PAD
RF
AMPLIFIER
3dB PAD IN
RF
OUT 3dB PAD
IEEE
VP GND
3dB PAD
AGILENT
11636A
R&S SMT-06
6dB PAD
IEEE
RF
SWITCH
MATRIX
VPOS CHAR BOARD
LO
I 6dB PAD
IEEE
6dB PAD
AGILENT E3631
PWER SUPPLY
RF
INPUT
AGILENT E8257D
SIGNAL GENERATOR
IEEE
PC CONTROLLER
IEEE
R&S FSEA30
SPECTRUM ANALYZER
Figure 62. General ADL5387 Characterization Setup
Rev. 0 | Page 22 of 28
HP 8508A
VECTOR VOLTMETER
06764-059
IEEE
Q 6dB PAD
ADL5387
IEEE
RF
GND
INPUT CHANNELS
A AND B
R&S SMT-06
ADL5387
EVALUATION BOARD
The ADL5387 evaluation board is available. The board can be
used for single-ended or differential baseband analysis. The default
configuration of the board is for single-ended baseband analysis.
T1
RFC
C1
23
22
21
20
19
CMRF
RFIP
RFIN
CMRF
VPX
R7
R6
VPB 18
VPOS
C8
C2
2 COM
VPB 17
3 BIAS
QHI 16
ADL5387
4 VPL
C9
R9
Q OUTPUT OR QHI
R14
R15
QLO 15
T2
C12
R3
5 VPL
IHI 14
6 VPL
ILO 13
R16
LOIP
LOIN
CML
CML
COM
C4
CML
7
8
9
10
11
12
QLO
R10
R11
R4
C5
I OUTPUT OR IHI
R5
T3
C13
R13
C6
R17
C7
ILO
R12
T4
06764-060
C3
L1
24
1 VPA
R2
VPOS
C10
CMRF
VPOS
R1
C11
L2
R8
LO
Figure 63. Evaluation Board Schematic
Rev. 0 | Page 23 of 28
ADL5387
Table 4. Evaluation Board Configuration Options
Component
VPOS, GND
R1, R3, R6
C1, C2, C3,
C4, C8, C9
C5, C6, C7,
C10, C11
R4, R5,
R9 to R16
L1, L2,
R7, R8
T2, T3
C12, C13
R17
T1
R2
Function
Power Supply and Ground Vector Pins.
Power Supply Decoupling. Shorts or power supply decoupling resistors.
The capacitors provide the required dc coupling up to 2 GHz.
AC Coupling Capacitors. These capacitors provide the required ac coupling from
50 MHz to 2 GHz.
Single-Ended Baseband Output Path. This is the default configuration of the evaluation
board. R14 to R16 and R4, R5, and R13 are populated for appropriate balun interface.
R9, R10 and R11, R12 are not populated. Baseband outputs are taken from QHI and IHI.
The user can reconfigure the board to use full differential baseband outputs. R9 to R12
provide a means to bypass the 9:1 TCM9-1 transformer to allow for differential baseband
outputs. Access the differential baseband signals by populating R9 to R12 with 0 Ω and
not populating R4, R5, R13 to R16. This way the transformer does not need to be removed.
The baseband outputs are taken from the SMAs of Q_HI, Q_LO, I_HI, and I_LO.
Input Biasing. Inductance and resistance sets the input biasing of the common base
input stage. Default value is 120 nH.
IF Output Interface. TCM9-1 converts a differential high impedance IF output to a singleended output. When loaded with 50 Ω, this balun presents a 450 Ω load to the device.
The center tap can be decoupled through a capacitor to ground.
Decoupling Capacitors. C12 and C13 are the decoupling capacitors used to reject noise
on the center tap of the TCM9-1.
LO Input Interface. The LO is driven as a single-ended signal. Although, there is no
performance change for a differential signal drive, the option is available by placing a
transformer (T4, ETC1-1-13) on the LO input path.
RF Input Interface. ETC1-1-13 is a 1:1 RF balun that converts the single-ended RF input
to differential signal.
RBIAS. Optional bias setting resistor. See the Bias Circuit section to see how to use this feature.
Rev. 0 | Page 24 of 28
Default Condition
Not Applicable
R1, R3, R6 = 0 Ω (0805)
C2, C4, C8 = 100 pF (0402)
C1, C3, C9 = 0.1 μF (0603)
C5, C6, C10, C11 = 1000 pF (0402),
C7 = Open
R4, R5, R13 to R16 = 0 Ω (0402),
R9 to R12 = Open
L1, L2 = 120 nH (0402)
R7, R8 = 0 Ω (0402)
T2, T3 = TCM9-1, 9:1 (Mini-Circuits)
C12, C13 = 0.1 μF (0402)
R17 = 0 Ω (0402)
T1 = ETC1-1-13, 1:1 (M/A COM)
R2 = Open
06764-166
06764-164
ADL5387
Figure 66. Evaluation Board Bottom Layer
06764-165
06764-167
Figure 64. Evaluation Board Top Layer
Figure 67. Evaluation Board Bottom Layer Silkscreen
Figure 65. Evaluation Board Top Layer Silkscreen
Rev. 0 | Page 25 of 28
ADL5387
OUTLINE DIMENSIONS
0.60 MAX
4.00
BSC SQ
PIN 1
INDICATOR
0.60 MAX
TOP
VIEW
0.50
BSC
3.75
BSC SQ
0.50
0.40
0.30
1.00
0.85
0.80
12° MAX
SEATING
PLANE
0.80 MAX
0.65 TYP
0.30
0.23
0.18
PIN 1
INDICATOR
19
18
24 1
*2.45
2.30 SQ
2.15
EXPOSED
PAD
(BOTTOMVIEW)
13
12
7
6
0.23 MIN
2.50 REF
0.05 MAX
0.02 NOM
0.20 REF
COPLANARITY
0.08
*COMPLIANT TO JEDEC STANDARDS MO-220-VGGD-2
EXCEPT FOR EXPOSED PAD DIMENSION
Figure 68. 24-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
4 mm × 4 mm Body, Very Thin Quad
(CP-24-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADL5387ACPZ-R7 1
ADL5387ACPZ-WP1
ADL5387-EVALZ1
1
Temperature Range
–40°C to +85°C
–40°C to +85°C
Package Description
24-Lead LFCSP_VQ, 7” Tape and Reel
24-Lead LFCSP_VQ, Waffle Pack
Evaluation Board
Z = RoHS Compliant Part.
Rev. 0 | Page 26 of 28
Package Option
CP-24-2
CP-24-2
Ordering Quantity
1,500
64
ADL5387
NOTES
Rev. 0 | Page 27 of 28
ADL5387
NOTES
©2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06764-0-10/07(0)
Rev. 0 | Page 28 of 28