LINER LTC1735CS High efficiency synchronous step-down switching regulator Datasheet

LTC1735
High Efficiency
Synchronous Step-Down
Switching Regulator
DESCRIPTIO
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FEATURES
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Dual N-Channel MOSFET Synchronous Drive
Synchronizable/Programmable Fixed Frequency
Wide VIN Range: 3.5V to 36V Operation
VOUT Range: 0.8V to 6V
OPTI-LOOPTM Compensation Minimizes COUT
±1% Output Voltage Accuracy
Internal Current Foldback
Output Overvoltage Crowbar Protection
Latched Short-Circuit Shutdown Timer
with Defeat Option
Very Low Dropout Operation: 99% Duty Cycle
Forced Continuous Control Pin
Optional Programmable Soft-Start
Remote Output Voltage Sense
Logic Controlled Micropower Shutdown: IQ < 25µA
LTC1435 Pin Compatible with
Minor Component Changes
Available in 16-Lead Narrow SSOP and SO Packages
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APPLICATIO S
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Notebook and Palmtop Computers, PDAs
Cellular Telephones and Wireless Modems
DC Power Distribution Systems
The operating frequency (synchronizable up to 500kHz) is
set by an external capacitor allowing maximum flexibility
in optimizing efficiency. A forced continuous control pin
reduces noise and RF interference and can assist secondary winding regulation by disabling Burst Mode operation
when the main output is lightly loaded.
Protection features include internal foldback current limiting, output overvoltage crowbar and optional shortcircuit shutdown. Soft-start is provided by an external
capacitor that can be used to properly sequence supplies.
The operating current level is user-programmable via an
external current sense resistor. Wide input supply range
allows operation from 3.5V to 30V (36V maximum).
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode and OPTI-LOOP are trademarks of Linear Technology Corporation.
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The LTC®1735 is a synchronous step-down switching
regulator controller that drives external N-channel power
MOSFETs using a fixed frequency architecture. Burst
ModeTM operation provides high efficiency at low load
currents. The precision 0.8V reference is compatible with
future microprocessor generations. OPTI-LOOP compensation allows the transient response to be optimized over
a wide range of output capacitance and ESR values.
TYPICAL APPLICATIO
CIN
22µF
50V
COSC
COSC
47pF
CSS
0.1µF
TG
RUN/SS
M1
FDS6680A
COUT: PANASONIC EEFUEOG181R
CIN: MARCON THCR70E1H226ZT
L1: PANASONIC ETQP6FZR0HFA
RSENSE: IRC LRF2010-01-R005J
BOOST
ITH
CC
330pF
RC
33k
CB
0.22µF
SW
LTC1735
CC2
100pF
VIN
5V TO 24V
L1
2µH
DB
CMDSH-3
VIN
RSENSE
0.005Ω
VOUT
1.6V
9A
SGND
100pF
INTVCC
VOSENSE
4.7µF
BG
SENSE –
1000pF
R1
20k
1%
+
M2
FDS6680A
D1
MBRS340T3
PGND
R2
20k
1%
+
COUT
180µF
4V
×4
SP
SENSE +
1735 F01
Figure 1. High Efficiency Step-Down Converter
1
LTC1735
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W W
W
ABSOLUTE
AXI U RATI GS
U
W
U
PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Supply Voltage (VIN).........................36V to – 0.3V
Topside Driver Supply Voltage (BOOST)....42V to – 0.3V
Switch Voltage (SW) ....................................36V to – 5V
EXTVCC Voltage ...........................................7V to – 0.3V
Boosted Driver Voltage (BOOST – SW) .......7V to – 0.3V
SENSE +, SENSE – Voltages .......... 1.1 (INTVCC) to – 0.3V
FCB Voltage ............................(INTVCC + 0.3V) to – 0.3V
ITH, VOSENSE Voltages ...............................2.7V to – 0.3V
RUN/SS Voltages .........................................7V to – 0.3V
Peak Driver Output Current <10µs (TG, BG) .............. 3A
INTVCC Output Current ......................................... 50mA
Operating Ambient Temperature Range
LTC1735C ............................................... 0°C to 85°C
LTC1735I ............................................ – 40°C to 85°C
Junction Temperature (Note 2) ............................. 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
COSC 1
RUN/SS 2
16 TG
15 BOOST
ITH 3
14 SW
FCB 4
13 VIN
SGND 5
12 INTVCC
VOSENSE 6
11 BG
SENSE –
10 PGND
7
SENSE + 8
GN PACKAGE
16-LEAD NARROW
PLASTIC SSOP
LTC1735CGN
LTC1735CS
LTC1735IGN
LTC1735IS
9
EXTVCC
GN PART MARKING
1735
1735I
S PACKAGE
16-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 130°C/W (GN)
TJMAX = 125°C, θJA = 110°C/W (S)
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
–4
– 25
nA
Main Control Loop
IVOSENSE
Feedback Current
(Note 3)
VOSENSE
Feedback Voltage
(Note 3)
∆VLINEREG
Reference Voltage Line Regulation
VIN = 3.6V to 30V (Note 3)
∆VLOADREG
Output Voltage Load Regulation
(Note 3)
Measured in Servo Loop; VITH = 0.7V
Measured in Servo Loop; VITH = 2V
●
0.792
●
●
98
0.808
V
0.02
%/V
0.1
– 0.1
0.3
– 0.3
%
%
DF Max
Maximum Duty Factor
gm
Transconductance Amplifier gm
VFCB
Forced Continuous Threshold
IFCB
Forced Continuous Current
VOVL
Feedback Overvoltage Lockout
IQ
Input DC Supply Current
Normal Mode
Shutdown
(Note 4)
VRUN/SS
Run Pin Start Threshold
VRUN/SS, Ramping Positive
VRUN/SS
Run Pin Begin Latchoff Threshold
VRUN/SS, Ramping Positive
IRUN/SS
Soft-Start Charge Current
VRUN/SS = 0V
– 0.7
– 1.2
ISCL
RUN/SS Discharge Current
Soft Short Condition, VOSENSE = 0.5V,
VRUN/SS = 4.5V
0.5
2
4
µA
UVLO
Undervoltage Lockout
Measured at VIN Pin (VIN Ramping Down)
●
3.5
3.9
V
∆VSENSE(MAX)
Maximum Current Sense Threshold
VOSENSE = 0.7V
●
75
85
mV
2
In Dropout
0.8
0.001
99.4
%
1.3
●
0.76
VFCB = 0.85V
●
0.84
VRUN/SS = 0V
1.0
60
mmho
0.8
0.84
V
– 0.17
– 0.3
µA
0.86
0.88
V
450
15
25
µA
µA
1.5
1.9
V
4.1
4.5
V
µA
LTC1735
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
ISENSE
Sense Pins Total Source Current
VSENSE–
60
80
µA
tON(MIN)
Minimum On-Time
Tested with a Square Wave (Note 6)
160
200
ns
TG tr
TG tf
TG Transition Time:
Rise Time
Fall Time
(Note 7)
CLOAD = 3300pF
CLOAD = 3300pF
50
50
90
90
ns
ns
BG tr
BG tf
BG Transition Time:
Rise Time
Fall Time
(Note 7)
CLOAD = 3300pF
CLOAD = 3300pF
50
40
90
80
ns
ns
TG/BG t1D
Top Gate Off to Synchronous
Gate On Delay Time
CLOAD = 3300pF Each Driver
100
ns
TG/BG t2D
Synchronous Gate Off to Top
Gate On Delay Time
CLOAD = 3300pF Each Driver
70
ns
= VSENSE+
MIN
= 0V
TYP
MAX
UNITS
Internal VCC Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 30V, VEXTVCC = 4V
5.2
5.4
V
VLDO(INT)
Internal VCC Load Regulation
ICC = 0 to 20mA, VEXTVCC = 4V
5.0
0.2
1
%
VLDO(EXT)
EXTVCC Drop Voltage
ICC = 20mA, VEXTVCC = 5V
130
200
mV
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, EXTVCC Ramping Positive
VEXTVCC(HYS)
EXTVCC Hysteresis
●
4.5
4.7
V
0.2
V
Oscillator
fOSC
Oscillator Frequency
fH/fOSC
Maximum Sync Frequency Ratio
fFCB(SYNC)
FCB Pin Threshold For Sync
COSC = 43pF (Note 5)
265
300
335
kHz
1.3
Ramping Negative
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC1735CS, LTC1735IS: TJ = TA + (PD • 110 °C/W)
LTC1735CGN, LTC1735IGN: TJ = TA + (PD • 130°C/W)
Note 3: The LTC1735 is tested in a feedback loop that servos VOSENSE to
the balance point for the error amplifier (VITH = 1.2V).
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
0.9
1.2
V
Note 5: Oscillator frequency is tested by measuring the COSC charge
current (IOSC) and applying the formula:
–1
 8.477(1011)   1
1 
fOSC = 
+

 COSC (pF) + 11  ICHG IDIS 


Note 6: The minimum on-time condition corresponds to an inductor peakto-peak ripple current ≥40% of IMAX (see Minimum On-Time
Considerations in the Applications Information section).
Note 7: Rise and fall times are measured using 10% and 90% levels.
Delay times are measured using 50% levels.
3
LTC1735
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Load Current
(3 Operating Modes)
Efficiency vs Load Current
100
EXTVCC OPEN
90
SYNC
EFFICIENCY (%)
EFFICIENCY (%)
90
BURST
80
70
CONT
60
50
40
VIN = 10V
VOUT = 3.3V
RS = 0.01Ω
fO = 300kHz
30
20
0.001
EXTVCC = 5V
FIGURE 1
0.1
0.01
1
LOAD CURRENT (A)
Efficiency vs Input Voltage
100
95
VIN = 5V
80
VIN = 24V
70
60
50
0.1
1
LOAD CURRENT (A)
FCB = 0V
VIN = 15V
FIGURE 1
IOUT = 5A
85
IOUT = 0.5A
80
–0.2
–0.3
–0.4
30
200
0
2
0
6
4
LOAD CURRENT (A)
8
0
10
2
4
6
LOAD CURRENT (A)
8
EXTVCC Switch Drop
vs INTVCC Load Current
INTVCC Line Regulation
100
6
60
200
40
SHUTDOWN
20
400
EXTVCC – INTVCC (mV)
300
5
INTVCC VOLTAGE (V)
80
SHUTDOWN CURRENT (µA)
400
500
1mA LOAD
EXTVCC OPEN
10
1735 G06
1735 G05
Input and Shutdown Currents
vs Input Voltage
INPUT CURRENT (µA)
300
100
1735 G04
100
RSENSE = 0.005Ω
VOUT = 5V – 5% DROP
400
75
500
30
–0.1
VIN – VOUT (mV)
NORMALIZED VOUT (%)
EFFICIENCY (%)
90
25
25
10
15
20
INPUT VOLTAGE (V)
5
VIN – VOUT Dropout Voltage
vs Load Current
500
0
EXTVCC OPEN
VOUT = 1.6V
95 FIGURE 1
10
15
20
INPUT VOLTAGE (V)
0
1735 G03
Load Regulation
100
5
IOUT = 0.5A
80
1735 G02
Efficiency vs Input Voltage
0
IOUT = 5A
85
70
10
1735 G01
70
90
75
40
0.01
10
EXTVCC = 5V
VOUT = 1.6V
FIGURE 1
VIN = 15V
EFFICIENCY (%)
100
4
3
2
300
200
100
1
EXTVCC = 5V
0
0
0
5
20
15
10
25
INPUT VOLTAGE (V)
30
35
1735 G07
4
0
0
5
20
15
25
10
INPUT VOLTAGE (V)
30
35
1735 G08
0
0
10
30
40
20
INTVCC LOAD CURRENT (mA)
50
1735 G09
LTC1735
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TYPICAL PERFOR A CE CHARACTERISTICS
70
60
50
40
30
20
10
0
50
25
75
NORMALIZED OUTPUT VOLTAGE (%)
80
VSENSE(CM) = 1.6V
60
40
20
0
0
100
1
2
3
4
5
70
60
50
40
30
20
10
0
–10
–20
1.5
VITH (V)
2
2.5
2.0
70
1.0
65
0.5
60
–40 –15
0
85
10
35
60
TEMPERATURE (°C)
110
135
ITH VOLTAGE (V)
CONTINUOUS
MODE
1.5
SYNCHRONIZED f = fO
1.0
Burst Mode
OPERATION
–50
0.5
0
6
1735 G16
1
2
5
3
4
VRUN/SS (V)
0
1
2
3
4
LOAD CURRENT (A)
6
1735 G15
AVERAGE OUTPUT CURRENT IOUT/IMAX (%)
2.0
VSENSE COMMON MODE VOLTAGE (V)
0
Output Current vs Duty Cycle
VIN = 10V
VOUT = 3.3V
RSENSE = 0.01Ω
fO = 300kHz
50
ISENSE (µA)
1.5
ITH Voltage vs Load Current
0
5
VOSENSE = 0.7V
VSENSE(CM) = 1.6V
2.5
4
1
3
4
2
COMMON MODE VOLTAGE (V)
0
VITH vs VRUN/SS
75
SENSE Pins Total Source Current
2
60
1735 G18
100
0
64
2.5
80
1735 G13
–100
68
1735 G12
VITH (V)
80
1
72
Maximum Current Sense Threshold
vs Temperature
MAXIMUM CURRENT SENSE THRESHOLD (mV)
MAXIMUM CURRENT SENSE THRESHOLD (mV)
90
0.5
76
1735 G11
Maximum Current Sense Threshold
vs ITH Voltage
0
6
80
VRUN/SS (V)
1735 G10
–30
Maximum Current Sense Threshold
vs Sense Common Mode Voltage
MAXIMUM CURRENT SENSE THRESHOLD (mV)
80
0
Maximum Current Sense Threshold
vs VRUN/SS
MAXIMUM CURRENT SENSE THRESHOLD (mV)
MAXIMUM CURRENT SENSE THRESHOLD (mV)
Maximum Current Sense Threshold
vs Normalized Output Voltage
(Foldback)
5
6
1735 G17
100
IOUT/IMAX (SYNC)
IOUT/IMAX
(FREE RUN)
80
60
40
20
fSYNC = fO
0
0
20
40
60
DUTY CYCLE (%)
80
100
1735 G14
5
LTC1735
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TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
vs Temperature
0
COSC = 47pF
280
270
260
250
–40
VRUN/SS = 0V
–1
RUN/SS CURRENT (µA)
FREQUENCY (kHz)
290
FCB Pin Current vs Temperature
0
–2
–3
–4
–15
60
35
10
85
TEMPERATURE (°C)
110
135
–0.4
–0.6
–0.8
–5
–40
–15
60
35
10
85
TEMPERATURE (°C)
110
1735 G19
135
–1.0
–40
–15
60
35
10
85
TEMPERATURE (°C)
1735 G20
ILOAD = 10mA
110
135
1735 G21
VOUT(RIPPLE)
(Burst Mode Operation)
VOUT(RIPPLE) (Synchronized)
Start-Up
VOUT
1V/DIV
VFCB = 0.85V
–0.2
FCB CURRENT (µA)
300
RUN/SS Pin Current
vs Temperature
FIGURE 1
VOUT
10mV/DIV
ILOAD = 50mA
FIGURE 1
VOUT
20mV/DIV
VRUN/SS
5V/DIV
IL
5A/DIV
IL
5A/DIV
VIN = 15V
VOUT = 1.6V
RLOAD = 0.16Ω
5ms/DIV
1735 G22
EXT SYNC f = fO
VIN = 15V
VOUT = 1.6V
VOUT(RIPPLE)
(Burst Mode Operation)
ILOAD = 1.5A
10µs/DIV
1735 G23
FIGURE 1
5µs/DIV
1735 G27
Load Step (Continuous Mode)
FIGURE 1
VOUT
50mV/DIV
IL
5A/DIV
IL
5A/DIV
IL
5A/DIV
1735 G24
50µs/DIV
FIGURE 1
VOUT
50mV/DIV
FCB = 5V
VIN = 15V
VOUT = 1.6V
FCB = 5V
VIN = 15V
VOUT = 1.6V
Load Step (Burst Mode Operation)
VOUT
20mV/DIV
6
IL
5A/DIV
10mA TO
9A LOAD STEP
FCB = 5V
VIN = 15V
VOUT = 1.6V
10µs/DIV
1735 G26
0A TO
9A LOAD STEP
FCB = 0V
VIN = 15V
VOUT = 1.6V
10µs/DIV
1735 G25
LTC1735
U
U
U
PI FU CTIO S
COSC (Pin 1): External capacitor COSC from this pin to
ground sets the operating frequency.
RUN/SS (Pin 2): Combination of Soft-Start and Run
Control Inputs. A capacitor to ground at this pin sets the
ramp time to full output current. The time is approximately
1.25s/µF. Forcing this pin below 1.5V causes the device to
be shutdown. In shutdown all functions are disabled.
Latchoff overcurrent protection is also invoked via this pin
as described in the Applications Information section.
ITH (Pin 3): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 2.4V.
FCB (Pin 4): Forced Continuous/Synchronization Input.
Tie this pin to ground for continuous synchronous operation, to a resistive divider from the secondary output when
using a secondary winding or to INTVCC to enable Burst
Mode operation at low load currents. Clocking this pin with
a signal above 1.5VP–P disables Burst Mode operation but
allows cycle-skipping at low load currents and synchronizes the internal oscillator with the external clock.
SGND (Pin 5): Small-Signal Ground. All small-signal
components such as COSC, CSS, the feedback divider plus
the loop compensation resistors and capacitor(s) should
single-point tie to this pin. This pin should, in turn, connect
to PGND.
VOSENSE (Pin 6): Receives the feedback voltage from an
external resistive divider across the output.
SENSE – (Pin 7): The (–) Input to the Current Comparator.
SENSE + (Pin 8): The (+) Input to the Current Comparator.
Built-in offsets between SENSE – and SENSE + pins in
conjunction with RSENSE set the current trip threshold.
EXTVCC (Pin 9): Input to the Internal Switch Connected to
INTVCC. This switch closes and supplies VCC power whenever EXTVCC is higher than 4.7V. See EXTVCC connection
in the Applications Information section. Do not exceed 7V
on this pin and ensure EXTVCC ≤ VIN.
PGND (Pin 10): Driver Power Ground. Connects to the
source of bottom N-channel MOSFET, the anode of the
Schottky diode, and the (–) terminal of CIN.
BG (Pin 11): High Current Gate Drive for Bottom
N-Channel MOSFET. Voltage swing at this pin is from
ground to INTVCC.
INTVCC (Pin 12): Output of the Internal 5.2V Regulator and
EXTVCC Switch. The driver and control circuits are powered from this voltage. Decouple to power ground with a
1µF ceramic capacitor placed directly adjacent to the IC
together with a minimum of 4.7µF tantalum or other low
ESR capacitor.
VIN (Pin 13): Main Supply Pin. Must be closely decoupled
to power ground.
SW (Pin 14): Switch Node Connection to Inductor and
Bootstrap Capacitor. Voltage swing at this pin is from a
Schottky diode (external) voltage drop below ground to
VIN.
BOOST (Pin 15): Supply to Topside Floating Driver. The
bootstrap capacitor is returned to this pin. Voltage swing
at this pin is from a diode drop below INTVCC to (VIN +
INTVCC).
TG (Pin 16): High Current Gate Drive for Top N-Channel
MOSFET. This is the output of a floating driver with a
voltage swing equal to INTVCC superimposed on the
switch node voltage SW.
7
LTC1735
W
FU CTIO AL DIAGRA
U
U
VIN
+
13 VIN
COSC
0.8V
REF
COSC 1
SGND 5
OSC
INTVCC
FC
C
–
+
1.2V
0.8V
BOOST
F
–
OV
S
R
–
0.86V
VSEC
CB
16
+
SW
BOT
0.55V
CSEC
14
SWITCH
LOGIC
TOP ON
Q
TG
TOP
DROP
OUT
DET
DB
15
+
FORCE BOT
+
UVL
4 FCB
0.17µA
SYNC
CIN
D1
B
+
–
2.4V
SD
2k
6
R1
R2
VFB
gm =1.3m
–
+
0.8V
Ω
VOSENSE
ICMP
I1
EA
0.86V
SD
6V
4(VFB)
CSS
2
–
+
+
–
+
BURST
DISABLE
FC
A
+
BOT
I2
BUFFERED
ITH
4.7V
30k
SLOPE COMP
COUT
INTVCC
12
VIN
3mV
30k
+
5.2V
LDO
REG
CINTVCC
BG
+
11
–
PGND
RC
RUN/SS
+
VOUT
IREV
–
INTVCC
RUN
SOFTSTART
+
OVERCURRENT
LATCHOFF
1.2µA
–
45k
45k
3 ITH
SENSE + 8
7 SENSE –
EXTVCC 9
10
CC
RSENSE
1735 FD
8
LTC1735
U
OPERATIO
(Refer to Functional Diagram)
Main Control Loop
The LTC1735 uses a constant frequency, current mode
step-down architecture. During normal operation, the top
MOSFET is turned on each cycle when the oscillator sets
the RS latch and turned off when the main current comparator I1 resets the RS latch. The peak inductor current at
which I1 resets the RS latch is controlled by the voltage on
Pin 3 (ITH), which is the output of error amplifier EA. Pin␣ 6
(VOSENSE), described in the pin functions, allows EA to
receive an output feedback voltage VFB from an external
resistive divider. When the load current increases, it
causes a slight decrease in VFB relative to the 0.8V reference, which in turn causes the ITH voltage to increase until
the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is
turned on until either the inductor current starts to reverse,
as indicated by current comparator I2, or the beginning of
the next cycle.
The top MOSFET driver is powered from a floating bootstrap capacitor CB. This capacitor is normally recharged
from INTVCC through an external diode when the top
MOSFET is turned off. As VIN decreases towards VOUT, the
converter will attempt to turn on the top MOSFET continuously (“dropout’’). A dropout counter detects this condition and forces the top MOSFET to turn off for about 500ns
every tenth cycle to recharge the bootstrap capacitor.
The main control loop is shut down by pulling Pin 2
(RUN/SS) low. Releasing RUN/SS allows an internal 1.2µA
current source to charge soft-start capacitor CSS. When
CSS reaches 1.5V, the main control loop is enabled with the
ITH voltage clamped at approximately 30% of its maximum
value. As CSS continues to charge, ITH is gradually released allowing normal operation to resume. If VOUT has
not reached 70% of its final value when CSS has charged
to 4.1V, latchoff can be invoked as described in the
Applications Information section.
The internal oscillator can be synchronized to an external
clock applied to the FCB pin and can lock to a frequency
between 90% and 130% of its nominal rate set by capacitor COSC.
An overvoltage comparator, OV, guards against transient
overshoots (>7.5%) as well as other more serious
conditions that may overvoltage the output. In this case,
the top MOSFET is turned off and the bottom MOSFET is
turned on until the overvoltage condition is cleared.
Foldback current limiting for an output shorted to ground
is provided by amplifier A. As VOSENSE drops below 0.6V,
the buffered ITH input to the current comparator is gradually pulled down to a 0.86V clamp. This reduces peak
inductor current to about 1/4 of its maximum value.
Low Current Operation
The LTC1735 has three low current modes controlled by
the FCB pin. Burst Mode operation is selected when the
FCB pin is above 0.8V (typically tied to INTVCC). In Burst
Mode operation, if the error amplifier drives the ITH voltage
below 0.86V, the buffered ITH input to the current comparator will be clamped at 0.86V. The inductor current
peak is then held at approximately 20mV/RSENSE (about
1/4 of maximum output current). If ITH then drops below
0.5V, the Burst Mode comparator B will turn off both
MOSFETs to maximize efficiency. The load current will be
supplied solely by the output capacitor until ITH rises
above the 60mV hysteresis of the comparator and switching is resumed. Burst Mode operation is disabled by
comparator F when the FCB pin is brought below 0.8V.
This forces continuous operation and can assist secondary winding regulation.
When the FCB pin is driven by an external oscillator, a low
noise cycle-skipping mode is invoked and the internal
oscillator is synchronized to the external clock by comparator C. In this mode the 25% minimum inductor
current clamp is removed, providing constant frequency
discontinuous operation over the widest possible output
current range. This constant frequency operation is not
quite as efficient as Burst Mode operation, but provides a
lower noise, constant frequency spectrum.
The FCB pin is tied to ground when forced continuous
operation is desired. This is the least efficient mode, but is
desirable in certain applications. The output can source or
sink current in this mode. When sinking current while in
forced continuous operation, current will be forced back
into the main power supply potentially boosting the input
supply to dangerous voltage levels—BEWARE.
9
LTC1735
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OPERATIO
(Refer to Functional Diagram)
Foldback Current, Short-Circuit Detection and
Short-Circuit Latchoff
The RUN/SS capacitor, CSS, is used initially to limit the
inrush current of the switching regulator. After the controller has been started and been given adequate time to
charge up the output capacitors and provide full load current, CSS is used as a short-circuit time-out circuit. If the
output voltage falls to less than 70% of its nominal output
voltage, CSS begins discharging on the assumption that
the output is in an overcurrent and/or short-circuit condition. If the condition lasts for a long enough period as
determined by the size of CSS, the controller will be shut
down until the RUN/SS pin voltage is recycled. This builtin latchoff can be overridden by providing a current >5µA
at a compliance of 5V to the RUN/SS pin. This current
shortens the soft-start period but also prevents net discharge of CSS during an overcurrent and/or short-circuit
condition. Foldback current limiting is activated when the
output voltage falls below 70% of its nominal level whether
or not the short-circuit latchoff circuit is enabled.
INTVCC/EXTVCC POWER
Power for the top and bottom MOSFET drivers and most
of the internal circuitry of the LTC1735 is derived from the
INTVCC pin. When the EXTVCC pin is left open, an internal
5.2V low dropout regulator supplies the INTVCC power
from VIN. If EXTVCC is raised above 4.7V, the internal
regulator is turned off and an internal switch connects
EXTVCC to INTVCC. This allows a high efficiency source,
such as the primary or a secondary output of the converter
itself, to provide the INTVCC power. Voltages up to 7V can
be applied to EXTVCC for additional gate drive capability.
To provide clean start-up and to protect the MOSFETs,
undervoltage lockout is used to keep both MOSFETs off
until the input voltage is above 3.5V.
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The basic LTC1735 application circuit is shown in Figure␣ 1
on the first page. External component selection is driven
by the load requirement and begins with the selection of
RSENSE. Once RSENSE is known, COSC and L can be chosen.
Next, the power MOSFETs and D1 are selected. The
operating frequency and the inductor are chosen based
largely on the desired amount of ripple current. Finally, CIN
is selected for its ability to handle the large RMS current
into the converter and COUT is chosen with low enough
ESR to meet the output voltage ripple and transient specifications. The circuit shown in Figure 1 can be configured
for operation up to an input voltage of 28V (limited by the
external MOSFETs).
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
The LTC1735 current comparator has a maximum threshold of 75mV/RSENSE and an input common mode range of
SGND to 1.1(INTVCC). The current comparator threshold
sets the peak of the inductor current, yielding a maximum
average output current IMAX equal to the peak value less
half the peak-to-peak ripple current, ∆IL.
10
Allowing a margin for variations in the LTC1735 and
external component values yields:
RSENSE =
50mV
IMAX
COSC Selection for Operating Frequency and
Synchronization
The choice of operating frequency and inductor value is a
trade-off between efficiency and component size. Low
frequency operation improves efficiency by reducing
MOSFET switching losses, both gate charge loss and
transition loss. However, lower frequency operation requires more inductance for a given amount of ripple
current.
The LTC1735 uses a constant frequency architecture with
the frequency determined by an external oscillator capacitor COSC. Each time the topside MOSFET turns on, the
voltage on COSC is reset to ground. During the on-time,
COSC is charged by a fixed current. When the voltage on the
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capacitor reaches 1.19V, COSC is reset to ground. The
process then repeats.
The value of COSC is calculated from the desired operating
frequency assuming no external clock input on the FCB
pin:
 1.61(107 ) 
 – 11
COSC (pF) = 
 Frequency 


Inductor Value Calculation
A graph for selecting COSC versus frequency is shown in
Figure 2. The maximum recommended switching frequency is 550kHz .
The internal oscillator runs at its nominal frequency (fO)
when the FCB pin is pulled high to INTVCC or connected to
ground. Clocking the FCB pin above and below 0.8V will
cause the internal oscillator to injection lock to an external
clock signal applied to the FCB pin with a frequency
between 0.9fO and 1.3fO. The clock high level must exceed
1.3V for at least 0.3µs and the clock low level must be less
than 0.3V for at least 0.3µs. The top MOSFET turn-on will
synchronize with the rising edge of the clock.
Attempting to synchronize to too high an external frequency (above 1.3fO) can result in inadequate slope compensation and possible loop instability. If this condition
exists simply lower the value of COSC so fEXT = fO according
to Figure 2.
When synchronized to an external clock, Burst Mode
operation is disabled but the inductor current is not
100.0
87.5
COSC VALUE (pF)
75.0
62.5
50.0
37.5
25.0
12.5
0
0
100
200
300
400
500
OPERATING FREQUENCY (kHZ)
allowed to reverse. The 25% minimum inductor current
clamp present in Burst Mode operation is removed,
providing constant frequency discontinuous operation
over the widest possible output current range. In this
mode the synchronous MOSFET is forced on once every
10 clock cycles to recharge the bootstrap capacitor. This
minimizes audible noise while maintaining reasonably
high efficiency.
600
1735 F02
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET gate charge losses. In addition to this basic trade
off, the effect of inductor value on ripple current and low
current operation must also be considered.
The inductor value has a direct effect on ripple current. The
inductor ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN or VOUT:
∆IL =
 V

1
VOUT 1 – OUT 
VIN 
( f)(L)

Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.3 to 0.4(IMAX). Remember,
the maximum ∆IL occurs at the maximum input voltage.
The inductor value also has an effect on low current
operation. The transition to low current operation begins
when the inductor current reaches zero while the bottom
MOSFET is on. Burst Mode operation begins when the
average inductor current required results in a peak current
below 25% of the current limit determined by RSENSE.
Lower inductor values (higher ∆IL) will cause this to occur
at higher load currents, which can cause a dip in efficiency
in the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
Figure 2. Timing Capacitor Value
11
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Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of
core size for a fixed inductor value, but it is very dependent
on inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will
increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more
difficult. However, designs for surface mount are available
that do not increase the height significantly.
Power MOSFET and D1 Selection
Two external power MOSFETs must be selected for use
with the LTC1735: An N-channel MOSFET for the top
(main) switch and an N-channel MOSFET for the bottom
(synchronous) switch.
The peak-to-peak gate drive levels are set by the INTVCC
voltage. This voltage is typically 5.2V during start-up (see
EXTVCC pin connection). Consequently, logic-level threshold MOSFETs must be used in most LTC1735 applications. The only exception is when low input voltage is
expected (VIN < 5V); then, sub-logic level threshold
MOSFETs (VGS(TH) < 3V) should be used. Pay close
attention to the BVDSS specification for the MOSFETs as
well; many of the logic level MOSFETs are limited to 30V
or less.
12
Selection criteria for the power MOSFETs include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1735 is operating in continuous mode the duty cycles
for the top and bottom MOSFETs are given by:
V
Main Switch Duty Cycle = OUT
VIN
V –V
Synchronous Switch Duty Cycle = IN OUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
( )( )
2
V
PMAIN = OUT IMAX 1 + δ RDS(ON) +
VIN
( ) (IMAX )(CRSS )(f)
k VIN
2
( )( )
2
V –V
PSYNC = IN OUT IMAX 1 + δ RDS(ON)
VIN
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the topside
N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For
VIN < 20V the high current efficiency generally improves
with larger MOSFETs, while for VIN > 20V the transition
losses rapidly increase to the point that the use of a higher
RDS(ON) device with lower CRSS actually provides higher
efficiency. The synchronous MOSFET losses are greatest
at high input voltage or during a short-circuit when the
duty cycle in this switch is nearly 100%.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ␣ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the
MOSFET characteristics. The constant k = 1.7 can be
used to estimate the contributions of the two terms in the
main switch dissipation equation.
Kool Mµ is a registered trademark of Magnetics, Inc.
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The Schottky diode D1 shown in Figure 1 conducts during the
dead-time between the conduction of the two power MOSFETs.
This prevents the body diode of the bottom MOSFET from
turning on and storing charge during the dead-time, which
could cost as much as 1% in efficiency. A 3A Schottky is
generally a good size for 10A to 12A regulators due to the
relatively small average current. Larger diodes can result in
additional transition losses due to their larger junction capacitance. The diode may be omitted if the efficiency loss can be
tolerated.
CIN Selection
In continuous mode, the source current of the top
N-channel MOSFET is a square wave of duty cycle VOUT/
VIN. To prevent large voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current is given by:
 V

V
IRMS ≅ IO(MAX ) OUT  IN – 1
VIN  VOUT 
1/ 2
This formula has a maximum at VIN = 2VOUT, where
IRMS␣ =␣ IO(MAX)/2. This simple worst case condition is commonly used for design because even significant deviations do
not offer much relief. Note that capacitor manufacturers’
ripple current ratings are often based on only 2000 hours of
life. This makes it advisable to further derate the capacitor or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult the
manufacturer if there is any question.
COUT Selection
The selection of COUT is primarily determined by the
effective series resistance (ESR) to minimize voltage
ripple. The output ripple (∆VOUT) in continuous mode is
determined by:

1 
∆VOUT ≈ ∆IL  ESR +

8 fCOUT 

Where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. The output
ripple is highest at maximum input voltage since ∆IL
increases with input voltage. Typically, once the ESR
requirement for COUT has been met, the RMS current
rating generally far exceeds the IRIPPLE(P–P) requirement.
With ∆IL = 0.3IOUT(MAX) and allowing 2/3 of the ripple due
to ESR the output ripple will be less than 50mV at max VIN
assuming:
COUT required ESR < 2.2 RSENSE
COUT > 1/(8fRSENSE)
The first condition relates to the ripple current into the ESR
of the output capacitance while the second term guarantees that the output capacitance does not significantly
discharge during the operating frequency period due to
ripple current. The choice of using smaller output capacitance increases the ripple voltage due to the discharging
term but can be compensated for by using capacitors of
very low ESR to maintain the ripple voltage at or below
50mV. The ITH pin OPTI-LOOP compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected.
The selection of output capacitors for CPU or other applications with large load current transients is primarily
determined by the voltage tolerance specifications of the
load. The resistive component of the capacitor, ESR,
multiplied by the load current change plus any output
voltage ripple must be within the voltage tolerance of the
load (CPU).
The required ESR due to a load current step is:
RESR < ∆V/∆I
where ∆I is the change in current from full load to zero load
(or minimum load) and ∆V is the allowed voltage deviation
(not including any droop due to finite capacitance).
The amount of capacitance needed is determined by the
maximum energy stored in the inductor. The capacitance
must be sufficient to absorb the change in inductor current
when a high current to low current transition occurs. The
opposite load current transition is generally determined by
the control loop OPTI-LOOP components, so make sure
not to over compensate and slow down the response. The
minimum capacitance to assure the inductors’ energy is
adequately absorbed is:
13
LTC1735
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L( ∆I)2
COUT >
2( ∆V)VOUT
where ∆I is the change in load current.
Manufacturers such as Nichicon, United Chemicon and
Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest (ESR)(size)
product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON capacitors is recommended to reduce the
inductance effects.
In surface mount applications multiple capacitors may
need to be used in parallel to meet the ESR, RMS current
handling and load step requirements of the application.
Aluminum electrolytic, dry tantalum and special polymer
capacitors are available in surface mount packages. Special polymer surface mount capacitors offer very low ESR
but have much lower capacitive density per unit volume
than other capacitor types. These capacitors offer a very
cost-effective output capacitor solution and are an ideal
choice when combined with a controller having high loop
bandwidth. Tantalum capacitors offer the highest capacitance density and are often used as output capacitors for
switching regulators having controlled soft-start. Several
excellent surge-tested choices are the AVX TPS, AVX
TPSV or the KEMET T510 series of surface mount
tantalums, available in case heights ranging from 2mm to
4mm. Aluminum electrolytic capacitors can be used in
cost-driven applications providing that consideration is
given to ripple current ratings, temperature and long-term
reliability. A typical application will require several to many
aluminum electrolytic capacitors in parallel. A combination of the above mentioned capacitors will often result in
maximizing performance and minimizing overall cost.
Other capacitor types include Nichicon PL series, NEC
Neocap, Panasonic SP and Sprague 595D series. Consult
manufacturers for other specific recommendations.
Like all components, capacitors are not ideal. Each capacitor has its own benefits and limitations. Combinations of different capacitor types have proven to be a very
cost effective solution. Remember also to include high
frequency decoupling capacitors. They should be placed
14
as close as possible to the power pins of the load. Any
inductance present in the circuit board traces negates
their usefulness.
INTVCC Regulator
An internal P-channel low dropout regulator produces the
5.2V supply that powers the drivers and internal circuitry
within the LTC1735. The INTVCC pin can supply a maximum RMS current of 50mA and must be bypassed to
ground with a minimum of 4.7µF tantalum, 10µF special
polymer or low ESR type electrolytic capacitor. A 1µF
ceramic capacitor placed directly adjacent to the INTVCC
and PGND IC pins is highly recommended. Good bypassing is required to supply the high transient currents
required by the MOSFET gate drivers.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC1735 to be
exceeded. The system supply current is normally dominated by the gate charge current. Additional loading of
INTVCC also needs to be taken into account for the power
dissipation calculations. The total INTVCC current can be
supplied by either the 5.2V internal linear regulator or by
the EXTVCC input pin. When the voltage applied to the
EXTVCC pin is less than 4.7V, all of the INTVCC current is
supplied by the internal 5.2V linear regulator. Power
dissipation for the IC in this case is highest: (VIN)(IINTVCC)
and overall efficiency is lowered. The gate charge is
dependent on operating frequency as discussed in the
Efficiency Considerations section. The junction temperature can be estimated by using the equations given in
Note␣ 2 of the Electrical Characteristics. For example, the
LTC1735CS is limited to less than 17mA from a 30V
supply when not using the EXTVCC pin as follows:
TJ = 70°C + (17mA)(30V)(110°C/W) = 126°C
Use of the EXTVCC input pin reduces the junction temperature to:
TJ = 70°C + (17mA)(5V)(110°C/W) = 79°C
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
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EXTVCC Connection
The LTC1735 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
Whenever the EXTVCC pin is above 4.7V the internal 5.2V
regulator shuts off, the switch closes and INTVCC power is
supplied via EXTVCC until EXTVCC drops below 4.5V. This
allows the MOSFET gate drive and control power to be
derived from the output or other external source during
normal operation. When the output is out of regulation
(start-up, short circuit) power is supplied from the internal
regulator. Do not apply greater than 7V to the EXTVCC pin
and ensure that EXTVCC ≤ VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of (Duty Cycle)/(Efficiency). For 5V regulators this
simply means connecting the EXTVCC pin directly to VOUT.
However, for 3.3V and other lower voltage regulators,
additional circuitry is required to derive INTVCC power
from the output.
MOSFET gate drive requirements. This is the typical case
as the 5V power is almost always present and is derived by
another high efficiency regulator.
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V output regulator and provides the
highest efficiency. For output voltages higher than 5V,
EXTVCC is required to connect to VOUT so the SENSE pins’
absolute maximum ratings are not exceeded.
3. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency gains
can still be realized by connecting EXTVCC to an outputderived voltage that has been boosted to greater than
4.7V. This can be done with either the inductive boost
winding as shown in Figure 3a or the capacitive charge
pump shown in Figure 3b. The charge pump has the
advantage of simple magnetics.
4. EXTVCC connected to an external supply. If an external
supply is available in the 5V to 7V range (EXTVCC ≤ VIN),
such as notebook main 5V system power, it may be used
to power EXTVCC providing it is compatible with the
+
CIN
1N4148
VIN
LTC1735
VSEC
6.8V
+
1µF
N-CH
TG
RSENSE
VOUT
EXTVCC
SW
FCB
BG
L1
1:N
R4
+
COUT
N-CH
R3
SGND
PGND
1735 F03a
Figure 3a. Secondary Output Loop and EXTVCC Connection
+
VIN
1µF
+
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5.2V regulator resulting in
an efficiency penalty of up to 10% at high input voltages.
VIN
OPTIONAL EXTVCC
CONNECTION
5V ≤ VSEC ≤ 7V
CIN
BAT85
VIN
0.22µF
BAT85
LTC1735
TG
BAT85
VN2222LL
N-CH
RSENSE
VOUT
EXTVCC
L1
SW
+
BG
COUT
N-CH
PGND
1735 F03b
Figure 3b. Capacitive Charge Pump for EXTVCC
Output Voltage Programming
The output voltage is set by an external resistive divider
according to the following formula:
 R2 
VOUT = 0.8 V 1 + 
 R1
The resistive divider is connected to the output as shown
in Figure 4 allowing remote voltage sensing.
15
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VOUT
R2
VOSENSE
47pF
LTC1735
R1
SGND
1735 F04
Figure 4. Setting the LTC1735 Output Voltage
Topside MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor CB connected to the
BOOST pin supplies the gate drive voltage for the topside
MOSFET. Capacitor CB in the Functional Diagram is charged
though external diode DB from INTVCC when the SW pin
is low. Note that the voltage across CB is about a diode
drop below INTVCC. When the topside MOSFET is to be
turned on, the driver places the CB voltage across the
gate-source of the MOSFET. This enhances the MOSFET
and turns on the topside switch. The switch node voltage
SW rises to VIN and the BOOST pin rises to VIN + INTVCC.
The value of the boost capacitor CB needs to be 100 times
greater than the total input capacitance of the topside
MOSFET. In most applications 0.1µF to 0.33µF is adequate. The reverse breakdown on DB must be greater
than VIN(MAX).
When adjusting the gate drive level, the final arbiter is the
total input current for the regulator. If you make a change
and the input current decreases, then you improved the
efficiency. If there is no change in input current, then there
is no change in efficiency.
SENSE +/SENSE – Pins
The common mode input range of the current comparator
is from 0V to 1.1(INTVCC). Continuous linear operation in
step-down applications is guaranteed throughout this
range allowing output voltages anywhere from 0.8V to 7V.
A differential NPN input stage is used and is biased with
internal resistors from an internal 2.4V source as shown
in the Functional Diagram. This causes current to either be
sourced or sunk by the sense pins depending on the
output voltage. If the output voltage is below 2.4V current
will flow out of both sense pins to the main output. This
forces a minimum load current that can be fulfilled by the
16
VOUT resistive divider. The maximum current flowing out
of the sense pins is:
ISENSE+ + ISENSE– = (2.4V – VOUT)/24k
Since VOSENSE is servoed to the 0.8V reference voltage, we
can choose R1 in Figure 4 to have a maximum value to
absorb this current:


0.8 V
R1(MAX ) = 24k 

 2.4V – VOUT 
Regulating an output voltage of 1.8V, the maximum value
of R1 should be 32k. Note that at output voltages above
2.4V no maximum value of R1 is necessary to absorb the
sense pin currents; however, R1 is still bounded by the
VOSENSE feedback current.
Soft-Start/Run Function
The RUN/SS pin is a multipurpose pin that provides a softstart function and a means to shut down the LTC1735.
Soft-start reduces surge currents from VIN by gradually
increasing the controller’s current limit ITH(MAX). This pin
can also be used for power supply sequencing.
Pulling the RUN/SS pin below 1.5V puts the LTC1735 into
a low quiescent current shutdown (IQ < 25µA). This pin can
be driven directly from logic as shown in Figure 5. Releasing the RUN/SS pin allows an internal 1.2µA current
source to charge up the external soft-start capacitor CSS.
If RUN/SS has been pulled all the way to ground there is
a delay before starting of approximately:
tDELAY =
1.5V
CSS = (1.25s /µF ) CSS
1.2µA
When the voltage on RUN/SS reaches 1.5V the LTC1735
begins operating with a current limit at approximately
25mV/RSENSE. As the voltage on the RUN/SS pin increases
from 1.5V to 3.0V, the internal current limit is increased
from 25mV/RSENSE to 75mV/RSENSE. The output current
limit ramps up slowly, taking an additional 1.25s/µF to
reach full current. The output current thus ramps up
slowly, reducing the starting surge current required from
the input power supply.
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Diode D1 in Figure 5 reduces the start delay while allowing
CSS to charge up slowly for the soft-start function. This
diode and CSS can be deleted if soft-start is not needed.
The RUN/SS pin has an internal 6V zener clamp (See
Functional Diagram).
3.3V OR 5V
RUN/SS
RUN/SS
D1
CSS
CSS
1735 F05
Figure 5. RUN/SS Pin Interfacing
Fault Conditions: Overcurrent Latchoff
The RUN/SS pin also provides the ability to shut off the
controller and latch off when an overcurrent condition is
detected. The RUN/SS capacitor, CSS, is used initially to
turn on and limit the inrush current of the controller. After
the controller has been started and given adequate time to
charge up the output capacitor and provide full load
current, CSS is used as a short-circuit timer. If the output
voltage falls to less than 70% of it’s nominal output voltage
after CSS reaches 4.1V, the assumption is made that the
output is in a severe overcurrent and/or short-circuit
condition and CSS begins discharging. If the condition
lasts for a long enough period as determined by the size of
CSS, the controller will be shut down until the RUN/SS pin
voltage is recycled.
This built-in latchoff can be overridden by providing a
current >5µA at a compliance of 5V to the RUN/SS pin as
shown in Figure␣ 6. This current shortens the soft-start
period but also prevents net discharge of the RUN/SS
VIN
INTVCC
3.3V OR 5V
RUN/SS
RSS
D1
RSS
D1
RUN/SS
CSS
CSS
(a)
(b)
1735 F06
Figure 6. RUN/SS Pin Interfacing with Latchoff Defeated
capacitor during a severe overcurrent and/or short-circuit
condition. When deriving the 5µA current from VIN as in
Figure␣ 6a, current latchoff is always defeated. Diode connecting this pull-up resistor to INTVCC , as in Figure␣ 6b,
eliminates any extra supply current during controller shutdown while eliminating the INTVCC loading from preventing controller start-up. If the voltage on CSS does not
exceed 4.1V the overcurrent latch is not armed and the
function is disabled.
Why should you defeat overcurrent latchoff? During the
prototyping stage of a design, there may be a problem with
noise pickup or poor layout causing the protection circuit
to latch off. Defeating this feature will easily allow troubleshooting of the circuit and PC layout. The internal shortcircuit and foldback current limiting still remains active,
thereby protecting the power supply system from failure.
After the design is complete, a decision can be made
whether to enable the latchoff feature.
The value of the soft-start capacitor CSS will need to be
scaled with output current, output capacitance and load
current characteristics. The minimum soft-start capacitance is given by:
CSS > (COUT )(VOUT)(10 – 4)(RSENSE)
The minimum recommended soft-start capacitor of
CSS␣ =␣ 0.1µF will be sufficient for most applications.
Fault Conditions: Current Limit and Current Foldback
The LTC1735 current comparator has a maximum sense
voltage of 75mV resulting in a maximum MOSFET current
of 75mV/RSENSE.
The LTC1735 includes current foldback to help further
limit load current when the output is shorted to ground.
The foldback circuit is active even when the overload
shutdown latch described above is defeated. If the output
falls by more than half, then the maximum sense voltage
is progressively lowered from 75mV to 30mV. Under
short-circuit conditions with very low duty cycle, the
LTC1735 will begin cycle skipping in order to limit the
short-circuit current. In this situation the bottom MOSFET
will be conducting the peak current. The short-circuit
ripple current is determined by the minimum on-time
17
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∆IL(SC) = tON(MIN)VIN/L
The resulting short-circuit current is:
ISC =
30mV 1
+ ∆IL(SC)
RSENSE 2
The current foldback function is always active and is not
effected by the current latchoff function.
Fault Conditions: Output Overvoltage Protection
(Crowbar)
The output overvoltage crowbar is designed to blow a
system fuse in the input lead when the output of the
regulator rises much higher than nominal levels. This
condition causes huge currents to flow, much greater than
in normal operation. This feature is designed to protect
against a shorted top MOSFET; it does not protect against
a failure of the controller itself.
The comparator (OV in the Functional Diagram) detects
overvoltage faults greater than 7.5% above the nominal
output voltage. When this condition is sensed the top
MOSFET is turned off and the bottom MOSFET is forced
on. The bottom MOSFET remains on continuously for as
long as the 0V condition persists; if VOUT returns to a safe
level, normal operation automatically resumes.
Note that dynamically changing the output voltage may
cause overvoltage protection to be momentarily activated
during programmed output voltage decreases. This will
not cause permanent latchoff nor will it disrupt the desired
voltage change. With soft-latch overvoltage protection,
dynamically changing the output voltage is allowed and
the overvoltage protection tracks the newly programmed
output voltage, always protecting the load.
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest amount of time
that the LTC1735 is capable of turning the top MOSFET on
and off again. It is determined by internal timing delays and
18
the gate charge required to turn on the top MOSFET. Low
duty cycle applications may approach this minimum ontime limit and care should be taken to ensure that:
V
t ON(MIN) < OUT
VIN( f)
If the duty cycle falls below what can be accommodated by
the minimum on-time, the LTC1735 will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple current and voltage will increase.
The minimum on-time for the LTC1735 in a properly
configured application is generally less than 200ns. However, as the peak sense voltage decreases, the minimum
on-time gradually increases as shown in Figure 7. This is
of particular concern in forced continuous applications
with low ripple current at light loads. If the duty cycle drops
below the minimum on-time limit in this situation, a
significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple.
If an application can operate close to the minimum ontime limit, an inductor must be chosen that is low enough
to provide sufficient ripple amplitude to meet the minimum on-time requirement. As a general rule keep the
inductor ripple current equal or greater than 30% of
IOUT(MAX) at VIN(MAX).
250
200
MINIMUM ON-TIME (ns)
tON(MIN) of the LTC1735 (approximately 200ns), the input
voltage and inductor value:
150
100
50
0
0
10
20
30
40
∆IL /IOUT(MAX) (%)
1735 F07
Figure 7. Minimum On-Time vs ∆IL
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FCB Pin Operation
When the FCB pin drops below its 0.8V threshold, continuous mode operation is forced. In this case, the top and
bottom MOSFETs continue to be driven synchronously
regardless of the load on the main output. Burst Mode
operation is disabled and current reversal is allowed in the
inductor.
In addition to providing a logic input to force continuous
synchronous operation and external synchronization, the
FCB pin provides a means to regulate a flyback winding
output. During continuous mode, current flows continuously in the transformer primary. The secondary winding(s)
draw current only when the bottom, synchronous switch
is on. When primary load currents are low and/or the VIN/
VOUT ratio is low, the synchronous switch may not be on
for a sufficient amount of time to transfer power from the
output capacitor to the secondary load. Forced continuous
operation will support secondary windings providing there
is sufficient synchronous switch duty factor. Thus, the
FCB input pin removes the requirement that power must
be drawn from the inductor primary in order to extract
power from the auxiliary windings. With the loop in
continuous mode, the auxiliary outputs may nominally be
loaded without regard to the primary output load.
The secondary output voltage VSEC is normally set as
shown in Figure␣ 3a by the turns ratio N of the transformer:
VSEC ≅ (N + 1)VOUT
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
then VSEC will droop. An external resistive divider from
VSEC to the FCB pin sets a minimum voltage VSEC(MIN):
 R4 
VSEC(MIN) ≈ 0.8 V 1 + 
 R3 
If VSEC drops below this level, the FCB voltage forces
continuous switching operation until VSEC is again above
its minimum.
In order to prevent erratic operation if no external connections are made to the FCB pin, the FCB pin has a 0.17µA
internal current source pulling the pin high. Remember to
include this current when choosing resistor values R3 and
R4.
The internal LTC1735 oscillator can be synchronized to an
external oscillator by applying and clocking the FCB pin
with a signal above 1.5VP–P. When synchronized to an
external frequency, Burst Mode operation is disabled but
cycle skipping is allowed at low load currents since current
reversal is inhibited. The bottom gate will come on every
10 clock cycles to assure the bootstrap cap is kept refreshed. The rising edge of an external clock applied to the
FCB pin starts a new cycle.
The range of synchronization is from 0.9fO to 1.3fO, with
fO set by COSC. Attempting to synchronize to a higher
frequency than 1.3fO can result in inadequate slope compensation and cause loop instability with high duty cycles
(duty cycle > 50%). If loop instability is observed while
synchronized, additional slope compensation can be obtained by simply decreasing COSC.
The following table summarizes the possible states available on the FCB pin:
Table 1
FCB Pin
Condition
DC Voltage: 0V to 0.7V
Burst Disabled/Forced Continuous
Current Reversal Enabled
DC Voltage: ≥ 0.9V
Burst Mode Operation,
No Current Reversal
Feedback Resistors
Regulating a Secondary Winding
Ext Clock: (0V to VFCBSYNC)
(VFCBSYNC > 1.5V)
Burst Mode Operation Disabled
No Current Reversal
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + …)
where L1, L2, etc. are the individual losses as a percentage
of input power.
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Although all dissipative elements in the circuit produce
losses, 4 main sources usually account for most of the
losses in LTC1735 circuits: 1) LTC1735 VIN current,
2)␣ INTVCC current, 3) I2R losses, 4) Topside MOSFET
transition losses.
4) Transition losses apply only to the topside MOSFET(s)
and only become significant when operating at high input
voltages (typically 12V or greater). Transition losses can
be estimated from:
1) The VIN current is the DC supply current given in the
electrical characteristics which excludes MOSFET driver
and control currents. VIN current results in a small (<0.1%)
loss that increases with VIN.
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these “system” level losses in the
design of a system. The internal battery and fuse resistance losses can be minimized by making sure that CIN has
adequate charge storage and very low ESR at the switching frequency. A 25W supply will typically require a
minimum of 20µF to 40µF of capacitance having a maximum of 0.01Ω to 0.02Ω of ESR. Other losses including
Schottky conduction losses during dead-time and inductor core losses generally account for less than 2% total
additional loss.
2) INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from INTVCC to
ground. The resulting dQ/dt is a current out of INTVCC that
is typically much larger than the control circuit current. In
continuous mode, IGATECHG = f(QT+QB), where QT and QB
are the gate charges of the topside and bottom-side
MOSFETs.
Supplying INTVCC power through the EXTVCC switch input
from an output-derived or other high efficiency source will
scale the VIN current required for the driver and control
circuits by a factor of (Duty Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current
results in approximately 3mA of VIN current. This reduces
the mid-current loss from 10% or more (if the driver was
powered directly from VIN) to only a few percent.
3) I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous mode
the average output current flows through L and RSENSE,
but is “chopped” between the topside main MOSFET and
the synchronous MOSFET. If the two MOSFETs have
approximately the same RDS(ON), then the resistance of
one MOSFET can simply be summed with the resistances
of L and RSENSE to obtain I2R losses. For example, if each
RDS(ON) = 0.03Ω, RL = 0.05Ω and RSENSE = 0.01Ω, then
the total resistance is 0.09Ω. This results in losses ranging
from 2% to 9% as the output current increases from 1A to
5A for a 5V output, or a 3% to 14% loss for a 3.3V output.
Effeciency varies as the inverse square of VOUT for the
same external components and output power level. I2R
losses cause the efficiency to drop at high output currents.
20
Transition Loss = (1.7) VIN2 IO(MAX) CRSS f
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in load current.
When a load step occurs, VOUT shifts by an amount equal
to ∆ILOAD (ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge
COUT generating the feedback error signal that forces the
regulator to adapt to the current change and return VOUT
to its steady-state value. During this recovery time VOUT
can be monitored for excessive overshoot or ringing,
which would indicate a stability problem. OPTI-LOOP
compensation allows the transient response to be optimized over a wide range of output capacitance and ESR
values. The availability of the ITH pin not only allows
optimization of control loop behavior but also provides a
DC coupled and AC filtered closed loop response test
point. The DC step, rise time and settling at this test point
truly reflects the closed loop response. Assuming a predominantly second order system, phase margin and/or
damping factor can be estimated using the percentage of
overshoot seen at this pin. The bandwidth can also be
estimated by examining the rise time at the pin. The ITH
external components shown in the Figure␣ 1 circuit will
provide an adequate starting point for most applications.
LTC1735
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The ITH series RC–CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
feedback factor gain and phase. An output current pulse of
20% to 100% of full load current having a rise time of 1µs
to 10µs will produce output voltage and ITH pin waveforms
that will give a sense of the overall loop stability without
breaking the feedback loop. The initial output voltage step
may not be within the bandwidth of the feedback loop, so
the standard second-order overshoot/DC ratio cannot be
used to determine phase margin. The gain of the loop will
be increased by increasing RC and the bandwidth of the
loop will be increased by decreasing CC. If RC is increased
by the same factor that CC is decreased, the zero frequency
will be kept the same, thereby keeping the phase the same
in the most critical frequency range of the feedback loop.
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance. For a detailed
explanation of optimizing the compensation components,
including a review of control loop theory, refer to Application Note 76.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than1:50, the switch rise time
should be controlled so that the load rise time is limited to
approximately (25)(CLOAD). Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current to
about 200mA.
Improve Transient Response and Reduce Output
Capacitance with Active Voltage Positioning
Fast load transient response, limited board space and low
cost are requirements of microprocessor power supplies.
Active voltage positioning improves transient response
and reduces the output capacitance required to power a
microprocessor where a typical load step can be from 0.2A
to 15A in 100ns or 15A to 0.2A in 100ns. The voltage at the
microprocessor must be held to about ±0.1V of nominal
in spite of these load current steps. Since the control loop
cannot respond this fast, the output capacitors must
supply the load current until the control loop can respond.
Capacitor ESR and ESL primarily determine the amount of
droop or overshoot in the output voltage. Normally, several capacitors in parallel are required to meet microprocessor transient requirements.
Active voltage positioning is a form of deregulation. It sets
the output voltage high for light loads and low for heavy
loads. When load current suddenly increases, the output
voltage starts from a level higher than nominal so the
output voltage can droop more and stay within the specified voltage range. When load current suddenly decreases
the output voltage starts at a level lower than nominal so
the output voltage can have more overshoot and stay
within the specified voltage range. Less output capacitance is required when voltage positioning is used because more voltage variation is allowed on the output
capacitors.
Active voltage positioning can be implemented using the
OPTI-LOOP architecture of the LTC1735 and two resistors
connected to the ITH pin. An input voltage offset is introduced when the error amplifier has to drive a resistive load.
This offset is limited to ±30mV at the input of the error
amplifier. The resulting change in output voltage is the
product of input offset and the feedback voltage divider
ratio.
Figure 8 shows a CPU-core-voltage regulator with active
voltage positioning. Resistors R1 and R4 force the input
voltage offset that adjusts the output voltage according to
the load current level. To select values for R1 and R4, first
determine the amount of output deregulation allowed. The
actual specification for a typical microprocessor allows
the output to vary ±0.112V. The LTC1735 reference accuracy is ±1%. Using 1% tolerance resistors, the total
feedback divider accuracy is about 1% because both
feedback resistors are close to the same value. The resulting setpoint accuracy is ±2% so the output transient
21
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R3
680k
R4
100k
R1
27k
C7
0.1µF
C1
39pF
1
C2
0.1µF
2
R2
100k
C3
100pF
3
C4
100pF
4
5
C5
47pF
6
C6
1000pF
COSC
TG
RUN/SS
BOOST
ITH
SW
U1
LTC1735
FCB
SGND
VOSENSE
VIN
INTVCC
BG
7
SENSE –
PGND
8
SENSE +
EXTVCC
16
M1
FDS6680A
15
C8
0.22µF
14
13
L1
1µH
D1
CMDSH-3
12
R5
0.003Ω
C9
1µF
10
+
C10
4.7µF
10V
C19
1µF
R6
10k
M2, M3
FDS6680A
×2
+
R7
11.5k
GND
VOUT
1.5V
15A
C11
330pF
D2
MBRS340
11
9
C9, C19: TAIYO YUDEN JMK107BJ105
C10: KEMET T494A475M010AS
C12 TO C14: TAIYO YUDEN GMK325F106
C15 TO C18: PANASONIC EEFUE0G181R
D1: CENTRAL SEMI CMDSH-3
D2: MOTOROLA MBRS340
L1: PANASONIC ETQP6F1R0SA
M1 TO M3: FAIRCHILD FDS6680A
R5: IRC LRF2512-01-R003-J
U1: LINEAR TECHNOLOGY LTC1735CS
C12 TO C14
10µF
35V
VIN
7.5V TO
24V
C15 TO
C18
180µF
4V
GND
5V (OPTIONAL)
1735 F08
Figure 8. CPU-Core-Voltage Regulator with Active Voltage Positioning
voltage cannot exceed ±0.082V. At VOUT = 1.5V, the
maximum output voltage change controlled by the ITH pin
would be:
Input Offset • VOUT
VREF
± 0.03V • 1.5
=
= ±56mV
0.8 V
∆VOSENSE =
With the optimum resistor values at the ITH pin, the output
voltage will swing from 1.55V at minimum load to 1.44V
at full load. At this output voltage, active voltage positioning provides an additional ±56mV to the allowable transient voltage on the output capacitors, a 68% improvement over the ±82mV allowed without active voltage
positioning.
The next step is to calculate the ITH pin voltage, VITH, scale
factor. The VITH scale factor reflects the ITH pin voltage
22
required for a given load current. VITH controls the peak
sense resistor voltage, which represents the DC output
current plus one half of the peak-to-peak inductor current.
The no load to full load VITH range is from 0.3V to 2.4V,
which controls the sense resistor voltage from 0V to the
∆VSENSE(MAX) voltage of 75mV. The calculated VITH scale
factor with a 0.003Ω sense resistor is:
VITH Scale Factor =
=
VITH Range • Sense Re sistor Value
∆VSENSE(MAX)
(2.4V – 0.3V) • 0.003
= 0.084V/A
0.075V
VITH at any load current is:


∆I 
VITH = IOUTDC + L  • VITH Scale Factor 
2 


+ VITH Offset
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At full load current:



5A
VITH(MAX) =  15A + P−P  • 0.084V/A  + 0.3V
2 


= 1.77 V
At minimum load current:



2A
VITH(MIN) =  0.2A + P−P  • 0.084V/A  + 0.3V
2 


= 0.40 V
In this circuit, VITH changes from 0.40V at light load to
1.77V at full load, a 1.37V change. Notice that ∆IL, the
peak-to-peak inductor current, changes from light load to
full load. Increasing the DC inductor current decreases the
permeability of the inductor core material, which decreases the inductance and increases ∆IL. The amount of
inductance change is a function of the inductor design.
To create the ±30mV input offset, the gain of the error
amplifier must be limited. The desired gain is:
AV =
∆VITH
1.37 V
=
= 22.8
Input Offset Error 2(0.03V)
Connecting a resistor to the output of the transconductance
error amplifier will limit the voltage gain. The value of this
resistor is:
RITH =
AV
22.8
=
= 17.54k
Error Amplifier gm 1.3ms
To center the output voltage variation, VITH must be
centered so that no ITH pin current flows when the output
voltage is nominal. VITH(NOM) is the average voltage between VITH at maximum output current and minimum
output current:
The Thevenin equivalent of the gain limiting resistance
value of 17.54k is made up of a resistor R4 that sources
current into the ITH pin and resistor R1 that sinks current
to SGND.
To calculate the resistor values, first determine the ratio
between them:
k=
VINTVCC – VITH(NOM) 5.2V – 1.085V
=
= 3.79
1.085V
VITH(NOM)
VINTVCC is equal to VEXTVCC or 5.2V if EXTVCC is not used.
Resistor R4 is:
R4 = (k + 1) • RITH = (3.79 + 1) • 17.54 = 84.0k
Resistor R1 is:
R1 =
(k + 1) • RITH (3.79 + 1) • 17.54k
=
= 22.17k
k
3.79
Unfortunately, PCB noise can add to the voltage developed
across the sense resistor, R5, causing the ITH pin voltage
to be slightly higher than calculated for a given output
current. The amount of noise is proportional to the output
current level. This PCB noise does not present a serious
problem but it does change the effective value of R5 so the
calculated values of R1 and R4 may need to be adjusted to
achieve the required results. Since PCB noise is a function
of the layout, it will be the same on all boards with the same
layout.
Figures 9 and 10 show the transient response before and
after active voltage positioning is implemented. Notice
that active voltage positioning reduced the transient response from almost 200mVP-P to a little over 100mVP-P.
Refer to Design Solutions 10 for more information about
active voltage positioning.
VITH(MAX) – VITH(MIN)
+ VITH(MIN)
2
1.77 V – 0.40 V
=
+ 0.40 V = 1.085V
2
VITH(NOM) =
23
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VIN = 12V
VOUT = 1.5V
1.5V
100mV/DIV
OUTPUT
VOLTAGE
FIGURE 8 CIRCUIT
15A
LOAD
CURRENT
10A/DIV
0A
50µs/DIV
1735 F09
Figure 9. Normal Transient Response (Without R1, R4)
The network shown in Figure␣ 11 is the most straight
forward approach to protect a DC/DC converter from the
ravages of an automotive power line. The series diode
prevents current from flowing during reverse-battery,
while the transient suppressor clamps the input voltage
during load-dump. Note that the transient suppressor
should not conduct during double-battery operation, but
must still clamp the input voltage below breakdown of the
converter. Although the LTC1735 has a maximum input
voltage of 36V, most applications will be limited to 30V by
the MOSFET BVDSS.
50A IPK RATING
FIGURE 8 CIRCUIT
VIN = 12V
VOUT = 1.5V
1.582V
100mV/DIV 1.5V
12V
OUTPUT
VOLTAGE
VIN
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
1.418V
15A
1735 F11
LOAD
CURRENT
10A/DIV
Figure 11. Plugging into the Cigarette Lighter
0A
50µs/DIV
1735 F10
Figure 10. Transient Response with Active Voltage Positioning
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation.
But before you connect, be advised: you are plugging
into the supply from hell. The main power line in an
automobile is the source of a number of nasty potential
transients, including load-dump, reverse-battery and
double-battery.
Load-dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
just what it says, while double-battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
24
LTC1735
Design Example
As a design example, assume VIN = 12V(nominal),
VIN = 22V(max), VOUT = 1.8V, IMAX = 5A and f = 300kHz.
RSENSE and COSC can immediately be calculated:
RSENSE = 50mV/5A = 0.01Ω
COSC = 1.61(107)/(300kHz) – 11pF = 43pF
Assume a 3.3µH inductor and check the actual value of the
ripple current. The following equation is used:
 V

V
∆IL = OUT  1 – OUT 
( f)(L) 
VIN 
The highest value of the ripple current occurs at the
maximum input voltage:
∆IL =
 1.8 V 
1.8 V
 1–
 = 2.3A
300kHz(3.3µH) 
22V 
The maximum ripple current is 33% of maximum output
current, which is about right.
LTC1735
U
W
U
U
APPLICATIO S I FOR ATIO
Next verify the minimum on-time of 200ns is not violated.
The minimum on-time occurs at maximum VIN:
tON(MIN) =
VOUT
VIN(MAX )f
=
1.8 V
= 273ns
22V(300kHz)
Since the output voltage is below 2.4V the output resistive
divider will need to be sized to not only set the output
voltage but also to absorb the sense pin current.
Choosing 1% resistors; R1 = 25.5k and R2 = 32.4k yields
an output voltage of 1.816V.
The power dissipation on the topside MOSFET can be
easily estimated. Choosing a Siliconix Si4412DY results in
RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
()[
](
2
+1.7(22V ) (5A )(100pF )(300kHz)
1.8 V 2
5 1 + (0.005)(50°C – 25°C) 0.042Ω
22V
)
= 220mW
Because the duty cycle of the bottom MOSFET is much
greater than the top, a larger MOSFET, Siliconix Si4410DY,
(RDS(ON) = 0.02Ω) is chosen. The power dissipation in the
bottom MOSFET, again assuming TA = 50°C, is:
( ) ( )(
2
22V – 1.8 V
5A 1.1 0.02Ω
22V
= 500mW
PSYNC =
VORIPPLE = RESR( ∆IL ) = 0.02Ω(2.3A) = 46mVP −P
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1735. These items are also illustrated graphically in
the layout diagram of Figure␣ 12. Check the following in
your layout:


0.8 V
R1(MAX ) = 24k 

 2.4V – VOUT 

0.8 V 
= 24K
 = 32k
 2.4V – 1.8 V 
PMAIN =
CIN is chosen for an RMS current rating of at least 2.5A at
temperature. COUT is chosen with an ESR of 0.02Ω for low
output ripple. The output ripple in continuous mode will be
highest at the maximum input voltage. The worst-case
output voltage ripple due to ESR is approximately:
)
Thanks to current foldback, the bottom MOSFET dissipation in short-circuit will be less than under full load
conditions.
1) Are the signal and power grounds segregated? The
LTC1735 PGND pin should tie to the ground plane close to
the input capacitor(s). The SGND pin should then connect
to PGND and all components that connect to SGND should
make a single point tie to the SGND pin. The synchronous
MOSFET source pins should connect to the input
capacitor(s) ground.
2) Does the VOSENSE pin connect directly to the feedback
resistors? The resistive divider R1, R2 must be connected
between the (+) plate of COUT and signal ground. The 47pF
to 100pF capacitor should be as close as possible to the
LTC1735. Be careful locating the feedback resistors too far
away from the LTC1735. The VOSENSE line should not be
routed close to any other nodes with high slew rates.
3) Are the SENSE – and SENSE + leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE + and SENSE – should be as close as possible to the
LTC1735. Ensure accurate current sensing with Kelvin
connections as shown in Figure 13. Series resistance can
be added to the SENSE lines to increase noise rejection.
4) Does the (+) terminal of CIN connect to the drain of the
topside MOSFET(s) as closely as possible? This capacitor
provides the AC current to the MOSFET(s).
5) Is the INTVCC decoupling capacitor connected closely
between INTVCC and the power ground pin? This capacitor carries the MOSFET driver peak currents. An additional 1µF ceramic capacitor placed immediately next to
25
LTC1735
U
U
W
U
APPLICATIO S I FOR ATIO
the INTVCC and PGND pins can help improve noise
performance.
feedback pins. All of these nodes have very large and fast
moving signals and therefore should be kept on the
“output side” (Pin 9 to Pin 16) of the LTC1735 and occupy
minimum PC trace area.
6) Keep the switching node (SW), top gate node (TG) and
boost node (BOOST) away from sensitive small-signal
nodes, especially from the voltage and current sensing
+
COSC
M1
1
COSC
TG
16
CSS
RC
RUN/SS
BOOST
CIN
15
+
2
CC
3
ITH
SW
14
VIN
LTC1735
CC2
4
5
FCB
SGND
VIN
INTVCC
13
47pF
6
VOSENSE
BG
DB
12
11
CB
D1
+
4.7µF
M2
7
SENSE –
PGND
8
SENSE +
EXTVCC
10
–
1000pF
9
L1
–
R1
COUT
VOUT
+
R2
RSENSE
+
1735 F12
Figure 12. LTC1735 Layout Diagram
HIGH CURRENT PATH
1735 F13
SENSE + SENSE –
CURRENT SENSE
RESISTOR
(RSENSE)
Figure 13. Kelvin Sensing RSENSE
26
LTC1735
U
TYPICAL APPLICATIO S
1.8V/5A Converter from Design Example with Burst Mode Operation Disabled
VIN
4.5V TO 22V
CIN
22µF
50V
CER
COSC
43pF
1
COSC
CSS
0.1µF
2
RUN/SS
CC
470pF
RC
33k
TG
3
BOOST
ITH
CC2
220pF
SW
16
M1
Si4412DY
CB
0.1µF
15
14
LTC1735
4
FCB
5
VIN
SGND
INTVCC
13
L1
3.3µH
DB
CMDSH-3
RSENSE
0.01Ω
R2
32.4k
1%
12
+
47pF
VOUT
1.8V
5A
+
4.7µF
6
7
VOSENSE
BG
SENSE –
PGND
11
M2
Si4410DY
R1
25.5k
1%
MBRS140T3
10
COUT
150µF
6.3V
×2
PANASONIC SP
SGND
1000pF
8
SENSE +
EXTVCC
9
OPTIONAL:
CONNECT TO 5V
COUT: PANASONIC EEFUEOG151R
CIN: MARCON THCR70LE1H226ZT
L1: PANASONIC ETQP6F3R3HFA
RSENSE: IRC LR 2010-01-R010F
1735 TA02
CPU Core Voltage Regulator for 2-Step Applications (VIN = 5V)
VIN
5V
100k*
COSC 39pF
1
CSS
0.1µF
2
RC
20k
CC
220pF
3
CC2
220pF
COSC
TG
RUN/SS
BOOST
ITH
SW
16
CB
0.22µF
15
14
LTC1735
4
5
FCB
SGND
VIN
INTVCC
13
DB
MBR0530
BG
11
1µF
7
SENSE –
PGND
RSENSE
0.004Ω
100pF
+
4.7µF
VOSENSE
L1
0.78µH
12
47pF
6
M1
FDS6680A
CIN
150µF
6.3V
×2
10
M2, M3
FDS6680A
×2
MBRD835L
R2
32.4k
1%
VOUT
1.5V
12A
+
R1
25.5k
1%
COUT
180µF
4V
×3
CO
47µF
10V
SGND
1000pF
8
SENSE +
EXTVCC
9
VIN
COUT: PANASONIC EEFUEOG181R
CIN: PANASONIC EEFUEOJ151R
CO: TAIYO YUDEN LMK550BJ476MM-B
L1: COILCRAFT 1705022P-781HC
RSENSE: IRC LRF 2512-01-R004-J
1735 TA03
*OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF
27
LTC1735
U
TYPICAL APPLICATIO S
Selectable Output Voltage Converter with Burst Mode Operation Disabled for CPU Power
0.1µF
VIN
4.5V TO 24V
4.7Ω
+
COSC
43pF
1
CSS
0.1µF
2
CC
RC 330pF
33k
3
CC2
47pF
COSC
16
TG
RUN/SS
15
BOOST
ITH
M1
FDS6680A
CB
0.22µF
CIN: MARCON THCR70EIH226ZT
COUT: KEMET T510X447M006AS
L1: PANASONIC ETQP6F1R2HFA
RSENSE: IRC LRF2512-01-R004F
14
SW
LTC1735
4
5
FCB
13
VIN
SGND
CIN
22µF
×2
CER
L1
1.2µH
DB
CMDSH-3
RSENSE
0.004Ω
12
INTVCC
R2
10k
1%
47pF
+
47pF
VOUT
1.35V/1.60V
12A
COUT
470µF
6.3V
×3
KEMET
+
4.7µF
6
7
VOSENSE
BG
SENSE –
PGND
11
10
M2
FDS6680A
×2
MBRS340T3
1µF
CER
R3
33.2k
1%
47pF
R1
14.3k
1%
VN2222
1000pF
8
SENSE
+
9
EXTVCC
OPTIONAL:
CONNECT TO 5V
ON: VOUT = 1.60V
OFF: VOUT = 1.35V
10k
10Ω
SGND
10Ω
1735 TA05
4V to 40V Input to 12V Flyback Converter
VIN
4V TO 40V
CMDSH-3
FMMT625
CIN
22µF
50V
×2
10k
6.2V
1
CSS
0.1µF
2
CC
2200pF RC
3.3k
3
CC2
100pF
COSC
TG
RUN/SS
BOOST
ITH
SW
16
6
7
R2
113k
1%
M2
Si4450DY
M1
IR2910
47Ω
MBRS1100
22Ω
15
14
RSENSE
0.004Ω
1nF
100V
R1
8.06k
1%
1nF
100V
LTC1735
4
5
FCB
SGND
VIN
INTVCC
13
12
47pF
6
VOSENSE
BG
7
SENSE –
PGND
8
SENSE +
EXTVCC
11
0.1µF
+
4.7µF
10
3300pF
100Ω
3
•
1M
COSC
150pF
9
CIN: MARCON THCR70EIH226ZT
COUT: AVX TPSV227M016R0150
T1: COILTRONICS VP5-0155
RSENSE: IRC LRF2512-01-R004F
1735 TA07
28
VOUT
12V
3A
T1
10
•
VOUT
+
COUT
470µF
16V
×4
LTC1735
U
TYPICAL APPLICATIO S
5V/3.5A Converter with 12V/200mA Auxiliary Output
VIN
5.5V TO 28V
COSC
51pF
1
CSS
0.1µF
2
RC
33k
CIN
22µF
30V
OS-CON
22Ω
+
CC
470pF
3
CC2
220pF
COSC
TG
RUN/SS
BOOST
ITH
SW
16
M1
IRF7803
CB
0.1µF
MBRS1100T3
14
+
LTC1735
4
5
FCB
VIN
SGND
INTVCC
1000pF
15
13
RSENSE
0.012Ω
DB
CMDSH-3
T1
1:1.8
10µH
12
+
100pF
CSEC
22µF
35V
AVX
R2
105k
1%
VOUT
5V
3.5A
COUT
100µF
10V
×3
AVX
+
4.7µF
6
7
VOSENSE
BG
SENSE –
PGND
11
M2
IRF7803
R1
20k
1%
MBRS140T3
10
SGND
1000pF
8
SENSE
+
EXTVCC
9
CIN: SANYO OS-CON 305C22M
COUT: AVX TPSD107M010R0068
T1: 1:8 DALE LPE6562-A262
100Ω
100Ω
10k
90.9k
1735 TA04
VOUT2
12V
120mA
UNREG
Dual Output 15W 3.3V/5V Power Supply
VIN
4.5V TO 28V
COSC
47pF
+
1
CSS
0.1µF
2
CC
470pF
TG
RUN/SS
BOOST
CB
0.1µF
CIN
22µF
50V
M1
Si4412DY
•
ITH
SW
0.01µF
CC2
100pF
FCB
VIN
SGND
INTVCC
4.7k
13
DB
CMDSH-3
•
T1A
MBRS140T3
RSENSE
0.01Ω
T1B
•
8 2
7
R2
62.6k
1%
12
+
100pF
VOSENSE
BG
7
SENSE –
PGND
8
SENSE +
EXTVCC
COUT2
100µF
10V
×2
VOUT1
3.3V
2.5A
+
4.7µF
6
+
VOUT2
5V
1.5A
14
1
5
6
M3
Si4412DY
15
LTC1735
4
T1C
3
CMDSH-3
3
RC
33k
COSC
16
11
M2
Si4412DY
R1
20k
1%
MBRS140T3
COUT1
100µF
10V
×2
10
1000pF
9
SGND
VOUT2
1735 TA08
CIN: MARCON THCR70EIH226ZT
COUT1, 2: AVX TPSD107M010R0065
T1: BI TECHNOLOGIES HM00-93839
RSENSE: IRC LRF2512-01-R010 F
29
LTC1735
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
GN Package
16-Lead Plastic SSOP (Narrow 0.150)
(LTC DWG # 05-08-1641)
0.189 – 0.196*
(4.801 – 4.978)
16 15 14 13 12 11 10 9
0.229 – 0.244
(5.817 – 6.198)
0.150 – 0.157**
(3.810 – 3.988)
1
0.015 ± 0.004
× 45°
(0.38 ± 0.10)
0.007 – 0.0098
(0.178 – 0.249)
0.053 – 0.068
(1.351 – 1.727)
2 3
4
5 6
7
8
0.004 – 0.0098
(0.102 – 0.249)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
* DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
30
0.009
(0.229)
REF
0.008 – 0.012
(0.203 – 0.305)
0.0250
(0.635)
BSC
GN16 (SSOP) 1098
LTC1735
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
S Package
16-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
16
15
14
13
12
11
10
9
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
2
3
4
5
6
0.053 – 0.069
(1.346 – 1.752)
0.008 – 0.010
(0.203 – 0.254)
0.014 – 0.019
(0.355 – 0.483)
TYP
8
0.004 – 0.010
(0.101 – 0.254)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
7
0.050
(1.270)
BSC
S16 1098
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC1735
U
TYPICAL APPLICATIO
3.3V to 2.5V/5A Converter with External Clock Synchronization Operating at 500kHz
VIN
3.3V
5V 0.1µF
COSC
20pF
+
1
CSS
0.1µF
2
CC
330pF
3
RC
33k
COSC
TG
RUN/SS
BOOST
ITH
SW
16
CB
0.1µF
M1
Si4410DY
15
CIN: SANYO OS-CON 10SL100M
COUT: AVX TPSD107M010R0065
L1: COILCRAFT DO3316P-152
RSENSE: IRC LR2010-01-R010-F
14
LTC1735
CC2
51pF
EXT
CLOCK
500kHz
4
5
FCB
SGND
VIN
INTVCC
CIN
100µF
10V
OS-CON
13
L1
1.5µH
DB
CMDSH-3
12
RSENSE
0.01Ω
47pF
+
47pF
R2
43.2k
1%
VOUT
2.5V
5A
+
4.7µF
6
7
VOSENSE
BG
SENSE –
PGND
11
10
M2
Si4410DY
R1
20k
1%
MBRS140T3
COUT
100µF
10V
AVX
×3
SGND
1000pF
8
SENSE +
EXTVCC
9
1735 TA06
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1147
High Efficiency Step-Down Controller
100% DC*, Burst Mode Operation, SO-8
LTC1148HV/LTC1148
High Efficiency Synchronous Step-Down Controller
100% DC*, Burst Mode Operation, VIN < 20V
LTC1149
High Efficiency Synchronous Step-Down Controller
100% DC*, Std Threshold MOSFETs, VIN < 48V
LTC1159
High Efficiency Synchronous Step-Down Controller
100% DC*, Logic Level MOSFETs, VIN < 40V
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High Efficiency Synchronous Step-Down Controller, N-Ch Drive
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1.5A 500kHz Step-Down Switching Regulator
High Efficiency, Monolithic, SO-8
LTC1435A
High Efficiency Synchronous Step-Down Controller, N-Ch Drive
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LTC1436/LTC1436-PLL
High Efficiency Low Noise Synchronous Step-Down Converter, N-Ch Drive
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100% DC*, 8-Pin MSOP, 10µA IQ
LTC1624
High Efficiency SO-8 N-Channel Switching Regulator Controller
95% DC*, 3.5V to 36V VIN
LTC1625/LTC1775
No RSENSETM Current Mode Synchronous Step-Down Controller
Burst Mode Operation, 16-Pin SSOP
LTC1627
Synchronous Monolithic 0.5A Step-Down Regulator
100% DC*, 2.6V to 8.5V VIN, SO-8
LTC1628
Dual High Efficiency 2-Phase Step-Down Controller
Antiphase Drive for Reduced Input Capacitance
LTC1702
550kHz Dual Output Synchronous Step-Down Controller
Antiphase Drive, 24-Pin SSOP
LTC1735-1
High Efficiency Step-Down Controller with Power Good
Output Fault Protection, 16-Pin SO/SSOP
LTC1736
High Efficiency Step-Down Controller with VID Control
Output Fault Protection, 24-Pin SSOP
LTC1772
SOT-23 Step-Down Controller
100% DC*, Up to 4A, 2.2V to 9.8V VIN
No RSENSE is trademark of Linear Technology Corporation. *DC = Duty Cycle
32
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
1735f LT/TP 1199 4K • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 1998
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