LINER LTC1629-6 Polyphase, synchronous step-down switching regulator Datasheet

LTC1629-6
PolyPhase, Synchronous
Step-Down Switching Regulator
DESCRIPTIO
U
FEATURES
■
■
■
■
■
■
■
■
■
■
■
■
■
■
■
■
Reduces Required Input Capacitance and Power
Supply Induced Noise
True Remote Sensing Differential Amplifier
0.6V, ±1% Voltage Reference Accuracy
PolyPhase Controller Operates from Two to Twelve
Phases
Current Mode Control Ensures Current Sharing
Phase-Lockable Fixed Frequency: 150kHz to 300kHz
1.8MHz Effective Switching Frequency Capability
OPTI-LOOP® Compensation Reduces COUT
Power Good Output Voltage Monitor
Wide VIN Range: 4V to 36V Operation
Very Low Dropout Operation: 99% Duty Cycle
Adjustable Soft-Start Current Ramping
Internal Current Foldback Plus Shutdown Timer
Overvoltage Soft-Latch Eliminates Nuisance Trips
Micropower Shutdown
Available in 28-Lead SSOP Package
U
APPLICATIO S
■
■
■
■
The LTC®1629-6 is a PolyPhaseTM synchronous stepdown
current mode switching regulator controller with 0.6V
reference that drives N-channel external power MOSFET
stages in a phase-lockable fixed frequency architecture.
The PolyPhase controller drives its two output stages out
of phase at frequencies up to 300kHz to minimize the RMS
ripple currents in both input and output capacitors. The
output clock signal allows expansion for up to 12 evenly
phased controllers for systems requiring 15A to 200A of
output current. The multiple phase technique effectively
multiplies the fundamental frequency by the number
of channels used, improving transient response while
operating each channel at an optimum frequency for
efficiency. Thermal design is also simplified.
An internal differential amplifier provides true remote
sensing of the regulated supply’s positive and negative
output terminals as required for high current applications.
A RUN/SS pin provides both soft-start and optional timed,
short-circuit shutdown. Current foldback limits MOSFET
dissipation during short-circuit conditions when the
overcurrent latchoff is disabled. OPTI-LOOP compensation allows the transient response to be optimized over a
wide range of output capacitance and ESR values. The
LTC1629-6 includes a power good output pin that
indicates when the output voltage is inside the ±10%
tolerance window.
Desktop Computers
Internet Servers
Large Memory Arrays
DC Power Distribution Systems
, LTC and LT are registered trademarks of Linear Technology Corporation.
OPTI-LOOP is a registered trademark of Linear Technology Corporation.
PolyPhase is a trademark of Linear Technology Corporation.
U
TYPICAL APPLICATIO
10Ω
10µF ×6
35V
CERAMIC
0.1µF
S
LTC1629-6
VIN
0.1µF
3300pF
BG1
PGOOD
PGND
M3
TG2
S
BOOST2
SW2
VDIFFOUT
S
BG2
EAIN
INTVCC
–
+
VOS +
SENSE2
SENSE2 –
COUT: T510E108K004AS
D1, D2: UPS840
0.002Ω
0.47µF
S
VOS
D1
SENSE1 –
10k
60.4k
M2
×2
0.002Ω
L1
1µH
SENSE1 +
SGND
S
0.47µF
S
RUN/SS
CLKOUT
S
M1
S
SW1
ITH
1.5k
TG1
BOOST1
VIN
5V TO 15V
M4
×2
D2
VOUT
0.7V/40A
L2
1µH
S
10µF CER
X5R
L1, L2: CEPH149-IROMC-H
+
M1, M3: Si7840DP
M2, M4: Si7440DP
Figure 1. High Current Dual Phase Step-Down Converter
COUT
270µF ×4
2V
1629 TA01
16296f
1
LTC1629-6
U
W W
W
ABSOLUTE
AXI U RATI GS
U
W
U
PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Supply Voltage (VIN).........................36V to – 0.3V
Topside Driver Voltages (BOOST1,2) .........42V to – 0.3V
Switch Voltage (SW1, 2) .............................36V to – 5 V
SENSE1+, SENSE2 +, SENSE1–,
SENSE2 – Voltages ........................ (1.1)INTVCC to – 0.3V
EAIN, VOS+, VOS–, EXTVCC, INTVCC,
RUN/SS, PGOOD Voltages ...........................7V to – 0.3V
Boosted Driver Voltage (BOOST-SW) ..........7V to – 0.3V
PLLFLTR, PLLIN, CLKOUT, PHASMD,
VDIFFOUT Voltages ................................ INTVCC to – 0.3V
ITH Voltage ................................................2.7V to – 0.3V
Peak Output Current <1µs(TGL1,2, BG1,2) ................ 5A
INTVCC RMS Output Current ................................ 50mA
Operating Ambient Temperature Range
LTC1629EG-6 (Note 6) ......................... – 40°C to 85°C
Junction Temperature (Note 2) ............................. 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
RUN/SS
1
28 CLKOUT
SENSE1 +
2
27 TG1
SENSE1 –
3
26 SW1
LTC1629EG-6
EAIN
4
25 BOOST1
PLLFLTR
5
24 VIN
PLLIN
6
23 BG1
PHASMD
7
22 EXTVCC
ITH
8
21 INTVCC
SGND
9
20 PGND
VDIFFOUT 10
19 BG2
VOS – 11
18 BOOST2
VOS +
12
17 SW2
SENSE2 – 13
16 TG2
SENSE2 + 14
15 PGOOD
G PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 95°C/W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
●
0.594
0.600
0.606
V
●
62
65
75
75
88
85
mV
mV
–5
– 50
nA
0.1
– 0.1
0.5
– 0.5
%
%
0.002
0.02
%/V
0.64
0.66
0.68
V
3
3.5
4
Main Control Loop
VEAIN
Regulated Feedback Voltage
(Note 3); ITH Voltage = 1.2V
– = 5V
VSENSEMAX
Maximum Current Sense Threshold
VSENSE
IINEAIN
Feedback Current
(Note 3)
VLOADREG
Output Voltage Load Regulation
(Note 3)
Measured in Servo Loop; ITH Voltage = 0.7V
Measured in Servo Loop; ITH Voltage = 2V
VREFLNREG
Reference Voltage Line Regulation
VIN = 3.6V to 30V (Note 3)
VOVL
Output Overvoltage Threshold
Measured at VEAIN
UVLO
Undervoltage Lockout
VIN Ramping Down
gm
Transconductance Amplifier gm
ITH = 1.2V; Sink/Source 5µA; (Note 3)
gmOL
Transconductance Amplifier Gain
IQ
●
●
●
V
3
mmho
ITH = 1.2V; (gmxZL; No Ext Load); (Note 3)
1.5
V/mV
Input DC Supply Current
Normal Mode
Shutdown
(Note 4)
EXTVCC Tied to VOUT; VOUT = 5V
VRUN/SS = 0V
470
20
IRUN/SS
Soft-Start Charge Current
VRUN/SS = 1.9V
– 0.5
– 1.2
VRUN/SS
RUN/SS Pin ON Threshold
VRUN/SS Rising
1.0
1.5
1.9
V
VRUN/SSLO
RUN/SS Pin Latchoff Arming
VRUN/SS Rising from 3V
4.1
4.5
V
40
µA
µA
µA
16296f
2
LTC1629-6
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
ISCL
RUN/SS Discharge Current
Soft Short Condition VEAIN = 0.4V; VRUN/SS = 4.5V
0.5
2
4
ISDLDO
Shutdown Latch Disable Current
VEAIN = 0.4V
1.6
5
ISENSE
Total Sense Pins Source Current
Each Channel; VSENSE1 –, 2 – = VSENSE1+, 2 + = 0V
– 85
– 60
µA
DFMAX
Maximum Duty Factor
In Dropout
98
99.5
%
TG1, 2 tr
TG1, 2 tf
Top Gate Transition Time:
Rise Time
Fall Time
CLOAD = 3300pF
CLOAD = 3300pF
30
40
90
90
ns
ns
BG1, 2 tr
BG1, 2 tf
Bottom Gate Transition Time:
Rise Time
Fall Time
CLOAD = 3300pF
CLOAD = 3300pF
30
20
90
90
ns
ns
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
90
ns
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
90
ns
Minimum On-Time
Tested with a Square Wave (Note 5)
180
ns
TG/BG t1D
BG/TG t2D
tON(MIN)
UNITS
µA
µA
Internal VCC Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 30V; VEXTVCC = 4V
VLDO INT
INTVCC Load Regulation
ICC = 0 to 20mA; VEXTVCC = 4V
VLDO EXT
EXTVCC Voltage Drop
ICC = 20mA; VEXTVCC = 5V
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, EXTVCC Ramping Positive
VLDOHYS
EXTVCC Switchover Hysteresis
ICC = 20mA, EXTVCC Ramping Negative
4.8
●
4.5
5.0
5.2
V
0.2
1.0
%
80
160
mV
4.7
V
0.2
V
Oscillator and Phase-Locked Loop
fNOM
Nominal Frequency
VPLLFLTR = 1.2V
190
220
250
kHz
fLOW
Lowest Frequency
VPLLFLTR = 0V
120
140
160
kHz
fHIGH
Highest Frequency
VPLLFLTR ≥ 2.4V
280
310
360
kHz
RPLLIN
PLLIN Input Resistance
IPLLFLTR
Phase Detector Output Current
Sinking Capability
Sourcing Capability
50
kΩ
fPLLIN < fOSC
fPLLIN > fOSC
– 15
15
µA
µA
RRELPHS
Controller 2-Controller 1 Phase
VPHASMD = 0V, Open
VPHASMD = 5V
180
240
Deg
Deg
CLKOUT
Phase (Relative to Controller 1)
VPHASMD = 0V
VPHASMD = Open
VPHASMD = 5V
60
90
120
Deg
Deg
Deg
CLKHIGH
Clock High Output Voltage
CLKLOW
Clock Low Output Voltage
4
V
0.2
V
PGOOD Output
VPGL
PGOOD Voltage Low
IPGOOD = 2mA
IPGOOD
PGOOD Leakage Current
VPGOOD = 5V
VPG
PGOOD Trip Level, Either Controller
VEAIN with Respect to Set Output Voltage
VEAIN Ramping Negative
VEAIN Ramping Positive
0.1
–8
8
– 10
10
0.3
V
±1
µA
– 12
12
%
%
16296f
3
LTC1629-6
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0.995
1
1.005
V/V
46
55
dB
80
kΩ
Differential Amplifier Gain Block (Note 5)
ADA
Gain
CMRRDA
Common Mode Rejection Ratio
0V < VCM < 5V
RIN
Input Resistance
Measured at VOS + Input
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 5: The minimum on-time condition corresponds to the on inductor
peak-to-peak ripple current ≥ 40% of IMAX (see Minimum On-Time
Considerations in the Applications Information section).
Note 6: The LTC1629EG-6 is guaranteed to meet performance
specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls.
Note 1: Absolute Maximum Ratings are those values beyond which the
life of a device may be impaired.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
TJ = TA + (PD • 95°C/W)
Note 3: The I.C. is tested in a feedback loop that servos VITH to a
specified voltage and measures the resultant VEAIN.
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Output Current
(Figure 12)
Efficiency vs Output Current
(Figure 12)
100
Efficiency vs Input Voltage
(Figure 12)
100
100
VOUT = 3.3V
VEXTVCC = 5V
IOUT = 20A
VEXTVCC = 5V
80
90
VIN = 12V
VIN = 20V
40
VOUT = 3.3V
VEXTVCC = 5V
IOUT = 20A
20
1
10
OUTPUT CURRENT (A)
80
70
90
80
60
VOUT = 3.3V
0
0.1
VEXTVCC = 0V
EFFICIENCY (%)
VIN = 8V
60
EFFICIENCY (%)
EFFICIENCY (%)
VIN = 5V
100
1629 G01
50
70
1
10
OUTPUT CURRENT (A)
100
5
10
15
20
VIN (V)
1629 G02
1629 G03
16296f
4
LTC1629-6
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Supply Current vs Input Voltage
and Mode
INTVCC and EXTVCC Switch
Voltage vs Temperature
EXTVCC Voltage Drop
1000
250
5.05
EXTVCC VOLTAGE DROP (mV)
SUPPLY CURRENT (µA)
800
EXTVCC PIN OPEN
600
400
200
INTVCC AND EXTVCC SWITCH VOLTAGE (V)
ON
200
150
100
50
SHUTDOWN
0
0
20
15
10
25
INPUT VOLTAGE (V)
5
30
0
35
10
0
30
20
CURRENT (mA)
40
1629 G04
4.85
4.80
EXTVCC SWITCHOVER THRESHOLD
4.75
50
25
75
0
TEMPERATURE (°C)
100
125
1629 G06
Maximum Current Sense Threshold
vs Percent of Nominal Output
Voltage (Foldback)
Maximum Current Sense Threshold
vs Duty Factor
75
80
ILOAD = 1mA
70
60
4.8
4.7
50
VSENSE (mV)
4.9
VSENSE (mV)
INTVCC VOLTAGE (V)
4.90
4.70
– 50 – 25
50
5.0
25
4.6
50
40
30
20
4.5
10
0
4.4
0
20
15
25
10
INPUT VOLTAGE (V)
5
30
0
35
20
40
60
DUTY FACTOR (%)
80
Maximum Current Sense Threshold
vs VRUN/SS (Soft-Start)
80
0
100
50
100
0
25
75
PERCENT ON NOMINAL OUTPUT VOLTAGE (%)
1629 G08
1629 G07
1629 G09
Current Sense Threshold
vs ITH Voltage
Maximum Current Sense Threshold
vs Sense Common Mode Voltage
80
VSENSE(CM) = 1.6V
90
80
70
76
60
40
VSENSE (mV)
60
VSENSE (mV)
VSENSE (mV)
4.95
1629 G05
Internal 5V LDO Line Reg
5.1
INTVCC VOLTAGE
5.00
72
68
50
40
30
20
10
20
0
64
–10
–20
60
0
0
1
2
3
4
5
6
VRUN/SS (V)
1629 G10
0
1
3
4
2
COMMON MODE VOLTAGE (V)
5
1629 G11
–30
0
0.5
1
1.5
VITH (V)
2
2.5
1629
G12
16296f
5
LTC1629-6
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Load Regulation
VITH vs VRUN/SS
2.5
FCB = 0V
VIN = 15V
FIGURE 1
100
VOSENSE = 0.7V
2.0
–0.2
50
ISENSE (µA)
–0.1
VITH (V)
NORMALIZED VOUT (%)
0.0
SENSE Pins Total Source Current
1.5
1.0
–0.3
0
–50
0.5
–0.4
0
10
30
20
LOAD CURRENT (A)
0
40
50
0
1
2
3
4
5
–100
6
VRUN/SS (V)
2
0
4
1629 G14
1629 G13
6
VSENSE COMMON MODE VOLTAGE (V)
Maximum Current Sense
Threshold vs Temperature
1629 G15
RUN/SS Current vs Temperature
80
1.8
78
1.4
RUN/SS CURRENT (µA)
VSENSE (mV)
1.6
76
74
72
1.2
1.0
0.8
0.6
0.4
0.2
70
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
125
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
1629 G17
100
125
1629 G19
Load Step (Figure 12)
Soft-Start Up (Figure 12)
VITH
1V/DIV
IOUT
O/30A
VITH
1V/DIV
VOUT
2V/DIV
VOUT
200mV/DIV
VRUNSS
2V/DIV
100ms/DIV
1629 G20
10µs/DIV
1629 G21
16296f
6
LTC1629-6
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Current Sense Pin Input Current
vs Temperature
EXTVCC Switch Resistance
vs Temperature
10
350
VOUT = 5V
VPLLFLTR = 2.4V
33
31
29
27
25
–50 –25
50
25
0
75
TEMPERATURE (°C)
100
300
8
6
4
200
VPLLFLTR = 0V
150
100
50
50
25
0
75
TEMPERATURE (°C)
1629 G23
100
0
– 50 – 25
125
50
25
75
0
TEMPERATURE (°C)
1629 G24
Undervoltage Lockout
vs Temperature
100
125
1629 G25
Shutdown Latch Thresholds
vs Temperature
4.5
SHUTDOWN LATCH THRESHOLDS (V)
3.50
UNDERVOLTAGE LOCKOUT (V)
250
2
0
–50 –25
125
FREQUENCY (kHz)
EXTVCC SWITCH RESISTANCE (Ω)
CURRENT SENSE INPUT CURRENT (µA)
35
Oscillator Frequency
vs Temperature
3.45
3.40
3.35
3.30
3.25
3.20
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1629 G26
LATCH ARMING
4.0
3.5
3.0
LATCHOFF
THRESHOLD
2.5
2.0
1.5
1.0
0.5
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
1629 G27
U
U
U
PI FU CTIO S
RUN/SS (Pin 1): Combination of Soft-Start, Run Control
Input and Short-Circuit Detection Timer. A capacitor to
ground at this pin sets the ramp time to full current output.
Forcing this pin below 0.8V causes the IC to shut down all
internal circuitry. All functions are disabled in shutdown.
SENSE1+, SENSE2+ (Pins 2,14): The (+) Input to the
Differential Current Comparators. The ITH pin voltage and
built-in offsets between SENSE– and SENSE+ pins in
conjunction with RSENSE set the current trip threshold.
SENSE1–, SENSE2– (Pins 3, 13): The (–) Input to the
Differential Current Comparators.
EAIN (Pin 4): Input to the Error Amplifier that compares
the feedback voltage to the internal 0.6V reference voltage.
This pin is normally connected to a resistive divider from
the output of the differential amplifier (DIFFOUT).
PLLFLTR (Pin 5): The Phase-Locked Loop’s Low Pass
Filter is tied to this pin. Alternatively, this pin can be driven
with an AC or DC voltage source to vary the frequency of
the internal oscillator.
16296f
7
LTC1629-6
U
U
U
PI FU CTIO S
PLLIN (Pin 6): External Synchronization Input to Phase
Detector. This pin is internally terminated to SGND with
50kΩ. The phase-locked loop will force the rising top gate
signal of controller 1 to be synchronized with the rising
edge of the PLLIN signal.
PHASMD (Pin 7): Control Input to Phase Selector which
determines the phase relationships between controller 1,
controller 2 and the CLKOUT signal.
ITH (Pin 8): Error Amplifier Output and Switching Regulator Compensation Point. Both current comparator’s thresholds increase with this control voltage. The normal voltage
range of this pin is from 0V to 2.4V.
SGND (Pin 9): Signal Ground, common to both controllers, must be routed separately from the input switched
current ground path to the common (–) terminal(s) of the
COUT capacitor(s).
VDIFFOUT (Pin 10): Output of a Differential Amplifier that
provides true remote output voltage sensing. This pin
normally drives an external resistive divider that sets the
output voltage.
VOS–, VOS+ (Pins 11, 12): Inputs to an Operational Amplifier. Internal precision resistors capable of being electronically switched in or out can configure it as a differential amplifier or an uncommitted Op Amp.
PGOOD (Pin 15): Open-Drain Logic Output. PGOOD is
pulled to ground when the voltage on the EAIN pin is not
within ±10% of its set point.
TG2, TG1 (Pins 16, 27): High Current Gate Drives for Top
N-Channel MOSFETS. These are the outputs of floating
drivers with a voltage swing equal to INTVCC superimposed on the switch node voltage SW.
SW2, SW1 (Pins 17, 26): Switch Node Connections to
Inductors. Voltage swing at these pins is from a Schottky
diode (external) voltage drop below ground to VIN.
BOOST2, BOOST1 (Pins 18, 25): Bootstrapped Supplies
to the Topside Floating Drivers. Capacitors are connected
between the Boost and Switch pins and Schottky diodes
are tied between the Boost and INTVCC pins. Voltage swing
at the Boost pins is from INTVCC to (VIN + INTVCC).
BG2, BG1 (Pins 19, 23): High Current Gate Drives for
Bottom Synchronous N-Channel MOSFETS. Voltage swing
at these pins is from ground to INTVCC.
PGND (Pin 20): Driver Power Ground. Connect to sources
of bottom N-channel MOSFETS and the (–) terminals of
CIN.
INTVCC (Pin 21): Output of the Internal 5V Linear Low
Dropout Regulator and the EXTVCC Switch. The driver and
control circuits are powered from this voltage source.
Decouple to power ground with a 1µF ceramic capacitor
placed directly adjacent to the IC and minimum of 4.7µF
additional tantalum or other low ESR capacitor.
EXTVCC (Pin 22): External Power Input to an Internal
Switch . This switch closes and supplies INTVCC, bypassing the internal low dropout regulator whenever EXTVCC is
higher than 4.7V. See EXTVCC Connection in the Applications Information section. Do not exceed 7V on this pin
and ensure VEXTVCC ≤ VINTVCC.
VIN (Pin 24): Main Supply Pin. Should be closely decoupled
to the IC’s signal ground pin.
CLKOUT (Pin 28): Output Clock Signal available to
daisychain other controller ICs for additional MOSFET
driver stages/phases.
16296f
8
LTC1629-6
W
FU CTIO AL DIAGRA
U
U
PLLIN
PHASE DET
FIN
50k
PLLLPF
RLP
CLKOUT
CLK1
CLP
OSCILLATOR
INTVCC
CLK2
DUPLICATE FOR
SECOND
CONTROLLER
CHANNEL
PHASMD
±2µA
DROP
OUT
DET
DIFFOUT
VOS
40k
–
S
Q
R
Q
BOT
CB
TG
TOP
+
CIN
FCB
SW
FORCE BOT
SWITCH
LOGIC
40k
INTVCC
BOT
BG
PGND
A1
VOS +
DB
BOOST
PHASE LOGIC
VIN
–
+
40k
SHDN
INTVCC
40k
I1
PGOOD
–
L
–
+
0.66V
+
–
+
30k SENSE
+
0.86V
5(VFB)
EAIN
SLOPE
COMP
0.54V
COUT
+
+
RSENSE
–
30k SENSE
–
45k
45k
VOUT
2.4V
EAIN
0.6V
VIN
–
EA
+
VREF
VIN
OV
4.7V
EXTVCC
+
–
5V
LDO
REG
RST
6V
SGND
INTERNAL
SUPPLY
R2
0.66V
ITH
CC
1.2µA
INTVCC
+
0.60V
+
–
SHDN
5V
R1
RUN
SOFT
START
RC
RUN/SS
CSS
1629 FBD
16296f
9
LTC1629-6
U
OPERATIO
(Refer to Functional Diagram)
Main Control Loop
Low Current Operation
The LTC1629-6 uses a constant frequency, current mode
step-down architecture. During normal operation, the top
MOSFET is turned on each cycle when the oscillator sets
the RS latch, and turned off when the main current
comparator, I1, resets the RS latch. The peak inductor
current at which I1 resets the RS latch is controlled by the
voltage on the ITH pin, which is the output of the error
amplifier EA. The differential amplifier, A1, produces a
signal equal to the differential voltage sensed across the
output capacitor but re-references it to the internal signal
ground (SGND) reference. The EAIN pin receives a portion
of this voltage feedback signal at the DIFFOUT pin which
is compared to the internal reference voltage by the EA.
When the load current increases, it causes a slight decrease in the EAIN pin voltage relative to the 0.6V reference, which in turn causes the ITH voltage to increase until
the average inductor current matches the new load current. After the top MOSFET has turned off, the bottom
MOSFET is turned on for the rest of the period.
The LTC1629-6 operates in a continuous, PWM control
mode. The resulting operation at low output currents
optimizes transient response at the expense of substantial
negative inductor current during the latter part of the
period. The level of ripple current is determined by the
inductor value, input voltage, output voltage, and frequency of operation.
The top MOSFET drivers are biased from floating bootstrap capacitor CB, which normally is recharged during
each off cycle through an external Schottky diode. When
VIN decreases to a voltage close to VOUT, however, the loop
may enter dropout and attempt to turn on the top MOSFET
continuously. A dropout detector detects this condition
and forces the top MOSFET to turn off for about 400ns
every 10th cycle to recharge the bootstrap capacitor.
The main control loop is shut down by pulling Pin 1 (RUN/
SS) low. Releasing RUN/SS allows an internal 1.2µA
current source to charge soft-start capacitor CSS. When
CSS reaches 1.5V, the main control loop is enabled with the
ITH voltage clamped at approximately 30% of its maximum
value. As CSS continues to charge, ITH is gradually released allowing normal operation to resume. When the
RUN/SS pin is low, all LTC1629-6 functions are shut
down. If VOUT has not reached 70% of its nominal value
when CSS has charged to 4.1V, an overcurrent latchoff can
be invoked as described in the Applications Information
section.
Frequency Synchronization
The phase-locked loop allows the internal oscillator to be
synchronized to an external source via the PLLIN pin. The
output of the phase detector at the PLLFLTR pin is also the
DC frequency control input of the oscillator that operates
over a 140kHz to 310kHz range corresponding to a DC
voltage input from 0V to 2.4V. When locked, the PLL aligns
the turn on of the top MOSFET to the rising edge of the
synchronizing signal. When PLLIN is left open, the PLLFLTR
pin goes low, forcing the oscillator to minimum frequency.
The internal master oscillator runs at a frequency twelve
times that of each controller’s frequency. The PHASMD
pin determines the relative phases between the internal
controllers as well as the CLKOUT signal as shown in
Table␣ 1. The phases tabulated are relative to zero phase
being defined as the rising edge of the top gate (TG1)
driver output of controller 1.
Table 1.
VPHASMD
GND
OPEN
INTVCC
Controller 2
180°
180°
240°
CLKOUT
60°
90°
120°
The CLKOUT signal can be used to synchronize additional
power stages in a multiphase power supply solution
feeding a single, high current output or separate outputs.
Input capacitance ESR requirements and efficiency losses
are substantially reduced because the peak current drawn
from the input capacitor is effectively divided by the
number of phases used and power loss is proportional to
the RMS current squared. A two stage, single output
voltage implementation can reduce input path power loss
by 75% and radically reduce the required RMS current
rating of the input capacitor(s).
16296f
10
LTC1629-6
U
OPERATIO
(Refer to Functional Diagram)
INTVCC/EXTVCC Power
Power Good (PGOOD)
Power for the top and bottom MOSFET drivers and most
of the IC circuitry is derived from INTVCC. When the
EXTVCC pin is left open, an internal 5V low dropout
regulator supplies INTVCC power. If the EXTVCC pin is
taken above 4.7V, the 5V regulator is turned off and an
internal switch is turned on connecting EXTVCC to INTVCC.
This allows the INTVCC power to be derived from a high
efficiency external source such as the output of the regulator itself or a secondary winding, as described in the
Applications Information section. An external Schottky
diode can be used to minimize the voltage drop from
EXTVCC to INTVCC in applications requiring greater than
the specified INTVCC current. Voltages up to 7V can be
applied to EXTVCC for additional gate drive capability.
The PGOOD pin is connected to the drain of an internal
MOSFET. The MOSFET turns on when the output is not
within ±10% of its nominal output level as determined by
the feedback divider. When the output is within ±10% of
its nominal value, the MOSFET is turned off within 10µs
and the PGOOD pin should be pulled up by an external
resistor to a source of up to 7V.
Differential Amplifier
This amplifier provides true differential output voltage
sensing. Sensing both VOUT + and VOUT – benefits regulation in high current applications and/or applications having electrical interconnection losses. The amplifier has an
output slew rate of 5V/µs and is capable of driving capacitive loads with an output RMS current typically up to
25mA. The amplifier is not capable of sinking current and
therefore must be resistively loaded to do so. The differential amplifier is configured as a unity-gain differencing
amplifier.
Short-Circuit Detection
The RUN/SS capacitor is used initially to limit the inrush
current from the input power source. Once the controllers
have been given time, as determined by the capacitor on
the RUN/SS pin, to charge up the output capacitors and
provide full load current, the RUN/SS capacitor is then
used as a short-circuit timeout circuit. If the output voltage
falls to less than 70% of its nominal output voltage the
RUN/SS capacitor begins discharging assuming that the
output is in a severe overcurrent and/or short-circuit
condition. If the condition lasts for a long enough period
as determined by the size of the RUN/SS capacitor, the
controller will be shut down until the RUN/SS pin voltage
is recycled. This built-in latchoff can be overidden by
providing a >5µA pull-up current at a compliance of 5V to
the RUN/SS pin. This current shortens the soft-start
period but also prevents net discharge of the RUN/SS
capacitor during a severe overcurrent and/or short-circuit
condition. Foldback current limiting is activated when the
output voltage falls below 70% of its nominal level whether
or not the short-circuit latchoff circuit is enabled.
U
W
U
U
APPLICATIO S I FOR ATIO
The basic LTC1629-6 application circuit is shown in Figure␣ 1 on the first page. External component selection is
driven by the load requirement, and begins with the
selection of RSENSE1, 2. Once RSENSE1, 2 are known, L1 and
L2 can be chosen. Next, the power MOSFETs and D1 and
D2 are selected. The operating frequency and the inductor
are chosen based mainly on the amount of ripple current.
Finally, CIN is selected for its ability to handle the input
ripple current (that PolyPhase operation minimizes) and
COUT is chosen with low enough ESR to meet the output
ripple voltage and load step specifications (also minimized
with PolyPhase). The circuit shown in Figure␣ 1 can be
configured for operation up to an input voltage of 28V
(limited by the external MOSFETs).
16296f
11
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
2.5
RSENSE Selection For Output Current
PLLFLTR PIN VOLTAGE (V)
RSENSE1, 2 are chosen based on the required output
current. The LTC1629-6 current comparator has a maximum threshold of 75mV/RSENSE and an input common
mode range of SGND to 1.1(INTVCC). The current comparator threshold sets the peak inductor current, yielding
a maximum average output current IMAX equal to the peak
value less half the peak-to-peak ripple current, ∆IL.
Allowing a margin for variations in the LTC1629-6 and
external component values yields:
The LTC1629-6 uses a constant frequency, phase-lockable architecture with the frequency determined by an
internal capacitor. This capacitor is charged by a fixed
current plus an additional current which is proportional to
the voltage applied to the PLLFLTR pin. Refer to PhaseLocked Loop and Frequency Synchronization in the Applications Information section for additional information.
A graph for the voltage applied to the PLLFLTR pin vs
frequency is given in Figure␣ 2. As the operating frequency
is increased the gate charge losses will be higher, reducing
efficiency (see Efficiency Considerations). The maximum
switching frequency is approximately 310kHz.
Inductor Value Calculation and Output Ripple Current
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
1.0
0.5
170
220
270
OPERATING FREQUENCY (kHz)
320
1629 F02
Figure 2. Operating Frequency vs VPLLFLTR
where N = number of stages.
Operating Frequency
1.5
0
120
RSENSE = (50mV/IMAX)N
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due to
internal slope compensation required to meet stability
criterion for buck regulators operating at greater than 50%
duty factor. A curve is provided to estimate this reduction
in peak output current level depending upon the operating
duty factor.
2.0
MOSFET gate charge and transition losses. In addition to
this basic tradeoff, the effect of inductor value on ripple
current and low current operation must also be considered. The PolyPhase approach reduces both input and
output ripple currents while optimizing individual output
stages to run at a lower fundamental frequency, enhancing
efficiency.
The inductor value has a direct effect on ripple current. The
inductor ripple current ∆IL per individual section, N,
decreases with higher inductance or frequency and increases with higher VIN or VOUT:
∆IL =
VOUT  VOUT 
 1−

fL 
VIN 
where f is the individual output stage operating frequency.
In a PolyPhase converter, the net ripple current seen by the
output capacitor is much smaller than the individual
inductor ripple currents due to the ripple cancellation. The
details on how to calculate the net output ripple current
can be found in Application Note 77.
Figure 3 shows the net ripple current seen by the output
capacitors for the different phase configurations. The
output ripple current is plotted for a fixed output voltage as
the duty factor is varied between 10% and 90% on the
x-axis. The output ripple current is normalized against the
inductor ripple current at zero duty factor. The graph can
be used in place of tedious calculations. As shown in
Figure␣ 3, the zero output ripple current is obtained when:
16296f
12
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
VOUT k
=
VIN
N
where k = 1, 2, …, N – 1
So the number of phases used can be selected to minimize
the output ripple current and therefore the output ripple
voltage at the given input and output voltages. In applications having a highly varying input voltage, additional
phases will produce the best results.
Accepting larger values of ∆IL allows the use of low
inductances, but can result in higher output voltage ripple.
A reasonable starting point for setting ripple current is ∆IL
= 0.4(IOUT)/N, where N is the number of channels and IOUT
is the total load current. Remember, the maximum ∆IL
occurs at the maximum input voltage. The individual
inductor ripple currents are constant determined by the
inductor, input and output voltages.
1-PHASE
2-PHASE
3-PHASE
4-PHASE
6-PHASE
0.9
0.8
0.7
Two external power MOSFETs must be selected for each
controller with the LTC1629-6: One N-channel MOSFET
for the top (main) switch, and one N-channel MOSFET for
the bottom (synchronous) switch.
0.6
VO/fL
∆IO(P-P)
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Because they lack a bobbin, mounting is more difficult.
However, designs for surface mount are available which
do not increase the height significantly.
Power MOSFET, D1 and D2 Selection
1.0
0.5
0.4
0.3
0.2
0.1
0
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
0.1
0.2
0.3 0.4 0.5 0.6 0.7
DUTY FACTOR (VOUT/VIN)
0.8
0.9
1629 F03
Figure 3. Normalized Peak Output Current vs
Duty Factor [IRMS ≈ 0.3 (∆IO(P–P))]
Inductor Core Selection
Once the values for L1 and L2 are known, the type of
inductor must be selected. High efficiency converters
generally cannot afford the core loss found in low cost
powdered iron cores, forcing the use of more expensive
ferrite, molypermalloy, or Kool Mµ® cores. Actual core
loss is independent of core size for a fixed inductor value,
but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately,
increased inductance requires more turns of wire and
therefore copper losses will increase.
Kool Mµ is a registered trademark of Magnetics, Inc.
The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5V during start-up (see
EXTVCC Pin Connection). Consequently, logic-level threshold MOSFETs must be used in most applications. The only
exception is if low input voltage is expected (VIN < 5V);
then, sublogic-level threshold MOSFETs (VGS(TH) < 3V)
should be used. Pay close attention to the BVDSS specification for the MOSFETs as well; most of the logic-level
MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS,
input voltage, and maximum output current. When the
LTC1629-6 is operating in continuous mode the duty
factors for the top and bottom MOSFETs of each output
stage are given by:
Main Switch Duty Cycle =
VOUT
VIN
V –V 
Synchronous Switch Duty Cycle =  IN OUT 


VIN
16296f
13
LTC1629-6
U
U
W
U
APPLICATIO S I FOR ATIO

2 I
k( VIN )  MAX  (CRSS )( f)
 N 
2
I

V –V
PSYNC = IN OUT  MAX  1 + δ RDS(ON)
VIN
 N 
( )
where δ is the temperature dependency of RDS(ON), k is a
constant inversely related to the gate drive current and N
is the number of stages.
Both MOSFETs have I2R losses but the topside N-channel
equation includes an additional term for transition losses,
which peak at the highest input voltage. For VIN < 20V the
high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CRSS actual provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during a
short-circuit when the synchronous switch is on close to
100% of the period.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs. Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the MOSFET characteristics. The constant k = 1.7 can be used to
estimate the contributions of the two terms in the main
switch dissipation equation.
The Schottky diodes, D1 and D2 shown in Figure 1 conduct
during the dead-time between the conduction of the two
large power MOSFETs. This helps prevent the body diode
of the bottom MOSFET from turning on, storing charge
during the dead-time, and requiring a reverse recovery
period which would reduce efficiency. A 1A to 3A (depending on output current) Schottky diode is generally a good
compromise for both regions of operation due to the
relatively small average current. Larger diodes result in
CIN and COUT Selection
In continuous mode, the source current of each top
N-channel MOSFET is a square wave of duty cycle VOUT/
VIN. A low ESR input capacitor sized for the maximum
RMS current must be used. The details of a close form
equation can be found in Application Note 77. Figure 4
shows the input capacitor ripple current for different
phase configurations with the output voltage fixed and
input voltage varied. The input ripple current is normalized
against the DC output current. The graph can be used in
place of tedious calculations. The minimum input ripple
current can be achieved when the product of phase number and output voltage, N(VOUT), is approximately equal to
the input voltage VIN or:
VOUT k
=
VIN
N
where k = 1, 2, …, N – 1
So the phase number can be chosen to minimize the input
capacitor size for the given input and output voltages.
In the graph of Figure 4, the local maximum input RMS
capacitor currents are reached when:
VOUT 2k − 1
=
VIN
2N
where k = 1, 2, …, N
0.6
0.5
DC LOAD CURRENT
2
V
I

PMAIN = OUT  MAX  (1 + δ )RDS(ON) +
VIN  N 
additional transition losses due to their larger junction
capacitance.
RMS INPUT RIPPLE CURRNET
The MOSFET power dissipations at maximum output
current are given by:
1-PHASE
2-PHASE
3-PHASE
4-PHASE
6-PHASE
0.4
0.3
0.2
0.1
0
0.1
0.2
0.3 0.4 0.5 0.6 0.7
DUTY FACTOR (VOUT/VIN)
0.8
0.9
1629 F04
Figure 4. Normalized Input RMS Ripple Current vs
Duty Factor for 1 to 6 Output Stages
16296f
14
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
These worst-case conditions are commonly used for
design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to
meet size or height requirements in the design. Always
consult the capacitor manufacturer if there is any question.
The graph shows that the peak RMS input current is
reduced linearly, inversely proportional to the number, N
of stages used. It is important to note that the efficiency
loss is proportional to the input RMS current squared and
therefore a 2-stage implementation results in 75% less
power loss when compared to a single phase design.
Battery/input protection fuse resistance (if used), PC
board trace and connector resistance losses are also
reduced by the reduction of the input ripple current in a
PolyPhase system. The required amount of input capacitance is further reduced by the factor, N, due to the
effective increase in the frequency of the current pulses.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically once the ESR requirement has been met, the RMS current rating generally far
exceeds the IRIPPLE(P-P) requirements. The steady state
output ripple (∆VOUT) is determined by:

1 
∆VOUT ≈ ∆IRIPPLE  ESR +


8NfCOUT 
Where f = operating frequency of each stage, N is the
number of phases, COUT = output capacitance, and
∆IRIPPLE = combined inductor ripple currents.
The output ripple varies with input voltage since ∆IL is a
function of input voltage. The output ripple will be less than
50mV at max VIN with ∆IL = 0.4IOUT(MAX)/N assuming:
COUT required ESR < 2N(RSENSE) and
COUT > 1/(8Nf)(RSENSE)
The emergence of very low ESR capacitors in small,
surface mount packages makes very physically small
implementations possible. The ability to externally compensate the switching regulator loop using the ITH pin(OPTI-
LOOP compensation) allows a much wider selection of
output capacitor types. OPTI-LOOP compensation effectively removes constraints on output capacitor ESR. The
impedance characteristics of each capacitor type are significantly different than an ideal capacitor and therefore
require accurate modeling or bench evaluation during
design.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo and the Panasonic SP
surface mount types have the lowest (ESR)(size) product
of any aluminum electrolytic at a somewhat higher price.
An additional ceramic capacitor in parallel with OS-CON
type capacitors is recommended to reduce the inductance
effects.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
surface mount configurations. New special polymer surface mount capacitors offer very low ESR also but have
much lower capacitive density per unit volume. In the case
of tantalum, it is critical that the capacitors are surge tested
for use in switching power supplies. Several excellent
choices are the AVX TPS, AVX TPSV or the KEMET T510
series of surface mount tantalums, available in case heights
ranging from 2mm to 4mm. Other capacitor types include
Sanyo OS-CON, Nichicon PL series and Sprague 595D
series. Consult the manufacturer for other specific recommendations. A combination of capacitors will often result
in maximizing performance and minimizing overall cost
and size.
INTVCC Regulator
An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. The INTVCC
regulator powers the drivers and internal circuitry of the
LTC1629-6. The INTVCC pin regulator can supply up to
50mA peak and must be bypassed to power ground with
a minimum of 4.7µF tantalum or electrolytic capacitor. An
additional 1µF ceramic capacitor placed very close to the
IC is recommended due to the extremely high instantaneous currents required by the MOSFET gate drivers.16296f
15
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC1629-6 to be
exceeded. The supply current is dominated by the gate
charge supply current, in addition to the current drawn
from the differential amplifier output. The gate charge is
dependent on operating frequency as discussed in the
Efficiency Considerations section. The supply current can
either be supplied by the internal 5V regulator or via the
EXTVCC pin. When the voltage applied to the EXTVCC pin
is less than 4.7V, all of the INTVCC load current is supplied
by the internal 5V linear regulator. Power dissipation for
the IC is higher in this case by (IIN)(VIN – INTVCC) and
efficiency is lowered. The junction temperature can be
estimated by using the equations given in Note 1 of the
Electrical Characteristics. For example, the LTC1629-6 VIN
current is limited to less than 24mA from a 24V supply:
TJ = 70°C + (24mA)(24V)(95°C/W) = 125°C
Use of the EXTVCC pin reduces the junction temperature
to:
TJ = 70°C + (24mA)(5V)(95°C/W) = 81.4°C
The input supply current should be measured while the
controller is operating in continuous mode at maximum
VIN and the power dissipation calculated in order to prevent the maximum junction temperature from being exceeded.
EXTVCC Connection
The LTC1629-6 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
When the voltage applied to EXTVCC rises above 4.7V, the
internal regulator is turned off and the switch closes,
connecting the EXTVCC pin to the INTVCC pin thereby
supplying internal and MOSFET gate driving power. The
switch remains closed as long as the voltage applied to
EXTVCC remains above 4.5V. This allows the MOSFET
driver and control power to be derived from the output
during normal operation (4.7V < VEXTVCC < 7V) and from
the internal regulator when the output is out of regulation
(start-up, short-circuit). Do not apply greater than 7V to
the EXTVCC pin and ensure that EXTVCC < VIN + 0.3V when
using the application circuits shown. If an external voltage
source is applied to the EXTVCC pin when the VIN supply is
not present, a diode can be placed in series with the
LTC1629-6’s VIN pin and a Schottky diode between the
EXTVCC and the VIN pin, to prevent current from backfeeding
VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by the
ratio: (Duty Factor)/(Efficiency). For 5V regulators this
means connecting the EXTVCC pin directly to VOUT. However, for 3.3V and other lower voltage regulators, additional circuitry is required to derive INTVCC power from the
output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting in
a significant efficiency penalty at high input voltages.
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
3. EXTVCC connected to an external supply. If an external
supply is available in the 5V to 7V range, it may be used to
power EXTVCC providing it is compatible with the MOSFET
gate drive requirements. VIN must be greater than or equal
to the voltage applied to the EXTVCC pin.
4. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency gains
can still be realized by connecting EXTVCC to an outputderived voltage which has been boosted to greater than
4.7V but less than 7V. This can be done with either the
inductive boost winding as shown in Figure 5a or the
capacitive charge pump shown in Figure 5b. The charge
pump has the advantage of simple magnetics.
16296f
16
LTC1629-6
U
U
W
U
APPLICATIO S I FOR ATIO
OPTIONAL EXTVCC CONNECTION
5V < VSEC < 7V
+
CIN
+
VIN
LTC1629-6
LTC1629-6
VIN
1N4148
VSEC
TG1
EXTVCC
N-CH
VIN
6.8V
BAT85
N-CH
1µF
BAT85
RSENSE
VOUT
SW1
COUT
BG1
VOUT
L1
+
+
BG1
BAT85
VN2222LL
EXTVCC
T1
0.22µF
TG1
+
RSENSE
SW1
+
VIN
CIN
COUT
N-CH
N-CH
PGND
PGND
1629 F05b
1629 F05a
Figure 5a. Secondary Output Loop and EXTVCC Connection
Topside MOSFET Driver Supply (CB,DB) (Refer to
Functional Diagram)
External bootstrap capacitors CB1 and CB2 connected to
the BOOST1 and BOOST2 pins supply the gate drive
voltages for the topside MOSFETs. Capacitor CB in the
Functional Diagram is charged though diode DB from
INTVCC when the SW pin is low. When the topside MOSFET
turns on, the driver places the CB voltage across the gatesource of the desired MOSFET. This enhances the MOSFET
and turns on the topside switch. The switch node voltage,
SW, rises to VIN and the BOOST pin rises to VIN + VINTVCC.
The value of the boost capacitor CB needs to be 30 to 100
times that of the total input capacitance of the topside
MOSFET(s). The reverse breakdown of DB must be greater
than VIN(MAX).
The final arbiter when defining the best gate drive amplitude level will be the input supply current. If a change is
made that decreases input current, the efficiency has
improved. If the input current does not change then the
efficiency has not changed either.
Output Voltage
The LTC1629-6 has a true remote voltage sense capablity.
The sensing connections should be returned from the load
back to the differential amplifier’s inputs through a common, tightly coupled pair of PC traces. The differential
amplifier rejects common mode signals capacitively or
inductively radiated into the feedback PC traces as well
Figure 5b. Capacitive Charge Pump for EXTVCC
as ground loop disturbances. The differential amplifier
output signal is divided down and compared with the
internal precision 0.6V voltage reference by the error
amplifier.
The output is an NPN emitter follower without any internal
pull-down current. A DC resistive load to ground is required in order to sink current. The output will swing from
0V to 10V. (VIN ≥ VDIFFOUT␣ +␣ 2V.)
Soft-Start/Run Function
The RUN/SS pin provides three functions: 1) Run/Shutdown, 2) soft-start and 3) a defeatable short-circuit latchoff
timer. Soft-start reduces the input power sources’ surge
currents by gradually increasing the controller’s current
limit ITH(MAX). The latchoff timer prevents very short,
extreme load transients from tripping the overcurrent
latch. A small pull-up current (>5µA) supplied to the RUN/
SS pin will prevent the overcurrent latch from operating.
The following explanation describes how the functions
operate.
An internal 1.2µA current source charges up the CSS
capacitor. When the voltage on RUN/SS reaches 1.5V, the
controller is permitted to start operating. As the voltage on
RUN/SS increases from 1.5V to 3.0V, the internal current
limit is increased from 25mV/RSENSE to 75mV/RSENSE.
The output current limit ramps up slowly, taking an
additional 1.4µs/µF to reach full current. The output
current thus ramps up slowly, reducing the starting surge
16296f
17
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
current required from the input power supply. If RUN/SS
has been pulled all the way to ground there is a delay before
starting of approximately:
tDELAY =
(
)
1.5V
CSS = 1.25s / µF CSS
1.2µA
The time for the output current to ramp up is then:
tRAMP =
3V − 1.5V
CSS = (1.25s / µF ) CSS
1.2µA
By pulling the RUN/SS pin below 0.8V the LTC1629-6 is
put into low current shutdown (IQ < 40µA). RUN/SS can be
driven directly from logic as shown in Figure 6. Diode D1
in Figure 6 reduces the start delay but allows CSS to ramp
up slowly providing the soft-start function. The RUN/SS
pin has an internal 6V zener clamp (see Functional Diagram).
Fault Conditions: Overcurrent Latchoff
The RUN/SS pin also provides the ability to latch off the
controllers when an overcurrent condition is detected. The
RUN/SS capacitor, CSS, is used initially to limit the inrush
current of both controllers. After the controllers have been
started and been given adequate time to charge up the
output capacitors and provide full load current, the RUN/
SS capacitor is used for a short-circuit timer. If the output
voltage falls to less than 70% of its nominal value after CSS
reaches 4.1V, CSS begins discharging on the assumption
that the output is in an overcurrent condition. If the
condition lasts for a long enough period as determined by
the size of CSS, the controller will be shut down until the
RUN/SS pin voltage is recycled. If the overload occurs
during start-up, the time can be approximated by:
tLO1 ≈ (CSS • 0.6V)/(1.2µA) = 5 • 105 (CSS)
If the overload occurs after start-up, the voltage on CSS will
continue charging and will provide additional time before
latching off:
tLO2 ≈ (CSS • 3V)/(1.2µA) = 2.5 • 106 (CSS)
This built-in overcurrent latchoff can be overridden by
providing a pull-up resistor, RSS, to the RUN/SS pin as
shown in Figure 6. This resistance shortens the soft-start
period and prevents the discharge of the RUN/SS capacitor during a severe overcurrent and/or short-circuit condition. When deriving the 5µA current from VIN as in the
figure, current latchoff is always defeated. Diodeconnecting this pull-up resistor to INTV CC, as in
Figure␣ 6, eliminates any extra supply current during shutdown while eliminating the INTVCC loading from preventing controller start-up.
Why should you defeat current latchoff? During the
prototyping stage of a design, there may be a problem with
noise pickup or poor layout causing the protection circuit
to latch off the controller. Defeating this feature allows
troubleshooting of the circuit and PC layout. The internal
short-circuit and foldback current limiting still remains
active, thereby protecting the power supply system from
failure. A decision can be made after the design is complete whether to rely solely on foldback current limiting or
to enable the latchoff feature by removing the pull-up
resistor.
The value of the soft-start capacitor CSS may need to be
scaled with output voltage, output capacitance and load
current characteristics. The minimum soft-start capacitance is given by:
CSS > (COUT )(VOUT)(10-4)(RSENSE)
The minimum recommended soft-start capacitor of CSS =
0.1µF will be sufficient for most applications.
INTVCC
VIN
3.3V OR 5V
D1
RUN/SS
RSS*
RSS*
D1*
RUN/SS
CSS
CSS
*OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF
1629 F06
Figure 6. RUN/SS Pin Interfacing
Phase-Locked Loop and Frequency Synchronization
The LTC1629-6 has a phase-locked loop comprised of an
internal voltage controlled oscillator and phase detector.
This allows the top MOSFET turn-on to be locked to the
rising edge of an external source. The frequency range of
the voltage controlled oscillator is ±50% around the
16296f
18
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
center frequency fO. A voltage applied to the PLLFLTR pin
of 1.2V corresponds to a frequency of approximately
220kHz. The nominal operating frequency range of the
LTC1629-6 is 140kHz to 310kHz.
The phase detector used is an edge sensitive digital type
which provides zero degrees phase shift between the
external and internal oscillators. This type of phase detector will not lock up on input frequencies close to the
harmonics of the VCO center frequency. The PLL hold-in
range, ∆fH, is equal to the capture range, ∆fC:
2.4V
PHASE
DETECTOR
RLP
10k
CLP
EXTERNAL
OSC
PLLFLTR
PLLIN
50k
DIGITAL
PHASE/
FREQUENCY
DETECTOR
∆fH = ∆fC = ±0.5 fO (150kHz-300kHz)
The output of the phase detector is a complementary pair
of current sources charging or discharging the external
filter network on the PLLFLTR pin. A simplified block
diagram is shown in Figure 7.
If the external frequency (fPLLIN) is greater than the oscillator frequency f0SC, current is sourced continuously,
pulling up the PLLFLTR pin. When the external frequency
is less than f0SC, current is sunk continuously, pulling
down the PLLFLTR pin. If the external and internal frequencies are the same but exhibit a phase difference, the
current sources turn on for an amount of time corresponding to the phase difference. Thus the voltage on the
PLLFLTR pin is adjusted until the phase and frequency of
the external and internal oscillators are identical. At this
stable operating point the phase comparator output is
open and the filter capacitor CLP holds the voltage. The
LTC1629-6 PLLIN pin must be driven from a low impedance source such as a logic gate located close to the pin.
When using multiple LTC1629-6’s for a phase-locked
system, the PLLFLTR pin of the master oscillator should
be biased at a voltage that will guarantee the slave
oscillator(s) ability to lock onto the master’s frequency. A
DC voltage of 1.6V to 1.7V applied to the master oscillator’s
PLLFLTR pin is recommended in order to meet this
requirement. The resultant operating frequency will be
approximately 300kHz.
The loop filter components (CLP, RLP) smooth out the
current pulses from the phase detector and provide a
stable input to the voltage controlled oscillator. The filter
components CLP and RLP determine how fast the loop
acquires lock. Typically RLP =10kΩ and CLP is 0.01µF to
0.1µF.
OSC
1629 F07
Figure 7. Phase-Locked Loop Block Diagram
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest time duration
that the LTC1629-6 is capable of turning on the top
MOSFET. It is determined by internal timing delays and the
gate charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that:
tON(MIN) <
VOUT
()
VIN f
If the duty cycle falls below what can be accommodated by
the minimum on-time, the LTC1629-6 will begin to skip
cycles resulting in nonconstant frequency operation. The
output voltage will continue to be regulated, but the ripple
current and ripple voltage will increase.
The minimum on-time for the LTC1629-6 is generally less
than 200ns. However, as the peak sense voltage decreases
the minimum on-time gradually increases. This is of
particular concern in forced continuous applications with
low ripple current at light loads. If the duty cycle drops
below the minimum on-time limit in this situation, a
significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple.
If an application can operate close to the minimum ontime limit, an inductor must be chosen that has a low
enough inductance to provide sufficient ripple amplitude
to meet the minimum on-time requirement. As a general
rule, keep the inductor ripple current of each phase equal
to or greater than 15% of IOUT(MAX)/N at VIN(MAX). 16296f
19
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1629-6 circuits: 1) LTC1629-6 VIN current
(including loading on the differential amplifier output),
2) INTVCC regulator current, 3) I2R losses and 4) Topside
MOSFET transition losses.
1) The VIN current has two components: the first is the
DC supply current given in the Electrical Characteristics
table, which excludes MOSFET driver and control currents; the second is the current drawn from the differential
amplifier output. VIN current typically results in a small
(<0.1%) loss.
2) INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from INTVCC to
ground. The resulting dQ/dt is a current out of INTVCC that
is typically much larger than the control circuit current. In
continuous mode, IGATECHG = (QT + QB), where QT and QB
are the gate charges of the topside and bottom side
MOSFETs.
Supplying INTVCC power through the EXTVCC switch input
from an output-derived source will scale the VIN current
required for the driver and control circuits by the ratio
(Duty Factor)/(Efficiency). For example, in a 20V to 5V
application, 10mA of INTVCC current results in approximately 3mA of VIN current. This reduces the mid-current
loss from 10% or more (if the driver was powered directly
from VIN) to only a few percent.
3) I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor,
and input and output capacitor ESR. In continuous mode
the average output current flows through L and RSENSE,
but is “chopped” between the topside MOSFET and the
synchronous MOSFET. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one
MOSFET can simply be summed with the resistances of L,
RSENSE and ESR to obtain I2R losses. For example, if each
RDS(ON)=10mΩ, RL=10mΩ, and RSENSE=5mΩ, then the
total resistance is 25mΩ. This results in losses ranging
from 2% to 8% as the output current increases from 3A to
15A per output stage for a 5V output, or a 3% to 12% loss
per output stage for a 3.3V output. Efficiency varies as the
inverse square of VOUT for the same external components
and output power level. The combined effects of increasingly lower output voltages and higher currents required
by high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!
4) Transition losses apply only to the topside MOSFET(s),
and only when operating at high input voltages (typically
20V or greater). Transition losses can be estimated from:
Transition Loss = (1.7) VIN2 IO(MAX) CRSS f
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these “system” level losses in the
design of a system. The internal battery and input fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and a very low ESR at the
switching frequency. A 50W supply will typically require a
minimum of 200µF to 300µF of capacitance having a
maximum of 10mΩ to 20mΩ of ESR. The LTC1629-6
PolyPhase architecture typically halves to quarters this
input capacitance requirement over competing solutions.
Other losses including Schottky conduction losses during
dead-time and inductor core losses generally account for
less than 2% total additional loss.
16296f
20
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD(ESR), where ESR is the effective
series resistance of COUT(∆ILOAD) also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. The
availability of the ITH pin not only allows optimization of
control loop behavior but also provides a DC coupled and
AC filtered closed loop response test point. The DC step,
rise time, and settling at this test point truly reflects the
closed loop response. Assuming a predominantly second
order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining
the rise time at the pin. The ITH external components
shown in the Figure 1 circuit will provide an adequate
starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.2 to 5 times their suggested values) to maximize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be decided
upon because the various types and values determine the
loop feedback factor gain and phase. An output current
pulse of 20% to 80% of full-load current having a rise time
of <2µs will produce output voltage and ITH pin waveforms
that will give a sense of the overall loop stability without
breaking the feedback loop. The initial output voltage step
resulting from the step change in output current may not
be within the bandwidth of the feedback loop, so this signal
cannot be used to determine phase margin. This is why it
is better to look at the Ith pin signal which is in the feedback
loop and is the filtered and compensated control loop
response. The gain of the loop will be increased by
increasing RC and the bandwidth of the loop will be
increased by decreasing CC. If RC is increased by the same
factor that CC is decreased, the zero frequency will be kept
the same, thereby keeping the phase shift the same in the
most critical frequency range of the feedback loop. The
output voltage settling behavior is related to the stability of
the closed-loop system and will demonstrate the actual
overall supply performance.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than1:50, the switch rise time
should be controlled so that the load rise time is limited to
approximately 25 • CLOAD. Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current to
about 200mA.
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation.
But before you connect, be advised: you are plugging into
the supply from hell. The main battery line in an automobile is the source of a number of nasty potential transients,
including load-dump, reverse-battery, and double-battery.
Load-dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
just what it says, while double-battery is a consequence of
tow truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 8 is the most straightforward
approach to protect a DC/DC converter from the ravages
of an automotive battery line. The series diode prevents
current from flowing during reverse-battery, while the
transient suppressor clamps the input voltage during
load-dump. Note that the transient suppressor should not
conduct during double-battery operation, but must still
clamp the input voltage below breakdown of the converter.
16296f
21
LTC1629-6
U
U
W
U
APPLICATIO S I FOR ATIO
Although the LTC1629-6 has a maximum input voltage of
36V, most applications will be limited to 30V by the
MOSFET BVDSS.
50A IPK RATING
Since the output voltage is below 2.4V the output resistive
divider will need to be sized to not only set the output
voltage but also to absorb the sense pin input current for
both channels.
VIN
12V
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
R1(MIN) =
LTC1629-6
1.8 V 

= 10k 
 = 30k
 2.4V − 1.8 V 
1629 F08
Figure 8. Automotive Application Protection
Design Example (Using Two Phases)
As a design example, assume VIN = 5V (nominal), VIN␣ =␣ 5.5V
(max), VOUT = 1.8V, IMAX = 20A, TA = 70°C and f␣ =␣ 300kHz.
The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQSET pin
to the INTVCC pin for 300kHz operation. The minimum
inductance for 30% ripple current is:
L≥
≥
Choosing 1% resistors: R1 = 10k and R2 = 20k yields an
output voltage of 1.80V and satisfies the above condition.
The power dissipation on the topside MOSFET can be
easily estimated. Using a Siliconix Si4420DY for example;
RDS(ON) = 0.013Ω, CRSS = 300pF. At maximum input
voltage with Tj (estimated) = 110°C at an elevated ambient
temperature:
PMAIN =
VOUT  VOUT 
 1−

f( ∆I) 
VIN 
1.8 V
 1.8 V 
 1−

(300kHz)(30%)(10A)  5.5V 
≥ 1.35µH
A 2µH inductor will produce 20% ripple current. The peak
inductor current will be the maximum DC value plus one
half the ripple current, or 11.5A. The minimum on-time
occurs at maximum VIN:
tON(MIN) =
VOUT
1.8 V
=
= 1.1µs
VINf (5.5V )(300kHz)
The RSENSE resistors value can be calculated by using the
maximum current sense voltage specification with some
accomodation for tolerances:
RSENSE =

20k 
VOUT


2  2.4V − VOUT 
( ) [1+ (0.005)(110°C − 25°C)]
2
0.013Ω + 1.7(5.5V ) (10 A )(300pF )
(310kHz)= 0.61W
1.8 V
10
5.5V
2
The worst-case power disipated by the synchronous
MOSFET under normal operating conditions at elevated
ambient temperature and estimated 50°C junction temperature rise is:
( ) (1.48)(0.013Ω)
5.5V − 1.8 V
10 A
5.5V
= 1.29W
PSYNC =
2
A short-circuit to ground will result in a folded back current
of:
ISC =
( )  = 5.28A

25mV
1 200ns 5.5V
+ 
0.005Ω 2 
2µH



60mV
≈ 0.005Ω
11.5A
16296f
22
LTC1629-6
U
U
W
U
APPLICATIO S I FOR ATIO
The worst-case power disipated by the synchronous
MOSFET under short-circuit conditions at elevated ambient temperature and estimated 50°C junction temperature
rise is:
(
5.5V − 1.8 V
5.28 A
5.5V
= 360mW
PSYNC =
) (1.48)(0.013Ω)
2
which is much less than normal, full-load conditions.
Incidentally, since the load no longer dissipates power in
the shorted condition, total system power dissipation is
decreased by over 99%.
The duty cycles when the peak RMS input current occurs
is at D = 0.25 and D = 0.75 according to Figure 4. Calculate
the worst-case required RMS input current rating at the
input voltage, which is 5.5V, that provides a duty cycle
nearest to the peak.
From Figure 4, CIN will require an RMS current rating of:
CIN requiredIRMS = (20 A )(0.23)
= 4.6 ARMS
The output capacitor ripple current is calculated by using
the inductor ripple already calculated for each inductor
and multiplying by the factor obtained from Figure␣ 3 along
with the calculated duty factor. The output ripple in continuous mode will be highest at the maximum input
voltage. From Figure 3, the maximum output current ripple
is:
VOUT
(0.34)
fL
1.8(0.34)
∆ICOUTMAX =
= 1A
(300kHz)(2µH)
∆ICOUT =
Note that the PolyPhase technique will have its maximum
benefit for input and output ripple currents when the
number of phases times the output voltage is approximately equal to or greater than the input voltage.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1629-6. These items are also illustrated graphically in
the layout diagram of Figure␣ 11. Check the following in
your layout:
1) Are the signal and power grounds segregated? The
LTC1629-6 signal ground pin should return to the (–) plate
of COUT separately. The power ground returns to the
sources of the bottom N-channel MOSFETs, anodes of the
Schottky diodes, and (–) plates of CIN, which should have
as short lead lengths as possible.
2) Does the LTC1629-6 VOS+ pin connect to the (+) plate(s)
of COUT? Does the LTC1629-6 VOS– pin connect to the
(–) plate(s) of COUT? The resistive divider R1, R2 must be
connected between the VDIFFOUT and signal ground and
any feedforward capacitor across R1 should be as close as
possible to the LTC1629-6.
3) Are the SENSE – and SENSE + leads routed together with
minimum PC trace spacing? The filter capacitors between
SENSE + and SENSE – pin pairs should be as close as
possible to the LTC1629-6. Ensure accurate current sensing with Kelvin connections to the sense resistors.
4) Do the (+) plates of CIN connect to the drains of the
topside MOSFETs as closely as possible? This capacitor
provides the AC current to the MOSFETs. Keep the input
current path formed by the input capacitor, top and bottom
MOSFETs, and the Schottky diode on the same side of the
PC board in a tight loop to minimize conducted and
radiated EMI.
5) Is the INTVCC 1µF ceramic decoupling capacitor connected closely between INTVCC and the power ground pin?
This capacitor carries the MOSFET driver peak currents. A
small value is used to allow placement immediately adjacent to the IC.
6) Keep the switching nodes, SW1 (SW2), away from
sensitive small-signal nodes. Ideally the switch
nodes should be placed at the furthest point from the
LTC1629-6.
7) Use a low impedance source such as a logic gate to drive
the PLLIN pin and keep the lead as short as possible.
16296f
23
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
8) Minimize the capacitive load on the CLKOUT pin to
minimize excess phase shift. Buffer if necessary with an
NPN emitter follower.
The diagram in Figure 9 illustrates all branch currents in a
2-phase switching regulator. It becomes very clear after
studying the current waveforms why it is critical to keep
the high-switching-current paths to a small physical size.
High electric and magnetic fields will radiate from these
“loops” just as radio stations transmit signals. The output
capacitor ground should return to the negative terminal of
the input capacitor and not share a common ground path
with any switched current paths. The left half of the circuit
gives rise to the “noise” generated by a switching regula-
SW1
tor. The ground terminations of the sychronous MOSFETs
and Schottky diodes should return to the bottom plate(s)
of the input capacitor(s) with a short isolated PC trace
since very high switched currents are present. A separate
isolated path from the bottom plate(s) of the input
capacitor(s) should be used to tie in the IC power ground
pin (PGND) and the signal ground pin (SGND). This
technique keeps inherent signals generated by high current pulses from taking alternate current paths that have
finite impedances during the total period of the switching
regulator. External OPTI-LOOP compensation allows overcompensation for PC layouts which are not optimized but
this is not the recommended design procedure.
L1
RSENSE1
D1
VIN
VOUT
RIN
CIN
+
+
SW2
BOLD LINES INDICATE
HIGH, SWITCHING
CURRENT LINES.
KEEP LINES TO A
MINIMUM LENGTH.
L2
COUT
RL
RSENSE2
D2
1629 F09
Figure 9. Instantaneous Current Path Flow in a Multiple Phase Switching Regulator
16296f
24
LTC1629-6
U
W
U
U
APPLICATIO S I FOR ATIO
Simplified Visual Explanation of How a 2-Phase
Controller Reduces Both Input and Output RMS Ripple
Current
A multiphase power supply significantly reduces the
amount of ripple current in both the input and output
capacitors. The RMS input ripple current is divided by, and
the effective ripple frequency is multiplied up by the
number of phases used (assuming that the input voltage
is greater than the number of phases used times the output
voltage). The output ripple amplitude is also reduced by,
and the effective ripple frequency is increased by the
number of phases used. Figure 10 graphically illustrates
the principle.
The worst-case RMS ripple current for a single stage
design peaks at twice the value of the output voltage . The
worst-case RMS ripple current for a two stage design
results in peaks at 1/4 and 3/4 of input voltage. When the
RMS current is calculated, higher effective duty factor
results and the peak current levels are divided as long as
the currents in each stage are balanced. Refer to Application Note 19 for a detailed description of how to calculate
RMS current for the single stage switching regulator.
Figures 3 and 4 help to illustrate how the input and output
currents are reduced by using an additional phase. The
input current peaks drop in half and the frequency is
doubled for a 2-phase converter. The input capacity requirement is reduced theoretically by a factor of four! A
ceramic input capacitor with its unbeatably low ESR
characteristic can be used.
Figure 4 illustrates the RMS input current drawn from the
input capacitance versus the duty cycle as determined by
the ratio of input and output voltage. The peak input RMS
current level of the single phase system is reduced by 50%
in a 2-phase solution due to the current splitting between
the two stages.
An interesting result of the multi-phase solution is that the
VIN which produces worst-case ripple current for the input
capacitor, VOUT = VIN/2, in the single phase design produces zero input current ripple in the 2-phase design.
The output ripple current is reduced significantly when
compared to the single phase solution using the same
inductance value because the VOUT/L discharge current
term from the stage(s) that has its bottom MOSFET on
subtracts current from the (VIN - VOUT)/L charging current
resulting from the stage which has its top MOSFET on. The
output ripple current is:
IRIPPLE =
( ) 

2VOUT  1 − 2D 1 − D
fL  1 − 2D + 1



where D is duty factor.
The input and output ripple frequency is increased by the
number of stages used, reducing the output capacity
requirements. When VIN is approximately equal to NVOUT
as illustrated in Figures 3 and 4, very low input and output
ripple currents result.
Again, the interesting result of 2-phase operation results
in no output ripple at VOUT = VIN/2. The addition of more
phases by phase locking additional controllers always
results in no net input or output ripple at VOUT/VIN ratios
equal to the number of stages implemented. Designing a
system with a multiple of stages close to the VOUT/VIN ratio
will significantly reduce the ripple voltage at the input and
outputs and thereby improve efficiency, physical size, and
heat generation of the overall switching power supply.
SINGLE PHASE
SW V
ICIN
ICOUT
DUAL PHASE
SW1 V
SW2 V
IL1
IL2
ICIN
ICOUT
RIPPLE
1629 F10
Figure 10. Single and PolyPhase Current Waveforms
16296f
25
LTC1629-6
U
TYPICAL APPLICATIO S
OPTIONAL
SYNC
CLOCK IN
L1
2
0.33µF
3
6.04k, 1%
4
5
100pF
0.33µF
6800pF
7
8
9
27.4k, 1%
10
11
12
100pF
13
14
SENSE1 +
TG1
SENSE1 –
SW1
EAIN
BOOST1
0.003Ω
28
27
0.47µF
26
25
M1
D7
24
VIN
PLLFLTR
23
PLLIN LTC1629-6 BG1
22
5V
PHASMD
EXTVCC
21
ITH
INTVCC
20
SGND
PGND
19
VDIFFOUT
BG2
18
–
VOS
BOOST2
17
+
SW2
VOS
16
TG2
SENSE2 –
15
+
PGOOD
SENSE2
M2
M3
D1
B320A
10Ω
1µF
1µF,25V
2X150µF
22µF
6.3V
16V
GND
D2
B320A
0.47µF
M4
M5
M6
L2
PGOOD
0.003Ω
75k
L3
2
3
4
47pF
10k
5
6
1nF
8
100pF 9
10
NC
11
12
13
14
CLKOUT
SENSE1 +
TG1
SENSE1 –
SW1
EAIN
BOOST1
0.003Ω
27
0.47µF
26
25
M7
D9
24
VIN
PLLFLTR
23
PLLIN LTC1629-6 BG1
22
5V
PHASMD
EXTVCC
21
ITH
INTVCC
20
SGND
PGND
19
VDIFFOUT
BG2
18
–
BOOST2
VOS
17
SW2
VOS +
16
–
TG2
SENSE2
15
+
PGOOD
SENSE2
M8
M9
D3
B320A
10Ω
1µF
3X470µF, 6.3V
+
7
RUN/SS
28
1µF,25V
2X150µF
22µF
6.3V
+
1
+
1000pF
0.01µF
3X470µF, 6.3V
KEMET CAP
VOUT1
3.3V/90A
D8
10k
1000pF
24k
CLKOUT
+
47k
6
RUN/SS
+
1
+
1000pF
16V
KEMET CAP
GND
D10
VIN
12V
D4
B320A
0.47µF
M10
M11
1000pF
M12
L4
0.003Ω
L5
2
3
4
47pF
10k
1nF
CLKOUT
SENSE1 +
TG1
SENSE1 –
SW1
EAIN
BOOST1
VIN
PLLFLTR
6
PLLIN LTC1629-6 BG1
7
PHASMD
EXTVCC
8
ITH
INTVCC
9
100pF
SGND
PGND
10
NC
V
BG2
11 DIFFOUT
BOOST2
V –
12 OS +
SW2
V
13 OS
TG2
SENSE2 –
14
PGOOD
SENSE2 +
28
27
0.47µF
26
25
M13
D11
24
23
22
21
5V
M14
10Ω
D5
B320A
1µF
3X470µF, 6.3V
1µF,25V
20
2X150µF
22µF
6.3V
16V
KEMET CAP
GND
19
18
17
D12
D6
B320A
16
15
0.47µF
M16
1000pF
VIN: 12V
VOUT: 3.3V/90A
SWITCHING FREQUENCY = 220kHz
M25
+
0.01µF
5
RUN/SS
+
1
+
1000pF
M17
M18
L6
MI – M18: IRF7811W
L1 – L6: 1µH PANASONIC ETQP6F1R0S
D7 – D12: CENTROL CMDSH-3TR
OUTPUT CAPACITORS: KEMET T510X477M006AS
1629 TA03
Figure 11. High Current 3.3V/90A 6-Phase Application
16296f
26
LTC1629-6
U
PACKAGE DESCRIPTIO
G Package
28-Lead Plastic SSOP (5.3mm)
(Reference LTC DWG # 05-08-1640)
10.07 – 10.33*
(.397 – .407)
28 27 26 25 24 23 22 21 20 19 18 17 16 15
7.65 – 7.90
(.301 – .311)
1
2
3
4
5
6
7
8
9 10 11 12 13 14
5.20 – 5.38**
(.205 – .212)
1.73 – 1.99
(.068 – .078)
0° – 8°
.13 – .22
(.005 – .009)
.55 – .95
(.022 – .037)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
.65
(.0256)
BSC
.25 – .38
(.010 – .015)
.05 – .21
(.002 – .008)
G28 SSOP 0501
3. DRAWING NOT TO SCALE
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
16296f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC1629-6
U
TYPICAL APPLICATIO
L1
0.1µF
1
2
17.8k
3
10k
4
5
6.04k
1%
100pF
1000pF
6
1000pF
3.3k
8
9
10
11
27.4k
1%
12
13
14
100pF
SENSE1 +
TG1
–
SW1
SENSE1
EAIN
BOOST1
PLLFLTR
VIN
BG1
PLLIN
PHASMD
EXTVCC
ITH
INTVCC
SGND
LTC1629-6
VDIFFOUT
PGND
BG2
VOS –
BOOST2
+
SW2
VOS
SENSE2 –
TG2
SENSE2 +
PGOOD
27
0.47µF
26
M1
M2
D1
UPS840
25
D3
24
10Ω
23
22
21
0.1µF
5V
CIN
1µF,25V
20
4.7µF
6.3V
COUT
19
18
D4
17
16
100k
0.47µF
15
POWER
GOOD
M3
M4
L2
MI, M3: FDS7764
M2, M4: FDS7764 ×2
L1, L2: 1µH SUMIDA CEP125-1R0MC
VIN
5V TO
16V
D2
UPS840
1000pF
VIN: 5V TO 16V
VOUT: 3.3V/30A
SWITCHING FREQUENCY = 250kHz
0.003Ω
+
7
CLKOUT
RUN/SS
28
+
1000pF
+
INTVCC
0.003Ω
CIN: OS CON 2-16SP270M
COUT: POS CAP THREE 4TPB470M
D3, D4: CENTRAL CMDSH-3TR
VOUT
3.3V/30A
1629 TA02
Figure 12. 3.3V/30A Power Supply with Active Voltage Positioning
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1435/LTC1435A
High Efficiency Synchronous Step-Down Switching Regulator
Burst Mode® Operation, 16-Pin Narrow SO
LTC1436A-PLL
High Efficiency Low Noise Synchronous Step-Down Switching Regulator
Adaptive PowerTM Mode, 24-Pin SSOP
LTC1438/LTC1439
Dual High Efficiency Low Noise Synchronous Step-Down Switching Regulators
POR, Auxiliary Regulator
LTC1438-ADJ
Dual Synchronous Controller with Auxiliary Regulator
POR, External Feedback Divider
LTC1538-AUX
Dual High Efficiency Low Noise Synchronous Step-Down Switching Regulator
Auxiliary Regulator, 5V Standby
LTC1539
Dual High Efficiency Low Noise Synchronous Step-Down Switching Regulator
5V Standby, POR, Low-Battery,
Aux Regulator
LTC1530
High Power Step-Down Switching Regulator Controller
High Efficiency 5V to 3.3V Conversion
at Up to 15A
LTC1628/LTC1628-PG Dual High Efficiency, PolyPhase Synchronous Step-Down Switching Regulator
Constant Frequency, Standby,
5V and 3.3V LDOs, PGOOD
LTC1702/LTC1703
Dual High Efficiency, PolyPhase Synchronous Step-Down Switching Regulator
500kHz, 25MHz GBW, Voltage Mode
LTC1708-PG
Dual High Efficiency, PolyPhase Synchronous Step-Down Switching Regulator
Constant Frequency, 5-Bit VID,
Standby 5V, 3.3V, PGOOD
LTC1709 Family
High Efficiency, 2-Phase Synchronous Step-Down Switching Regulator
with 5-Bit VID (Various Code Tables Available)
Current Mode Ensures
Accurate Current Sharing, 3.5V ≤ VIN ≤ 36V
LTC1735
High Efficiency Synchronous Step-Down Controller
Burst Mode Operation, 16-Pin Narrow SSOP,
Fault Protection, 3.5V ≤ VIN ≤ 36V
LTC1736
High Efficiency Synchronous Step-Down Controller with 5-Bit VID
Output Fault Protection, Power Good,
24-Pin SSOP, 3.5V ≤ VIN ≤ 36V
LTC1929
High Efficiency, 2-Phase Synchronous Step-Down Switching Regulator
Low Cost, 2-Phase Current Mode
Adaptive Power and Burst Mode are trademarks of Linear Technology Corporation.
16296f
28
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507 ● www.linear.com
LT/TP 1201 2K • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 1999
Similar pages