MC34166, MC33166 3.0 A, Step-Up/Down/ Inverting Switching Regulators The MC34166, MC33166 series are high performance fixed frequency power switching regulators that contain the primary functions required for dc–to–dc converters. This series was specifically designed to be incorporated in step–down and voltage–inverting configurations with a minimum number of external components and can also be used cost effectively in step–up applications. These devices consist of an internal temperature compensated reference, fixed frequency oscillator with on–chip timing components, latching pulse width modulator for single pulse metering, high gain error amplifier, and a high current output switch. Protective features consist of cycle–by–cycle current limiting, undervoltage lockout, and thermal shutdown. Also included is a low power standby mode that reduces power supply current to 36 µA. • Output Switch Current in Excess of 3.0 A • Fixed Frequency Oscillator (72 kHz) with On–Chip Timing • Provides 5.05 V Output without External Resistor Divider • Precision 2% Reference • 0% to 95% Output Duty Cycle • Cycle–by–Cycle Current Limiting • Undervoltage Lockout with Hysteresis • Internal Thermal Shutdown • Operation from 7.5 V to 40 V • Standby Mode Reduces Power Supply Current to 36 µA • Economical 5–Lead TO–220 Package with Two Optional Leadforms • Also Available in Surface Mount D2PAK Package • Moisture Sensitivity Level (MSL) Equals 1 http://onsemi.com x A WL Y WW = 3 or 4 = Assembly Location = Wafer Lot = Year = Work Week TO–220 TH SUFFIX CASE 314A MARKING DIAGRAMS MC 3x166T AWLYWW 1 5 TO–220 TV SUFFIX CASE 314B 1 MC 3x166T AWLYWW 5 Heatsink surface connected to Pin 3 Vin MC 3x166T AWLYWW TO–220 T SUFFIX CASE 314D 4 ILIMIT 1 Oscillator S Q 5 2 R Pin PWM UVLO Thermal L 1. 2. 3. 4. 5. Voltage Feedback Input Switch Output Ground Input Voltage/VCC Compensation/Standby Reference EA 1 VO 5.05 V/ 3.0 A 3 5 D2PAK D2T SUFFIX CASE 936A 1 5 Heatsink surface (shown as terminal 6 in case outline drawing) is connected to Pin 3 MC 3x166T AWLYWW 1 5 This device contains 143 active transistors. ORDERING INFORMATION Figure 1. Simplified Block Diagram (Step Down Application) Semiconductor Components Industries, LLC, 2002 January, 2002 – Rev. 6 See detailed ordering and shipping information in the package dimensions section on page 17 of this data sheet. 1 Publication Order Number: MC34166/D MC34166, MC33166 MAXIMUM RATINGS (Note 2) Symbol Rating Power Supply Input Voltage Value Unit VCC 40 V VO(switch) –1.5 to + Vin V VFB, VComp –1.0 to + 7.0 V Power Dissipation Case 314A, 314B and 314D (TA = +25°C) Thermal Resistance, Junction–to–Ambient Thermal Resistance, Junction–to–Case Case 936A (D2PAK) (TA = +25°C) Thermal Resistance, Junction–to–Ambient Thermal Resistance, Junction–to–Case PD θJA θJC PD θJA θJC Internally Limited 65 5.0 Internally Limited 70 5.0 W °C/W °C/W W °C/W °C/W Operating Junction Temperature TJ +150 °C Operating Ambient Temperature (Note 3) MC34166 MC33166 TA Storage Temperature Range Tstg Switch Output Voltage Range Voltage Feedback and Compensation Input Voltage Range °C 0 to + 70 – 40 to + 85 1. Maximum package power dissipation limits must be observed to prevent thermal shutdown activation. 2. This device series contains ESD protection and exceeds the following tests: Human Body Model 2000 V per MIL–STD–883, Method 3015. Machine Model Method 200 V. Thigh = + 70°C for MC34166 3. Tlow = 0°C for MC34166 = – 40°C for MC33166 = + 85°C for MC33166 http://onsemi.com 2 – 65 to +150 °C MC34166, MC33166 ELECTRICAL CHARACTERISTICS (VCC = 12 V, for typical values TA = +25°C, for min/max values TA is the operating ambient temperature range that applies [Notes 4, 5], unless otherwise noted.) Symbol Min Typ Max Unit TA = +25°C TA = Tlow to Thigh fOSC 65 62 72 – 79 81 kHz TA = +25°C TA = Tlow to Thigh VFB(th) 4.95 4.85 5.05 – 5.15 5.2 V Regline – 0.03 0.078 %/V Characteristic OSCILLATOR Frequency (VCC = 7.5 V to 40 V) ERROR AMPLIFIER Voltage Feedback Input Threshold Line Regulation (VCC = 7.5 V to 40 V, TA = +25°C) IIB – 0.15 1.0 µA PSRR 60 80 – dB VOH VOL 4.2 – 4.9 1.6 – 1.9 DC(max) DC(min) 92 0 95 0 100 0 Vsat – (VCC –1.5) (VCC –1.8) V Isw(off) – 0 100 µA Ipk(switch) 3.3 4.3 6.0 Input Bias Current (VFB = VFB(th) + 0.15 V) Power Supply Rejection Ratio (VCC = 10 V to 20 V, f = 120 Hz) Output Voltage Swing High State (ISource = 75 µA, VFB = 4.5 V) Low State (ISink = 0.4 mA, VFB = 5.5 V) V PWM COMPARATOR Duty Cycle Maximum (VFB = 0 V) Minimum (VComp = 1.9 V) % SWITCH OUTPUT Output Voltage Source Saturation (VCC = 7.5 V, ISource = 3.0 A) Off–State Leakage (VCC = 40 V, Pin 2 = Gnd) Current Limit Threshold Switching Times (VCC = 40 V, Ipk = 3.0 A, L = 375 µH, TA = +25°C) Output Voltage Rise Time Output Voltage Fall Time A ns tr tf – – 100 50 200 100 Startup Threshold (VCC Increasing, TA = +25°C) Vth(UVLO) 5.5 5.9 6.3 V Hysteresis (VCC Decreasing, TA = +25°C) VH(UVLO) 0.6 0.9 1.2 V – – 36 31 100 55 µA mA UNDERVOLTAGE LOCKOUT TOTAL DEVICE Power Supply Current (TA = +25°C ) Standby (VCC = 12 V, VComp < 0.15 V) Operating (VCC = 40 V, Pin 1 = Gnd for maximum duty cycle) ICC 4. Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible. 5. Tlow = 0°C for MC34166 Thigh = + 70°C for MC34166 = – 40°C for MC33166 = + 85°C for MC33166 http://onsemi.com 3 MC34166, MC33166 100 VCC = 12 V VFB(th) Max = 5.15 V 5.17 5.09 IIB, INPUT BIAS CURRENT (nA) V FB(th), VOLTAGE FEEDBACK INPUT THRESHOLD (V) 5.25 VFB(th) Typ = 5.05 V 5.01 VFB(th) Min = 4.95 V 4.93 4.85 -55 -25 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 100 VCC = 12 V VFB = VFB(th) 80 60 40 20 0 -55 125 80 Gain 60 VCC = 12 V VComp = 3.25 V RL = 100 k TA = +25°C 40 0 30 60 90 Phase φ, EXCESS PHASE (DEGREES) A VOL, OPEN LOOP VOLTAGE GAIN (dB) 100 20 120 0 150 -20 10 100 1.0 k 10 k 100 k f, FREQUENCY (Hz) 180 10 M 1.0 M DC, SWITCH OUTPUT DUTY CYCLE (%) ∆ f OSC , OSCILLATOR FREQUENCY CHANGE (%) VCC = 12 V -4.0 -8.0 -25 0 25 50 75 100 TA, AMBIENT TEMPERATURE (°C) 125 1.6 1.2 0.8 VCC = 12 V VFB = 5.5 V TA = +25°C 0.4 0 0 0.4 0.8 1.2 1.6 ISink, OUTPUT SINK CURRENT (mA) 2.0 Figure 5. Error Amp Output Saturation versus Sink Current 4.0 -12 -55 100 2.0 Figure 4. Error Amp Open Loop Gain and Phase versus Frequency 0 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) Figure 3. Voltage Feedback Input Bias Current versus Temperature Vsat , OUTPUT SATURATION VOLTAGE (V) Figure 2. Voltage Feedback Input Threshold versus Temperature -25 125 100 80 VCC = 12 V TA = +25°C 60 40 20 0 1.5 Figure 6. Oscillator Frequency Change versus Temperature 2.0 2.5 3.0 3.5 4.0 VComp, COMPENSATION VOLTAGE (V) Figure 7. Switch Output Duty Cycle versus Compensation Voltage http://onsemi.com 4 4.5 0 0 VCC -0.5 Vsw, SWITCH OUTPUT VOLTAGE (V) Vsat , SWITCH OUTPUT SOURCE SATURATION (V) MC34166, MC33166 TA = +25°C -1.0 -1.5 -2.0 -2.5 -3.0 0 1.0 2.0 3.0 4.0 ISource, SWITCH OUTPUT SOURCE CURRENT (A) VCC = 12 V Pin 5 = 2.0 V Pins 1, 3 = Gnd Pin 2 Driven Negative -0.2 -0.4 Isw = 100 µA -0.6 -0.8 Isw = 10 mA -1.0 -1.2 -55 5.0 VCC = 12 V Pins 1, 2, 3 = Gnd 100 125 4.5 4.3 4.1 -25 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 100 Pin 4 = VCC Pins 1, 3, 5 = Gnd Pin 2 Open TA = +25°C 120 80 40 0 0 125 Figure 10. Switch Output Current Limit Threshold versus Temperature 10 20 30 VCC, SUPPLY VOLTAGE (V) 40 Figure 11. Standby Supply Current versus Supply Voltage 6.5 40 Startup Threshold VCC Increasing 6.0 I CC, SUPPLY CURRENT (mA) Vth(UVLO) , UNDERVOLTAGE LOCKOUT THRESHOLD (V) 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 160 4.7 5.5 Turn-Off Threshold VCC Decreasing 5.0 4.5 4.0 -55 -25 Figure 9. Negative Switch Output Voltage versus Temperature I CC, SUPPLY CURRENT ( µ A) I pk(switch) , CURRENT LIMIT THRESHOLD (A) Figure 8. Switch Output Source Saturation versus Source Current 3.9 -55 Gnd -25 0 25 50 75 TA, AMBIENT TEMPERATURE (°C) 100 30 20 0 125 Pin 4 = VCC Pins 1, 3 = Gnd Pins 2, 5 Open TA = +25°C 10 0 Figure 12. Undervoltage Lockout Threshold versus Temperature 10 20 30 VCC, SUPPLY VOLTAGE (V) Figure 13. Operating Supply Current versus Supply Voltage http://onsemi.com 5 40 MC34166, MC33166 Vin Current Sense + S Switch Output Q R Pulse Width Modulator L 5.05 V Reference + + Error Amp 100 µA Compensation 5 Voltage Feedback Input 1 120 3 2 Undervoltage Lockout PWM Latch Thermal Shutdown Gnd Input Voltage/VCC Cin Oscillator CT 4 CF RF = Sink Only Positive True Logic Figure 14. MC34166 Representative Block Diagram 4.1 V Timing Capacitor CT Compensation 2.3 V ON Switch Output OFF Figure 15. Timing Diagram http://onsemi.com 6 R2 R1 CO VO MC34166, MC33166 INTRODUCTION at 4.3 A. Figure 10 illustrates switch output current limit threshold versus temperature. The MC34166, MC33166 series are monolithic power switching regulators that are optimized for dc–to–dc converter applications. These devices operate as fixed frequency, voltage mode regulators containing all the active functions required to directly implement step–down and voltage–inverting converters with a minimum number of external components. They can also be used cost effectively in step–up converter applications. Potential markets include automotive, computer, industrial, and cost sensitive consumer products. A description of each section of the device is given below with the representative block diagram shown in Figure 14. Error Amplifier and Reference A high gain Error Amplifier is provided with access to the inverting input and output. This amplifier features a typical dc voltage gain of 80 dB, and a unity gain bandwidth of 600 kHz with 70 degrees of phase margin (Figure 4). The noninverting input is biased to the internal 5.05 V reference and is not pinned out. The reference has an accuracy of ± 2.0% at room temperature. To provide 5.0 V at the load, the reference is programmed 50 mV above 5.0 V to compensate for a 1.0% voltage drop in the cable and connector from the converter output. If the converter design requires an output voltage greater than 5.05 V, resistor R1 must be added to form a divider network at the feedback input as shown in Figures 14 and 19. The equation for determining the output voltage with the divider network is: Oscillator The oscillator frequency is internally programmed to 72 kHz by capacitor CT and a trimmed current source. The charge to discharge ratio is controlled to yield a 95% maximum duty cycle at the Switch Output. During the discharge of CT, the oscillator generates an internal blanking pulse that holds the inverting input of the AND gate high, disabling the output switch transistor. The nominal oscillator peak and valley thresholds are 4.1 V and 2.3 V respectively. Vout 5.05 RR2 1 1 External loop compensation is required for converter stability. A simple low–pass filter is formed by connecting a resistor (R2) from the regulated output to the inverting input, and a series resistor–capacitor (RF, CF) between Pins 1 and 5. The compensation network component values shown in each of the applications circuits were selected to provide stability over the tested operating conditions. The step–down converter (Figure 19) is the easiest to compensate for stability. The step–up (Figure 21) and voltage–inverting (Figure 23) configurations operate as continuous conduction flyback converters, and are more difficult to compensate. The simplest way to optimize the compensation network is to observe the response of the output voltage to a step load change, while adjusting RF and CF for critical damping. The final circuit should be verified for stability under four boundary conditions. These conditions are minimum and maximum input voltages, with minimum and maximum loads. By clamping the voltage on the error amplifier output (Pin 5) to less than 150 mV, the internal circuitry will be placed into a low power standby mode, reducing the power supply current to 36 µA with a 12 V supply voltage. Figure 11 illustrates the standby supply current versus supply voltage. The Error Amplifier output has a 100 µA current source pull–up that can be used to implement soft–start. Figure 18 shows the current source charging capacitor CSS through a series diode. The diode disconnects CSS from the feedback loop when the 1.0 M resistor charges it above the operating range of Pin 5. Pulse Width Modulator The Pulse Width Modulator consists of a comparator with the oscillator ramp voltage applied to the noninverting input, while the error amplifier output is applied into the inverting input. Output switch conduction is initiated when CT is discharged to the oscillator valley voltage. As CT charges to a voltage that exceeds the error amplifier output, the latch resets, terminating output transistor conduction for the duration of the oscillator ramp–up period. This PWM/Latch combination prevents multiple output pulses during a given oscillator clock cycle. Figures 7 and 15 illustrate the switch output duty cycle versus the compensation voltage. Current Sense The MC34166 series utilizes cycle–by–cycle current limiting as a means of protecting the output switch transistor from overstress. Each on–cycle is treated as a separate situation. Current limiting is implemented by monitoring the output switch transistor current buildup during conduction, and upon sensing an overcurrent condition, immediately turning off the switch for the duration of the oscillator ramp–up period. The collector current is converted to a voltage by an internal trimmed resistor and compared against a reference by the Current Sense comparator. When the current limit threshold is reached, the comparator resets the PWM latch. The current limit threshold is typically set http://onsemi.com 7 MC34166, MC33166 Switch Output functional before the output stage is enabled. The internal 5.05 V reference is monitored by the comparator which enables the output stage when VCC exceeds 5.9 V. To prevent erratic output switching as the threshold is crossed, 0.9 V of hysteresis is provided. The output transistor is designed to switch a maximum of 40 V, with a minimum peak collector current of 3.3 A. When configured for step–down or voltage–inverting applications, as in Figures 19 and 23, the inductor will forward bias the output rectifier when the switch turns off. Rectifiers with a high forward voltage drop or long turn–on delay time should not be used. If the emitter is allowed to go sufficiently negative, collector current will flow, causing additional device heating and reduced conversion efficiency. Figure 9 shows that by clamping the emitter to 0.5 V, the collector current will be in the range of 100 µA over temperature. A 1N5822 or equivalent Schottky barrier rectifier is recommended to fulfill these requirements. Thermal Protection Internal Thermal Shutdown circuitry is provided to protect the integrated circuit in the event that the maximum junction temperature is exceeded. When activated, typically at 170°C, the latch is forced into a ‘reset’ state, disabling the output switch. This feature is provided to prevent catastrophic failures from accidental device overheating. It is not intended to be used as a substitute for proper heatsinking. The MC34166 is contained in a 5–lead TO–220 type package. The tab of the package is common with the center pin (Pin 3) and is normally connected to ground. Undervoltage Lockout An Undervoltage Lockout comparator has been incorporated to guarantee that the integrated circuit is fully DESIGN CONSIDERATIONS tight component layout is recommended. Capacitors CIN, CO, and all feedback components should be placed as close to the IC as physically possible. It is also imperative that the Schottky diode connected to the Switch Output be located as close to the IC as possible. Do not attempt to construct a converter on wire–wrap or plug–in prototype boards. Special care should be taken to separate ground paths from signal currents and ground paths from load currents. All high current loops should be kept as short as possible using heavy copper runs to minimize ringing and radiated EMI. For best operation, a http://onsemi.com 8 MC34166, MC33166 + Error Amp 100 µA + Error Amp 100 µA 120 1 120 Compensation Compensation 1 5 R1 5 R1 I = Standby Mode VShutdown = VZener + 0.7 Figure 17. Over Voltage Shutdown Circuit Figure 16. Low Power Standby Circuit + Error Amp 100 µA 1 120 Compensation D2 Vin 1.0 M 5 R1 D1 Css tSoft-Start ≈ 35,000 Css Figure 18. Soft–Start Circuit http://onsemi.com 9 MC34166, MC33166 Vin 12 V + 4 ILIMIT + Oscillator Cin 330 S Q1 Q R 2 PWM D1 1N5822 UVLO L 190 µH Thermal Reference + + EA R2 1 5 3 Test CF RF 0.1 68 k CO 2200 6.8 k + VO 5.05 V/3.0 A R1 Conditions Results Line Regulation Vin = 8.0 V to 36 V, IO = 3.0 A 5.0 mV = ± 0.05% Load Regulation Vin = 12 V, IO = 0.25 A to 3.0 A 2.0 mV = ± 0.02% Output Ripple Vin = 12 V, IO = 3.0 A 10 mVpp Short Circuit Current Vin = 12 V, RL = 0.1 Ω 4.3 A Efficiency Vin = 12 V, IO = 3.0 A 82.8% L = Coilcraft M1496–A or General Magnetics Technology GMT–0223, 42 turns of #16 AWG on Magnetics Inc. 58350–A2 core. Heatsink = AAVID Engineering Inc. 5903B, or 5930B. The Step–Down Converter application is shown in Figure 19. The output switch transistor Q1 interrupts the input voltage, generating a squarewave at the LCO filter input. The filter averages the squarewaves, producing a dc output voltage that can be set to any level between Vin and Vref by controlling the percent conduction time of Q1 to that of the total oscillator cycle time. If the converter design requires an output voltage greater than 5.05 V, resistor R1 must be added to form a divider network at the feedback input. Figure 19. Step–Down Converter + - + ÉÉÉÉÉ ÉÉÉÉÉ ÉÉÉÉÉ ÉÉÉÉÉ ÉÉÉ ÉÉÉÉÉ + R2 - VO CO Vin 1.9 ″ + (Bottom View) R1 L CF D1 RF Cin MC34166 STEP–DOWN 3.0″ (Top View) Figure 20. Step–Down Converter Printed Circuit Board and Component Layout http://onsemi.com 10 MC34166, MC33166 Vin 12 V + 4 ILIMIT + Oscillator Cin 330 S D1 1N5822 Q1 Q R 2 PWM L 190 µH UVLO *RG 620 D4 1N4148 Thermal Q2 MTP3055EL Reference + D3 1N967A + EA 1 5 3 D2 1N5822 R2 CF RF 0.47 4.7 k CO 1000 6.8 k VO 28 V/0.6 A + R1 1.5 k *Gate resistor RG, zener diode D3, and diode D4 are required only when Vin is greater than 20 V. Test Conditions Results Line Regulation Vin = 8.0 V to 24 V, IO = 0.6 A 23 mV = ± 0.41% Load Regulation Vin = 12 V, IO = 0.1 A to 0.6 A 3.0 mV = ± 0.005% Output Ripple Vin = 12 V, IO = 0.6 A 100 mVpp Short Circuit Current Vin = 12 V, RL = 0.1 Ω 4.0 A Efficiency Vin = 12 V, IO = 0.6 A 82.8% L = Coilcraft M1496–A or General Magnetics Technology GMT–0223, 42 turns of #16 AWG on Magnetics Inc. 58350–A2 core. Heatsink = AAVID Engineering Inc. MC34166: 5903B, or 5930B MTP3055EL: 5925B Figure 21 shows that the MC34166 can be configured as a step–up/down converter with the addition of an external power MOSFET. Energy is stored in the inductor during the on–time of transistors Q1 and Q2. During the off–time, the energy is transferred, with respect to ground, to the output filter capacitor and load. This circuit configuration has two significant advantages over the basic step–up converter circuit. The first advantage is that output short–circuit protection is provided by the MC34166, since Q1 is directly in series with Vin and the load. Second, the output voltage can be programmed to be less than Vin. Notice that during the off–time, the inductor forward biases diodes D1 and D2, transferring its energy with respect to ground rather than with respect to Vin. When operating with Vin greater than 20 V, a gate protection network is required for the MOSFET. The network consists of components RG, D3, and D4. Figure 21. Step–Up/Down Converter ÎÎÎÎÎ ÎÎÎÎÎ ÎÎÎÎÎ ÎÎÎÎÎ ÎÎÎ ÎÎÎÎÎ + CO + (Bottom View) CF R1 + D1 Cin RF D2 (Top View) Figure 22. Step–Up/Down Converter Printed Circuit Board and Component Layout http://onsemi.com 11 ÎÎ Î ÎÎ Î ÎÎ Î ÎÎ ÎÎ Q2 - - D3 R2 Vin VO L + 1.9″ MC34166 STEP–UP/DOWN 3.45″ RG MC34166, MC33166 Vin 12 V + 4 ILIMIT + Oscillator Cin 330 S Q1 Q R 2 PWM L 190 µH UVLO D1 1N5822 Thermal Reference + + EA R1 1 Test CF RF 0.47 4.7 k + 5 3 2.4 k C1 R2 3.3 k Conditions VO -12 V/1.0 A CO 2200 0.047 Results Line Regulation Vin = 8.0 V to 24 V, IO = 1.0 A 3.0 mV = ± 0.01% Load Regulation Vin = 12 V, IO = 0.1 A to 1.0 A 4.0 mV = ± 0.017% Output Ripple Vin = 12 V, IO = 1.0 A 80 mVpp Short Circuit Current Vin = 12 V, RL = 0.1 Ω 3.74 A Efficiency Vin = 12 V, IO = 1.0 A 81.2% L = Coilcraft M1496–A or General Magnetics Technology GMT–0223, 42 turns of #16 AWG on Magnetics Inc. 58350–A2 core. Heatsink = AAVID Engineering Inc. 5903B, or 5930B. Two potential problems arise when designing the standard voltage–inverting converter with the MC34166. First, the Switch Output emitter is limited to –1.5 V with respect to the ground pin and second, the Error Amplifier’s noninverting input is internally committed to the reference and is not pinned out. Both of these problems are resolved by connecting the IC ground pin to the converter’s negative output as shown in Figure 23. This keeps the emitter of Q1 positive with respect to the ground pin and has the effect of reversing the Error Amplifier inputs. Note that the voltage drop across R1 is equal to 5.05 V when the output is in regulation. Figure 23. Voltage–Inverting Converter 3.0″ Cin (Bottom View) + CO CF L RF + + ÎÎÎÎÎ ÎÎÎÎÎ ÎÎÎÎÎ ÎÎÎÎÎ ÎÎÎ ÎÎÎÎÎ R2 - VO D1 Vin - R1 C1 + (Top View) Figure 24. Voltage–Inverting Converter Printed Circuit Board and Component Layout http://onsemi.com 12 + + 1.9 ″ MC34166 VOLTAGE-INVERTING + + MC34166, MC33166 Vin 24 V + 4 ILIMIT + Oscillator 1000 S Q R 2 PWM UVLO 1N5822 MUR110 + Thermal T1 Reference + MUR110 VO3 1000 -12 V/100 mA VO2 + 1000 12 V/300 mA + EA 6.8 k 1 1000 + VO1 5.05 V/2.0 A 5 3 0.1 Tests 68 k Conditions Results Line Regulation 5.0 V 12 V –12 V Vin = 15 V to 30 V, IO1 = 2.0 A, IO2 = 300 mA, IO3 = 100 mA 4.0 mV = ± 0.04% 450 mV = ±1.9% 350 mV = ±1.5% Load Regulation 5.0 V 12 V –12 V Vin = 24 V, IO1 = 500 mA to 2.0 A, IO2 = 300 mA, IO3 = 100 mA Vin = 24 V, IO1 = 2.0 A, IO2 = 100 mA to 300 mA, IO3 = 100 mA Vin = 24 V, IO1 = 2.0 A, IO2 = 300 mA, IO3 = 30 mA to 100 mA 2.0 mV = ± 0.02% 420 mV = ±1.7% 310 mV = ±1.3% Output Ripple 5.0 V 12 V –12 V Vin = 24 V, IO1 = 2.0 A, IO2 = 300 mA, IO3 = 100 mA 50 mVpp 25 mVpp 10 mVpp Short Circuit Current 5.0 V 12 V –12 V Vin = 24 V, RL = 0.1 Ω 4.3 A 1.83 A 1.47 A Vin = 24 V, IO1 = 2.0 A, IO2 = 300 mA, IO3 = 100 mA 83.3% Efficiency TOTAL T1 = Primary: Coilcraft M1496-A or General Magnetics Technology GMT–0223, 42 turns of #16 AWG on Magnetics Inc. 58350-A2 core. T1 = Secondary: VO2 – 65 turns of #26 AWG T1 = Secondary: VO3 – 96 turns of #28 AWG Heatsink = AAVID Engineering Inc. 5903B, or 5930B. Multiple auxiliary outputs can easily be derived by winding secondaries on the main output inductor to form a transformer. The secondaries must be connected so that the energy is delivered to the auxiliary outputs when the Switch Output turns off. During the OFF time, the voltage across the primary winding is regulated by the feedback loop, yielding a constant Volts/Turn ratio. The number of turns for any given secondary voltage can be calculated by the following equation: # TURNS(SEC) VO(SEC) VF(SEC) VO(PRI)VF(PRI) #TURNS(PRI) Note that the 12 V winding is stacked on top of the 5.0 V output. This reduces the number of secondary turns and improves lead regulation. For best auxiliary regulation, the auxiliary outputs should be less than 33% of the total output power. Figure 25. Triple Output Converter http://onsemi.com 13 MC34166, MC33166 + 4 ILIMIT Oscillator 22 0.01 1N5822 S Q1 Q R L Reference 5 Vin -12 V 0.22 + Z1 1 2N3906 470 k R2 5.1 k 0.002 1000 + + VO +36 V/0.25 A R1 36 k MTP 3055E EA 3 MUR415 R1 D1 Thermal + 2 UVLO PWM VO 5.05 R1 0.7 R2 *Gate resistor RG, zener diode D3, and diode D4 are required only when Vin is greater than 20 V. Test Conditions Results Line Regulation Vin = –10 V to – 20 V, IO = 0.25 A 250 mV = ± 0.35% Load Regulation Vin = –12 V, IO = 0.025 A to 0.25 A 790 mV = ±1.19% Output Ripple Vin = –12 V, IO = 0.25 A 80 mVpp Efficiency Vin = –12 V, IO = 0.25 A 79.2% L = Coilcraft M1496–A or ELMACO CHK1050, 42 turns of #16 AWG on Magnetics Inc. 58350–A2 core. Heatsink = AAVID Engineering Inc. 5903B or 5930B Figure 26. Negative Input/Positive Output Regulator + 4 ILIMIT Oscillator + Vin 18 V 1000 S Q R UVLO 2 PWM Brush Motor Thermal Reference + EA 5 3 Test 0.1 Conditions 1N5822 + 1 5.6 k + 1.0 k 47 56 k Results Low Speed Line Regulation Vin = 12 V to 24 V 1760 RPM ±1% High Speed Line Regulation Vin = 12 V to 24 V 3260 RPM ± 6% Figure 27. Variable Motor Speed Control with EMF Feedback Sensing http://onsemi.com 14 50 k Faster 1000 MC34166, MC33166 0.001 T1 MBR20100CT 0.001 1N5404 115 VAC + 1000 MC34166 Step-Down Converter + Output 1 0.001 RFI Filter + 220 MJE13005 MBR20100CT 0.047 1N4937 100k T2 + 1000 0.01 50 0.001 MC34166 Step-Down Converter + Output 2 0.001 3.3 1N4003 + 100 MBR20100CT + 1000 0.001 T1 = T1 = T1 = T1 = T1 = Core and Bobbin - Coilcraft PT3595 Primary - 104 turns #26 AWG Base Drive - 3 turns #26 AWG Secondaries - 16 turns #16 AWG Total Gap - 0.002″ MC34166 Step-Down Converter + Output 3 T2 = Core - TDK T6 x 1.5 x 3 H5C2 T2 = 14 turns center tapped #30 AWG T2 = Heatsink = AAVID Engineering Inc. T2 = MC34166 and MJE13005 - 5903B T2 = MBR20100CT - 5925B The MC34166 can be used cost effectively in off–line applications even though it is limited to a maximum input voltage of 40 V. Figure 28 shows a simple and efficient method for converting the AC line voltage down to 24 V. This preconverter has a total power rating of 125 W with a conversion efficiency of 90%. Transformer T1 provides output isolation from the AC line and isolation between each of the secondaries. The circuit self–oscillates at 50 kHz and is controlled by the saturation characteristics of T2. Multiple MC34166 post regulators can be used to provide accurate independently regulated outputs for a distributed power system. JUNCTIONTOAIR (° C/W) R θ JA, THERMAL RESISTANCE 80 70 3.0 Free Air Mounted Vertically 60 ÎÎÎÎ ÎÎÎÎ ÎÎÎÎ ÎÎÎÎ 2.0 oz. Copper L Minimum Size Pad 50 2.5 2.0 L 1.5 40 RθJA 30 3.5 PD(max) for TA = +50°C 0 5.0 10 15 20 25 30 1.0 L, LENGTH OF COPPER (mm) Figure 29. D2PAK Thermal Resistance and Maximum Power Dissipation versus P.C.B. Copper Length http://onsemi.com 15 PD, MAXIMUM POWER DISSIPATION (W) Figure 28. Off–Line Preconverter MC34166, MC33166 Table 1. Design Equations Calculation Step–Down Step–Up/Down Voltage–Inverting ton toff (Notes 1, 2) Vout VF Vin Vsat Vout Vout VF1 VF2 Vin VsatQ1 VsatQ2 |Vout| VF Vin Vsat ton ton toff ton fosc 1 toff ton toff ton fosc 1 toff ton toff ton fosc 1 toff Duty Cycle (Note 3) ton fosc ton fosc ton fosc IL avg Iout t Iout on 1 toff t Iout on 1 toff Ipk(switch) I IL avg L 2 I IL avg L 2 I IL avg L 2 L Vin VIsatL Voutton VsatQ2 Vin VsatQ1 ton IL Vin ILVsatton Vripple(pp) IL 1 2 (ESR)2 8foscCo tton 1 1 2 (ESR)2 foscCo off tton 1 1 2 (ESR)2 foscCo off R R R Vref 2 1 Vref 2 1 Vref 2 1 R1 R1 R1 1. Vsat – Switch Output source saturation voltage, refer to Figure 8. 2. VF – Output rectifier forward voltage drop. Typical value for 1N5822 Schottky barrier rectifier is 0.5 V. 3. Duty cycle is calculated at the minimum operating input voltage and must not exceed the guaranteed minimum DC(max) specification of 0.92. Vout The following converter characteristics must be chosen: Vout – Desired output voltage. Iout – Desired output current. ∆IL – Desired peak–to–peak inductor ripple current. For maximum output current especially when the duty cycle is greater than 0.5, it is suggested that ∆IL be chosen to be less than 10% of the average inductor current IL avg. This will help prevent Ipk(switch) from reaching the guaranteed minimum current limit threshold of 3.3 A. If the design goal is to use a minimum inductance value, let ∆IL = 2 (IL avg). This will proportionally reduce the converter’s output current capability. Vripple(pp) – Desired peak–to–peak output ripple voltage. For best performance, the ripple voltage should be kept to less than 2% of Vout. Capacitor CO should be a low equivalent series resistance (ESR) electrolytic designed for switching regulator applications. http://onsemi.com 16 MC34166, MC33166 ORDERING INFORMATION Device Operating Temperature Range Package MC33166D2T D2PAK (Surface Mount) MC33166D2TR4 D2PAK (Surface Mount) MC33166T TO–220 (Straight Lead) TA= –40° to +85°C MC33166TH TO–220 (Horizontal Mount) MC33166TV TO–220 (Vertical Mount) MC34166D2T D2PAK (Surface Mount) MC34166D2TR4 D2PAK (Surface Mount) MC34166T TO–220 (Straight Lead) TA= 0° to +70°C MC34166TH TO–220 (Horizontal Mount) MC34166TV TO–220 (Vertical Mount) http://onsemi.com 17 Shipping 50 Units/Rail MC34166, MC33166 PACKAGE DIMENSIONS TO–220 TH SUFFIX CASE 314A–03 ISSUE E –T– B –P– Q C E OPTIONAL CHAMFER A U F L K G 5X J S D 5X NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSION D DOES NOT INCLUDE INTERCONNECT BAR (DAMBAR) PROTRUSION. DIMENSION D INCLUDING PROTRUSION SHALL NOT EXCEED 0.043 (1.092) MAXIMUM. SEATING PLANE 0.014 (0.356) M T P M DIM A B C D E F G J K L Q S U INCHES MIN MAX 0.572 0.613 0.390 0.415 0.170 0.180 0.025 0.038 0.048 0.055 0.570 0.585 0.067 BSC 0.015 0.025 0.730 0.745 0.320 0.365 0.140 0.153 0.210 0.260 0.468 0.505 MILLIMETERS MIN MAX 14.529 15.570 9.906 10.541 4.318 4.572 0.635 0.965 1.219 1.397 14.478 14.859 1.702 BSC 0.381 0.635 18.542 18.923 8.128 9.271 3.556 3.886 5.334 6.604 11.888 12.827 TO–220 TV SUFFIX CASE 314B–05 ISSUE J Q OPTIONAL CHAMFER E A U K NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSION D DOES NOT INCLUDE INTERCONNECT BAR (DAMBAR) PROTRUSION. DIMENSION D INCLUDING PROTRUSION SHALL NOT EXCEED 0.043 (1.092) MAXIMUM. C B –P– S L W V F 5X G 5X 0.24 (0.610) M J T H D 0.10 (0.254) M T P N M –T– http://onsemi.com 18 SEATING PLANE DIM A B C D E F G H J K L N Q S U V W INCHES MIN MAX 0.572 0.613 0.390 0.415 0.170 0.180 0.025 0.038 0.048 0.055 0.850 0.935 0.067 BSC 0.166 BSC 0.015 0.025 0.900 1.100 0.320 0.365 0.320 BSC 0.140 0.153 --0.620 0.468 0.505 --0.735 0.090 0.110 MILLIMETERS MIN MAX 14.529 15.570 9.906 10.541 4.318 4.572 0.635 0.965 1.219 1.397 21.590 23.749 1.702 BSC 4.216 BSC 0.381 0.635 22.860 27.940 8.128 9.271 8.128 BSC 3.556 3.886 --- 15.748 11.888 12.827 --- 18.669 2.286 2.794 MC34166, MC33166 PACKAGE DIMENSIONS TO–220 T SUFFIX CASE 314D–04 ISSUE E –T– –Q– SEATING PLANE NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSION D DOES NOT INCLUDE INTERCONNECT BAR (DAMBAR) PROTRUSION. DIMENSION D INCLUDING PROTRUSION SHALL NOT EXCEED 10.92 (0.043) MAXIMUM. C B E A U L K J H G D DIM A B C D E G H J K L Q U 1234 5 5 PL 0.356 (0.014) T Q M M INCHES MIN MAX 0.572 0.613 0.390 0.415 0.170 0.180 0.025 0.038 0.048 0.055 0.067 BSC 0.087 0.112 0.015 0.025 0.990 1.045 0.320 0.365 0.140 0.153 0.105 0.117 MILLIMETERS MIN MAX 14.529 15.570 9.906 10.541 4.318 4.572 0.635 0.965 1.219 1.397 1.702 BSC 2.210 2.845 0.381 0.635 25.146 26.543 8.128 9.271 3.556 3.886 2.667 2.972 D2PAK D2T SUFFIX CASE 936A–02 ISSUE B –T– OPTIONAL CHAMFER A TERMINAL 6 E U S K B V H NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. TAB CONTOUR OPTIONAL WITHIN DIMENSIONS A AND K. 4. DIMENSIONS U AND V ESTABLISH A MINIMUM MOUNTING SURFACE FOR TERMINAL 6. 5. DIMENSIONS A AND B DO NOT INCLUDE MOLD FLASH OR GATE PROTRUSIONS. MOLD FLASH AND GATE PROTRUSIONS NOT TO EXCEED 0.025 (0.635) MAXIMUM. 1 2 3 4 5 M D 0.010 (0.254) M T L P N G R C http://onsemi.com 19 DIM A B C D E G H K L M N P R S U V INCHES MIN MAX 0.386 0.403 0.356 0.368 0.170 0.180 0.026 0.036 0.045 0.055 0.067 BSC 0.539 0.579 0.050 REF 0.000 0.010 0.088 0.102 0.018 0.026 0.058 0.078 5 REF 0.116 REF 0.200 MIN 0.250 MIN MILLIMETERS MIN MAX 9.804 10.236 9.042 9.347 4.318 4.572 0.660 0.914 1.143 1.397 1.702 BSC 13.691 14.707 1.270 REF 0.000 0.254 2.235 2.591 0.457 0.660 1.473 1.981 5 REF 2.946 REF 5.080 MIN 6.350 MIN MC34166, MC33166 ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. 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