AD AD8652ARZ-REEL7 50 mhz, precision, low distortion, low noise cmos amplifier Datasheet

50 MHz, Precision, Low Distortion,
Low Noise CMOS Amplifiers
AD8651/AD8652
V– 4
TOP VIEW
(Not to Scale)
APPLICATIONS
Optical communications
Laser source drivers/controllers
Broadband communications
High speed ADC and DAC
Microwave link interface
Cell phone PA control
Video line driver
Audio
+IN 3
The AD8651 is a rail-to-rail input and output amplifier with a
gain bandwidth of 50 MHz and a typical voltage offset of
100 µV across common mode from a 5 V supply. It also features
low noise—4.5 nV/√Hz.
The AD8651 can be used in communications applications, such
as cell phone transmission power control, fiber optic
networking, wireless networking, and video line drivers.
OUT A 1
V+
–IN A 2
6
OUT
+IN A 3
5
NC
Figure 1. 8-Lead MSOP (RM-8)
NC 1
The AD8651 is a high precision, low noise, low distortion, railto-rail CMOS operational amplifier that runs from a singlesupply voltage of 2.7 V to 5 V.
NC
7
NC = NO CONNECT
–IN 2
GENERAL DESCRIPTION
8
AD8651
V– 4
NC
OUT A 1
7
V+
–IN A 2
NC = NO CONNECT
Figure 3. 8-Lead SOIC (R-8)
TOP VIEW
(Not to Scale)
8
V+
7
OUT B
6
–IN B
5
+IN B
Figure 2. 8-Lead MSOP (RM-8)
8
6 OUT
TOP VIEW
V– 4 (Not to Scale) 5 NC
AD8652
AD8652
8
V+
7
OUT B
+IN A 3
6 –IN B
TOP VIEW
V– 4 (Not to Scale) 5 +IN B
03301-B-004
+IN 3
AD8651
03301-0-001
NC 1
–IN 2
03301-0-002
Bandwidth: 50 MHz @ 5 V
Low Noise: 4.5 nV/√Hz
Offset voltage: 100 µV typ, specified over
entire common-mode range
41 V/µs slew rate
Rail-to-rail input and output swing
Input bias current: 1 pA
Single-supply operation: 2.7 V to 5.5 V
Space-saving MSOP and SOIC packaging
03301-B-003
PIN CONFIGURATIONS
FEATURES
Figure 4. 8-Lead SOIC (R-8)
The AD8651 features the newest generation of DigiTrim®
in-package trimming. This new generation measures and
corrects the offset over the entire input common-mode range,
providing less distortion from VOS variation than is typical of
other rail-to-rail amplifiers. Offset voltage and CMRR are both
specified and guaranteed over the entire common-mode range
as well as over the extended industrial temperature range.
The AD8651 is offered in the 8-lead SOIC package and the
8-lead MSOP package. It is specified over the extended industrial temperature range (−40°C to +125°C).
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.326.8703
© 2004 Analog Devices, Inc. All rights reserved.
AD8651/AD8652
TABLE OF CONTENTS
Electrical Characteristics ................................................................. 3
Layout, Grounding, and Bypassing considerations ............... 15
Electrical Characteristics ................................................................. 4
Power Supply Bypassing........................................................ 15
Absolute Maximum Ratings............................................................ 5
Grounding............................................................................... 15
ESD Caution.................................................................................. 5
Leakage Currents.................................................................... 15
Typical Performance Characteristics ............................................. 6
Input Capacitance .................................................................. 15
Applications..................................................................................... 14
Output Capacitance ............................................................... 16
Theory of Operation .................................................................. 14
Settling Time........................................................................... 16
Rail-to-Rail Output Stage...................................................... 14
THD Readings vs. Common-Mode Voltage ...................... 16
Rail-to-Rail Input Stage ......................................................... 14
Driving a 16-Bit ADC............................................................ 17
Input Protection ..................................................................... 15
Outline Dimensions ....................................................................... 18
Overdrive Recovery ............................................................... 15
Ordering Guide .......................................................................... 18
REVISION HISTORY
9/04—Data Sheet Changed from Rev. A to Rev. B
Added AD8652 ....................................................................Universal
Change to General Description ....................................................... 1
Changes to Electrical Characteristics ............................................. 3
Changes to Absolute Maximum Ratings ........................................ 5
Change to Figure 23 .......................................................................... 9
Change to Figure 26 .......................................................................... 9
Change to Figure 36 ........................................................................ 11
Change to Figure 42 ........................................................................ 12
Change to Figure 49 ........................................................................ 13
Change to Figure 51 ........................................................................ 13
Inserted Figure 52............................................................................ 13
Change to Theory of Operation section....................................... 14
Change to Input Protection section .............................................. 15
Changes to Ordering Guide ........................................................... 20
6/04—Changed from REV. 0 to REV. A
Change to Figure 18 .............................................................................8
Change to Figure 21 .............................................................................9
Change to Figure 29 .............................................................................10
Change to Figure 30 .............................................................................10
Change to Figure 43 .............................................................................12
Change to Figure 44 .............................................................................12
Change to Figure 47 .............................................................................13
Change to Figure 57 .............................................................................17
10/03 Revision 0: Initial Version
Rev. B | Page 2 of 20
AD8651/AD8652
ELECTRICAL CHARACTERISTICS
Table 1. V+ = 2.7 V, V– = 0 V, VCM = V+/2, TA = 25°C, unless otherwise specified
Parameter
INPUT CHARACTERISTICS
Offset Voltage
AD8651
Symbol
Min
Typ
Max
Unit
100
350
1.4
1.6
300
1.3
μV
mV
mV
μV
mV
μV/°C
pA
pA
pA
pA
pA
V
VOS
0 ≤ VCM ≤ 2.7 V
–40°C ≤ TA ≤ +85°C, 0 ≤ VCM ≤ 2.7 V
–40°C ≤ TA ≤ +125°C, 0 ≤ VCM ≤ 2.7 V
0 ≤ VCM ≤ 2.7 V
–40°C ≤ TA ≤ +125°C, 0 ≤ VCM ≤ 2.7 V
AD8652
Offset Voltage Drift
Input Bias Current
Conditions
90
0.4
4
1
IB
–40°C ≤ TA ≤ +125°C
Input Offset Current
IOS
1
–40°C ≤ TA ≤ +85°C
–40°C ≤ TA ≤ +125°C
Input Voltage Range
Common-Mode Rejection Ratio
AD8651
VCM
CMRR
AD8652
Large Signal Voltage Gain
OUTPUT CHARACTERISTICS
Output Voltage High
Output Voltage Low
Short Circuit Limit
Output Current
POWER SUPPLY
Power Supply Rejection Ratio
Supply Current
AD8651
AVO
VOH
VOL
ISC
Current Noise Density
V+ = 2.7 V, –0.1 V < VCM < +2.8 V
–40°C ≤ TA ≤ +85°C, –0.1 V < VCM < +2.8 V
–40°C ≤ TA ≤ +125°C, –0.1 V < VCM < +2.8 V
V+ = 2.7 V, –0.1 V < VCM < +2.8 V
–40°C ≤ TA ≤ +125°C, –0.1 V < VCM < +2.8 V
RL = 1 kΩ, 200 mV < VO < 2.5 V
RL = 1 kΩ, 200 mV < VO < 2.5 V, TA = +85°C
RL = 1 kΩ, 200 mV < VO < 2.5 V, TA = +125°C
75
70
65
77
73
100
100
95
IL = 250 μA, –40°C ≤ TA ≤ +125°C
IL = 250 μA, –40°C ≤ TA ≤ +125°C
Sourcing
Sinking
2.67
PSRR
VS = 2.7 V to 5.5 V, VCM = 0 V
–40°C ≤ TA ≤ +125°C
76
74
95
88
85
95
90
115
114
108
dB
dB
dB
dB
dB
dB
dB
dB
80
80
+40
V
mV
mA
mA
mA
94
93
dB
dB
30
IO
ISY
IO = 0
–40°C ≤ TA ≤ +125°C
IO = 0
–40°C ≤ TA ≤ +125°C
AD8652
INPUT CAPACITANCE
Differential
Common-Mode
DYNAMIC PERFORMANCE
Slew Rate
Gain Bandwidth Product
Settling Time, 0.01%
Overload Recovery Time
Total Harmonic Distortion + Noise
NOISE PERFORMANCE
Voltage Noise Density
–0.1
10
600
10
30
600
+2.8
9
17.5
12
14.5
19.5
22.5
mA
mA
mA
mA
CIN
SR
GBP
THD + N
en
in
G = 1, RL = 10 kΩ
G=1
G = ±1, 2 V Step
VIN × G = 1.48 V+
G = 1, RL = 600 Ω, f = 1 kHz, VIN = 2 V p-p
6
9
pF
pF
41
50
0.2
0.1
0.0006
V/μs
MHz
μs
μs
%
nV/√Hz
nV/√Hz
fA/√Hz
f = 10 kHz
5
f = 100 kHz
4.5
f = 10 kHz
4
Rev. B | Page 3 of 20
AD8651/AD8652
ELECTRICAL CHARACTERISTICS
Table 2. V+ = 5 V, V– = 0 V, VCM = V+/2, TA = 25°C, unless otherwise specified
Parameter
INPUT CHARACTERISTICS
Offset Voltage
AD8651
Symbol
Min
Typ
Max
Unit
100
350
1.4
1.7
300
1.4
μV
mV
mV
μV
mV
μV/°C
pA
pA
pA
pA
pA
pA
V
VOS
0 ≤ VCM ≤ 5 V
–40°C ≤ TA ≤ +85°C, 0 ≤ VCM ≤ 5 V
–40°C ≤ TA ≤ +125°C, 0 ≤ VCM ≤ 5 V
0 ≤ VCM ≤ 5 V
–40°C ≤ TA ≤ +125°C, 0 ≤ VCM ≤ 5 V
AD8652
Offset Voltage Drift
Input Bias Current
Conditions
90
0.4
4
1
IB
–40°C ≤ TA ≤ +85°C
–40°C ≤ TA ≤ +125°C
Input Offset Current
IOS
1
–40°C ≤ TA ≤ +85°C
–40°C ≤ TA ≤ +125°C
Input Voltage Range
Common-Mode Rejection Ratio
AD8651
VCM
CMRR
AD8652
Large Signal Voltage Gain
OUTPUT CHARACTERISTICS
Output Voltage High
Output Voltage Low
Short Circuit Limit
Output Current
POWER SUPPLY
Power Supply Rejection Ratio
Supply Current
AD8651
AVO
VOH
VOL
ISC
Current Noise Density
0.1 V < VCM < 5.1 V
–40°C ≤ TA ≤ +85°C, 0.1 V < VCM < 5.1 V
–40°C ≤ TA ≤ +125°C, 0.1 V < VCM < 5.1 V
0.1 V < VCM < 5.1 V
–40°C ≤ TA ≤ +125°C, 0.1 V < VCM < 5.1 V
RL = 1 kΩ, 200 mV < VO < 4.8 V
RL = 1 kΩ, 200 mV < VO < 4.8 V, TA = +85°C
RL = 1 kΩ, 200 mV < VO < 4.8 V, TA = +125°C
80
75
70
84
76
100
98
95
IL = 250 µA, –40°C ≤ TA ≤ +125°C
IL = 250 µA, –40°C ≤ TA ≤ +125°C
Sourcing
Sinking
4.97
PSRR
VS = 2.7 V to 5.5 V, VCM = 0 V
–40°C ≤ TA ≤ +125°C
76
74
95
94
90
100
95
115
114
111
dB
dB
dB
dB
dB
dB
dB
dB
80
80
+40
V
mV
mA
mA
mA
94
93
dB
dB
30
IO
ISY
IO = 0
–40°C ≤ TA ≤ +125°C
IO = 0
–40°C ≤ TA ≤ +125°C
AD8652
INPUT CAPACITANCE
Differential
Common-Mode
DYNAMIC PERFORMANCE
Slew Rate
Gain Bandwidth Product
Settling Time, 0.01%
Overload Recovery Time
Total Harmonic Distortion + Noise
NOISE PERFORMANCE
Voltage Noise Density
–0.1
10
30
600
10
30
600
+5.1
9.5
17.5
14.0
15
20.0
23.5
mA
mA
mA
mA
CIN
SR
GBP
THD + N
en
In
G = 1, RL = 10 kΩ
G=1
G = ±1, 2 V Step
VIN × G = 1.2 V+
G = 1, RL = 600 Ω, f = 1 kHz, VIN = 2 V p-p
6
9
pF
pF
41
50
0.2
0.1
0.0006
V/µs
MHz
μs
μs
%
nV/√Hz
nV/√Hz
fA/√Hz
f = 10 kHz
5
f = 100 kHz
4.5
f = 10 kHz
4
Rev. B | Page 4 of 20
AD8651/AD8652
ABSOLUTE MAXIMUM RATINGS
Absolute maximum ratings apply at 25°C, unless otherwise noted.
Table 3.
Parameter
Supply Voltage
Input Voltage
Differential Input Voltage
Output Short-Circuit Duration to GND
Electrostatic Discharge (HBM)
Storage Temperature Range
RM, R Package
Operating Temperature Range
Junction Temperature Range
RM, R Package
Lead Temperature (Soldering, 10 s)
Table 4.
Rating
6.0 V
GND to VS + 0.3 V
±6.0 V
Indefinite
4000 V
Package Type
8-Lead MSOP (RM)
8-Lead SOIC (R)
1
−65°C to +150°C
−40°C to +125°C
θJA1
210
158
θJC
45
43
Unit
°C/W
°C/W
θJA is specified for the worst-case conditions, i.e., θJA is specified for device
soldered in circuit board for surface-mount packages.
−65°C to +150°C
300°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. B | Page 5 of 20
AD8651/AD8652
TYPICAL PERFORMANCE CHARACTERISTICS
100
60
VS = 5V
80
40
60
30
40
03301-B-005
200
160
120
–80
80
–20
40
0
0
0
–40
10
–120
20
–160
20
VOS (µV)
0
Figure 5. Input Offset Voltage Distribution
5
6
Figure 8. Input Offset Voltage vs. Common-Mode Voltage
VS = ±2.5V
VS = ±2.5V
VCM = 0V
200
0
–100
–300
–50
0
50
TEMPERATURE (°C)
100
150
03301-B-006
–200
300
200
100
0
0
20
40
60
80
100
TEMPERATURE (°C)
120
140
03301-B-009
INPUT BAIS CURRENT (pA)
400
100
Figure 9. Input Bias Current vs. Temperature
Figure 6. Input Offset Voltage vs. Temperature
10
60
VS = ±2.5V
VCM = 0V
T: –40°C TO 125°C
50
SUPPLY CURRENT (mA)
8
40
30
20
6
4
2
0
0
1
2
3
4
5
6
7
TCVOS (µV/°C)
8
9
10
11
03301-B-007
10
0
0
1
2
3
4
SUPPLY VOLTAGE (V)
5
Figure 10. Supply Current vs. Supply Voltage
Figure 7. TCVOS Distribution
Rev. B | Page 6 of 20
6
03301-B-010
VOS (µV)
2
3
4
COMMON-MODE VOLTAGE (V)
500
300
NUMBER OF AMPLIFIERS
1
03301-B-008
VOS (µV)
50
–200
NUMBER OF AMPLIFIERS
VS = ±2.5V
VCM = 0V
AD8651/AD8652
2.50
12
VS = 5V
IL = 250µA
VS = ±2.5V
2.00
10
9
8
6
–50
0
50
TEMPERATURE (°C)
100
150
03301-B-011
7
1.50
1.00
0.50
0
–50
0
50
TEMPERATURE (°C)
100
Figure 14. Output Voltage Swing Low vs. Temperature
Figure 11. Supply Current vs. Temperature
100
500
VS = ±2.5V
VS = ±2.5V
80
400
300
CMRR (dB)
VOH
200
60
40
VOL
20
0
0
20
40
60
CURRENT LOAD (mA)
80
100
03301-B-012
100
0
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
03301-B-015
VSY – VOUT (mV)
150
03301-B-014
OUTPUT SWING LOW (mV)
SUPPLY CURRENT (mA)
11
Figure 15. CMRR vs. Frequency
Figure 12. Output Voltage to Supply Rail vs. Load Current
110
4.997
VS = ±2.5V
VS = 5V
IL = 250µA
4.996
CMRR (dB)
4.994
4.993
4.992
100
95
4.990
–50
0
50
TEMPERATURE (°C)
100
150
90
–50
0
50
TEMPERATURE (°C)
100
Figure 16. CMRR vs. Temperature
Figure 13. Output Voltage Swing High vs. Temperature
Rev. B | Page 7 of 20
150
03301-B-016
4.991
03301-B-013
OUTPUT SWING HIGH (V)
105
4.995
AD8651/AD8652
100
100
VS = ±2.5V
91
88
82
–50
0
50
TEMPERATURE (°C)
100
150
03301-B-017
85
10
1
10
Figure 17. CMRR vs. Temperature
100
1k
FREQUENCY (Hz)
10k
Figure 20. Voltage Noise Density vs. Frequency
100
80
VS = ±2.5V
CURRENT NOISE DENSITY (fA/√Hz)
VS = ±2.5V
80
+PSRR
PSRR (dB)
100k
60
–PSRR
40
0
1
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
100M
03301-B-018
20
60
40
20
0
100
1k
10k
100k
FREQUENCY (Hz)
Figure 18. PSRR vs. Frequency
03301-B-021
CMRR (dB)
94
03301-B-020
VOLTAGE NOISE DENSITY (nV/√Hz)
97
Figure 21. Current Noise Density vs. Frequency
100
VS = ±2.5V
VIN = 6.4V
VS = ±2.5V
VIN
VOLTAGE (1V/DIV)
90
VOUT
0
80
–50
0
50
TEMPERATURE (°C)
100
150
TIME (200µs/DIV)
Figure 22. No Phase Reversal
Figure 19. PSRR vs. Temperature
Rev. B | Page 8 of 20
03301-B-022
85
03301-B-019
PSRR (dB)
95
AD8651/AD8652
0
140
60
VS = ±2.5V
RL = 1MΩ
CL = 47pF
VS = ±2.5V
120
–90
60
40
PHASE (Degrees)
OPEN-LOOP GAIN (dB)
80
–135
20
CLOSED-LOOP GAIN (dB)
40
–45
100
G = 100
20
G = 10
0
G=1
–20
–180
100
1k
10k
100k
1M
FREQUENCY (Hz)
10M
100M
–40
5k
Figure 23. Open-Loop Gain and Phase vs. Frequency
50k
500k
5M
FREQUENCY (Hz)
50M
300M
03301-B-026
–20
10
03301-B-023
0
Figure 26. Closed-Loop Gain vs. Frequency
6
117
VS = ±2.5V
RL = 1kΩ
5
114
112
–50
0
50
TEMPERATURE (°C)
100
150
03301-B-024
113
4
3
VS = 2.7V
2
1
0
100k
1M
10M
FREQUENCY (Hz)
Figure 27. Maximum Output Swing vs. Frequency
Figure 24. Open-Loop Gain vs. Temperature
140
IL = 250µA
130
VS = ±2.5V
CL = 47pF
AV = 1
VS = ±2.5V
2.5mA
120
VOLTAGE (1V/DIV)
4.2mA
110
100
90
80
70
60
0
50
100
150
200
OUTPUT VOLTAGE SWING FROM THE RAILS (mV)
250
03301-B-025
OPEN-LOOP GAIN (dB)
100M
TIME (100µs/DIV)
Figure 28. Large Signal Response
Figure 25. Open-Loop Gain vs. Output Voltage Swing
Rev. B | Page 9 of 20
03301-B-028
AVO (dB)
115
VS = 5V
03301-B-027
MAXIMUM OUTPUT SWING (V)
116
AD8651/AD8652
VS = ±2.5V
VIN = 200mV
AV = 1
VS = ±2.5V
VIN = 200mV
GAIN = –15
0V
VOLTAGE (100mV/DIV)
OUTPUT
–2.5V
200mV
INPUT
TIME (10µs/DIV)
TIME (200ns/DIV)
Figure 29. Small Signal Response
Figure 32. Positive Overload Recovery Time
30
40
VS = ±2.5V
25
OUTPUT IMPEDANCE (Ω)
VS = ±2.5V
VIN = 200mV
AV = 1
20
–OS
15
+OS
10
30
20
GAIN = 10
GAIN = 1
10
5
0
10
20
30
40
CAPACITANCE (pF)
50
60
70
0
10
100
Figure 30. Small Signal Overshoot vs. Load Capacitance
10000
100000
60
VS = ±2.5V
VIN = 200mV
GAIN = –15
VS = ±1.35V
VCM = 0V
50
NUMBER OF AMPLIFIERS
2.5V
1000
FREQUENCY (Hz)
Figure 33. Output Impedance vs. Frequency
0V
0V
–200mV
40
30
20
Figure 34. Input Offset Voltage Distribution
Figure 31. Negative Overload Recovery Time
Rev. B | Page 10 of 20
03301-B-034
VOS (µV)
200
160
120
80
40
0
–40
–80
–120
0
–200
TIME (200ns/DIV)
03301-B-031
10
–160
0
03301-B-033
GAIN = 100
03301-B-030
SMALL SIGNAL OVERSHOOT (%)
03301-B-032
03301-B-029
0V
AD8651/AD8652
300
500
VS = ±1.35V
VS = ±1.35V
VCM = 0V
200
400
VSY – VOUT (mV)
0
–100
300
VOH
200
VOL
100
–300
–50
0
50
TEMPERATURE (°C)
100
150
03301-B-035
–200
0
0
Figure 35. Input Offset Voltage vs. Temperature
80
20
40
60
CURRENT LOAD (mA)
80
Figure 38. Output Voltage to Supply Rail vs. Load Current
2.697
VS = 2.7V
VS = 2.7V
IL = 250µA
2.696
60
OUTPUT SWING HIGH (V)
INPUT OFFSET VOLTAGE (µV)
100
03301-B-038
VOS (µV)
100
40
20
2.695
2.694
2.693
2.692
0
0
1
2
INPUT COMMON-MODE VOLTAGE (V)
3
2.690
–50
Figure 36. Input Offset Voltage vs. Common-Mode Voltage
0
50
TEMPERATURE (°C)
100
150
03301-B-039
–20
03301-B-036
2.691
Figure 39. Output Voltage Swing High vs. Temperature
3.00
11
VS = 2.7V
IL = 250µA
VS = ±1.35V
2.50
OUTPUT SWING LOW (mV)
9
8
7
1.50
1.00
0
50
TEMPERATURE (°C)
100
150
0
–50
0
50
TEMPERATURE (°C)
100
Figure 40. Output Voltage Swing Low vs. Temperature
Figure 37. Supply Current vs. Temperature
Rev. B | Page 11 of 20
150
03301-B-040
6
–50
2.00
0.50
03301-B-037
SUPPLY CURRENT (mA)
10
AD8651/AD8652
30
VS = ±1.35V
AV = 1
TIME (200µs/DIV)
25
20
15
–OS
10
+OS
5
0
0
Figure 41. No Phase Reversal
10
20
30
40
CAPACITANCE (pF)
50
60
70
03301-B-044
03301-B-041
VOLTAGE (1V/DIV)
SMALL SIGNAL OVERSHOOT (%)
VS = ±1.35V
VIN = 200mV
Figure 44. Small Signal Overshoot vs. Load Capacitance
VS = ±1.35V
CL = 47pF
AV = 1
VS = ±1.35V
VIN = 200mV
GAIN = –10
VOLTAGE (500mV/DIV)
1.35V
0V
0V
TIME (100µs/DIV)
03301-B-045
03301-B-042
–200mV
TIME (200ns/DIV)
Figure 42. Large Signal Response
Figure 45. Negative Overload Recovery Time
VS = ±1.35V
VIN = 200mV
CL = 47pF
AV = 1
VS = ±1.35V
VIN = 200mV
GAIN = –10
VOLTAGE (100mV/DIV)
0V
–1.35V
200mV
TIME (200ns/DIV)
Figure 46. Positive Overload Recovery Time
Figure 43. Small Signal Response
Rev. B | Page 12 of 20
03301-B-046
TIME (10µs/DIV)
03301-B-043
0V
AD8651/AD8652
120
100
VS = ±1.35V
VS = ±1.35V
RL = 1kΩ
118
80
AVO (dB)
CMRR (dB)
116
60
40
114
112
20
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
108
–50
Figure 47. CMRR vs. Frequency
0
50
TEMPERATURE (°C)
100
Figure 50. Open-Loop Gain vs. Temperature
100
60
VS = ±1.35V
80
40
60
CLOSED-LOOP GAIN (dB)
–PSRR
40
20
20
0
VS = ±1.35V
RL = 1MΩ
CL = 47pF
G = 100
G = 10
G=1
1
10
100
1k
10k
FREQUENCY (Hz)
100k
1M
10M
–40
5k
Figure 48. PSRR vs. Frequency
50k
500k
5M
FREQUENCY (Hz)
300M
Figure 51. Closed-Loop Gain vs. Frequency
140
0
0
R1
10kΩ
VS = ±1.35V
120
+2.5V
80
60
–90
40
20
PHASE (Degrees)
–45
–135
CHANNEL SEPARATION (dB)
–20
100
VIN
–40
V+
28mV p-p
V–
–60
R2
100Ω
V–
VOUT
V+
–2.5V
–80
–100
–120
0
–180
100
1k
10k
100k
1M
FREQUENCY (Hz)
10M
100M
03301-B-050
OPEN-LOOP GAIN (dB)
50M
03301-B-052
0
03301-B-048
–20
–140
100
VS = ±2.5V
1k
10k
100k
FREQUENCY (Hz)
Figure 52. Channel Separation
Figure 49. Open-Loop Gain and Phase vs. Frequency
Rev. B | Page 13 of 20
1M
10M
03301-B-062
PSRR (dB)
+PSRR
–20
10
150
03301-B-051
0
10
03301-B-047
110
AD8651/AD8652
APPLICATIONS
output voltage swing is proportional to the output current, and
larger currents will limit how close the output voltage can get to
the proximity of the output voltage to the supply rail. This is a
characteristic of all rail-to-rail output amplifiers. With 40 mA of
output current, the output voltage can reach within 5 mV of the
positive and negative rails. At light loads of >100 kΩ, the output
swings within ~1 mV of the supplies.
THEORY OF OPERATION
The AD8651 amplifier is a voltage feedback, rail-to-rail input
and output precision CMOS amplifier that operates from 2.7 V
to 5.0 V of power supply voltage. This amplifier uses Analog
Devices’ DigiTrim technology to achieve a higher degree of
precision than is available from most CMOS amplifiers.
DigiTrim technology, used in a number of ADI amplifiers, is a
method of trimming the offset voltage of the amplifier after it
has been assembled. The advantage of post-package trimming is
that it corrects any offset voltages caused by the mechanical
stresses of assembly.
Rail-to-Rail Input Stage
The input common-mode voltage range of the AD8651 extends
to both positive and negative supply voltages. This maximizes
the usable voltage range of the amplifier, an important feature
for single-supply and low voltage applications. This rail-to-rail
input range is achieved by using two input differential pairs, one
NMOS and one PMOS, placed in parallel. The NMOS pair is
active at the upper end of the common-mode voltage range,
and the PMOS pair is active at the lower end of the commonmode range.
The AD8651 is available in standard op amp pinout, making
DigiTrim completely transparent to the user. The input stage of
the amplifier is a true rail-to-rail architecture, allowing the
input common-mode voltage range of the op amp to extend to
both positive and negative supply rails. The open-loop gain of
the AD8651/AD8652 with a load of 1 kΩ is typically 115 dB.
The NMOS and PMOS input stages are separately trimmed
using DigiTrim to minimize the offset voltage in both differential pairs. Both NMOS and PMOS input differential pairs are
active in a 500 mV transition region when the input commonmode voltage is approximately 1.5 V below the positive supply
voltage. A special design technique improves the input offset
voltage in the transition region that traditionally exhibits a
slight VOS variation. As a result, the common-mode rejection
ratio is improved within this transition band. Compared to the
Burr Brown OPA350 amplifier, shown in Figure 53 (A), the
AD8651, shown in Figure 53 (B), exhibits much lower offset
voltage shift across the entire input common-mode range,
including the transition region.
Rail-to-Rail Output Stage
600
400
400
200
200
VOS (µV)
600
0
0
–200
–200
–400
–400
–600
0
1
2
3
4
COMMON-MODE VOLTAGE (V)
5
6
03301-B-053
VOS (µV)
The voltage swing of the output stage is rail-to-rail and is
achieved by using an NMOS and PMOS transistor pair connected in a common source configuration. The maximum
–600
0
(A) OPA350 VOS vs. VCM
1
2
3
4
COMMON-MODE VOLTAGE (V)
(B) AD8651 VOS vs. VCM
Figure 53. Input Offset Distribution over Common-Mode Voltage
Rev. B | Page 14 of 20
5
6
03301-B-054
The AD8651 can be used in any precision op amp application.
The amplifier does not exhibit phase reversal for commonmode voltages within the power supply. With voltage noise of
4.5 nV/√Hz and –105 dB distortion for 10 kHz, 2 V p-p signals,
the AD8651/AD8652 is a great choice for high resolution data
acquisition systems. Its low noise, sub-pA input bias current,
precision offset, and high speed make it a superb preamp
for fast photodiode applications. The speed and output
drive capability of the AD8651 also make it useful in
video applications.
AD8651/AD8652
Input Protection
As with any semiconductor device, if a condition could exist for
the input voltage to exceed the power supply, the device’s input
overvoltage characteristic must be considered. The inputs of the
AD8651 are protected with ESD diodes to either power supply.
Excess input voltage will energize internal PN junctions in the
AD8651, allowing current to flow from the input to the
supplies. This results in an input stage with picoamps of input
current that can withstand up to 4000 V ESD events (human
body model) with no degradation.
Excessive power dissipation through the protection devices will
destroy or degrade the performance of any amplifier. Differential voltages greater than 7 V will result in an input current of
approximately (|VCC – VEE| – 0.7 V)/RI, where RI is the
resistance in series with the inputs. For input voltages beyond
the positive supply, the input current will be approximately (VI
– VCC – 0.7)/RI. For input voltages beyond the negative supply,
the input current will be about (VI – VEE + 0.7)/RI. If the inputs
of the amplifier sustain differential voltages greater than 7 V or
input voltages beyond the amplifier power supply, limit the
input current to 10 mA by using an appropriately sized input
resistor (RI), as shown in Figure 54.
(| VCC – VEE | – 0.7V)
RI >
30mA
FOR LARGE | VCC – VEE |
+
AD8651
–
– VI +
RI >
(VI – VEE – 0.7V)
(VI – VEE + 0.7V)
30mA
FOR VI BEYOND
SUPPLY VOLTAGES
+ VO
RI
Grounding
A ground plane layer is important for densely packed PC
boards to spread the current-minimizing parasitic inductances.
However, an understanding of where the current flows in a
circuit is critical to implementing effective high speed circuit
design. The length of the current path is directly proportional to
the magnitude of parasitic inductances and, therefore, the high
frequency impedance of the path. High speed currents in an
inductive ground return will create an unwanted voltage noise.
The length of the high frequency bypass capacitor leads is
critical. A parasitic inductance in the bypass grounding will
work against the low impedance created by the bypass capacitor.
Place the ground leads of the bypass capacitors at the same
physical location. Because load currents also flow from the
supplies, the ground for the load impedance should be at the
same physical location as the bypass capacitor grounds. For the
larger value capacitors, intended to be effective at lower frequencies, the current return path distance is less critical.
Leakage Currents
30mA
03301-B-055
RI >
(X7R or NPO) are critical and should be as close as possible to
the amplifier package. The 4.7 µF tantalum capacitor is less
critical for high frequency bypassing, and, in most cases, only
one is needed per board at the supply inputs.
Figure 54. Input Protection Method
Overdrive Recovery
Overdrive recovery is defined as the time it takes for the output
of an amplifier to come off the supply rail after an overload
signal is initiated. This is usually tested by placing the amplifier
in a closed-loop gain of 15 with an input square wave of
200 mV p-p while the amplifier is powered from either 5 V or
3 V. The AD8651 has excellent recovery time from overload
conditions (see Figure 31 and Figure 32). The output recovers
from the positive supply rail within 200 ns at all supply voltages.
Recovery from the negative rail is within 100 ns at 5 V supply.
LAYOUT, GROUNDING, AND BYPASSING
CONSIDERATIONS
Power Supply Bypassing
Power supply pins can act as inputs for noise, so care must be
taken that a noise-free, stable dc voltage is applied. The purpose
of bypass capacitors is to create low impedances from the supply
to ground at all frequencies, thereby shunting or filtering most
of the noise. Bypassing schemes are designed to minimize the
supply impedance at all frequencies with a parallel combination
of capacitors of 0.1 µF and 4.7 µF. Chip capacitors of 0.1 µF
Poor PC board layout, contaminants, and the board insulator
material can create leakage currents that are much larger than
the input bias current of the AD8651/AD8652. Any voltage
differential between the inputs and nearby traces will set up
leakage currents through the PC board insulator, for example,
1 V/100 G = 10 pA. Similarly, any contaminants on the board
can create significant leakage (skin oils are a common problem).
To significantly reduce leakages, put a guard ring (shield)
around the inputs and input leads that are driven to the same
voltage potential as the inputs. This ensures that there is no
voltage potential between the inputs and the surrounding area
to set up any leakage currents. To be effective, the guard ring
must be driven by a relatively low impedance source and should
completely surround the input leads on all sides, above and
below, using a multilayer board.
Another effect that can cause leakage currents is the charge
absorption of the insulator material itself. Minimizing the
amount of material between the input leads and the guard
ring will help to reduce the absorption. Also, low absorption
materials, such as Teflon® or ceramic, may be necessary in
some instances.
Input Capacitance
Along with bypassing and ground, high speed amplifiers can be
sensitive to parasitic capacitance between the inputs and
ground. A few picofarads of capacitance will reduce the input
impedance at high frequencies, which in turn increases the
amplifier’s gain, causing peaking in the frequency response or
Rev. B | Page 15 of 20
AD8651/AD8652
oscillations. With the AD8651, additional input damping is
required for stability with capacitive loads greater than 47 pF
with direct input to output feedback (see the next section).
V+
+
Output Capacitance
1) As shown in Figure 55, place a small value resistor (RS) in
series with the output to isolate the load capacitor from the
amplifier’s output. Heavy capacitive loads can reduce the phase
margin of an amplifier and cause the amplifier response to peak
or become unstable. The AD8651 is able to drive up to 47 pF in
a unity gain buffer configuration without oscillation or external
compensation. However, if an application will require a higher
capacitive load drive when the AD8651 is in unity gain, then
the use of external isolation networks can be used. The effect
produced by this resistor is to isolate the op amp output from
the capacitive load. The required amount of series resistance has
been tabulated in Table 5 for different capacitive load. While
this technique will improve the overall capacitive load drive for
the amplifier, its biggest drawback is that it reduces the output
swing of the overall circuit.
VCC
V+
AD8651
03301-B-057
Figure 56. Snubber Network
Settling Time
The settling time of an amplifier is defined as the time it takes
for the output to respond to a step change of input and enter
and remain within a defined error band, as measured relative to
the 50% point of the input pulse. This parameter is especially
important in measurements and control circuits where amplifiers are used to buffer A/D inputs or DAC outputs. The design of
the AD8651 combines a high slew rate and a wide gain bandwidth product to produce an amplifier with very fast settling
time. The AD8651 is configured in the noninverting gain of 1
with a 2 V p-p step applied to its input. The AD8651 has a
settling time of about 130 ns to 0.01% (2 mV). The output is
monitored with a 10×, 10 M, 11.2 pF scope probe.
THD Readings vs. Common-Mode Voltage
Total harmonic distortion of the AD8651 is well below 0.0004%
with any load down to 600 Ω. The distortion is a function of the
circuit configuration, the voltage applied, and the layout, in
addition to other factors. The AD8651 outperforms its
competitor for distortion, especially at frequencies below
20 kHz, as shown in Figure 57.
0.1
RS
VOUT
VSY = +3.5V/–1.5V
VOUT = 2.0V p-p
0.05
V–
RL
0
0
0.02
0.01
Figure 55. Driving Large Capacitive Loads
Table 5. Optimum Values for Driving Large Capacitive Loads
RS
50 Ω
35 Ω
25 Ω
0.005
0.002
OPA350
0.001
0.0005
AD8651
0.0002
0.0001
20
2) Another way to stabilize an op amp driving a large capacitive
load is to use a snubber network, as shown in Figure 56.
Because there is not any isolation resistor in the signal path, this
method has the significant advantage of not reducing the output
swing. The exact values of RS and CS are derived experimentally.
In Figure 56, an optimum RS and CS combination for a
capacitive load drive ranging from 50 pF to 1 nF was chosen.
For this, RS = 3 Ω and CS = 10 nF were chosen.
50
100
500
1k
2k
FREQUENCY (Hz)
5k
Figure 57. Total Harmonic Distortion
3.5V
–
VOUT
AD8651
600Ω
+
VIN
2V p-p
47pF
–1.5V
Figure 58. THD + N Test Circuit
Rev. B | Page 16 of 20
20k
03301-B-058
CL
0
CL
100 pF
500 pF
1.0 nF
RL
03301-B-059
–
CL
V–
200mV
THD + NOISE (%)
2
RS
CS
U1
+
03301-B-056
3
VOUT
V–
–
When using high speed amplifiers, it is important to consider
the effects of the capacitive loading on the amplifier’s stability.
Capacitive loading interacts with the output impedance of the
amplifier, causing reduction of the BW as well as peaking and
ringing of the frequency response. To reduce the effects of the
capacitive loading and allow higher capacitive loads, there are
two commonly used methods:
VIN
V+
AD8651
AD8651/AD8652
5V
Driving a 16-Bit ADC
The AD8651 is configured in an inverting gain of 1 with a 5 V
single supply. Input of 45 kHz is applied, and the ADC samples
at 250 kSPS. The results of this solution are listed in Table 6.
The advantage of this circuit is that the amplifier and ADC can
be powered with the same power supply. For the case of a
noninverting gain of 1, the input common-mode voltage
encompasses both supplies.
0
3
10kΩ
U1
+
V+
VCC
33Ω
AD8651
1µF
2
–
IN
V–
AD7685
2.7nF
1kΩ
VIN
0V – 5V
fIN = 45kHz
1kΩ
Figure 60. AD8651 Driving a 16-Bit ADC
Table 6. Data Acquisition Solution of Figure 60
Parameter
THD + N
SFDR
2nd Harmonics
3rd Harmonics
fSAMPLE = 250kSPS
fIN = 45kHz
INPUT RANGE = 0 TO 5V
–20
Reading (dB)
105.2
106.6
107.7
113.6
–40
–60
1
–80
–100
For more information about the AD7685 data converter, go to
http://www.analog.com/Analog_Root/productPage/productHome/0%2C21
21%2CAD7685%2C00.html
–120
–140
–160
0
10
20
30 40
50 60 70 80
FREQUENCY (kHz)
90
100 110 120
03301-B-060
AMPLITUDE (dB of Full Scale)
10kΩ
03301-B-061
The AD8651 is an excellent choice for driving high speed, high
precision ADCs. The driver amplifier for this type of
application needs to have low THD + N as well as quick settling
time. Figure 60 shows a complete single-supply data acquisition
solution. The AD8651 drives the AD7685, a 250 kSPS, 16-bit
data converter.1
Figure 59. Frequency Response of AD8651 Driving a 16-Bit ADC
Rev. B | Page 17 of 20
OUTLINE DIMENSIONS
3.00
BSC
8
5
4.90
BSC
3.00
BSC
4
PIN 1
0.65 BSC
1.10 MAX
0.15
0.00
0.38
0.22
COPLANARITY
0.10
0.80
0.60
0.40
8°
0°
0.23
0.08
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-187AA
Figure 61. 8-Lead Mini Small Outline Package [MSOP]
(RM-8)
Dimensions shown in millimeters
5.00 (0.1968)
4.80 (0.1890)
8
5
4.00 (0.1574)
3.80 (0.1497) 1
4
1.27 (0.0500)
BSC
0.25 (0.0098)
0.10 (0.0040)
6.20 (0.2440)
5.80 (0.2284)
1.75 (0.0688)
1.35 (0.0532)
0.51 (0.0201)
COPLANARITY
SEATING 0.31 (0.0122)
0.10
PLANE
0.50 (0.0196)
× 45°
0.25 (0.0099)
8°
0.25 (0.0098) 0° 1.27 (0.0500)
0.40 (0.0157)
0.17 (0.0067)
COMPLIANT TO JEDEC STANDARDS MS-012AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Figure 62. 8-Lead Standard Small Outline Package [SOIC]
(R-8)
Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Model
AD8651ARM-REEL
AD8651ARM-R2
AD8651AR
AD8651AR-REEL
AD8651AR-REEL7
AD8652ARMZ-R2*
AD8652ARMZ-REEL*
AD8652ARZ*
AD8652ARZ-REEL*
AD8652ARZ-REEL7*
*
Temperature Range
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
Package Description
8-Lead MSOP
8-Lead MSOP
8-Lead SOIC
8-Lead SOIC
8-Lead SOIC
8-Lead MSOP
8-Lead MSOP
8-Lead SOIC
8-Lead SOIC
8-Lead SOIC
Z = Pb-free part.
Rev. B | Page 18 of 20
Package Option
RM-8
RM-8
R-8
R-8
R-8
RM-8
RM-8
R-8
R-8
R-8
Branding
BEA
BEA
A05
A05
AD8651/AD8652
NOTES
Rev. B | Page 19 of 20
NOTES
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C03301-0-9/04(B)
Rev. B | Page 20 of 20