IRF IR3841WMPBF Highly efficient integrated 8a synchronous buck regulator Datasheet

PD-97658
IR3841WMPbF
SupIRBuck
HIGHLY EFFICIENT
TM
INTEGRATED 8A SYNCHRONOUS BUCK REGULATOR
Features
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Description
Greater than 96% Maximum Efficiency
Wide Input Voltage Range 1.5V to 16V
Wide Output Voltage Range 0.7V to 0.9*Vin
Continuous 8A Load Capability
Integrated Bootstrap-diode
High Bandwidth E/A for excellent transient
performance
Programmable Switching Frequency up to 1.5MHz
Programmable Over Current Protection
PGood output
Hiccup Current Limit
Precision Reference Voltage (0.7V, +/-1%)
Programmable Soft-Start
Enable Input with Voltage Monitoring Capability
Enhanced Pre-Bias Start-up
Seq input for Tracking applications
-40oC to 125oC operating junction temperature
Thermal Protection
Pin compatible option for 4A,12A, and 14A devices
5mm x 6mm Power QFN Package, 0.9 mm height
Lead-free, halogen-free and RoHS compliant
The IR3841W SupIRBuckTM is an easy-to-use,
fully integrated and highly efficient DC/DC
synchronous Buck regulator. The MOSFETs copackaged with the on-chip PWM controller make
IR3841W a space-efficient solution, providing
accurate power delivery for low output voltage
applications.
IR3841W is a versatile regulator which offers
programmability of start up time, switching
frequency and current limit while operating in
wide input and output voltage range.
The switching frequency is programmable from
250kHz to 1.5MHz for an optimum solution.
It also features important protection functions,
such as Pre-Bias startup, hiccup current limit and
thermal shutdown to give required system level
security in the event of fault conditions.
Applications
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Server Applications
Storage Applications
Embedded Telecom Systems
Distributed Point of Load Power Architectures
Netcom Applications
Computing Peripheral Voltage Regulators
General DC-DC Converters
1.5V <Vin<16V
4.5V <Vcc<5.5V
Seq
Enable
Vin
Boot
Vo
Vcc
SW
PGood
PGood
OCSet
Fb
Rt
SS/ SD
Gnd
PGnd
Comp
Fig. 1. Typical application diagram
Rev 8.0
1
PD-97658
IR3841WMPbF
ABSOLUTE MAXIMUM RATINGS
(Voltages referenced to GND unless otherwise specified)
•
Vin ……………………………………………………. -0.3V to 25V
•
Vcc ……………….….…………….……..……….…… -0.3V to 8V (Note2)
•
Boot
……………………………………..……….…. -0.3V to 33V
•
SW
…………………………………………..……… -0.3V to 25V(DC), -4V to 25V(AC, 100ns)
•
Boot to SW
•
OCSet
•
Input / output Pins
•
PGND to GND ……………...………………………….. -0.3V to +0.3V
•
Storage Temperature Range ................................... -55°C To 150°C
•
Junction Temperature Range ................................... -40°C To 150°C (Note2)
•
ESD Classification …………………………… ……… JEDEC Class 1C
•
Moisture sensitivity level………………...………………JEDEC Level 2@260 °C (Note5)
……..…………………………….…..….. -0.3V to Vcc+0.3V (Note1)
………………………………………….……. -0.3V to 30V, 30mA
……………………………….. ... -0.3V to Vcc+0.3V (Note1)
Note1: Must not exceed 8V
Note2: Vcc must not exceed 7.5V for Junction Temperature between -10oC and -40oC
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the
device. These are stress ratings only and functional operation of the device at these or any other
conditions beyond those indicated in the operational sections of the specifications are not implied.
PACKAGE INFORMATION
SW
5mm x 6mm POWER QFN
VIN
11
12
10
PGnd
θJA = 35 o C / W
θJ -PCB = 2 o C / W
Boot
13
Enable
14
1
ORDERING INFORMATION
Rev 8.0
Seq
2
15
Gnd
3
4
5
FB COMP Gnd Rt
6
9
VCC
8
PGood
7
SS OCSet
PACKAGE
DESIGNATOR
PACKAGE
DESCRIPTION
PIN COUNT
PARTS PER
REEL
M
IR3841WMTRPbF
15
4000
M
IR3841WMTR1PbF
15
750
2
PD-97658
IR3841WMPbF
Block Diagram
Fig. 2. Simplified block diagram of the IR3841W
Rev 8.0
3
PD-97658
IR3841WMPbF
Pin Description
Pin
Name
Description
1
Seq
Sequence pin. Use two external resistors to set Simultaneous Power up
sequencing. If this pin is not used connect to Vcc.
2
Fb
3
Comp
4
Gnd
5
Rt
Set the switching frequency. Connect an external resistor from this pin
to Gnd to set the switching frequency.
6
SS/SD
¯¯
Soft start / shutdown. This pin provides user programmable soft-start
function. Connect an external capacitor from this pin to Gnd to set the
start up time of the output voltage. The converter can be shutdown by
pulling this pin below 0.3V.
7
OCSet
Current limit set point. A resistor from this pin to SW pin will set the
current limit threshold.
8
PGood
Power Good status pin. Output is open drain. Connect a pull up resistor
from this pin to Vcc. If unused, it can be left open.
9
VCC
This pin powers the internal IC and the drivers. A minimum of 1uF high
frequency capacitor must be connected from this pin to the power
ground (PGnd).
10
PGnd
Power Ground. This pin serves as a separated ground for the MOSFET
drivers and should be connected to the system’s power ground plane.
11
SW
Switch node. This pin is connected to the output inductor.
12
VIN
Input voltage connection pin.
13
Boot
14
Enable
15
Gnd
Rev 8.0
Inverting input to the error amplifier. This pin is connected directly to the
output of the regulator via resistor divider to set the output voltage and
provide feedback to the error amplifier.
Output of error amplifier. An external resistor and capacitor network is
typically connected from this pin to Fb pin to provide loop
compensation.
Signal ground for internal reference and control circuitry.
Supply voltage for high side driver. A 0.1uF capacitor must be
connected from this pin to SW.
Enable pin to turn on and off the device. Use two external resistors to
set the turn on threshold (see Enable section). Connect this pin to Vcc if
it is not used.
Signal ground for internal reference and control circuitry.
4
PD-97658
IR3841WMPbF
Recommended Operating Conditions
Symbol
Vin
Vcc
Boot to SW
Vo
Io
Fs
Tj
Definition
Input Voltage
Supply Voltage
Supply Voltage
Output Voltage
Output Current
Switching Frequency
Junction Temperature
Min
Max
1.5
4.5
4.5
0.7
0
225
-40
16
5.5
5.5
0.9*Vin
8
1650
125
Units
V
A
kHz
o
C
Electrical Specifications
Unless otherwise specified, these specification apply over 4.5V< Vcc<5.5V, Vin=12V, 0oC<Tj< 125oC.
Typical values are specified at Ta = 25oC.
Parameter
Symbol
Test Condition
Min
TYP
MAX
Units
Power Loss
Power Loss
Ploss
Vcc=5V, Vin=12V, Vo=1.8V, Io=8A,
Fs=600kHz, L=1uH, Note4
1.5
W
MOSFET Rds(on)
Top Switch
Rds(on)_Top
Bottom Switch
Rds(on)_Bot
o
VBoot -Vsw =5V, ID=10A, Tj=25 C
o
Vcc=5V, ID=10A, Tj=25 C
17.8
26.5
8.5
10.7
mΩ
Reference Voltage
Feedback Voltage
VFB
0.7
o
Accuracy
o
0 C<Tj<125 C
o
o
-40 C<Tj<125 C, Note3
V
-1.0
+1.0
-2.0
+2.0
%
Supply Current
VCC Supply Current (Standby)
ICC(Standby)
Vcc Supply Current (Dyn)
ICC(Dyn)
SS=0V, No Switching, Enable low
500
SS=3V, Vcc=5V, Fs=500kHz
Enable high
12
μA
mA
Under Voltage Lockout
VCC-Start-Threshold
VCC_UVLO_Start
Vcc Rising Trip Level
3.95
4.15
4.35
VCC-Stop-Threshold
VCC_UVLO_Stop
Vcc Falling Trip Level
3.65
3.85
4.05
Enable-Start-Threshold
Enable_UVLO_Start
Supply ramping up
1.14
1.2
1.36
Enable-Stop-Threshold
Enable_UVLO_Stop
Supply ramping down
0.9
1.0
1.06
Enable leakage current
Ien
Enable=3.3V
Rev 8.0
15
V
μA
5
PD-97658
IR3841WMPbF
Electrical Specifications (continued)
Unless otherwise specified, these specification apply over 4.5V< Vcc<5.5V, Vin=12V, 0oC<Tj< 125oC.
Typical values are specified at Ta = 25oC.
Parameter
Symbol
Test Condition
Min
TYP
MAX
Units
0.665
0.7
0.735
V
Oscillator
Rt Voltage
Frequency
FS
Rt=59K
225
250
275
Rt=28.7K
450
500
550
Rt=9.31K, Note4
1350
1500
1650
Accuracy
-10
+10
kHz
%
Vramp
Note4
1.8
Vp-p
Ramp Offset
Ramp (os)
Note4
0.6
V
Min Pulse Width
Dmin(ctrl)
Note4
50
Ramp Amplitude
Note4
Fixed Off Time
Max Duty Cycle
130
Dmax
Fs=250kHz
92
Vos
Vfb-Vseq,
Vseq=0.8V
-10
200
ns
%
Error Amplifier
Input Offset Voltage
0
+10
Input Bias Current
IFb(E/A)
-1
+1
Input Bias Current
IVp(E/A)
-1
+1
Isink(E/A)
0.40
0.85
1.2
Isource(E/A)
8
10
13
Sink Current
Source Current
Slew Rate
Gain-Bandwidth Product
DC Gain
mV
μA
mA
SR
Note4
7
12
20
V/μs
GBWP
Note4
20
30
40
MHz
Gain
Note4
100
110
120
dB
Vcc=4.5V
3.4
3.5
3.75
V
120
220
mV
1
V
μA
Maximum Voltage
Vmax(E/A)
Minimum Voltage
Vmin(E/A)
Note4
Common Mode Voltage
0
Soft Start/SD
Soft Start Current
Soft Start Clamp Voltage
Shutdown Output Threshold
ISS
Source
Vss(clamp)
14
20
26
2.7
3.0
3.3
SD
0.3
V
Over Current Protection
OCSET Current
OC Comp Offset Voltage
SS off time
IOCSET
VOFFSET
Fs=250kHz
20.8
23.6
26.4
Fs=500kHz
43
48.8
54.6
Fs=1500kHz
136
154
172
Note4
-10
0
+10
SS_Hiccup
4096
μA
mV
Cycles
Bootstrap Diode
Forward Voltage
I(Boot)=30mA
180
260
470
mV
5
10
30
ns
Deadband
Deadband time
Rev 8.0
Note4
6
PD-97658
IR3841WMPbF
Electrical Specifications (continued)
Unless otherwise specified, these specification apply over 4.5V< Vcc<5.5V, Vin=12V, 0oC<Tj< 125oC.
Typical values are specified at Ta = 25oC.
Parameter
SYM
Test Condition
Min
TYP
MAX
Units
Thermal Shutdown
Thermal Shutdown
Note4
140
Hysteresis
Note4
20
o
C
Power Good
Power Good upper
Threshold
Upper Threshold
Delay
Power Good lower
Threshold
Lower Threshold
Delay
Delay Comparator
Threshold
Delay Comparator
Hysteresis
PGood Voltage Low
Leakage Current
VPG(upper)
Fb Rising
VPG(upper)_Dly
Fb Rising
VPG(lower)
Fb Falling
VPG(lower)_Dly
Fb Falling
PG(Delay)
Relative to charge voltage, SS rising
Delay(hys)
Note4
PG(voltage)
0.770
0.805
256/Fs
0.560
0.595
V
s
0.630
256/Fs
V
s
2
2.1
2.3
V
260
300
340
mV
IPGood=-5mA
I leakage
0.840
0
0.5
V
10
μA
Switch Node
SW Bias Current
SW=0V, Enable=0V
Isw
SW=0V,Enable=high,SS=3V,Vseq=0V
, Note4
6
μA
Note3: Cold temperature performance is guaranteed via correlation using statistical quality control. Not tested in production.
Note4: Guaranteed by Design but not tested in production.
Note5: Upgrade to industrial/MSL2 level applies from date codes 1227 (marking explained on application note AN1132 page 2).
Products with prior date code of 1227 are qualified with MSL3 for Consumer market.
Rev 8.0
7
PD-97658
IR3841WMPbF
Typical Efficiency and Power Loss Curves
Vin=12V, Vcc=5V, Io=1A-8A, Fs=600kHz, Room Temperature, No Air Flow
The table below shows the inductors used for each of the output voltages
in the efficiency measurement.
Vo (V)
0.7
0.8
0.9
1.0
1.1
1.2
1.5
1.8
2.5
3.3
5.0
L (uH)
0.4
0.4
0.4
0.5
0.5
0.6
0.9
0.9
1.0
1.2
1.5
P/N
DCR (mOhm)
59PR9875N
0.29
59PR9875N
0.29
59PR9875N
0.29
59PR9876N
0.29
59PR9876N
0.29
MPL104-0R6
1.50
MPC1040LR88
2.30
MPC1040LR88
2.30
MPL105-1R0IR
2.30
MPL105-1R0IR
2.30
MPL105-1R0IR
2.30
98
96
Efficiency (%)
94
92
90
88
86
84
82
80
1
2
3
4
5
6
7
8
Load Current (A)
0.9V
1.0V
1.1V
1.2V
1.5V
1.8V
2.5V
3.3V
5.0V
2.1
1.9
Power Loss (W)
1.7
1.5
1.3
1.1
0.9
0.7
0.5
0.3
0.1
1
2
3
4
5
6
7
8
Load Current (A)
0.9V
Rev 8.0
1.0V
1.1V
1.2V
1.5V
1.8V
2.5V
3.3V
5.0V
8
PD-97658
IR3841WMPbF
Typical Efficiency and Power Loss Curves
Vin=5V, Vcc=5V, Io=1A-8A, Fs=600kHz, Room Temperature, No Air Flow
For all the output voltages, L=0.51uH (DCR=0.29 mΩ, P/N: 59PR9876N)
97
95
Efficiency (%)
93
91
89
87
85
83
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
7.0
7.5
8.0
Load Current (A)
0.7V
0.75V
0.9V
1.0V
1.1V
1.2V
1.5V
1.8V
2.5V
3.3V
1.7
1.5
Power Loss (W)
1.3
1.1
0.9
0.7
0.5
0.3
0.1
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
7.0
7.5
8.0
Load Current (A)
0.7V
Rev 8.0
0.75V
0.9V
1.0V
1.1V
1.2V
1.5V
1.8V
2.5V
3.3V
9
PD-97658
IR3841WMPbF
TYPICAL OPERATING CHARACTERISTICS (-40oC - 125oC) Fs=500 kHz
Icc(Standby)
Ic(Dyn)
12.5
290
12.4
12.3
270
12.2
250
[mA]
[uA]
12.1
230
12.0
11.9
210
11.8
190
11.7
170
11.6
11.5
150
-40
-20
0
20
40
60
80
100
120
-40
o
-20
0
20
Temp[ C]
52.0
520
51.0
510
50.0
[uA]
[kHz]
120
80
100
120
80
100
120
80
100
120
80
100
120
53.0
530
500
49.0
48.0
47.0
480
46.0
470
45.0
460
44.0
43.0
450
-40
-20
0
20
40
60
80
100
-40
120
-20
0
20
60
Temp[ C]
Vcc(UVLO) Start
4.46
40
o
o
Temp[ C]
Vcc(UVLO) Stop
4.16
4.41
4.11
4.36
4.06
4.31
4.01
[V]
[V]
100
IOCSET(500kHz)
490
4.26
3.96
4.21
3.91
4.16
3.86
4.11
3.81
4.06
3.76
-40
-20
0
20
40
60
80
100
120
-40
-20
0
20
Temp[ oC]
40
60
Temp[ oC]
Enable(UVLO) Start
1.36
Enable(UVLO) Stop
1.06
1.34
1.04
1.32
1.30
1.02
1.28
1.00
1.26
[V]
[V]
80
54.0
540
1.24
0.98
1.22
0.96
1.20
0.94
1.18
0.92
1.16
1.14
0.90
-40
-20
0
20
40
60
80
100
120
-40
-20
0
20
60
Temp[ C]
ISS
26.0
40
ο
Temp[ oC]
Vfb
711
24.0
706
[mV]
22.0
[uA]
60
o
FREQUENCY
550
40
Temp[ C]
20.0
701
18.0
696
16.0
691
686
14.0
-40
-20
0
20
40
o
60
Temp[ C]
Rev 8.0
80
100
120
-40
-20
0
20
40
60
o
Temp[ C]
10
PD-97658
IR3841WMPbF
Rdson of MOSFETs Over Temperature at Vcc=5V
28
26
Resistance [mΩ]
24
22
20
18
16
14
12
10
8
6
-40
-20
0
20
40
60
80
100
120
140
Temperature [°C]
Sync-FET
Rev 8.0
Ctrl-FET
11
PD-97658
IR3841WMPbF
Circuit Description
THEORY OF OPERATION
Introduction
The IR3841W uses a PWM voltage mode control
scheme with external compensation to provide
good noise immunity and maximum flexibility in
selecting inductor values and capacitor types.
The switching frequency is programmable from
250kHz to 1.5MHz and provides the capability of
optimizing the design in terms of size and
performance.
IR3841W provides precisely regulated output
voltage programmed via two external resistors
from 0.7V to 0.9*Vin.
If the input to the Enable pin is derived from the
bus voltage by a suitably programmed resistive
divider, it can be ensured that the IR3841W does
not turn on until the bus voltage reaches the
desired level. Only after the bus voltage reaches
or exceeds this level will the voltage at Enable
pin exceed its threshold, thus enabling the
IR3841W. Therefore, in addition to being a logic
input pin to enable the IR3841W, the Enable
feature, with its precise threshold, also allows the
user to implement an Under-Voltage Lockout for
the bus voltage Vin. This is desirable particularly
for high output voltage applications, where we
might want the IR3841W to be disabled at least
until Vin exceeds the desired output voltage level.
The IR3841W operates with an external bias
supply from 4.5V to 5.5V, allowing an extended
operating input voltage range from 1.5V to 16V.
The device utilizes the on-resistance of the low
side MOSFET as current sense element, this
method enhances the converter’s efficiency and
reduces cost by eliminating the need for external
current sense resistor.
IR3841W includes two low Rds(on) MOSFETs
using IR’s HEXFET technology. These are
specifically designed for high efficiency
applications.
Under-Voltage Lockout and POR
The under-voltage lockout circuit monitors the
input supply Vcc and the Enable input. It assures
that the MOSFET driver outputs remain in the off
state whenever either of these two signals drop
below the set thresholds. Normal operation
resumes once Vcc and Enable rise above their
thresholds.
The POR (Power On Ready) signal is generated
when all these signals reach the valid logic level
(see system block diagram). When the POR is
asserted the soft start sequence starts (see soft
start section).
Enable
The Enable features another level of flexibility for
start up. The Enable has precise threshold which
is internally monitored by Under-Voltage Lockout
(UVLO) circuit. Therefore, the IR3841W will turn
on only when the voltage at the Enable pin
exceeds this threshold, typically, 1.2V.
Rev 8.0
Fig. 3a. Normal Start up, Device turns on
when the Bus voltage reaches 10.2V
Figure 3b. shows the recommended start-up
sequence for the non-sequenced operation of
IR3841W, when Enable is used as a logic input.
Fig. 3b. Recommended startup sequence,
Non-Sequenced operation
12
PD-97658
IR3841WMPbF
Figure 3c. shows the recommended startup
sequence for sequenced operation of IR3841W
with Enable used as logic input.
Fig. 5. Pre-Bias startup pulses
Soft-Start
Fig. 3c. Recommended startup sequence,
Sequenced operation
Pre-Bias Startup
IR3841W is able to start up into pre-charged
output,
which
prevents
oscillation
and
disturbances of the output voltage.
The output starts in asynchronous fashion and
keeps the synchronous MOSFET off until the first
gate signal for control MOSFET is generated.
Figure 4 shows a typical Pre-Bias condition at
start up.
The IR3841W has a programmable soft-start to
control the output voltage rise and to limit the
current surge at the start-up. To ensure correct
start-up, the soft-start sequence initiates when
the Enable and Vcc rise above their UVLO
thresholds and generate the Power On Ready
(POR) signal. The internal current source
(typically 20uA) charges the external capacitor
Css linearly from 0V to 3V. Figure 6 shows the
waveforms during the soft start.
The start up time can be estimated by:
Tstart =
(1.4 - 0.7) * CSS
20μA
- - - - - - - - - - - - - - - - - - - - (1)
During the soft start the OCP is enabled to
protect the device for any short circuit and over
current condition.
The synchronous MOSFET always starts with a
narrow pulse width and gradually increases its
duty cycle with a step of 25%, 50%, 75% and
100% until it reaches the steady state value. The
number of these startup pulses for the
synchronous MOSFET is internally programmed.
Figure 5 shows a series of 32, 16, 8 startup
pulses.
Fig. 6. Theoretical operation waveforms
during soft-start
Fig. 4. Pre-Bias startup
Rev 8.0
13
PD-97658
IR3841WMPbF
Operating Frequency
The switching frequency can be programmed
between 250kHz – 1500kHz by connecting an
external resistor from Rt pin to Gnd. Table 1
tabulates the oscillator frequency versus Rt.
Table 1. Switching Frequency and IOCSet vs.
External Resistor (Rt)
Rt (kΩ)
47.5
35.7
28.7
23.7
20.5
17.8
15.8
14.3
12.7
11.5
10.7
9.76
9.31
Fs (kHz)
300
400
500
600
700
800
900
1000
1100
1200
1300
1400
1500
Iocset (μA)
29.4
39.2
48.7
59.07
68.2
78.6
88.6
97.9
110.2
121.7
130.8
143.4
150.3
Shutdown
The IR3841W can be shutdown by pulling the
Enable pin below its 1 V threshold. This will tristate both, the high side driver as well as the low
side driver. Alternatively, the output can be
shutdown by pulling the soft-start pin below 0.3V.
Normal operation is resumed by cycling the
voltage at the Soft Start pin.
Over-Current Protection
The over current protection is performed by
sensing current through the RDS(on) of low side
MOSFET. This method enhances the converter’s
efficiency and reduces cost by eliminating a
current sense resistor. As shown in figure 7, an
external resistor (ROCSet) is connected between
OCSet pin and the switch node (SW) which sets
the current limit set point.
An internal current source sources current (IOCSet
) out of the OCSet pin. This current is a function
of the switching frequency and hence, of Rt.
Rev 8.0
I OCSet (μA ) =
1400
.......... .......... ...............( 2)
R t (kΩ)
Table 1. shows IOCSet at different switching
frequencies. The internal current source
develops a voltage across ROCSet. When the low
side MOSFET is turned on, the inductor current
flows through the Q2 and results in a voltage at
OCSet which is given by:
VOCSet = ( IOCSet ∗ ROCSet ) − ( RDS(on) ∗ I L ) ...........(3)
Fig. 7. Connection of over current sensing resistor
An over current is detected if the OCSet pin goes
below ground. Hence, at the current limit
threshold, VOCset=0. Then, for a current limit
setting ILimit, ROCSet is calculated as follows:
ROCSet =
R
DS (on)
* I Limit
IOCSet
........................ (4)
An overcurrent detection trips the OCP
comparator, latches OCP signal and cycles the
soft start function in hiccup mode.
The hiccup is performed by shorting the soft-start
capacitor to ground and counting the number of
switching cycles. The Soft Start pin is held low
until 4096 cycles have been completed. The
OCP signal resets and the converter recovers.
After every soft start cycle, the converter stays in
this mode until the overload or short circuit is
removed.
The OCP circuit starts sampling current typically
160 ns after the low gate drive rises to about 3V.
This delay functions to filter out switching noise.
14
PD-97658
IR3841WMPbF
Thermal Shutdown
Temperature sensing is provided inside
IR3841W. The trip threshold is typically set to
140oC. When trip threshold is exceeded, thermal
shutdown turns off both MOSFETs and
discharges the soft start capacitor.
Automatic restart is initiated when the sensed
temperature drops within the operating range.
There is a 20oC hysteresis in the thermal
shutdown threshold.
Output Voltage Sequencing
The
IR3841W
can
accommodate
user
programmable sequencing options using Seq,
Enable and Power Good pins.
1.5V <Vin<16V
4.5V <Vcc<5.5V
Enable
Vo(master)
Vin
Boot
Vcc
Vo(slave)
SW
PGood
Vo1
RE
PGood
OCSet
Seq
Vo2
RF
Fb
Rt
SS/ SD
RC
RD
Gnd
PGnd
Comp
Simultaneous Powerup
Fig. 8a. Simultaneous Power-up of the slave
with respect to the master.
Through these pins, voltage sequencing such as
simultaneous
and
sequential
can
be
implemented. Figure 8. shows simultaneous
sequencing configurations. In simultaneous
power-up, the voltage at the Seq pin of the slave
reaches 0.7V before the Fb pin of the master. For
RE/RF =RC/RD, therefore, the output voltage of
the slave follows that of the master until the
voltage at the Seq pin of the slave reaches 0.7 V.
After the voltage at the Seq pin of the slave
exceeds 0.85V, the internal 0.7V reference of
the slave dictates its output voltage.
Rev 8.0
Fig. 8b. Application Circuit for Simultaneous
Sequencing
Power Good Output
The IC continually monitors the output voltage via
Feedback (Fb pin). The feedback voltage forms
an input to a window comparator whose upper
and lower thresholds are 0.805V and 0.595V
respectively. Hence, the Power Good signal is
flagged when the Fb pin voltage is within the
PGood window, i. e. between 0.595V to 0.805V,
as shown in Fig .9. The PGood pin is open drain
and it needs to be externally pulled high. High
state indicates that output is in regulation. Fig. 9a
shows the PGood timing diagram for nontracking operation. In this case, during startup,
PGood goes high after the SS voltage reaches
2.1V if the Fb voltage is within the PGood
comparator window. Fig. 9a. and Fig 9.b. also
show a 256 cycle delay between the Fb voltage
entering within the thresholds defined by the
PGood window and PGood going high.
15
PD-97658
IR3841WMPbF
TIMING DIAGRAM OF PGOOD FUNCTION
Fig.9a IR3841W Non-Tracking Operation (Seq=Vcc)
Fig.9b IR3841W Tracking Operation
Rev 8.0
16
PD-97658
IR3841WMPbF
Minimum on time Considerations
Maximum Duty Ratio Considerations
The minimum ON time is the shortest amount of
time for which the Control FET may be reliably
turned on, and this depends on the internal
timing delays. For the IR3841W, the typical
minimum on-time is specified as 50 ns.
Any design or application using the IR3841W
must ensure operation with a pulse width that is
higher than this minimum on-time and preferably
higher than 100 ns. This is necessary for the
circuit to operate without jitter and pulseskipping, which can cause high inductor current
ripple and high output voltage ripple.
A fixed off-time of 200 ns maximum is specified
for the IR3841W. This provides an upper limit on
the operating duty ratio at any given switching
frequency. It is clear, that higher the switching
frequency, the lower is the maximum duty ratio at
which the IR3841W can operate. To allow a
margin of 50ns, the maximum operating duty
ratio in any application using the IR3841W
should still accommodate about 250 ns off-time.
Fig 10. shows a plot of the maximum duty ratio
v/s the switching frequency, with 250 ns off-time.
=
M a x Duty Cycle
D
Fs
Vout
Vin × Fs
In any application that uses the IR3841W, the
following condition must be satisfied:
t on (min) ≤ t on
∴ t on (min) ≤
Vout
t on(min)
The minimum output voltage is limited by the
reference voltage and hence Vout(min) = 0.7 V.
Therefore, for Vout(min) = 0.7 V,
∴ Vin × Fs ≤
95
90
85
80
75
70
65
60
55
250
Vout
Vin × Fs
∴ Vin × Fs ≤
Max D uty C ycle (% )
t on =
450
650
850
1050
1250
1450
1650
S w itching Frequency (kH z )
Fig. 10. Maximum duty cycle v/s switching
frequency.
Vout (min)
∴ Vin × Fs ≤
t on(min)
0.7 V
= 7 × 10 6 V/s
100 ns
Therefore, at the maximum recommended input
voltage 16V and minimum output voltage, the
converter should be designed at a switching
frequency that does not exceed 440 kHz.
Conversely, for operation at the maximum
recommended operating frequency 1.65 MHz
and minimum output voltage, any voltage above
4.2 V may not be stepped down without pulseskipping.
Rev 8.0
17
PD-97658
IR3841WMPbF
When an external resistor divider is connected to
the output as shown in figure 11.
Equation (5) can be rewritten as:
Application Information
Design Example:
The following example is a typical application for
IR3841W. The application circuit is shown on
page 23.
Vin = 12 V ( 13.2V max)
⎛ V
R9 = R8 ∗ ⎜⎜ ref
⎝ V o−Vref
⎞
⎟ ..................................(8)
⎟
⎠
For the calculated values of R8
feedback compensation section.
Vo = 1.8 V
Io = 8 A
VOUT
ΔVo ≤ 54mV
IR3841W
IR3624
Fs = 600 kHz
R9
As explained earlier, the precise threshold of
the Enable lends itself well to implementation of
a UVLO for the Bus Voltage.
V in
IR3841W
Enable
R1
R2
For a typical Enable threshold of VEN = 1.2 V
R2
= VEN = 1.2 .......... (5)
R1 + R 2
VEN
R 2 = R1
.......... (6)
Vin( min ) − VEN
For a Vin (min)=10.2V, R1=49.9K and R2=7.5K is a
good choice.
Programming the frequency
For Fs = 600 kHz, select Rt = 23.7 kΩ, using
Table. 1.
Output Voltage Programming
Output voltage is programmed by reference
voltage and external voltage divider. The Fb pin
is the inverting input of the error amplifier, which
is internally referenced to 0.7V. The divider is
ratioed to provide 0.7V at the Fb pin when the
output is at its desired value. The output voltage
is defined by using the following equation:
⎛ R ⎞
Vo = Vref ∗ ⎜⎜1 + 8 ⎟⎟ ...................................(7)
⎝ R9 ⎠
Rev 8.0
R8
Fb
Enabling the IR3841W
Vin(min) *
and R9 see
Fig. 11. Typical application of the IR3841W for
programming the output voltage
Soft-Start Programming
The soft-start timing can be programmed by
selecting the soft-start capacitance value. From
(1), for a desired start-up time of the converter,
the soft start capacitor can be calculated by
using:
C SS (μF) = Tstart ( ms ) × 0.02857 .......... (9)
Where Tstart is the desired start-up time (ms).
For a start-up time of 3.5ms, the soft-start
capacitor will be 0.099μF. Choose a 0.1μF
ceramic capacitor.
Bootstrap Capacitor Selection
To drive the Control FET, it is necessary to
supply a gate voltage at least 4V greater than
the voltage at the SW pin, which is connected
the source of the Control FET . This is achieved
by using a bootstrap configuration, which
comprises the internal bootstrap diode and an
external bootstrap capacitor (C6) as shown in
Fig. 12. The operation of the circuit is as follows:
When the lower MOSFET is turned on, the
capacitor node connected to SW is pulled down
to ground. The capacitor charges towards Vcc
through the internal bootstrap diode, which has a
forward voltage drop VD. The voltage Vc across
the bootstrap capacitor C6 is approximately
given as
Vc ≅ Vcc − VD .......... .......... ...... (10)
When the upper MOSFET turns on in the next
cycle, the capacitor node connected to SW rises
to the bus voltage Vin. However, if the value of
C6 is appropriately chosen,
18
PD-97658
IR3841WMPbF
the voltage Vc across C6 remains approximately
unchanged and the voltage at the Boot pin
becomes
VBoot ≅ Vin + Vcc − VD ........................................ (11)
Inductor Selection
The inductor is selected based on output power,
operating frequency and efficiency requirements.
A low inductor value causes large ripple current,
resulting in the smaller size, faster response to a
load transient but poor efficiency and high output
noise. Generally, the selection of the inductor
value can be reduced to the desired maximum
ripple current in the inductor (Δi ). The optimum
point is usually found between 20% and 50%
ripple of the output current.
For the buck converter, the inductor value for the
desired operating ripple current can be
determined using the following relation:
Δi
1
; Δt = D ∗
Δt
Fs
.......... .......... .......... . (14)
Vo
L = (Vin − Vo ) ∗
Vin ∗ Δi * Fs
Vin − Vo = L ∗
Fig. 12. Bootstrap circuit to generate
Vc voltage
A bootstrap capacitor of value 0.1uF is suitable
for most applications.
Where:
Vin = Maximum input voltage
Vo = Output Voltage
Input Capacitor Selection
The ripple current generated during the on time of
the upper MOSFET should be provided by the
input capacitor. The RMS value of this ripple is
expressed by:
IRMS = Io ∗ D ∗ (1 − D) ........................(12)
Vo
................................ (13)
Vin
Where:
Δi = Inductor ripple current
F s = Switching frequency
Δt = Turn on time
D = Duty cycle
If Δi ≈ 35%(Io), then the output inductor is
calculated to be 0.91μH. Select L=1 μH.
The MPL105-1R0 from Delta provides a compact,
low profile inductor suitable for this application.
D=
D is the Duty Cycle
IRMS is the RMS value of the input capacitor
current.
Io is the output current.
For Io=8A and D = 0.15, the IRMS = 2.86A.
Ceramic capacitors are recommended due to
their peak current capabilities. They also feature
low ESR and ESL at higher frequency which
enables better efficiency. For this application, it is
advisable to have 4x10uF 16V ceramic capacitors
ECJ-3YX1C106K from Panasonic. In addition to
these, although not mandatory, a 1X330uF, 25V
SMD capacitor EEV-FK1E331P may also be
used as a bulk capacitor and is recommended if
the input power supply is not located close to the
converter.
.
Rev 8.0
19
PD-97658
IR3841WMPbF
Output Capacitor Selection
The voltage ripple and transient requirements
determine the output capacitors type and values.
The criteria is normally based on the value of the
Effective Series Resistance (ESR). However the
actual capacitance value and the Equivalent Series
Inductance (ESL)
are other contributing
components. These components can be described
as
ΔVo = ΔVo( ESR ) + ΔVo( ESL ) + ΔVo(C )
ΔVo(C ) =
⎞
⎟⎟ * ESL
⎠
voltage
ΔI L = Inductor
ripple
2 ∗ π Lo ∗ Co
................................ (16)
Phase
Gain
0 dB
00
-40dB/decade
ripple
current
Since the output capacitor has a major role in the
overall performance of the converter and
determines the result of transient response,
selection of the capacitor is critical. The
IR3841W can perform well with all types of
capacitors.
As a rule, the capacitor must have low enough
ESR to meet output ripple and load transient
requirements.
The goal for this design is to meet the voltage
ripple requirement in the smallest possible
capacitor size. Therefore it is advisable to select
ceramic capacitors due to their low ESR and ESL
and small size. Six of the Panasonic ECJ2FB0J226ML (22uF, 6.3V, 3mOhm) capacitors is
a good choice.
Feedback Compensation
The IR3841W is a voltage mode controller. The
control loop is a single voltage feedback path
including error amplifier and error comparator. To
achieve fast transient response and accurate
output regulation, a compensation circuit is
necessary. The goal of the compensation
network is to provide a closed-loop transfer
function with the highest 0 dB crossing frequency
and adequate phase margin (greater than 45o).
Rev 8.0
1
.......... .......... ..... (15)
ΔI L
8 * C o * Fs
ΔVo = Output
FLC =
Figure 13 shows gain and phase of the LC filter.
Since we already have 180o phase shift from the
output filter alone, the system runs the risk of
being unstable.
ΔVo( ESR ) = ΔI L * ESR
⎛ V − Vo
ΔVo( ESL ) = ⎜⎜ in
L
⎝
The output LC filter introduces a double pole,
–40dB/decade gain slope above its corner
resonant frequency, and a total phase lag of 180o
(see figure 13). The resonant frequency of the LC
filter is expressed as follows:
-1800
FLC
Frequency
FLC Frequency
Fig. 13. Gain and Phase of LC filter
The IR3841W uses a voltage-type error amplifier
with high-gain (110dB) and wide-bandwidth. The
output of the amplifier is available for DC gain
control and AC phase compensation.
The error amplifier can be compensated either in
type II or type III compensation.
Local feedback with Type II compensation is
shown in Fig. 14.
This method requires that the output capacitor
should have enough ESR to generate a zero
typically at 5kHz to 50kHz, which is essential for
an acceptable phase margin and to satisfy
stability requirements.
FESR =
1
.......... .......... ....... (17)
2 ∗ π*ESR*C o
20
PD-97658
IR3841WMPbF
VOUT
Z IN
C POLE
R3
C4
R8
Zf
Fb
R9
Gain(dB)
E/A
Comp
Ve
Fz = 0.75*
FZ
F
POLE
Frequency
Fig. 14. Type II compensation network
and its asymptotic gain plot
The transfer function (Ve/Vo) is given by:
Ve
1 + sR 3C4
Z
= H(s) = − f = −
.....(18)
ZIN
sR 8C4
Vo
The (s) indicates that the transfer function varies
as a function of frequency. This configuration
introduces a gain and zero, expressed by:
Fz =
To cancel one of the LC filter poles, place the
zero before the LC filter resonant frequency pole:
Fz = 75%FLC
VREF
H(s) dB
H(s) =
Where:
Vin = Maximum Input Voltage
Vosc = Oscillator Ramp Voltage
Fo = Crossover Frequency
FESR = Zero Frequency of the Output Capacitor
FLC = Resonant Frequency of the Output Filter
R8 = Feedback Resistor
R3
......................................(19)
R8
1
............................(20)
2π * R3 * C4
First select the desired zero-crossover frequency
(Fo):
Fo > FESR and Fo ≤ (1/5 ~ 1/10) * Fs
Vosc * Fo * FESR * R8
2
Vin * FLC
...........................(21)
FP =
.....................................(22)
1
.................................(23)
C *C
2π * R3 * 4 POLE
C4 + CPOLE
The pole sets to one half of the switching
frequency which results in the capacitor CPOLE:
CPOLE =
1
π*R3*Fs −
1
C4
≅
1
......................(24)
π*R3*Fs
For a general solution for unconditional stability
for any type of output capacitors, and a wide
range of ESR values, we should implement local
feedback with a type III compensation network.
The typically used compensation network for
voltage-mode controller is shown in figure 15.
Again, the transfer function is given by:
Ve
Z
= H(s) = − f
Vo
ZIN
By replacing Zin and Zf according to figure 15,
the transfer function can be expressed as:
H(s) = −
Rev 8.0
2π Lo * Co
Use equations (20), (21) and (22) to calculate
C4.
One more capacitor is sometimes added in
parallel with C4 and R3. This introduces one
more pole which is mainly used to suppress the
switching noise.
The additional pole is given by:
Use the following equation to calculate R3:
R3 =
1
(1 + sR3C4 )[1 + sC7 (R8 + R10 )]
⎡
⎛ C * C3 ⎞⎤
⎟⎥(1 + sR10C7 )
sR8 (C4 + C3 )⎢1 + sR3 ⎜⎜ 4
⎟
⎢⎣
⎝ C4 + C3 ⎠⎥⎦
21
....(25)
PD-97658
IR3841WMPbF
VOUT
ZIN
C3
C7
R3
R10
C4
R8
Zf
Fb
R9
Gain(dB)
E/A
Comp
Ve
VREF
H(s) dB
FZ1
FZ2
FP2
FP3
Frequency
Fig.15. Type III Compensation network and
its asymptotic gain plot
The compensation network has three poles and
two zeros and they are expressed as follows:
FP1 = 0 ..................................................................(26)
FP 2 =
1
...............................................(27)
2π * R10 * C7
1
FP3 =
FZ1
≅
1
...............(28)
2π * R3 * C3
⎛ C * C3 ⎞
⎟
2π * R3 ⎜⎜ 4
⎟
⎝ C4 + C3 ⎠
1
=
.............................................(29)
2π * R3 * C4
FZ 2 =
1
1
≅
..........(30)
2π * C7 * (R8 + R10 ) 2π * C7 * R8
Cross over frequency is expressed as:
Fo = R3 * C7 *
Vin
1
*
................................(31)
Vosc 2π * Lo * Co
Based on the frequency of the zero generated by
the output capacitor and its ESR, relative to
crossover frequency, the compensation type can
be different. The table below shows the
compensation types and location of the
crossover frequency.
Rev 8.0
Compensator
Type
FESR vs Fo
Output
Capacitor
Type II
FLC<FESR<Fo<Fs/2
Electrolytic
Tantalum
Type III
FLC<Fo<FESR
Tantalum
Ceramic
The higher the crossover frequency, the
potentially faster the load transient response.
However, the crossover frequency should be low
enough to allow attenuation of switching noise.
Typically, the control loop bandwidth or crossover
frequency is selected such that
Fo ≤ (1/5 ~ 1/10) * Fs
The DC gain should be large enough to provide
high DC-regulation accuracy. The phase margin
should be greater than 45o for overall stability.
For this design we have:
Vin=12V
Vo=1.8V
Vosc=1.8V
Vref=0.7V
Lo=1uH
Co=6x22uF, ESR=3mOhm each
It must be noted here that the value of the
capacitance used in the compensator design
must be the small signal value. For instance, the
small signal capacitance of the 22uF capacitor
used in this design is 12uF at 1.8 V DC bias and
600 kHz frequency. It is this value that must be
used for all computations related to the
compensation. The small signal value may be
obtained from the manufacturer’s datasheets,
design tools or SPICE models. Alternatively, they
may also be inferred from measuring the power
stage transfer function of the converter and
measuring the double pole frequency FLC and
using equation (16) to compute the small signal
Co.
These result to:
FLC=18.76 kHz
FESR=4.4 MHz
Fs/2=300 kHz
Select crossover frequency: Fo=100 kHz
Since FLC<Fo<Fs/2<FESR, TypeIII is selected to
place the pole and zeros.
22
PD-97658
IR3841WMPbF
Detailed calculation of compensation TypeIII
Desired Phase Margin Θ = 70o
FZ2 = Fo
1 − sin Θ
= 17.63kHz
1 + sin Θ
FP2 = Fo
1 + sin Θ
= 567.1kHz
1 − sin Θ
Programming the Current-Limit
The Current-Limit threshold can be set by
connecting a resistor (ROCSET) from the SW pin
to the OCSet pin. The resistor can be calculated
by using equation (4). This resistor ROCSET
must be placed close to the IC.
The RDS(on) has a positive temperature
coefficient and it should be considered for the
worst case operation.
ISET = IL (critical) =
Select: FZ1 = 0.5 * FZ2 = 8.82 kHz and
FP3 = 0.5* Fs = 300 kHz
ROCSet∗ IOCSet
.......................(32)
RDS(on)
RDS( on) = 8.5 mΩ *1.25 = 10.625 mΩ
ISET ≅ Io( LIM) = 8 A *1.5 = 12 A
(50% over nominal output current )
Select: C7 = 2.2nF
IOCSet = 59.07 μA (at Fs = 600kHz)
ROCSet = 2.15 kΩ Select R7 = 2.15 kΩ
Calculate R3, C3 and C4 :
2π * Fo * Lo * Co * Vosc
;R3 = 3.05 kΩ
C7 * Vin
R3 =
Select: R3 = 3.01kΩ
C4 =
1
; C4 = 6 nF, Select: C4 = 10 nF
2π * FZ1 * R 3
C3 =
1
; C3 = 176.25 pF, Select: C3 = 150 pF
2π * FP3 * R3
Setting the Power Good Threshold
A window comparator internally sets a lower
Power Good threshold at 0.6V and an upper
Power Good threshold at 0.8V. When the voltage
at the FB pin is within the window set by these
thresholds, PGood is asserted.
The PGood is an open drain output. Hence, it is
necessary to use a pull up resistor RPG from
PGood pin to Vcc. The value of the pull-up
resistor must be chosen such as to limit the
current flowing into the PGood pin, when the
output voltage is not in regulation, to less than 5
mA. A typical value used is 10kΩ.
Calculate R10, R8 and R9 :
R10 =
1
; R10 = 128 Ω, Select: R10 = 130 Ω
2π * C7 * FP2
R8 =
1
- R10; R8 = 3.97 kΩ,
2π * C7 * FZ2
Select: R8 = 4.02 kΩ
R9 =
Vref
* R8; R9 = 2.56 kΩ Select: R9 = 2.55 kΩ
Vo - Vref
Rev 8.0
23
PD-97658
IR3841WMPbF
Application Diagram:
Fig. 16. Application circuit diagram for a 12V to 1.8 V, 8 A Point Of Load Converter
Suggested Bill of Materials for the application circuit:
Part Reference
Cin
Lo
Co
R1
R2
Rt
Rocset
RPG
Css
R3
C3
C4
R8
R9
R10
C7
CVcc
U1
Rev 8.0
Quantity
1
4
1
1
6
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
Value
330uF
10uF
0.1uF
1.0uH
22uF
49.9k
7.5k
23.7k
2.15k
10k
0.1uF
3.01k
150pF
10nF
4.02k
2.55k
130
2200pF
1.0uF
IR3841W
Description
SMD Elecrolytic, Fsize, 25V, 20%
1206, 16V, X5R, 20%
0603, 25V, X7R, 10%
11.5x10x5mm, 20%, 2.3mOhm
0805, 6.3V, X5R, 20%
Thick Film, 0603,1/10 W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
0603, 25V, X7R, 10%
Thick Film, 0603,1/10W,1%
50V, 0603, NPO, 5%
0603, 50V, X7R, 10%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
0603, 50V, X7R, 10%
0603, 16V, X5R, 20%
SupIRBuck, 8A, PQFN 5x6mm
Manufacturer
Panasonic
TDK
Panasonic
Delta
Panasonic
Rohm
Rohm
Rohm
Rohm
Rohm
Panasonic
Rohm
Panasonic
Panasonic
Rohm
Rohm
Rohm
Panasonic
Panasonic
International Rectifier
Part Number
EEV-FK1E331P
C3216X5R1E106M
ECJ-1VB1E104K
MPL105-1R0
ECJ-2FB0J226ML
MCR03EZPFX4992
MCR03EZPFX7501
MCR03EZPFX2372
MCR03EZPFX2151
MCR03EZPFX1002
ECJ-1VB1E104K
MCR03EZPFX3011
ECJ-1VC1H151J
ECJ-1VB1H103K
MCR03EZPFX4021
MCR03EZPFX2551
ERJ-3EKF1300V
ECJ-1VB1H222K
ECJ-BVB1C105M
IR3841WMPbF
24
PD-97658
IR3841WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12.0V, Vcc=5V, Vo=1.8V, Io=0-8A, Room Temperature, No Air Flow
Fig. 17: Start up at 8A Load
Ch1:Vin, Ch2:Vout, Ch3:Vss, Ch4:Enable
Fig. 18: Start up at 8A Load,
Ch1:Vin, Ch2:Vout, Ch3:Vss, Ch4:VPGood
Fig. 19: Start up with 1.62V Pre
Bias, 0A Load, Ch2:Vout, Ch3:VSS
Fig. 20: Output Voltage Ripple, 8A
load Ch2: Vout
Fig. 21: Inductor node at 8A load
Ch3:LX
Fig. 22: Short (Hiccup) Recovery
Ch2:Vout , Ch3:Vss
Rev 8.0
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PD-97658
IR3841WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12V, Vcc=5V, Vo=1.8V, Io=4-8A, Room Temperature, No Air Flow
Fig. 23: Transient Response, 4A to 8A step 2.5A/us
Ch2:Vout, Ch4:Iout
Rev 8.0
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PD-97658
IR3841WMPbF
TYPICAL OPERATING WAVEFORMS
Vin=12V, Vcc=5V, Vo=1.8V, Io=8A, Room Temperature, No Air Flow
Fig. 24: Bode Plot at 8A load shows a bandwidth of 91kHz and phase margin of 55 degrees
Rev 8.0
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PD-97658
IR3841WMPbF
Simultaneous Tracking at Power Up and Power Down
Vin=12V, Vo=1.8V, Io=8A, Room Temperature, No Air Flow
VOUT
3.3V
4.02K
R s1
2..55K
Rs2
IR3841W
IR3624
Seq
R8 4.02K
Fb
R9 2.55K
Fig. 25: Simultaneous Tracking a 3.3V input at power-up and shut-down
Ch2: Vout Ch3:SS Ch4: SEQ
Rev 8.0
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PD-97658
IR3841WMPbF
Layout Considerations
The layout is very important when designing high
frequency switching converters. Layout will affect
noise pickup and can cause a good design to
perform with less than expected results.
Make all the connections for the power
components in the top layer with wide, copper
filled areas or polygons. In general, it is desirable
to make proper use of power planes and
polygons for power distribution and heat
dissipation.
The inductor, output capacitors and the IR3841W
should be as close to each other as possible.
This helps to reduce the EMI radiated by the
power traces due to the high switching currents
through them. Place the input capacitor directly at
the Vin pin of IR3841W.
The feedback part of the system should be kept
away from the inductor and other noise sources.
The critical bypass components such as
capacitors for Vcc should be close to their
respective pins. It is important to place the
feedback components including feedback
resistors and compensation components close to
Fb and Comp pins.
The connection between the OCSet resistor and
the Sw pin should not share any trace with the
connection between the bootstrap capacitor and
the Sw pin. Instead, it is recommended to use a
Kelvin connection of the trace from the OCSet
resistor and the trace from the bootstrap
Vin
capacitor at the Sw pin. PGnd
In a multilayer PCB use one layer as a power
ground plane and have a control circuit ground
(analog ground), to which all signals are
referenced. The goal is to localize the high
current path to a separate loop that does not
Vout
interfere
with the more sensitive
analog control
AGnd
function. These two grounds must be connected
together on the PC board layout at a single point.
The Power QFN is a thermally enhanced
package. Based on thermal performance it is
recommended to use at least a 4-layers PCB. To
effectively remove heat from the device the
exposed pad should be connected to the ground
plane using vias. Figure 26 illustrates the
implementation of the layout guidelines outlined
above, on the IRDC3841W 4 layer demoboard.
Vin
PGnd
Vout
AGnd
Enough copper &
minimum length
ground path between
Input and Output
PGnd
Compensation parts
should be placed as close
as possible to
the Comp pin.
Vin
Vin
PGnd
All bypass caps
should be placed as
close as possible to
their connecting
pins.
AGnd
Resistors Rt and
Rocset should be
placed as close as
possible to their pins.
AGnd
VoutVout
Fig.26a. IRDC3841W demoboard layout
considerations – Top Layer
Rev 8.0
29
PD-97658
IR3841WMPbF
PGnd
Feedback
trace should
be kept
away form
noise
sources
Fig.26b. IRDC3841W demoboard layout
considerations – Bottom Layer
Analog
Ground
plane
Power
Vin
Ground
Plane
Single point
connection between
AGND & PGND,
should be close to
the SupIRBuck, kept
away from noise
sources.
AGnd
Fig.26c. IRDC3841W demoboard layout
considerations – Mid Layer 1
Use separate traces
for connecting Boot
cap and Rocset to the
switch node and with
the minimum length
traces. Avoid big
loops.
Fig.26d. IRDC3841W demoboard layout
considerations – Mid Layer 2
Rev 8.0
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PD-97658
IR3841WMPbF
PCB Metal and Components Placement
Lead lands (the 11 IC pins) width should be equal to nominal part lead width. The
minimum lead to lead spacing should be ≥ 0.2mm to minimize shorting.
Lead land length should be equal to maximum part lead length + 0.3 mm outboard
extension. The outboard extension ensures a large and inspectable toe fillet.
Pad lands (the 4 big pads other than the 11 IC pins) length and width should be equal to
maximum part pad length and width. However, the minimum metal to metal spacing
should be no less than 0.17mm for 2 oz. Copper; no less than 0.1mm for 1 oz. Copper
and no less than 0.23mm for 3 oz. Copper.
Rev 8.0
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PD-97658
IR3841WMPbF
Solder Resist
It is recommended that the lead lands are Non Solder Mask Defined (NSMD). The solder
resist should be pulled away from the metal lead lands by a minimum of 0.025mm to ensure
NSMD pads.
The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder
resist onto the copper of 0.05mm to accommodate solder resist mis-alignment.
Ensure that the solder resist in-between the lead lands and the pad land is ≥ 0.15mm due
to the high aspect ratio of the solder resist strip separating the lead lands from the pad land.
Rev 8.0
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PD-97658
IR3841WMPbF
Stencil Design
•
•
The Stencil apertures for the lead lands should be approximately 80% of the area of
the lead lads. Reducing the amount of solder deposited will minimize the
occurrences of lead shorts. If too much solder is deposited on the center pad the part
will float and the lead lands will be open.
The maximum length and width of the land pad stencil aperture should be equal to
the solder resist opening minus an annular 0.2mm pull back to decrease the
incidence of shorting the center land to the lead lands when the part is pushed into
the solder paste.
Rev 8.0
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PD-97658
IR3841WMPbF
BOTTOM VIEW
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105
TAC Fax: (310) 252-7903
This product has been designed and qualified for the Industrial market (Note5)
Visit us at www.irf.com for sales contact information
Data and specifications subject to change without notice. 08/12
Rev 8.0
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