Intersil HIP6021EVAL1 Advanced pwm and triple linear power controller Datasheet

HIP6021A
TM
Data Sheet
December 2001
Advanced PWM and Triple Linear Power
Controller
The HIP6021A provides the power control and protection for
four output voltages in high-performance, graphics intensive
microprocessor and computer applications. The IC
integrates a voltage-mode PWM controller and three linear
controllers, as well as the monitoring and protection
functions into a 28 lead SOIC package.
The synchronous-rectified buck converter includes an Intelcompatible, TTL 5-input digital-to-analog converter (DAC)
that adjusts the core PWM output voltage from 1.3VDC to
2.05VDC in 0.05V steps and from 2.1VDC to 3.5VDC in 0.1V
increments. The precision reference and voltage-mode
control provide ±1% static regulation. A TTL-compatible
signal applied to the SELECT pin dictates which method of
control is used for the AGP bus power: a low state results in
linear control of the AGP bus to 1.5V, while a high state
transitions the output through a linearly controlled softstart to
3.3V, followed by full enhancement of the external MOSFET
to pass the input voltage. The other two linear regulators
provide fixed output voltages of 1.5V GTL bus power and
1.8V power for the North/South Bridge core and/or cache
memory. These levels are user-adjustable by means of an
external resistor divider and pulling the FIX pin low. All linear
controllers can employ either N-Channel MOSFETs or
bipolar NPNs for the pass transistor.
The HIP6021A monitors all the output voltages. A single
Power Good signal is issued when the core is within ±10% of
the DAC setting and all other outputs are above their undervoltage levels. Additional built-in over-voltage protection for
the core output uses the lower MOSFET to prevent output
voltages above 115% of the DAC setting. The PWM
controller’s over-current function monitors the output current
by using the voltage drop across the upper MOSFET’s
rDS(ON).
Ordering Information
PART NUMBER
HIP6021ACB
HIP6021EVAL1
TEMP.
RANGE (oC)
0 to 70
PACKAGE
28 Ld SOIC
Evaluation Board
PKG.
NO.
M28.3
Features
• Provides 4 Regulated Voltages
- Microprocessor Core, AGP Bus, Memory, and GTL Bus
Power
• Drives N-Channel MOSFETs
• Linear Regulator Drives Compatible with both MOSFET
and Bipolar Series Pass Transistors
• Fixed or Externally Resistor-Adjustable Linear Outputs
• Simple Single-Loop Control Design
- Voltage-Mode PWM Control
• Fast PWM Converter Transient Response
- High-Bandwidth Error Amplifier
- Full 0% to 100% Duty Ratio
• Excellent Output Voltage Regulation
- Core PWM Output: ±1% Over Temperature
- Other Outputs: ±3% Over Temperature
• TTL-Compatible 5-Bit DAC Core Output Voltage Selection
- Shutdown Feature Removed When All Inputs High
- Wide Range 1.3VDC to 3.5VDC
• Power-Good Output Voltage Monitor
• Over-Voltage and Over-Current Fault Monitors
- Switching Regulator Does Not Require Extra Current
Sensing Element, Uses Upper MOSFET’s rDS(ON)
• Small Converter Size
- Constant Frequency Operation
- 200kHz Free-Running Oscillator; Programmable From
50kHz to Over 1MHz
- Small External Component Count
Applications
• Motherboard Power Regulation for Computers
Pinout
HIP6021A (SOIC)
TOP VIEW
DRIVE2 1
28 VCC
FIX 2
27 UGATE
VID4 3
26 PHASE
VID3 4
25 LGATE
VID2 5
24 PGND
VID1 6
23 OCSET
VID0 7
22 VSEN1
PGOOD 8
21 FB
SD 9
20 COMP
VSEN2 10
19 VSEN3
SELECT 11
SS 12
FAULT/RT 13
VSEN4 14
1
FN4793.1
18 DRIVE3
17 GND
16 VAUX
15 DRIVE4
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2001. All Rights Reserved
2
SELECT
VSEN2
DRIVE2
-
+
+
-
DRIVE4
VSEN4
+
x 0.75
1.5V
OR
3.3VIN
+
DRIVE3
+
VAUX
-
+
1.26V
x 0.75
LUV
FIX
SD
FAULT / RT
SS
OV
28μA
VCC
SOFTINHIBIT
START
AND FAULT
FAULT
LOGIC
LINEAR
UNDERVOLTAGE
OSCILLATOR
+
-
+
-
VSEN3
4.5V
DACOUT
FB
ERROR
AMP1
x 1.15
x 0.90
x 1.10
+
-
+
-
+
-
+
-
VSEN1
COMP
OC1
PWM1
VID1
POWER-ON
VCC
SYNCH
DRIVE
GATE
CONTROL
DRIVE1
RESET (POR)
VID4
VID3
VID2
TTL D/A
CONVERTER
(DAC)
PWM
COMP1
VID0
+
-
+
-
200μA
OCSET
VCC
VCC
GND
PGND
LGATE
PHASE
UGATE
PGOOD
VAUX
HIP6021A
Block Diagram
HIP6021A
Simplified Power System Diagram
+5VIN
+3.3VIN
Q1
LINEAR
CONTROLLER
Q3
VOUT1
PWM
CONTROLLER
Q2
VOUT2
HIP6021A
Q4
VOUT3
LINEAR
CONTROLLER
LINEAR
CONTROLLER
Q5
VOUT4
Typical Application
+12VIN
+5VIN
LIN
CIN
VCC
OCSET
+3.3VIN
POWERGOOD
PGOOD
Q3
VOUT2
DRIVE2
1.5V OR 3.3VIN
UGATE
VSEN2
COUT2
LGATE
PGND
SELECT
TYPEDET
VSEN1
VAUX
HIP6021A
Q4
VOUT3
1.5V
DRIVE3
FB
COMP
VSEN3
COUT3
FIX
FAULT / RT
VID0
DRIVE4
Q5
VOUT4
1.8V
VID1
VID2
VSEN4
VID3
SS
COUT4
VID4
CSS
GND
3
Q1
LOUT1
PHASE
Q2
COUT1
VOUT1
1.3V TO 3.5V
HIP6021A
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +15V
PGOOD, RT/FAULT, DRIVE, PHASE,
and GATE Voltage . . . . . . . . . . . . . . . GND - 0.3V to VCC + 0.3V
Input, Output or I/O Voltage . . . . . . . . . . . . . . . . . . GND -0.3V to 7V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 1
Thermal Resistance (Typical, Note 1)
θJA (oC/W)
SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . .
70
Maximum Junction Temperature (Plastic Package) . . . . . . . 150oC
Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC
(SOIC - Lead Tips Only)
Operating Conditions
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10%
Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . 0oC to 70oC
Junction Temperature Range . . . . . . . . . . . . . . . . . . 0oC to 125oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on an evaluation PC board in free air.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System
Diagrams, and Typical Application Schematic
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
UGATE, LGATE, DRIVE2, DRIVE3, and
DRIVE4 Open
-
9
-
mA
Rising VCC Threshold
VOCSET = 4.5V
-
-
10.4
V
Falling VCC Threshold
VOCSET = 4.5V
8.2
-
-
V
Rising VAUX Threshold
VOCSET = 4.5V
-
2.5
-
V
VAUX Threshold Hysteresis
VOCSET = 4.5V
-
0.5
-
V
-
1.26
-
V
RT = OPEN
185
200
215
kHz
6kΩ < RT to GND < 200kΩ
-15
-
+15
%
-
1.9
-
VP-P
DAC(VID0-VID4) Input Low Voltage
-
-
0.8
V
DAC(VID0-VID4) Input High Voltage
2.0
-
-
V
DACOUT Voltage Accuracy
-1.0
-
+1.0
%
-
1.265
-
V
-2.5
-
+2.5
%
Except OUT2 when SELECT > 2.0V
-
3
-
%
SELECT < 0.8V
-
1.5
-
V
VCC SUPPLY CURRENT
Nominal Supply Current
ICC
POWER-ON RESET
Rising VOCSET Threshold
OSCILLATOR
Free Running Frequency
FOSC
Total Variation
ΔVOSC
Ramp Amplitude
RT = Open
DAC AND BANDGAP REFERENCE
Bandgap Reference Voltage
VBG
Bandgap Reference Tolerance
LINEAR REGULATORS (OUT2, OUT3, AND OUT4)
Regulation (All Linears)
VSEN2 Regulation Voltage
VREG2
VSEN3 Regulation Voltage
VREG3
-
1.5
-
V
VSEN4 Regulation Voltage
VREG4
-
1.8
-
V
VSEN Rising
-
75
-
%
Under-Voltage Hysteresis (VSEN/VREG)
VSEN Falling
-
7
-
%
Output Drive Current (All Linears)
VAUX-VDRIVE > 0.6V
20
40
-
mA
Under-Voltage Level (VSEN/VREG)
4
VSENUV
HIP6021A
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System
Diagrams, and Typical Application Schematic (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
-
88
-
dB
-
15
-
MHz
COMP = 10pF
-
6
-
V/μs
SYNCHRONOUS PWM CONTROLLER ERROR AMPLIFIER
DC Gain
Gain-Bandwidth Product
GBWP
Slew Rate
SR
PWM CONTROLLER GATE DRIVER
UGATE Source
IUGATE
VCC = 12V, VUGATE = 6V
-
1
-
A
UGATE Sink
RUGATE
VGATE-PHASE = 1V
-
1.7
3.5
Ω
LGATE Source
ILGATE
VCC = 12V, VLGATE = 1V
-
1
-
A
LGATE Sink
RLGATE
VLGATE = 1V
-
1.4
3.0
Ω
VSEN1 Rising
-
115
120
%
PROTECTION
VSEN1 Over-Voltage (VSEN1/DACOUT)
FAULT Sourcing Current
IOVP
VFAULT/RT = 2.0V
-
8.5
-
mA
OCSET1 Current Source
IOCSET
VOCSET = 4.5VDC
170
200
230
μA
-
28
-
μA
Soft-Start Current
ISS
POWER GOOD
VSEN1 Upper Threshold
(VSEN1/DACOUT)
VSEN1 Rising
108
-
110
%
VSEN1 Under-Voltage
(VSEN1/DACOUT)
VSEN1 Rising
92
-
94
%
VSEN1 Hysteresis (VSEN1/DACOUT)
Upper/Lower Threshold
-
2
-
%
IPGOOD = -4mA
-
-
0.8
V
PGOOD Voltage Low
VPGOOD
Typical Performance Curve
RESISTANCE (kΩ)
1000
RT PULLUP
TO +12V
100
10
RT PULLDOWN TO VSS
10
100
SWITCHING FREQUENCY (kHz)
FIGURE 1. RT RESISTANCE vs FREQUENCY
5
1000
HIP6021A
Functional Pin Descriptions
VCC (Pin 28)
Provide a 12V bias supply for the IC to this pin. This pin also
provides the gate bias charge for all the MOSFETs
controlled by the IC. The voltage at this pin is monitored for
Power-On Reset (POR) purposes.
GND (Pin 17)
Signal ground for the IC. All voltage levels are measured
with respect to this pin.
PGND (Pin 24)
This is the power ground connection. Tie the synchronous
PWM converter’s lower MOSFET source to this pin.
VAUX (Pin 16)
This pin provides boost current for the linear regulators’
output drives in the event bipolar NPN transistors (instead
of N-Channel MOSFETs) are employed as pass elements.
The voltage at this pin is monitored for power-on reset
(POR) purposes.
SS (Pin 12)
Connect a capacitor from this pin to ground. This capacitor,
along with an internal 28μA current source, sets the softstart
interval of the converter.
FAULT / RT (Pin 13)
This pin provides oscillator switching frequency adjustment.
By placing a resistor (RT) from this pin to GND, the nominal
200kHz switching frequency is increased according to the
following equation:
6
5 × 10
Fs ≈ 200kHz + --------------------R T ( kΩ )
(RT to GND)
Conversely, connecting a resistor from this pin to VCC
reduces the switching frequency according to the following
equation:
7
4 × 10
Fs ≈ 200kHz – --------------------R T ( kΩ )
(RT to 12V)
Nominally, the voltage at this pin is 1.26V. In the event of an
over-voltage or over-current condition, this pin is internally
pulled to VCC.
PGOOD (Pin 8)
the soft-start capacitor, disabling all the outputs. Dedicated
internal circuitry insures the core output voltage does not go
negative during this process. When re-enabled, the IC
undergoes a new soft-start cycle. Left open, this pin is pulled
low by an internal pull-down resistor, enabling operation.
FIX (Pin 2)
Grounding this pin bypasses the internal resistor dividers
that set the output voltage of the 1.5V and 1.8V linear
regulators. This way, the output voltage of the two regulators
can be adjusted from 1.26V up to the input voltage (+3.3V or
+5V) by way of an external resistor divider connected at the
corresponding VSEN pin. The new output voltage set by the
external resistor divider can be determined using the
following formula:
R OUT ⎞
⎛
V OUT = 1.265V × ⎜ 1 + -----------------⎟
R GND⎠
⎝
where ROUT is the resistor connected from VSEN to the
output of the regulator, and RGND is the resistor connected
from VSEN to ground. Left open, the FIX pin is pulled high,
enabling fixed output voltage operation.
VID0, VID1, VID2, VID3, VID4 (Pins 7, 6, 5, 4 and 3)
VID0-4 are the TTL-compatible input pins to the 5-bit DAC.
The logic states of these five pins program the internal
voltage reference (DACOUT). The level of DACOUT sets the
microprocessor core converter output voltage, as well as the
corresponding PGOOD and OVP thresholds.
OCSET (Pin 23)
Connect a resistor from this pin to the drain of the respective
upper MOSFET. This resistor, an internal 200μA current
source, and the upper MOSFET’s on-resistance set the
converter over-current trip point. An over-current trip cycles
the soft-start function.
The voltage at this pin is monitored for power-on reset
(POR) purposes and pulling this pin low with an open drain
device will shutdown the IC.
PHASE (Pin 26)
Connect the PHASE pin to the PWM converter’s upper
MOSFET source. This pin represents the gate drive return
current path and is used to monitor the voltage drop across
the upper MOSFET for over-current protection.
PGOOD is an open collector output used to indicate the
status of the output voltages. This pin is pulled low when the
synchronous regulator output is not within ±10% of the
DACOUT reference voltage or when any of the other outputs
are below their under-voltage thresholds.
UGATE (Pin 27)
The PGOOD output is open for ‘11111’ VID code.
Connect LGATE to the PWM converter’s lower MOSFET
gate. This pin provides the gate drive for the lower MOSFET.
SD (Pin 9)
This pin shuts down all the outputs. A TTL-compatible, logic
level high signal applied at this pin immediately discharges
6
Connect UGATE pin to the PWM converter’s upper
MOSFET gate. This pin provides the gate drive for the upper
MOSFET.
LGATE (Pin 25)
HIP6021A
COMP and FB (Pin 20 and 21)
COMP and FB are the available external pins of the PWM
converter error amplifier. The FB pin is the inverting input of the
error amplifier. Similarly, the COMP pin is the error amplifier
output. These pins are used to compensate the voltage-mode
control feedback loop of the synchronous PWM converter.
VSEN1 (Pin 22)
This pin is connected to the PWM converter’s output voltage.
The PGOOD and OVP comparator circuits use this signal to
report output voltage status and for over-voltage protection.
DRIVE2 (Pin 1)
Connect this pin to the gate of an external MOSFET. This
pin provides the drive for the AGP regulator’s pass
transistor.
VSEN2 (Pin 10)
Connect this pin to the output of the AGP linear regulator.
The voltage at this pin is regulated to the level
predetermined by the logic-level status of the SELECT pin.
This pin is also monitored for under-voltage events.
SELECT (Pin 11)
This pin determines the output voltage of the AGP bus linear
regulator. A low TTL input sets the output voltage to 1.5V
and the linear controller regulates this voltage to within ±3%.
A TTL high input turns Q3 on continuously, providing a DC
current path from the input (+3.3VIN) to the output (VOUT2)
of the AGP controller.
DRIVE3 (Pin 18)
Connect this pin to the gate of an external MOSFET. This
pin provides the drive for the 1.5V regulator’s pass
transistor.
VSEN3 (Pin 19)
Connect this pin to the output of the 1.5V linear regulator.
This pin is monitored for under-voltage events.
DRIVE4 (Pin 15)
Connect this pin to the gate of an external MOSFET. This
pin provides the drive for the 1.8V regulator’s pass
transistor.
VSEN4 (Pin 14)
Connect this pin to the output of the linear 1.8V regulator.
This pin is monitored for under voltage events.
Description
Operation
The HIP6021A monitors and precisely controls 4 output
voltage levels (Refer to Block and Simplified Power System
Diagrams, and Typical Application Schematic). It is
designed for microprocessor computer applications with
3.3V, 5V, and 12V bias input from an ATX power supply.
7
The microprocessor core voltage (VOUT1) is controlled in a
synchronous-rectified buck converter configuration. The
PWM controller regulates the microprocessor core voltage
to a level programmed by the 5-bit digital-to-analog
converter (DAC).
The AGP bus voltage (VOUT2) is set using the SELECT
pin to either a 1.5V linear regulated output or to the 3.3VIN
through a pass device. Selection of either output voltage is
set depending on the logic level of the SELECT pin.
The two remaining linear controllers supply the 1.5V GTL
bus power (VOUT3) and the 1.8V memory power (VOUT4).
These output voltages are user adjustable. All linear
controllers are designed to employ an external pass
transistor.
Initialization
The HIP6021A automatically initializes upon receipt of input
power. Special sequencing of the input supplies is not
necessary. The Power-On Reset (POR) function continually
monitors the input supply voltages. The POR monitors the
bias voltage (+12VIN) at the VCC pin, the 5V input voltage
(+5VIN) on the OCSET pin, and the 3.3V input voltage
(+3.3VIN) at the VAUX pin. The normal level on OCSET is
equal to +5VIN less a fixed voltage drop (see over-current
protection). The POR function initiates soft-start operation
after all supply voltages exceed their POR thresholds.
Soft-Start
The POR function initiates the soft-start sequence. Initially,
the voltage on the SS pin rapidly increases to approximately
1V (this minimizes the softstart interval). Then an internal
28μA current source charges an external capacitor (CSS) on
the SS pin to 4.5V. The PWM error amplifier reference input
(+ terminal) and output (COMP pin) are clamped to a level
proportional to the SS pin voltage. As the SS pin voltage
slews from 1V to 4V, the output clamp allows generation of
PHASE pulses of increasing width that charge the output
capacitor(s). After the output voltage increases to
approximately 70% of the set value, the reference input
clamp slows the output voltage rate-of-rise and provides a
smooth transition to the final set voltage. Additionally, all
linear regulators’ reference inputs are clamped to a voltage
proportional to the SS pin voltage. This method provides a
rapid and controlled output voltage rise.
Figure 2 shows the soft-start sequence for the typical
application. At T0 the SS voltage rapidly increases to
approximately 1V. At T1, the SS pin and error amplifier
output voltage reach the valley of the oscillator’s triangle
wave. The oscillator’s triangular waveform is compared to
the clamped error amplifier output voltage. As the SS pin
voltage increases, the pulse width on the PHASE pin
increases. The interval of increasing pulse width continues
until each output reaches sufficient voltage to transfer
control to the input reference clamp. If we consider the
2.5V core output (VOUT1) in Figure 2, this time occurs at
HIP6021A
T2. During the interval between T2 and T3, the error
amplifier reference ramps to the final value and the
converter regulates the output a voltage proportional to the
SS pin voltage. At T3 the input clamp voltage exceeds the
reference voltage and the output voltage is in regulation.
linear output (VSEN2, VSEN3, or VSEN4) is ignored until
after the soft-start interval (T4 in Figure 2). This allows
VOUT2 , VOUT3 , and VOUT4 to increase without fault at startup. Cycling the bias input voltage (+12VIN on the VCC pin off
then on) resets the counter and the fault latch.
LUV
OVERCURRENT
LATCH
PGOOD
0V
S Q
OC1
SOFT-START
(1V/DIV)
R
0.15V
VOUT2 (= 3.3VIN)
0V
SS
+
+
4V
COUNTER
-
-
R
VOUT1 (DAC = 2.5V)
VOUT4 (= 1.8V)
OUTPUT
VOLTAGES
(0.5V/DIV)
VOUT3 (= 1.5V)
0V
T2
TIME
T3
T4
FIGURE 2. SOFT-START INTERVAL
The remaining outputs are also programmed to follow the
SS pin voltage. The PGOOD signal toggles ‘high’ when all
output voltage levels have exceeded their under-voltage
levels. The waveform for VOUT2 represents the case where
SELECT is held ‘high’. The AGP bus voltage is controlled in
the same manner as the other linear regulators during the
softstart sequence. Once the softstart sequence is
complete (T4), the gate of the external pass device is fully
enhanced and VOUT2 tracks the 3.3VIN voltage. See the
Soft-Start Interval section under Applications Guidelines for
a procedure to determine the soft-start interval.
Fault Protection
All four outputs are monitored and protected against extreme
overload. A sustained overload on any output or an overvoltage on VOUT1 output (VSEN1) disables all outputs and
drives the FAULT/RT pin to VCC.
Figure 3 shows a simplified schematic of the fault logic. An
over-voltage detected on VSEN1 immediately sets the fault
latch. A sequence of three over-current fault signals also
sets the fault latch. The over-current latch is set dependent
upon the states of the over-current (OC), linear undervoltage (LUV) and the soft-start signals. A window
comparator monitors the SS pin and indicates when CSS is
fully charged to 4V (UP signal). An under-voltage on either
8
FAULT
LATCH
VCC
S Q
UP
POR
T0 T1
INHIBIT
R
FAULT
OV
FIGURE 3. FAULT LOGIC - SIMPLIFIED SCHEMATIC
Over-Voltage Protection
During operation, a short on the upper MOSFET of the PWM
regulator (Q1) causes VOUT1 to increase. When the output
exceeds the over-voltage threshold of 115% of DACOUT,
the over-voltage comparator trips to set the fault latch and
turns Q2 on. This blows the input fuse and reduces VOUT1.
The fault latch raises the FAULT/RT pin to VCC.
A separate over-voltage circuit provides protection during
the initial application of power. For voltages on the VCC pin
below the power-on reset (and above ~4V), the output level
is monitored for voltages above 1.3V. Should VSEN1
exceed this level, the lower MOSFET, Q2 is driven on.
Over-Current Protection
All outputs are protected against excessive over-currents.
The PWM controller uses the upper MOSFET’s
on-resistance, rDS(ON) to monitor the current for protection
against shorted output. All linear controllers monitor their
respective VSEN pins for under-voltage events to protect
against excessive currents.
Figure 4 illustrates the over-current protection with an
overload on OUT1. The overload is applied at T0 and the
current increases through the inductor (LOUT1). At time T1,
the OVER-CURRENT comparator trips when the voltage
across Q1 (iD • rDS(ON)) exceeds the level programmed by
ROCSET. This inhibits all outputs, discharges the soft-start
capacitor (CSS) with a 10mA current sink, and increments
the counter. CSS recharges at T2 and initiates a soft-start
cycle with the error amplifiers clamped by soft-start. With
OUT1 still overloaded, the inductor current increases to trip
the over-current comparator. Again, this inhibits all outputs,
but the soft-start voltage continues increasing to 4V before
discharging. The counter increments to 2. The soft-start
cycle repeats at T3 and trips the over-current comparator.
HIP6021A
The SS pin voltage increases to 4V at T4 and the counter
increments to 3. This sets the fault latch to disable the
converter. The fault is reported on the FAULT/RT pin.
The linear controllers operate in the same way as the PWM
in response to over-current faults. The differentiating factor
for the linear controllers is that they monitor the VSEN pins
for under-voltage events. Should excessive currents cause
the voltage at the VSEN pins to fall below the linear undervoltage threshold, the LUV signal sets the over-current
latch if CSS is fully charged. Blanking the LUV signal during
the CSS charge interval allows the linear outputs to build
above the under-voltage threshold during normal operation.
Cycling the bias input power off then on resets the counter
and the fault latch.
OVER-CURRENT TRIP:
V
>V
DS
SET
i ×r
D
DS ( ON ) > I OCSET × R OCSET
VIN = +5V
ROCSET
OCSET
IOCSET
200μA
OVERCURRENT
OC
VSET +
iD
VCC
DRIVE
+
UGATE
+
VDS
PHASE
-
V
PWM
GATE
CONTROL
= V –V
PHASE
IN
DS
V
= V –V
OCSET
IN
SET
FAULT/RT
FIGURE 5. OVER-CURRENT DETECTION
FAULT
REPORTED
10V
0V
INDUCTOR CURRENT
SOFT-START
COUNT
=1
COUNT
=2
COUNT
=3
The OC trip point varies with MOSFET’s rDS(ON)
temperature variations. To avoid over-current tripping in the
normal operating load range, determine the ROCSET
resistor from the equation above with:
4V
1. The maximum rDS(ON) at the highest junction temperature.
2V
2. The minimum IOCSET from the specification table.
0V
3. Determine IPEAK for IPEAK > IOUT(MAX) + (ΔI)/2, where
ΔI is the output inductor ripple current.
OVERLOAD
APPLIED
For an equation for the ripple current see the section under
component guidelines titled ‘PWM Output Inductor
Selection’.
OUT1 Voltage Program
0A
T0 T1
T2
T3
T4
TIME
FIGURE 4. OVER-CURRENT OPERATION
A resistor (ROCSET) programs the over-current trip level for
the PWM converter. As shown in Figure 5, the internal
200μA current sink, IOCSET develops a voltage across
ROCSET (VSET) that is referenced to VIN . The DRIVE
signal enables the over-current comparator (OVERCURRENT). When the voltage across the upper MOSFET
(VDS) exceeds VSET , the over-current comparator trips to
set the over-current latch. Both VSET and VDS are
referenced to VIN and a small capacitor across ROCSET
helps VOCSET track the variations of VIN due to MOSFET
switching. The over-current function will trip at a peak
inductor current (IPEAK) determined by:
I OCSET × R OCSET
I PEAK = ---------------------------------------------------r DS ( ON )
9
The output voltage of the PWM converter is programmed to
discrete levels between 1.3VDC and 3.5VDC . This output
(OUT1) is designed to supply the core voltage of Intel’s
advanced microprocessors. The voltage identification (VID)
pins program an internal voltage reference (DACOUT) with a
TTL-compatible 5-bit digital-to-analog converter. The level of
DACOUT also sets the PGOOD and OVP thresholds.
Table 1 specifies the DACOUT voltage for the different
combinations of connections on the VID pins. The VID pins
can be left open for a logic 1 input, because they are
internally pulled up to an internal voltage of about 5V by a
10μA current source. Changing the VID inputs during
operation is not recommended, as it could toggle the
PGOOD signal and exercise the over-voltage protection.
HIP6021A
TABLE 1. OUT1 VOLTAGE PROGRAM
PIN NAME
VID4
VID3
VID2
VID1
VID0
NOMINAL
DACOUT
VOLTAGE
0
1
1
1
1
1.30
0
1
1
1
0
1.35
0
1
1
0
1
1.40
0
1
1
0
0
1.45
0
1
0
1
1
1.50
0
1
0
1
0
1.55
0
1
0
0
1
1.60
0
1
0
0
0
1.65
0
0
1
1
1
1.70
0
0
1
1
0
1.75
0
0
1
0
1
1.80
0
0
1
0
0
1.85
0
0
0
1
1
1.90
0
0
0
1
0
1.95
0
0
0
0
1
2.00
0
0
0
0
0
2.05
1
1
1
1
1
2.00
circuitry. Left open, the SELECT pin is internally pulled ‘high’
and the AGP voltage is regulated to 3.3V during the softstart
sequence. Once complete, the gate drive is increased and
the regulator becomes a simple pass circuit for the 3.3V
input voltage.
OUT3 and OUT4 Voltage Adjustability
The GTL bus voltage (1.5V, OUT3) and the chip set and/or
cache memory voltage (1.8V, OUT4) are internally set for
simple, low-cost implementation in typical Intel motherboard
architectures. However, if different voltage settings are
desired for these two outputs, the FIX pin provides the
necessary adaptability. Left open (NC), this pin sets the fixed
output voltages described above. Grounding this pin allows
both output voltages to be set by means of external resistor
dividers as shown in Figure 6.
VAUX
+3.3VIN
Q4
DRIVE3
VOUT3
VSEN3
RS3
RP3
COUT3
HIP6021A
DRIVE4
Q5
VOUT4
1
1
1
1
0
2.1
1
1
1
0
1
2.2
1
1
1
0
0
2.3
1
1
0
1
1
2.4
1
1
0
1
0
2.5
1
1
0
0
1
2.6
1
1
0
0
0
2.7
1
0
1
1
1
2.8
1
0
1
1
0
2.9
Application Guidelines
1
0
1
0
1
3.0
Soft-Start Interval
1
0
1
0
0
3.1
1
0
0
1
1
3.2
1
0
0
1
0
3.3
1
0
0
0
1
3.4
1
0
0
0
0
3.5
Initially, the soft-start function clamps the error amplifier’s
output of the PWM converter. This generates PHASE pulses
of increasing width that charge the output capacitor(s). After
the output voltage increases to approximately 70% of the set
value, the reference input of the error amplifier is clamped to
a voltage proportional to the SS pin voltage. The resulting
output voltages start-up as shown in Figure 2.
NOTE: 0 = connected to GND, 1 = open or connected to 5V through
pull-up resistors.
OUT2 Voltage Selection
The AGP output voltage is internally set to one of two levels,
based on the status of the SELECT pin. Grounding the
SELECT pin enables the internal 1.5V regulator control
10
VSEN4
RS4
COUT4
RP4
FIX
R ⎞
⎛
S
V OUT = V BG × ⎜ 1 + --------⎟
R P⎠
⎝
FIGURE 6. ADJUSTING THE OUTPUT VOLTAGE OF
OUTPUTS 3 AND 4
The soft-start function controls the output voltage rate of rise
to limit the current surge at start-up. The soft-start interval and
the surge current are programmed by the soft-start capacitor,
CSS. Programming a faster soft-start interval increases the
peak surge current. The peak surge current occurs during the
initial output voltage rise to 70% of the set value.
HIP6021A
Shutdown
The HIP6021A features a dedicated shutdown pin (SD). A
TTL-compatible, logic high signal applied to this pin shuts
down (disables) all four outputs and discharges the soft-start
capacitor. Following a shutdown, a logic low signal
re-enables the outputs through initiation of a new soft-start
cycle. Left open this pin will asses a logic low state, due to its
internal pull-down resistor, thus enabling normal operation of
all outputs.
The PWM output does not switch until the soft-start voltage
(VSS) exceeds the oscillator’s valley voltage. The references
on each linear’s error amplifier are clamped to the soft-start
voltage. Holding the SS pin low (with an open drain or
collector signal) turns off all four regulators.
Layout Considerations
MOSFETs switch very fast and efficiently. The speed with
which the current transitions from one device to another
causes voltage spikes across the interconnecting
impedances and parasitic circuit elements. The voltage
spikes can degrade efficiency, radiate noise into the circuit,
and lead to device over-voltage stress. Careful component
layout and printed circuit design minimizes the voltage
spikes in the converter. Consider, as an example, the turnoff transition of the upper PWM MOSFET. Prior to turn-off,
the upper MOSFET was carrying the full load current.
During the turn-off, current stops flowing in the upper
MOSFET and is picked up by the lower MOSFET or
Schottky diode. Any inductance in the switched current
path generates a large voltage spike during the switching
interval. Careful component selection, tight layout of the
critical components, and short, wide circuit traces minimize
the magnitude of voltage spikes. See Application Note
AN9836 for evaluation board drawings of the component
placement and the printed circuit board layout of a typical
application.
There are two sets of critical components in a DC-DC
converter using a HIP6021A controller. The switching power
components are the most critical because they switch large
amounts of energy, and as such, they tend to generate
equally large amounts of noise. The critical small signal
components are those connected to sensitive nodes or
those supplying critical bypass current.
The power components and the controller IC should be
placed first. Locate the input capacitors, especially the highfrequency ceramic decoupling capacitors, close to the power
switches. Locate the output inductor and output capacitors
between the MOSFETs and the load. Locate the PWM
controller close to the MOSFETs.
The critical small signal components include the bypass
capacitor for VCC and the soft-start capacitor, CSS . Locate
these components close to their connecting pins on the
control IC. Minimize any leakage current paths from SS
node, since the internal current source is only 28μA.
11
A multi-layer printed circuit board is recommended.
Figure 7 shows the connections of the critical components
in the converter. Note that the capacitors CIN and COUT
each represent numerous physical capacitors. Dedicate
one solid layer for a ground plane and make all critical
component ground connections with vias to this layer.
Dedicate another solid layer as a power plane and break
this plane into smaller islands of common voltage levels.
The power plane should support the input power and
output power nodes. Use copper filled polygons on the top
and bottom circuit layers for the PHASE nodes, but do not
unnecessarily oversize these particular islands. Since the
PHASE nodes are subjected to very high dV/dt voltages,
the stray capacitor formed between these islands and the
surrounding circuitry will tend to couple switching noise.
Use the remaining printed circuit layers for small signal
wiring. The wiring traces from the control IC to the
MOSFET gate and source should be sized to carry 2A peak
currents.
PWM Controller Feedback Compensation
The PWM controller uses voltage-mode control for output
regulation. This section highlights the design consideration
for a PWM voltage-mode controller. Apply the methods and
considerations only to the PWM controller.
Figure 8 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(VOUT) is regulated to the Reference voltage level. The
reference voltage level is the DAC output voltage
(DACOUT). The error amplifier (Error Amp) output (VE/A) is
compared with the oscillator (OSC) triangular wave to
provide a pulse-width modulated (PWM) wave with an
amplitude of VIN at the PHASE node. The PWM wave is
smoothed by the output filter (LO and CO).
The modulator transfer function is the small-signal transfer
function of VOUT/VE/A . This function is dominated by a DC
Gain, given by VIN/VOSC , and shaped by the output filter,
with a double pole break frequency at FLC and a zero
at FESR .
Modulator Break Frequency Equations
1
F LC = ---------------------------------------2π × L O × C O
1
F ESR = ----------------------------------------2π × ESR × C O
The compensation network consists of the error amplifier
(internal to the HIP6021A) and the impedance networks ZIN
and ZFB . The goal of the compensation network is to provide
a closed loop transfer function with high 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180 degrees. The equations below relate the compensation
network’s poles, zeros and gain to the components (R1, R2,
R3, C1, C2, and C3) in Figure 7. Use these guidelines for
locating the poles and zeros of the compensation network:
HIP6021A
1. Pick Gain (R2/R1) for desired converter bandwidth
VIN
2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC)
OSC
3. Place 2ND Zero at Filter’s Double Pole
DRIVER
PWM
COMP
4. Place 1ST Pole at the ESR Zero
5. Place 2ND Pole at Half the Switching Frequency
LO
-
DRIVER
+
Δ VOSC
PHASE
6. Check Gain against Error Amplifier’s Open-Loop Gain
7. Estimate Phase Margin - Repeat if Necessary
ZFB
ZIN
-
ERROR
AMP
+12V
VCC GND
OCSET1
+3.3VIN
Q3
VOUT2
COCSET1
DRIVE2
UGATE1
DETAILED COMPENSATION COMPONENTS
Q1
LOUT1
PHASE1
COUT2
SS
CSS
VOUT3
LGATE1
C1
VOUT1
COUT1
CR1
Q2
PGND
Q4
COUT4
Q5
VOUT
ZIN
C3
R2
R3
R1
COMP
FB
-
HIP6021A
DRIVE3 DRIVE4
ZFB
C2
+
VOUT4
COUT3
REFERENCE
ROCSET1
LOAD
CVCC
LOAD
ESR
(PARASITIC)
+
CIN
LOAD
CO
VE/A
LIN
HIP6021A
LOAD
+5VIN
VOUT
DACOUT
FIGURE 8. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN
+3.3VIN
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 7. PRINTED CIRCUIT BOARD POWER PLANES AND
ISLANDS
Figure 9 shows an asymptotic plot of the DC-DC converter’s
gain vs frequency. The actual Modulator Gain has a high gain
peak dependent on the quality factor (Q) of the output filter,
which is not shown in Figure 8. Using the above guidelines
should yield a Compensation Gain similar to the curve plotted.
The gain. Check the compensation gain at FP2 with the
capabilities of the error amplifier. The Closed Loop Gain is
constructed on the log-log graph of Figure 9 by adding the
Modulator Gain (in dB) to the Compensation Gain (in dB). This
is equivalent to multiplying the modulator transfer function to
the compensation transfer function and plotting the gain.
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW)
overall loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.
12
Compensation Break Frequency Equations
1
F Z1 = ----------------------------------2π × R 2 × C1
1
F P1 = ------------------------------------------------------C1 × C2
2π × R 2 × ⎛ ----------------------⎞
⎝ C1 + C2⎠
1
F Z2 = ------------------------------------------------------2π × ( R1 + R3 ) × C3
1
F P2 = ----------------------------------2π × R 3 × C3
FZ1
FZ2
FP1
FP2
100
OPEN LOOP
ERROR AMP GAIN
⎛ V IN ⎞
20 log ⎜ ------------⎟
⎝ V PP⎠
80
60
GAIN (dB)
KEY
40
COMPENSATION
GAIN
20
0
-20
-40
-60
R2
20 log ⎛ --------⎞
⎝ R1⎠
MODULATOR
GAIN
10
100
FLC
1K
CLOSED LOOP
GAIN
FESR
10K
100K
1M
10M
FREQUENCY (Hz)
FIGURE 9. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
HIP6021A
Component Selection Guidelines
Output Capacitors
The output capacitors for each output have unique
requirements. In general, the output capacitors should be
selected to meet the dynamic regulation requirements.
Additionally, the PWM converters require an output
capacitor to filter the current ripple. The load transient for the
microprocessor core requires high quality capacitors to
supply the high slew rate (di/dt) current demands.
PWM Output
Modern microprocessors produce transient load rates above
1A/ns. High frequency capacitors initially supply the
transient current and slow the load rate-of-change seen by
the bulk capacitors. The bulk filter capacitor values are
generally determined by the ESR (effective series
resistance) and voltage rating requirements rather than
actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR determines the output ripple voltage and
the initial voltage drop following a high slew-rate transient’s
edge. An aluminum electrolytic capacitor’s ESR value is
related to the case size with lower ESR available in larger
case sizes. However, the equivalent series inductance (ESL)
of these capacitors increases with case size and can reduce
the usefulness of the capacitor to high slew-rate transient
loading. Unfortunately, ESL is not a specified parameter.
Work with your capacitor supplier and measure the
capacitor’s impedance with frequency to select a suitable
component. In most cases, multiple electrolytic capacitors of
small case size perform better than a single large case
capacitor.
Linear Output Capacitors
The output capacitors for the linear regulators provide
dynamic load current. The linear controllers use dominant
pole compensation integrated into the error amplifier and are
insensitive to output capacitor selection. Output capacitors
should be selected for transient load regulation.
PWM Output Inductor
The PWM converter requires an output inductor. The output
inductor is selected to meet the output voltage ripple
requirements and sets the converter’s response time to a
load transient. The inductor value determines the converter’s
ripple current and the ripple voltage is a function of the ripple
current. The ripple voltage and current are approximated by
the following equations:
13
V IN – V OUT V OUT
ΔI = -------------------------------- × ---------------FS × L
V IN
ΔV OUT = ΔI × ESR
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values increase
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
HIP6021A will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
interval required to slew the inductor current from an initial
current value to the post-transient current level. During this
interval the difference between the inductor current and the
transient current level must be supplied by the output
capacitor(s). Minimizing the response time can minimize the
output capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
L O × I TRAN
t RISE = -------------------------------V IN – V OUT
L O × I TRAN
t FALL = ------------------------------V OUT
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. Be sure to check both
of these equations at the minimum and maximum output
levels for the worst case response time.
Input Capacitors
The important parameters for the bulk input capacitors are
the voltage rating and the RMS current rating. For reliable
operation, select bulk input capacitors with voltage and
current ratings above the maximum input voltage and largest
RMS current required by the circuit. The capacitor voltage
rating should be at least 1.25 times greater than the
maximum input voltage and a voltage rating of 1.5 times is a
conservative guideline. The RMS current rating requirement
for the input capacitor of a buck regulator is approximately
1/2 of the summation of the DC load current.
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use ceramic capacitance
for the high frequency decoupling and bulk capacitors to
supply the RMS current. Small ceramic capacitors can be
placed very close to the upper MOSFET to suppress the
voltage induced in the parasitic circuit impedances.
For a through-hole design, several electrolytic capacitors
(Panasonic HFQ series or Nichicon PL series or Sanyo MV-GX
or equivalent) may be needed. For surface mount designs,
solid tantalum capacitors can be used, but caution must be
exercised with regard to the capacitor surge current rating.
These capacitors must be capable of handling the surgecurrent at power-up. The TPS series available from AVX, and
the 593D series from Sprague are both surge current tested.
HIP6021A
MOSFET Considerations
The HIP6021A requires 5 external transistors. Two
N-Channel MOSFETs are used in the synchronous-rectified
buck topology of PWM1 converter. It is recommended that
the AGP linear regulator pass element be a N-Channel
MOSFET as well. The GTL and memory linear controllers
can also each drive a MOSFET or a NPN bipolar as a pass
transistor. All these transistors should be selected based
upon rDS(ON) , current gain, saturation voltages, gate supply
requirements, and thermal management considerations.
+5V OR LESS
+12V
VCC
HIP6021A
UGATE
-
+
LGATE
PGND
PWM MOSFETs
In high-current PWM applications, the MOSFET power
dissipation, package selection and heatsink are the
dominant design factors. The power dissipation includes two
loss components; conduction loss and switching loss. These
losses are distributed between the upper and lower
MOSFETs according to duty factor (see the equations
below). The conduction losses are the main component of
power dissipation for the lower MOSFETs. Only the upper
MOSFET has significant switching losses, since the lower
device turns on and off into near zero voltage.
The equations below assume linear voltage-current
transitions and do not model power loss due to the reverserecovery of the lower MOSFET’s body diode. The gatecharge losses are dissipated by the HIP6021A and don't
heat the MOSFETs. However, large gate-charge increases
the switching time, tSW which increases the upper MOSFET
switching losses. Ensure that both MOSFETs are within their
maximum junction temperature at high ambient temperature
by calculating the temperature rise according to package
thermal-resistance specifications. A separate heatsink may
be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.
2
I O × r DS ( ON ) × V OUT I O × V IN × t SW × F S
P UPPER = ------------------------------------------------------------ + ---------------------------------------------------V IN
2
2
I O × r DS ( ON ) × ( V IN – V OUT )
P LOWER = --------------------------------------------------------------------------------V IN
The rDS(ON) is different for the two equations above even if
the same device is used for both. This is because the gate
drive applied to the upper MOSFET is different than the
lower MOSFET. Figure 10 shows the gate drive where the
upper MOSFET’s gate-to-source voltage is approximately
VCC less the input supply. For +5V main power and
+12VDC for the bias, the gate-to-source voltage of Q1 is 7V.
The lower gate drive voltage is +12VDC. A logic-level
MOSFET is a good choice for Q1 and a logic-level MOSFET
can be used for Q2 if its absolute gate-to-source voltage
rating exceeds the maximum voltage applied to VCC.
14
Q1
PHASE
GND
NOTE:
VGS ≈ VCC -5V
Q2
CR1
NOTE:
VGS ≈ VCC
FIGURE 10. UPPER GATE DRIVE - DIRECT VCC DRIVE OPTION
Rectifier CR1 is a clamp that catches the negative inductor
swing during the dead time between the turn off of the lower
MOSFET and the turn on of the upper MOSFET. The diode
must be a Schottky type to prevent the lossy parasitic
MOSFET body diode from conducting. It is acceptable to
omit the diode and let the body diode of the lower MOSFET
clamp the negative inductor swing, but efficiency could drop
one or two percent as a result. The diode's rated reverse
breakdown voltage must be greater than the maximum input
voltage.
Linear Controller Transistors
The main criteria for selection of transistors for the linear
regulators is package selection for efficient removal of heat.
The power dissipated in a linear regulator is:
P LINEAR = I O × ( V IN – V OUT )
Select a package and heatsink that maintains the junction
temperature below the rating with a the maximum expected
ambient temperature.
When selecting bipolar NPN transistors for use with the
linear controllers, insure the current gain at the given
operating VCE is sufficiently large to provide the desired
output load current when the base is fed with the minimum
driver output current.
HIP6021A
HIP6021A DC-DC Converter Application Circuit
memory voltage (VOUT4) from +3.3V, +5VDC, and +12VDC.
For detailed information on the circuit, including a Bill-ofMaterials and circuit board description, see Application Note
AN9836. Also see Intersil’s web page (http://www.intersil.com).
Figure 11 shows an application circuit of a power supply for a
microprocessor computer system. The power supply provides
the microprocessor core voltage (VOUT1), the AGP bus
voltage (VOUT2), the GTL bus voltage (VOUT3), and the
+12VIN
L1
+5VIN
1μH
GND
C1-6 +
6x1000μF
C7
1μF
C8
1000pF
C9
1μF
VCC
FAULT/RT
+3.3VIN
VAUX
DRIVE2
VOUT2
28
R1
23
13
16
8
1
27
26
(3.3VIN OR 1.5V)
+
C10, 11
2x1000μF
VSEN2
Q3
HUF76121D3S
10
25
SELECT
11
22
U1
HIP6021A
Q4
HUF76107D3S
DRIVE3
VOUT3
(1.5V)
VSEN3
+
18
21
20
VOUT4
VSEN4
(1.8V)
+
C25, 26
2x1000μF
SD
FIX
Q1, 2
2xHUF76143S3S
UGATE
L2
VOUT1
(1.3V-3.5V)
PHASE
C12-19 +
8x1000μF
LGATE
PGND
R2
10.2K
VSEN1
FB
COMP
R3
1.62K
C21
10pF
C20
0.22μF
19
7
DRIVE4
POWERGOOD
PGOOD
C23, 24
2x1000μF
Q5
HUF76107D3S
1.0K
4.2μH
24
TYPEDET
OCSET
R4
150K
R5
499K
6 VID1
VID2
5
4 VID3
15
14
3
9
2
VID0
C22
2.7nF
12
17
VID4
SS
C27
0.1μF
GND
FIGURE 11. POWER SUPPLY APPLICATION CIRCUIT FOR A MICROPROCESSOR COMPUTER SYSTEM
15
HIP6021A
Small Outline Plastic Packages (SOIC)
M28.3 (JEDEC MS-013-AE ISSUE C)
N
28 LEAD WIDE BODY SMALL OUTLINE PLASTIC PACKAGE
INDEX
AREA
0.25(0.010) M
H
B M
INCHES
E
SYMBOL
-B-
1
2
3
L
SEATING PLANE
-A-
h x 45o
A
D
-C-
e
A1
B
C
0.10(0.004)
0.25(0.010) M
C A M
B S
MILLIMETERS
MIN
MAX
NOTES
A
0.0926
0.1043
2.35
2.65
-
0.0040
0.0118
0.10
0.30
-
B
0.013
0.0200
0.33
0.51
9
C
0.0091
0.0125
0.23
0.32
-
D
0.6969
0.7125
17.70
18.10
3
E
0.2914
0.2992
7.40
7.60
4
0.05 BSC
10.00
h
0.01
0.029
0.25
0.75
5
L
0.016
0.050
0.40
1.27
6
8o
0o
28
0o
10.65
-
0.394
α
0.419
1.27 BSC
H
N
NOTES:
MAX
A1
e
µα
MIN
28
-
7
8o
Rev. 0 12/93
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2
of Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010
inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual
index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch)
10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
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