LT3795 110V LED Controller with Spread Spectrum Frequency Modulation Features n n n n n n n n n n n n n n n n Description 3000:1 True Color PWM™ Dimming Wide Input Voltage Range: 4.5V to 110V Input and Output Current Reporting PMOS Switch Driver for PWM and Output Disconnect Internal Spread Spectrum Frequency Modulation ±2% Constant Voltage Regulation ±3% Constant Current Regulation: 0V ≤ VOUT ≤ 110V Programmable Input Current limit CTRL Inputs Linearly Adjust LED Current Adjustable Frequency: 100kHz to 1MHz Programmable Open LED Protection with OPENLED Flag Short-Circuit Protection and SHORTLED Flag Programmable Undervoltage Lockout with Hysteresis Soft-Start with Programmable Fault Restart Timer C/10 Detection for Battery Charging Available in 28-Lead TSSOP Package Applications n n n High Power LED, High Voltage LED Battery Chargers Accurate Current Limited Voltage Regulators The LT®3795 is a DC/DC controller designed to regulate a constant-current or constant-voltage and is ideal for driving LEDs. It drives a low side external N-channel power MOSFET from an internal regulated 7.7V supply. The fixed frequency and current mode architecture result in stable operation over a wide range of supply and output voltages. Spread spectrum frequency modulation (SSFM) can be activated for improved electromagnetic compatibility (EMC) performance. The ground referred voltage FB pin serves as the input for several LED protection features, and also allows the converter to operate as a constantvoltage source. The maximum output current is set by an external resistor, and the output current amplifier has a railto-rail common mode range. The LT3795 also includes a separate input current sensing amplifier that is used to limit input current. The TG pin inverts and level shifts the PWM signal to drive the gate of the external PMOS. The PWM input provides LED dimming ratios of up to 3000:1, and the CTRL inputs provide additional analog dimming capability. L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 7199560, 7321203, 7746300. Typical Application Short-Circuit Robust Boost LED Driver with Spread Spectrum Frequency Modulation 2.2µF ×3 1M EN/UVLO VIN IVINP IVINN GATE OVLO 47nF 100k GND LT3795 PWM IVINCOMP 10nF FB ISP OPENLED SHORTLED ISN OPENLED SHORTLED VC RT 10k TG INTVCC ISMON 90 85 80 4.7µF SS 70 0 10 20 30 VIN (V) 40 50 60 3795 TA01b INTVCC 31.6k 250kHz 0.1µF 95 75 620mΩ RAMP 100k 6.8nF 13.3k 15mΩ CTRL1 CTRL2 VREF PWM 2.2µF ×4 SENSE 12.4k INTVCC 100 499k 115k Efficiency vs VIN 22µH 15mΩ EFFICIENCY (%) VIN 8V TO 60V 110V (TRANSIENT) 63V OVLO 87V LED 400mA 0.1µF 3795 TA01a 3795f For more information www.linear.com/LT3795 1 LT3795 Absolute Maximum Ratings Pin Configuration (Note 1) VIN ........................................................................... 110V EN/UVLO.................................................................. 110V ISP, ISN.................................................................... 110V TG, GATE...............................................................Note 2 IVINP, IVINN............................................................. 110V VIN - IVINN.................................................... –0.3V to 4V INTVCC (Note 3)......................................8.6V, VIN + 0.3V PWM, SHORTLED, OPENLED......................................12V FB, RAMP, OVLO..........................................................8V CTRL1, CTRL2............................................................15V SENSE.......................................................................0.5V ISMON, IVINCOMP......................................................5V VC, VREF, SS.................................................................3V RT................................................................................2V Operating Junction Temperature Range (Note 4) LT3795E/LT3795I................................... –40 to 125°C LT3795H................................................. –40 to 150°C Storage Temperature Range.......................–65 to 150°C TOP VIEW ISP 1 28 IVINCOMP ISN 2 27 IVINP TG 3 26 IVINN GND 4 25 OVLO ISMON 5 24 EN/UVLO CTRL2 6 FB 7 VC 8 CTRL1 9 23 VIN 29 GND 22 GND 21 GND 20 INTVCC VREF 10 19 GATE SS 11 18 SENSE RT 12 17 GND RAMP 13 16 OPENLED PWM 14 15 SHORTLED FE PACKAGE 28-LEAD PLASTIC TSSOP TJMAX = 150°C, θJA = 30°C/W, θJC = 10°C/W EXPOSED PAD (PIN 29) IS GND, MUST BE SOLDERED TO PCB Order Information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3795EFE#PBF LT3795EFE#TRPBF LT3795FE 28-Lead Plastic TSSOP –40°C to 125°C LT3795IFE#PBF LT3795IFE#TRPBF LT3795FE 28-Lead Plastic TSSOP –40°C to 125°C LT3795HFE#PBF LT3795HFE#TRPBF LT3795FE 28-Lead Plastic TSSOP –40°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 24V, EN/UVLO = 24V, CTRL1 = CTRL2 = 2V, PWM = 5V, unless otherwise noted. PARAMETER CONDITIONS VIN Minimum Operating Voltage VIN Tied to INTVCC 4.5 V VIN Shutdown IQ EN/UVLO = 0V, PWM = 0V EN/UVLO = 1.15V, PWM = 0V 10 22 µA µA VIN Operating IQ (Not Switching) RT = 82.5k to GND, FB = 1.5V 2.9 3.5 mA 2.015 2.06 VREF Voltage –100µA ≤ IREF ≤ 10µA VREF Pin Line Regulation 4.5V < VIN < 110V VREF Pin Load Regulation –100µA ≤ IREF ≤ 0µA SENSE Current Limit Threshold 2 MIN l 1.97 TYP MAX 1.5 100 117 V m%/V 10 l UNITS m%/µA 125 mV 3795f For more information www.linear.com/LT3795 LT3795 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 24V, EN/UVLO = 24V, CTRL1 = CTRL2 = 2V, PWM = 5V, unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS SENSE Input Bias Current Current Out of Pin 65 µA SS Sourcing Current SS = 0V 28 µA SS Sinking Current ISP – ISN = 1V, SS = 2V 2.8 µA Error Amplifier Full Scale LED Current Sense Threshold (V(ISP-ISN)) ISP = 48V, CTRL1 ≥ 1.2V, CTRL2 ≥ 1.2V ISP = 0V, CTRL1 ≥ 1.2V, CTRL2 ≥ 1.2V l l 243 243 250 250 257 257 mV mV 9/10th LED Current Sense Threshold (V(ISP-ISN)) ISP = 48V, CTRL1 = 1V, CTRL2 = 1.2V ISP = 0V, CTRL1 = 1V, CTRL2 = 1.2V l l 220 220 225 225 230 230 mV mV 1/2 LED Current Sense Threshold (V(ISP-ISN)) ISP = 48V, CTRL1 = 0.6V, CTRL2 = 1.2V ISP = 0V, CTRL1 = 0.6V, CTRL2 = 1.2V l l 119 119 125 125 130 130 mV mV 1/10th LED Current Sense Threshold (V(ISP-ISN)) ISP = 48V, CTRL1 = 0.2V, CTRL2 = 1.2V ISP = 0V, CTRL1 = 0.2V, CTRL2 = 1.2V l l 16 16 25 25 32 32 mV mV ISP/ISN Current Monitor Voltage (VISMON) V(ISP-ISN) = 250mV, ISP = 48V, –50µA ≤ IISMON ≤ 0 µA V(ISP-ISN) = 250mV, ISP = 0V, –50µA ≤ IISMON ≤ 0 µA l l 0.96 0.96 1 1 1.04 1.04 V V ISP/ISN Overcurrent Protection Threshold (V(ISP-ISN)) ISN = 48V ISN = 0V l l 360 360 375 375 390 390 mV mV CTRL1, CTRL2 Input Bias Current Current Out of Pin, CTRL = 1V 50 200 nA 110 V 0.1 µA µA ISP/ISN Current Sense Amplifier Input Common Mode Range 0 ISP/ISN Input Current Bias Current (Combined) PWM = 5V (Active), ISP = 48V PWM = 0V (Standby), ISP = 48V 700 0 ISP/ISN Current Sense Amplifier gm V(ISP-ISN) = 250mV 350 VC Output Impedance 2000 VC Standby Input Bias Current PWM = 0V FB Regulation Voltage (VFB) ISP = ISN = 48V ISP = ISN = 48V –20 l 1.230 1.238 1.250 1.250 kΩ 20 nA 1.270 1.264 V V 600 FB Amplifier gm FB Pin Input Bias Current µS Current Out of Pin, FB = VFB FB Open LED Threshold OPENLED Falling, ISP = ISN = 48V C/10 Comparator Threshold (V(ISP-ISN)) OPENLED Falling, FB = 1.25V, ISP = 48V OPENLED Falling, FB = 1.25V, ISN = 0V FB Overvoltage Threshold TG Rising 40 µS 200 VFB – 62mV VFB – 52mV VFB – 42mV 25 25 V mV mV VFB + 35mV VFB + 50mV VFB + 60mV VC Current Mode Gain (∆VVC/∆VSENSE) nA 4.2 V V/V FB SHORTLED Threshold SHORTLED Falling VC Pin Source Current VC = 1.2V 10 µA VC Pin Sink Current VC = 1.2V, FB = 1.4V 30 µA 300 l 350 mV Input Current Sense Amplifier Input Current Sense Amplifier Input Voltage Common Range (VIVINP/VIVINN) l 2.5 V 63 mV Input Current Sense Threshold (VIVINP - VIVINN) VIVINP = 48V, VIN = 48V l 57 Input Current Monitor V(IVINCOMP) VIVINP-VIVINN = 50mV l 0.94 1 1.06 V Input Bias Current (I(IVINN)) VIVINP-VIVINN = 50mV 100 1000 nA Input Current Sense Amplifier gm VIVINP-VIVINN = 60mV 3400 µS 1 µs 15 kΩ Input Step Response (to 50% of Output Step) ∆VSENSE = 60mV Step IVINCOMP Pin Resistance to GND 60 110 3795f For more information www.linear.com/LT3795 3 LT3795 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 24V, EN/UVLO = 24V, CTRL1 = CTRL2 = 2V, PWM = 5V, unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX 7.4 7.7 8 UNITS Linear Regulator INTVCC Regulation Voltage l Dropout (VIN -INTVCC) IINTVCC = –10mA, VIN = 4.5V INTVCC Current Limit VIN = 110V, INTVCC = 6V VIN = 12V, INTVCC = 6V INTVCC Shutdown Bias Current if Externally Driven to 7V EN/UVLO = 0V, INTVCC = 7V 550 mV 18 85 INTVCC Undervoltage Lockout 3.8 INTVCC Undervoltage Lockout Hysteresis V mA mA 13 17 4 4.1 200 µA V mV Oscillator Switching Frequency RT = 82.5k RT = 19.6k RT = 6.65k Minimum Off-Time (Note 5) 160 ns Minimum On-Time (Note 5) 210 ns Switching Frequency Modulation VRAMP = 2V 70 % l l l 85 340 900 105 400 1000 125 480 1150 kHz kHz kHz RAMP Input Low Threshold 1 V RAMP Input High Threshold 2 V RAMP Pin Source Current RAMP = 0.4V 12 µA RAMP Pin Sink Current RAMP = 1.6V 12 µA LOGIC Input/Outputs PWM Input Threshold Rising l 0.96 1 l 1.185 1.220 PWM Pin Bias Current 1.04 10 EN/UVLO Threshold Voltage Falling EN/UVLO Rising Hysteresis µA 1.25 20 EN/UVLO Input Low Voltage IVIN Drops Below 1µA 0.4 EN/UVLO Pin Bias Current Low EN/UVLO = 1.15V 2.5 EN/UVLO Pin Bias Current High EN/UVLO = 1.30V OPENLED OUTPUT Low IOPENLED = 0.5mA SHORTLED OUTPUT Low ISHORTLED = 0.5mA OVLO Threshold Voltage Rising 1.215 V mV V 3 10 l V 1.25 3.8 µA 100 nA 300 mV 300 mV 1.28 V OVLO Falling Hysteresis 28 mV OVLO Pin Input Current 150 nA ns Gate Driver tr NMOS GATE Driver Output Rise Time CL = 3300pF, 10% to 90% 20 tf NMOS GATE Driver Output Fall Time CL = 3300pF, 10% to 90% 18 NMOS GATE Output Low (VOL) ns 0.05 NMOS GATE Output High (VOH) INTVCC – 0.05 V V tr Top GATE Driver Output Rise Time CL = 300pF 50 ns tf Top GATE Driver Output Fall Time CL = 300pF 100 ns Top Gate On Voltage (VISP -VTG) ISP = 48V 7 8 V Top Gate Off Voltage (VISP -VTG) PWM = 0V, ISP = 48V 0 0.3 V 4 3795f For more information www.linear.com/LT3795 LT3795 Electrical Characteristics Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Do not apply a positive or negative voltage source to TG and GATE pins, otherwise permanent damage may occur. Note 3: Operating maximum for INTVCC is 8V. Note 4: The LT3795E is guaranteed to meet specified performance from 0°C to 125°C. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3795I is guaranteed to meet performance specifications over the –40°C to 125°C operating junction temperature range. The LT3795H is guaranteed over the full –40°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. Note 5: See Duty Cycle Considerations in the Applications Information section. Typical Performance Characteristics 253 200 150 100 50 0 0 0.2 0.4 0.6 0.8 VCTRL (V) 1.0 1.2 251 250 249 248 127 250 126 125 124 123 75 250 249 248 247 0 40 20 60 80 246 –50 –25 120 100 0 100 125 150 V(ISP-ISN) Threshold vs FB Voltage 3795 G03a VFB vs Temperature 1.27 1.26 200 150 1.25 100 1.24 50 0 1.1 1.15 1.2 1.25 1.3 VFB (V) TEMPERATURE (°C) 25 50 75 100 125 150 TEMPERATURE (°C) 3795 G03 VFB (V) V(ISP-ISN) THRESHOLD (mV) V(ISP-ISN) (mV) 300 50 251 3795 G02 128 25 252 VISP (V) V(ISP-ISN) Threshold at CTRL1 = 0.6V, CTRL2 = 1.2V vs Temperature 0 ISP = 48V CTRL1= CTRL2 = 2V 253 3795 G01 122 –50 –25 254 252 247 1.4 V(ISP-ISN) Full-Scale Threshold vs Temperature V(ISP-ISN) Threshold vs VISP V(ISP-ISN) THRESHOLD (mV) 250 V(ISP-ISN) THRESHOLD (mV) V(ISP-ISN) THRESHOLD (mV) 300 V(ISP-ISN) Threshold vs VCTRL TA = 25°C, unless otherwise noted. 1.23 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3795 G04 3795 G05 3795f For more information www.linear.com/LT3795 5 LT3795 Typical Performance Characteristics ISP/ISN Overcurrent Protection Threshold vs Temperature ISP/ISN Input Bias Current vs VISP , VISN VREF Voltage vs Temperature 900 380 2.05 800 376 374 372 2.04 ISP 700 600 2.02 500 400 0 25 50 75 200 1.98 0 100 125 150 1.97 ISN 0 20 40 60 80 100 1.96 –50 –25 120 VREF vs VIN 100 100 125 150 Switching Frequency vs Temperature 440 RT = 19.6k 430 SWITCHING FREQUENCY (kHz) 2.02 RT (kΩ) VREF (V) 75 3795 G08 RT vs Switching Frequency (kHz) 2.01 10 2.00 1.99 420 410 400 390 380 370 1.98 0 20 40 60 80 100 1 120 VIN (V) 25 50 75 100 125 150 TEMPERATURE (°C) 3795 G11 VISMON vs V(ISP-ISN) Quiescent Current vs VIN 3.0 RT = 19.6k 400 0 3795 G10 Switching Frequency vs SS Voltage 450 360 –50 –25 0 100 200 300 400 500 600 700 800 900 1000 SWITCHING FREQUENCY (kHz) 3795 G09 2000 1800 2.5 300 250 200 150 100 1600 1400 2.0 VISMON (mV) 350 VIN CURRENT (mA) SWITCHING FREQUENCY (kHz) 50 3795 G07 2.03 1.5 1.0 1200 1000 800 600 400 0.5 50 200 0 200 400 600 800 1000 1200 SS VOLTAGE (mV) 0 0 20 40 60 80 120 100 0 0 100 200 300 400 500 V(ISP-ISN) (mV) VIN (V) 3795 G11a 6 25 TEMPERATURE (°C) 2.04 0 0 VISP, VISN (V) 3795 G06 1.97 IREF = –100µA 2.00 1.99 TEMPERATURE (°C) 2.05 2.01 300 100 370 –50 –25 IREF = 0µA 2.03 VREF (V) 378 ISP, ISN BIAS CURRENT (µA) ISP/ISN OVERCURRENT THRESHOLD (mV) TA = 25°C, unless otherwise noted. 3795 G12 3795 G13 3795f For more information www.linear.com/LT3795 LT3795 Typical Performance Characteristics EN/UVLO Falling/Rising Threshold vs Temperature EN/UVLO Hysteresis Current vs Temperature 3.0 EN/UVLO (V) 2.5 2.0 1.5 SENSE Current Limit Threshold vs Temperature 1.28 118 1.27 117 1.26 116 SENSE THRESHOLD (mV) 3.5 EN/UVLO HYSTERESIS CURRENT (µA) TA = 25°C, unless otherwise noted. EN/UVLO RISING THRESHOLD 1.25 1.24 1.23 1.22 1.0 EN/UVLO FALLING THRESHOLD 1.21 0.5 0 25 50 75 100 125 150 0 25 50 75 105 80 VIN = 24V 80 60 40 20 0 20 60 40 DUTY CYCLE (%) 80 100 VIN = 48V 70 60 50 –50 –25 120 0 8 7 1400 125°C 3795 G20 7.8 75°C 25°C 800 0°C 600 0 7.7 7.6 7.5 400 7.4 200 0 10 20 30 40 50 60 70 80 90 100 110 VIN (V) 7.9 150°C 1000 1 100 125 150 8.0 VIN = 6V 1200 2 75 INTVCC vs Temperature INTVCC (V) 1800 1600 INTVCC DROPOUT (mV) 9 3 50 3795 G19 INTVCC Dropout Voltage vs Current, Temperature 4 25 3795 G18 INTVCC vs VIN INTVCC (V) 80 TEMPERATURE (°C) 3795 G17 0 90 VIN (V) 5 100 125 150 100 100 0 100 6 75 INTVCC Current Limit vs Temperature INTVCC CURRENT LIMIT (mA) INTVCC CURRENT LIMIT (mA) SENSE THRESHOLD (mV) 110 50 3795 G16 INTVCC Current Limit vs VIN 115 25 TEMPERATURE (°C) 120 60 0 3795 G15 120 40 111 TEMPERATURE (°C) SENSE Current Limit Threshold vs Duty Cycle 20 112 108 –50 –25 100 125 150 3795 G14 0 113 109 1.19 –50 –25 TEMPERATURE (°C) 100 114 110 1.20 0 –50 –25 115 –55°C –40°C 0 5 10 15 20 INTVCC LOAD (mA) 7.3 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3795 G21 3795 G22 3795f For more information www.linear.com/LT3795 7 LT3795 Typical Performance Characteristics V(IVINP-IVINN) Threshold vs Temperature V(IVINP-IVINN) Threshold vs VIN 64 1.6 63 63 1.4 62 62 V(IVINP-IVINN) (mV) VIN = 24V 61 60 59 58 VIVINCOMP (V) 1.8 64 61 60 59 58 0.8 0.6 0.4 56 56 0.2 25 50 75 55 100 125 150 0 20 TEMPERATURE (°C) 40 60 80 100 SOURCING TIME (ns) 10 160 800 700 120 600 100 RISE TIME 80 60 FALL TIME 50 75 100 125 150 0 0 10 20 30 40 85V 75V RISE TIME 50 0 0 1 2 100ns/DIV PMOS VISHAY SILICONIX Si7113DN 3 4 5 6 7 8 9 10 CAPACITANCE (nF) 3795 G27 Top Gate Driver Rising Edge TG 300 CAPACITANCE (nF) 3795 G26 5V 0V FALL TIME 400 100 TEMPERATURE (°C) PWM 500 200 20 25 80 Top Gate (PMOS) Rise/Fall Time vs Capacitance 140 40 9 0 60 3795 G25 TIME (ns) SINKING 11 40 V(IVINP-IVINN) (mV) NMOS Gate Rise/Fall Time vs Capacitance 13 8 –50 –25 20 0 3795 G24 RAMP Pin Sourcing and Sinking Current vs Temperature 12 0 120 VIN (V) 3795 G23 8 1.0 57 0 VIN = 24V 1.2 57 55 –50 –25 CURRENT (µA) VIVINCOMP vs V(IVINP-IVINN) 65 65 V(IVINP-IVINN) (mV) TA = 25°C, unless otherwise noted. 3795 G28 Top Gate Driver Falling Edge PWM 5V 0V TG 85V 75V 3795 G29 100ns/DIV 3795 G30 PMOS VISHAY SILICONIX Si7113DN 3795f For more information www.linear.com/LT3795 LT3795 Pin Functions ISP (Pin 1): Connection Point for the Positive Terminal of the Current Feedback Resistor (RLED). Also serves as positive rail for TG pin driver. ISN (Pin 2): Connection Point for the Negative Terminal of the Current Feedback Resistor (RLED). TG (Pin 3): Top Gate Driver Output. An inverted PWM signal drives the gate of a series PMOS device between VISP and (VISP – 7V) if VISP > 7V. An internal 7V clamp protects the PMOS gate by limiting VGS. Leave TG unconnected if not used. GND (Pins 4, 17, 21, 22, Exposed Pad Pin 29): Ground. These pins also serve as current sense input for control loop, sensing the negative terminal of the current sense resistor in the source of the N-channel MOSFET. Solder the exposed pad directly to the ground plane. ISMON (Pin 5): ISP/ISN Current Report Pin. The LED current sensed by ISP/ISN inputs is reported as VISMON = ILED • RLED • 4. Leave ISMON pin unconnected if not used. When PWM is low, ISMON is driven to ground. Bypass with a 47nF capacitor or higher if needed. CTRL2 (Pin 6): Current Sense Threshold Adjustment Pin 2. This pin has identical functions as CTRL1. The V(ISP-ISN) threshold is regulated by the internal 1.1V reference voltage, CTRL1 or CTRL2. Whichever is the lowest takes precedence. Regulating threshold V(ISP-ISN) is 0.25 • VCTRLX less an offset for 0.1V < VCTRLX < 1V. For VCTRLX > 1.2V the current sense threshold is constant at the full-scale value of 250mV. For 1V < VCTRLX < 1.2V, the dependence of the current sense threshold upon VCTRLX transitions from a linear function to a constant value, reaching 98% of full-scale value by VCTRLX = 1.1V. Connect CTRL1 and CTRL2 to VREF for the 250mV default current threshold. Do not leave this pin open. Connect either CTRL pin to GND for zero LED current. FB (Pin 7): Voltage Loop Feedback Pin. FB is intended for constant-voltage regulation or for LED protection/open LED detection. The internal transconductance amplifier with output VC regulates FB to 1.25V (nominal) through the DC/DC converter. If the FB input is regulating the loop, and V(ISP-ISN) is less than 25mV (typical), the OPENLED pull-down is asserted. This action may signal an open LED fault. If FB is driven above the 1.3V (by an external power supply spike, for example), the GATE pin is pulled low to turn off the external N-channel MOSFET and the TG pin is driven high to protect the LEDs from an overcurrent event. Do not tie this pin to GND as the SHORTLED will be asserted and the part will be shut down. VC (Pin 8): Transconductance Error Amplifier Output Pin. Used to stabilize the control loop with an RC network. This pin is high impedance when PWM is low, a feature that stores the demand current state variable for the next PWM high transition. Connect a capacitor between this pin and GND; a resistor in series with the capacitor is recommended for fast transient response. Do not leave this pin open. CTRL1 (Pin 9): Current Sense Threshold Adjustment Pin 1. This pin has an identical function as CTRL2. Please refer to the CTRL2 pin description. VREF (Pin 10): Voltage Reference Output Pin. Typically 2.015V. This pin drives a resistor divider for the CTRL pins, either for analog dimming or for temperature limit/ compensation of the LED load. It can supply up to 100μA. SS (Pin 11): Soft-Start Pin. This pin modulates oscillator frequency and compensation pin voltage (VC) clamp. The soft-start interval is set with an external capacitor. The pin has a 28μA (typical) pull-up current source to an internal 2.5V rail. This pin can be used as fault timer. Provided the SS pin has exceeded 1.7V to complete a blanking period at start-up, the pull-up current source is disabled and a 2.8µA pull-down current enabled when any one of the following fault conditions happen: 1. LED overcurrent (ISP-ISN > 0.375V) 2. INTVCC undervoltage 3. Output short (FB < 0.3V after start-up) 4. Thermal limit The SS pin must be discharged below 0.2V to reinitiate a soft-start cycle. Switching is disabled until SS begins to recharge. It is important to select a capacitor large enough that FB can exceed 0.3V under normal load conditions before SS exceeds 1.7V. Do not leave this pin open. RT (Pin 12): Switching Frequency Adjustment Pin. Set the frequency using a resistor to GND (for resistor values, see the Typical Performance curve or Table 2). Do not leave the RT pin open. 3795f For more information www.linear.com/LT3795 9 LT3795 Pin Functions RAMP (Pin 13): The RAMP pin is used for spread spectrum frequency modulation. The internal switching frequency is spread out to 70% of the original value, where the modulation frequency is set by 12µA/(2 • 1V • CRAMP). If not used, tie this pin to GND. INTVCC (Pin 20): Regulated Supply for Internal Loads and GATE Driver. Supplied from VIN and regulates to 7.7V (typical). INTVCC must be bypassed with a 4.7μF capacitor placed close to the pin. Connect INTVCC directly to VIN if VIN is always less than or equal to 8V. PWM (Pin 14): PWM Input Signal Pin. A low signal turns off switching, idles the oscillator, disconnects the VC pin from all internal loads, and drives TG to the ISP level. PWM has an internal 500k pull-down resistor. If not used, connect to VREF. VIN (Pin 23): Input Supply Pin. Must be locally bypassed with a 0.22μF (or larger) capacitor placed close to the IC. SHORTLED (Pin 15): An open-collector pull-down on SHORTLED asserts when any of the following conditions happen: 1. FB < 0.3V after SS pin reaches 1.7V at start-up. 2. LED overcurrent (V(ISP-ISN) > 375mV). To function, the pin requires an external pull-up resistor. SHORTLED status is only updated during PWM high state and latched during PWM low state. SHORTLED remains asserted until the SS pin is discharged below 0.2V. OPENLED (Pin 16): An open-collector pull-down on OPENLED asserts if the FB input is above 1.20V (typical), and V(ISP-ISN) is less than 25mV (typical). To function, the pin requires an external pull-up resistor. OPENLED status is updated only during PWM high state and latched during PWM low state. SENSE (Pin 18): The current sense input for the control loop. Kelvin connect this pin to the positive terminal of the switch current sense resistor, RSENSE, in the source of the N-channel MOSFET. The negative terminal of the current sense resistor should be Kelvin connected to the GND plane of the IC. GATE (Pin 19): N-Channel MOSFET Gate Driver Output. Switches between INTVCC and GND. It is driven to GND during shutdown, fault or idle states. 10 EN/UVLO (Pin 24): Enable and Undervoltage Lockout Pin. An accurate 1.22V falling threshold with externally programmable hysteresis detects when power is OK to enable switching. Rising hysteresis is generated by the external resistor divider and an accurate internal 3μA pull-down current. Above the threshold, EN/UVLO input bias current is sub-μA. Below the falling threshold, a 3μA pull-down current is enabled so the user can define the hysteresis with the external resistor selection. An undervoltage condition resets soft-start. Tie to 0.4V, or less, to disable the device. OVLO (Pin 25): Input Overvoltage Lockout Pin. An accurate 1.25V rising threshold detects when power is OK to enable switching. IVINN (Pin 26): Connection Point for the Negative Terminal of the Input Current Sense Resistor (RINSNS). The input current can be programmed by IIN = 60mV/RINSNS. IVINP (Pin 27): Connection Point for the Positive Terminal of the Input Current Sense Resistor. IVINCOMP (Pin 28): Input Current Sense Amplifier Output Pin. The voltage at IVINCOMP pin is proportional to IIN as VIVINCOMP = IIN • RINSNS • 20. A 10nF or larger capacitor to GND is required at this pin to compensate the input current loop. Do not leave this pin open, and do not load this pin with a current. 3795f For more information www.linear.com/LT3795 LT3795 Block Diagram EN/UVLO FB A1 – SHDN 1.22V A3 7.7V + INTVCC + SHORT-CIRCUIT DETECT – 1.5V TGOFFB SCILM A4 + ×1 A5 + 100mV –+ ×4 ISN VIN LDO – 1.25V gm 1.25V ISP PWM ISP-7V ISP 10µA AT FB = 1.25V A2 – + 3µA ISMON TG VC VLED 1.1V – CTRL1 2.5V gm EAMP 10µA AT A6+ = A6– – + + + FAULTB + 10µA R A7 A6 – S GATE DRIVER Q PWM COMPARATOR ILIM + CTRL2 113mV A9 – 10µA AT IVINCOMP = 1.2V gm R1 IVINP A8 + 1.2V ISENSE – R2 = R1 2.5V A11 + 28µA THERMAL SHDN IVINCOMP 5.5V SENSE – GND A10 – IVINN + PWM R3 = R1 × 20 RAMP GENERATOR SS OPENLED 100kHz TO 1MHz OSCILLATOR TGOFFB SS AND LOGIC FAULTB SHORTLED – 1.20V A14 A7 SCILM 1mA 2.8µA FB + VLED – VINOV OVFB 1.25V FB + OVLO EN – A16 0.3V INTVCC SCILM A17 FB + 1.3V – A21 + 200mV C/10 COMPARATOR WITH 200mV HYSTERESIS OVFB COMPARATOR + – A15 A7 INTVCC 12µA 12µA 2V INTVCC – A7 A19 + 100µA – – A7 A20 A18 + 2.015V 1V R S Q + PWM VREF SS RT RAMP 3796 BD 3795f For more information www.linear.com/LT3795 11 LT3795 Operation The LT3795 is a constant-frequency, current mode controller with a low side NMOS gate driver. The operation of the LT3795 is best understood by referring to the Block Diagram of the IC. In normal operation, with the PWM pin low, the GATE pin is driven to GND, the TG pin is pulled high to ISP to turn off the PMOS disconnect switch, the VC pin goes high impedance to store the previous switching state on the external compensation capacitor, and the ISP and ISN pin bias currents are reduced to leakage levels. When the PWM pin transitions high, the TG pin transitions low after a short delay. At the same time, the internal oscillator wakes up and generates a pulse to set the PWM latch, turning on the external power N-channel MOSFET switch (GATE goes high). A voltage input proportional to the switch current, sensed by an external current sense resistor between the SENSE and GND input pins, is added to a stabilizing slope compensation ramp and the resulting switch current sense signal is fed into the negative terminal of the PWM comparator. The current in the external inductor increases steadily during the time the switch is on. When the switch current sense voltage exceeds the output of the error amplifier, labeled VC, the latch is reset and the switch is turned off. During the switch off phase, the inductor current decreases. At the completion of each oscillator cycle, internal signals such as slope compensation return to their starting points and a new cycle begins with the set pulse from the oscillator. Through this repetitive action, the PWM control algorithm establishes a switch duty cycle to regulate a current or voltage in the load. The VC signal is integrated over many switching cycles and is an amplified version of the difference between the LED current sense voltage, measured between ISP and ISN, and the target difference voltage set by the CTRL1 or CTRL2 pin. In this manner, the error amplifier sets the correct peak switch current level to keep the LED current in regulation. If the error amplifier output increases, more current is demanded in the switch; if it decreases, less current is demanded. The switch current is monitored during the on phase and the voltage across the SENSE pin is not allowed to exceed the current limit threshold of 113mV (typical). If the SENSE pin exceeds the current limit threshold, the SR latch is reset regardless of the output state of the PWM comparator. Likewise, any fault condition, i.e. FB overvoltage (FB > 1.3V), output short (FB < 0.3V) after start-up, input overvoltage (OVLO > 12 1.25V) LED overcurrent, or INTVCC undervoltage (INTVCC < 4V), the GATE pin is pulled down to GND immediately. In voltage feedback mode, the operation is similar to that described above, except the voltage at the VC pin is set by the amplified difference of the internal reference of 1.25V (nominal) and the FB pin. If FB is lower than the reference voltage, the switch current increases; if FB is higher than the reference voltage, the switch demand current decreases. The LED current sense feedback interacts with the voltage feedback so that FB does not exceed the internal reference and the voltage between ISP and ISN does not exceed the threshold set by either of the CTRL pins. For accurate current or voltage regulation, it is necessary to be sure that under normal operating conditions, the appropriate loop is dominant. To deactivate the voltage loop entirely, FB can be set between 0.4V and 1V through a resistor network to VREF pin. To deactivate the LED current loop entirely, the ISP and ISN should be tied together and CTRL1 and CTRL2 tied to VREF. Two LED specific functions featured on the LT3795 are controlled by the voltage feedback FB pin. First, when the FB pin exceeds a voltage 52mV lower (–4%) than the FB regulation voltage and V(ISP-ISN) is less than 25mV (typical), the pull-down driver on the OPENLED pin is activated. This function provides a status indicator that the load may be disconnected and the constant-voltage feedback loop is taking control of the switching regulator. When the FB pin drops below 0.3V after start-up, the SHORTLED pin is asserted by comparator A16. A blanking period occurs during start-up for the SHORTLED protection feature from the EN/UVLO toggle until the SS pin reaches 1.7V. LT3795 features a PMOS disconnect switch driver. The PMOS disconnect switch can be used to improve the PWM dimming ratio, and operate as fault protection as well. Once a fault condition is detected, the TG pin is pulled high to turnoff the PMOS switch. The action isolates the LED array from the power path, preventing excessive current from damaging the LEDs. A standalone input current sense amplifier is integrated in the LT3795. The input current sense amplifier A11 senses the input current and converts it to a voltage signal at the IVINCOMP pin. When the voltage potential at the IVINCOMP pin moves close to 1.2V, the amplifier A8 starts to interact with the VC pin, and thus reduces the regulated LED current. In this way, the input current is limited. For more information www.linear.com/LT3795 3795f LT3795 Applications Information INTVCC Regulator Bypassing and Operation The INTVCC pin requires a capacitor for stable operation and to store the charge for the large GATE switching currents. Choose a 10V rated low ESR, X7R or X5R ceramic capacitor for best performance. A 4.7μF ceramic capacitor is adequate for many applications. Place the capacitor close to the IC to minimize the trace length to the INTVCC pin and also to the IC ground. An internal current limit on the INTVCC output protects the LT3795 from excessive on-chip power dissipation. The minimum value of this current limit should be considered when choosing the switching N-channel MOSFET and the operating frequency. IINTVCC can be calculated from the following equation: VIN EN/UVLO R2 3795 F01 Figure 1. Programming the Overvoltage Lockout Threshold with the OVLO Pin The input overvoltage lockout protection feature can be implemented by a resistor from the VIN to OVLO pins as shown in Figure 2. The following equations should be used to determine the values of the resistors: VIN,OVLO = 1.25 • IINTVCC = QG • fOSC Careful choice of a lower QG MOSFET allows higher switching frequencies, leading to smaller magnetics. The INTVCC pin has its own undervoltage disable (UVLO) set to 4V (typical) to protect the external FETs from excessive power dissipation caused by not being fully enhanced. If the INTVCC pin drops below the UVLO threshold, the GATE pin is forced to 0V, TG pin is pulled high and the soft-start pin will be reset. If the input voltage, VIN, will not exceed 8V, then the INTVCC pin should be connected to the input supply. Be aware that a small current (typically 13μA) loads the INTVCC in shutdown. If VIN is normally above, but occasionally drops below the INTVCC regulation voltage, then the minimum operating VIN is close to 4.5V. This value is determined by the dropout voltage of the linear regulator and the 4V INTVCC undervoltage lockout threshold mentioned above. Programming the Turn-On and Turn-Off Thresholds with the EN/UVLO Pin The falling UVLO value can be accurately set by the resistor divider. A small 3μA pull-down current is active when EN/ UVLO is below the threshold. The purpose of this current is to allow the user to program the rising hysteresis. The following equations should be used to determine the values of the resistors: R1+R2 VIN(FALLING) = 1.22 • R2 VIN(RISING) = VIN(FALLING) + 3µA •R1 R1 LT3795 R3+R4 R4 VIN R3 LT3795 OVLO R4 3795 F02 Figure 2. LED Current Programming The LED current is programmed by placing an appropriate value current sense resistor RLED between the ISP and ISN pins. For best fault protection provided by the high side PMOS disconnect switch, sensing of the current should be done at the top of the LED string. If this option is not available, then the current may be sensed at the bottom of the string. Both the CTRL pins should be tied to a voltage higher than 1.2V to get the full-scale 250mV (typical) threshold across the sense resistor. Either CTRL pin can also be used to dim the LED current to zero, although relative accuracy decreases with the decreasing voltage sense threshold. The two CTRL pins have identical functions. Whichever is the lowest takes precedence. When the lower CTRL pin voltage is less than 1V, the LED current is: V – 100mV ILED = CTRL , 0.1V< VCTRL < 1V RLED • 4 ILED = 0, VCTRL = 0V 3795f For more information www.linear.com/LT3795 13 LT3795 Applications Information When the lower CTRL pin voltage is between 1V and 1.2V, the LED current varies with CTRL, but departs from the previous equation by an increasing amount as the CTRL voltage increases. Ultimately above 1.2V, the LED current no longer varies with CTRL. The typical V(ISP-ISN) threshold vs CTRL is listed in Table 1. Table 1. V(ISP-ISN) Threshold vs CTRL VCTRL (V) V(ISP-ISN) (mV) 1 225 1.05 236 1.1 244.5 1.15 248.5 1.2 250 When both the CTRL pins are higher than 1.2V, the LED current is regulated to: ILED = FB R6 3795 F03 Figure 3. Feedback Resistor Connections for Boost and SEPIC Applications For an LED driver of buck mode or a buck-boost mode configuration, the FB voltage is typically level shifted to a signal with respect to GND as illustrated in Figure 4. The output can be expressed as: VOUT = 1.25 • R7 +V R8 BE(Q1) R7 The LT3795 has a voltage feedback pin FB that can be used to program a constant-voltage output. In addition, FB programming determines the output voltage that will cause OPENLED and SHORTLED to assert. For a boost LED driver, the output voltage can be programmed by selecting the values of R5 and R6 (see Figure 3) according to the following equation: R5+R6 R6 + VOUT Programming Output Voltage (Constant-Voltage Regulation) and Output Voltage Open LED and Shorted LED Thresholds VOUT = 1.25 • R5 LT3795 250mV RLED The CTRL pins should not be left open (tie to VREF if not used). Either CTRL pin can also be used in conjunction with a thermistor to provide overtemperature protection for the LED load, or with a resistor divider to VIN to reduce output power and switching current when VIN is low. The presence of a time varying differential voltage signal (ripple) across ISP and ISN at the switching frequency is expected. The amplitude of this signal is increased by high LED load current, low switching frequency and/or a smaller value output filter capacitor. For best accuracy, the amplitude of this ripple should be less than 25mV. 14 VOUT LT3795 RSENSE LED ARRAY – Q1 FB 3795 F04 R8 Figure 4. Feedback Resistor Connection for Buck Mode or Buck-Boost Mode LED Driver If the open LED clamp voltage is programmed correctly using the resistor divider, then the FB pin should never exceed 1.2V when LEDs are connected. To detect both open-circuit and short-circuit conditions at the output, the LT3795 monitors both output voltage and current. When FB exceeds VFB - 52mV, OPENLED is asserted if V(ISP-ISN) is less than 25mV. OPENLED is deasserted when V(ISP-ISN) is higher than 70mV (typical) or FB drops below VFB - 62mV (typical). The SHORTLED pin is asserted if V(ISP-ISN) > 375mV or the FB pin falls below 300mV (typical) after initial start-up and SS reaches 1.7V. The ratio between the FB OPENLED threshold of 1.2V and the SHORTLED threshold of 0.3V can limit the range of VOUT. The range of VOUT using the maximum SHORTLED threshold of 0.35V is 3.5:1. The range of VOUT can be made wider using the circuits shown 3795f For more information www.linear.com/LT3795 LT3795 Applications Information in Figure 5 and Figure 6. For a VOUT range that is greater than 8:1, consult factory applications. R10 FB R12 = 1000 R12 VREF R11 3795 F05 Figure 5. Feedback Resistor Connection for Wide Range Output in Boost and SEPIC Applications + RSENSE VOUT – Q1 LT3795 LED ARRAY R15 FB VREF R14 3795 F06 Figure 6. Feedback Resistor Connection for Wide Range Output in Buck Mode and Buck-Boost Mode Applications The equations to widen the range of VOUT are derived using a SHORTLED threshold of 0.35V, an OPENLED threshold of 1.2V and a reference voltage VREF of 2V. The resistor values for R11 and R12 in Figure 5 can be calculated as shown below. See the example that follows for a suggested R10 value. R11= R10• R12 = R10• 1.7 1.65• VHOUT – 0.8• VL OUT – 1.7 1.7 0.35• VHOUT – 1.2• VL OUT R15 = R13• 1.7 1.65• VHOUT – 0.8• VL OUT – 0.85• VBE (Q1) 1.7 0.35• VHOUT – 1.2• VL OUT + 0.85• VBE (Q1) Example: Calculate the resistor values required to increase the VOUT range of a buck-boost mode LED driver to 7.5:1 and have OPENLED occur when VOUT is 43.5V. Use VBE(Q1) = 0.7V: Step 1: Choose R13 = 357k Step 2: VLOUT = 43.5/7.5 = 5.8 Step 3: R14 = 357 1.7 = 9.12, (1.65) 43.5 –(0.8) 5.8 –(0.85) 0.7 Use R14 = 9.09kΩ R15 = 357 1.7 = 68.5, (0.35) 43.5 –(1.2) 5.8 +(0.85) 0.7 Use R15 = 68.1kΩ LED Overcurrent Protection Feature Example: Calculate the resistor values required to increase the VOUT range of a boost LED driver to 5:1 and have OPENLED occur when VOUT is 91.4V: Step 1: Choose R10 = 1M Step 2: VLOUT = 91.4/5 = 18.3 Step 3: 1.7 = 169kΩ (0.35) 91.4 –(1.2) 18.3 The resistor values for R14 and R15 in Figure 6 can be calculated as shown below. See the example that follows for a suggested R13 value. R14 = R13• R13 1.7 = 12.64, (1.65) 91.4 –(0.8) 18.3 – 1.7 Use R11= 12.7kΩ VOUT LT3795 R11= 1000 The ISP and ISN pins have a short-circuit protection feature independent of the LED current sense feature. This feature prevents the development of excessive switching currents and protects the power components. The short-circuit protection threshold (375mV, typ) is designed to be 50% higher than the default LED current sense threshold. Once the LED overcurrent is detected, the GATE pin is driven to GND to stop switching, TG pin is pulled high to disconnect the LED array from the power path, and fault protection is initiated via the SS pin. 3795f For more information www.linear.com/LT3795 15 LT3795 Applications Information D1 L1 VIN C1 VIN C2 M1 GATE SENSE LT3795 RSNS ISP RLED ISN TG M2 LED+ LED STRING D2 GND (BOOST) OR VIN (BUCK-BOOST MODE) 3795 F07 Figure 7. The Simplified LED Short-Circuit Protection Schematic for Boost or Buck-Boost Mode Converter LED 50V/DIV IM2 5A/DIV SHORTLED 5V/DIV 3795 F08 Figure 8. Short-Circuit Current VIN RLED M2 LED+ C2 LED STRING VIN ISP ISN TG LT3795 GATE D3 LED– L1 M1 D2 D1 C1 SENSE 3795 F09 RSNS Figure 9. The Simplified LED Short-Circuit Protection Schematic for Buck Mode Converter 16 Schottky or UltraFast recovery diodes D2 and D3 are recommended to protect against a short circuit for the buck mode circuit shown in Figure 9. PWM Dimming Control for Brightness There are two methods to control the LED current for dimming using the LT3795. One method uses the CTRL pins to adjust the current regulated in the LEDs. A second method uses the PWM pin to modulate the LED current between zero and full current to achieve a precisely programmed average current, without the possibility of color shift that occurs at low current in LEDs. To make PWM dimming more accurate, the switch demand current is stored on the VC node during the quiescent phase when PWM is low. This feature minimizes recovery time when the PWM signal goes high. To further improve the recovery time, a disconnect switch should be used in the LED current path to prevent the output capacitor from discharging during the PWM signal low phase. The minimum PWM on or off time depends on the choice of operating frequency set by the RT input. For best current accuracy, the minimum PWM high time should be at least three switching cycles (3μs for fSW = 1MHz). + 500ns/DIV A typical LED short-circuit protection scheme for a boost or buck-boost mode converter is shown in Figure 7. The Schottky or ultrafast diode D2 should be put close to the drain of M2 on the board. It protects the LED+ node from swinging well below ground when being shorted to ground through a long cable. Usually, the internal protection loop takes about 100ns to respond as shown in Figure 8. Refer to the Short-Circuit Robust Boost LED Driver with Input Current Limit and Spread Spectrum Frequency Modulation application circuit for the test schematic. Note that the impedance of the short-circuit cable affects the peak current. A low duty cycle PWM signal can cause excessive startup times if it were allowed to interrupt the soft-start sequence. Therefore, once start-up is initiated by PWM > 1V, the LT3795 will ignore a logical disable by the external PWM input signal. The device will continue to soft-start with switching and TG enabled until either the voltage at SS reaches the 1.0V level, or the output current reaches one-fourth of the full-scale current. At this point the device 3795f For more information www.linear.com/LT3795 LT3795 Applications Information will begin following the dimming control as designated by PWM. If at any time an output overcurrent is detected, GATE and TG will be disabled even as SS continues to charge. the Boost LED Driver with Input Current Limit and Spread Spectrum Frequency Modulation application circuit). 100 RAMP PIN GROUNDED 90 Programming the Switching Frequency AMPLITUDE (dBµV) 80 The RT frequency adjust pin allows the user to program the switching frequency from 100kHz to 1MHz to optimize efficiency/performance or external component size. Higher frequency operation yields smaller component size but increases switching losses and gate driving current, and may not allow sufficiently high or low duty cycle operation. Lower frequency operation gives better performance at the cost of larger external component size. For an appropriate RT resistor value see Table 2. An external resistor from the RT pin to GND is required—do not leave this pin open. 70 47nF AT RAMP PIN 60 50 40 30 20 EMI AVERAGE SA DETECTOR 10 0 150 250 200 300 FREQUENCY (kHz) 350 3796 F10 Figure 10. Input Noise Spectrum Comparison Table 2. Typical Switching Frequency vs RT Value (1% Resistor) RT(kΩ) 1000 6.65 900 7.50 800 8.87 700 10.2 600 12.4 500 15.4 400 19.6 300 26.1 200 39.2 100 82.5 Spread Spectrum Frequency Modulation Switching regulators can be particularly troublesome for applications where electromagnetic interference (EMI) is a concern. To improve the EMI performance, the LT3795 includes a spread spectrum frequency feature. If there is a capacitor (CRAMP)at the RAMP pin, a triangle wave sweeping between 1V and 2V is generated. This signal is then fed into the internal oscillator to modulate the switching frequency between 70% of the base frequency and the base frequency, which is set by the RT resistor. The modulation frequency is set by 12µA/(2 • 1V • CRAMP).Figure 10 shows the noise spectrum comparison between a conventional boost switching converter (with the LT3795 RAMP pin tied to GND) and a spread spectrum modulation enabled boost switching converter with 47nF at the RAMP pin (refer to Duty Cycle Considerations Switching duty cycle is a key variable defining converter operation, therefore, its limits must be considered when programming the switching frequency for a particular application. The fixed minimum on-time and minimum off-time (see Figure 11) and the switching frequency define the minimum and maximum duty cycle of the switch, respectively. The following equations express the minimum/ maximum duty cycle: Minimum Duty Cycle = minimum on-time • switching frequency Maximum Duty Cycle = 1 – minimum off-time • switching frequency 350 300 250 TIME (ns) fOSC(kHz) MINIMUM ON-TIME 200 MINIMUM OFF-TIME 150 100 50 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3796 F11 Figure 11. Typical Minimum On- and Off-Time vs Temperature 3795f For more information www.linear.com/LT3795 17 LT3795 Applications Information When calculating the operating limits, the typical values for on/off-time in the data sheet should be increased by at least 100ns to allow margin for PWM control latitude, GATE rise/fall times and SW node rise/fall times. Thermal Considerations Setting Input Current Limit The LT3795 has a standalone input current sense amplifier to limit the input current. The input current IIN shown in Figure 12 is converted to a voltage output at the IVINCOMP pin. When the IVINCOMP voltage exceeds 1.2V the GATE is pulled low, and the converter stops switching. The input current limit is calculated as follows: IIN = 60mV RINSNS IIN VIN COPT A low QG power MOSFET should always be used when operating at high input voltages, and the switching frequency should also be chosen carefully to ensure that the IC does not exceed a safe junction temperature. The internal junction temperature, TJ of the IC can be estimated by: TO LOAD RIN(OPT) IVINN TJ = TA + [VIN • (IQ + fSW • QG) •θJA] LT3795 IVINCOMP 3795 F12 CFILT Figure 12. Setting Input Current Limit Filter capacitor CFILT shown in Figure 12 filters the voltage at the IVINCOMP pin to minimize ripple due to the input current. CFILT also compensates the input current regulation loop, and is selected based on the loop response in addition to the intended voltage ripple on IVINCOMP. The IVINCOMP pin resistance to ground and CFILT form a second pole in the input current regulation loop in addition to the dominant pole at VC pin. Suggested values for CFILT of 10nF - 0.1µF will usually provide a second pole in the input current regulation loop that results in stable loop response and is equivalent to the second pole in the ISP/ISN regulation loop, which consists of the output capacitance COUT and the dynamic resistance of the LED load. For buck mode applications, filter components, RIN(OPT) and COPT, 18 The LT3795 is rated to a maximum input voltage of 110V. Careful attention must be paid to the internal power dissipation of the IC at higher input voltages to ensure that a junction temperature of 150°C is not exceeded. This junction limit is especially important when operating at high ambient temperatures. The majority of the power dissipation in the IC comes from the supply current needed to drive the gate capacitance of the external power N-channel MOSFET. This gate drive current can be calculated as: IGATE = fSW • QG RINSNS IVINP can be placed close to LT3795 to suppress substantial transient signal or noise at the IVINN and IVINP pins. For boost and buck-boost mode applications, RIN(OPT) and COPT are not required. where TA is the ambient temperature, IQ is the quiescent current of the part (2.9mA typical) and θJA is the package thermal impedance (30°C/W for the TSSOP package). For example, an application with TA(MAX) = 85°C, VIN(MAX) = 60V, fSW = 400kHz, and having a N-channel MOSFET with QG = 20nC, the maximum IC junction temperature will be approximately: TJ = 85°C + [60V • (2.9mA + 400kHz • 20nC) • 30°C/W] ≈ 104.6°C The exposed pad on the bottom of the package must be soldered to a ground plane. This ground should then be connected to an internal copper ground plane with thermal vias placed directly under the package to spread out the heat dissipated by the IC. It is best if the copper plane is extended on either the top or bottom layer of the PCB to have the maximum exposure to air. Internal ground layers do not dissipate thermals as much as top and bottom layer copper does. See the recommended layout as an example. 3795f For more information www.linear.com/LT3795 LT3795 Applications Information Input Capacitor Selection Table 3. Recommended Ceramic Capacitor Manufacturers The input capacitor supplies the transient input current for the power inductor of the converter and must be placed and sized according to the transient current requirements. The switching frequency, output current and tolerable input voltage ripple are key inputs to estimating the capacitor value. An X7R type ceramic capacitor is usually the best choice since it has the least variation with temperature and DC bias. Typically, boost and SEPIC converters require a lower value capacitor than a buck mode converter. Assuming that a 100mV input voltage ripple is acceptable, the required capacitor value for a boost converter can be estimated as follows (TSW = 1/fOSC): CIN(µF) =ILED(A)• VLED 1µF • TSW(µs)• VIN A •µs • 2.8 Therefore, a 2.2µF capacitor is an appropriate selection for a 400kHz boost regulator with 12V input, 48V output and 500mA load. With the same VIN voltage ripple of less than 100mV, the input capacitor for a buck mode converter can be estimated as follows: CIN(µF)=ILED(A)• VLED(VIN – V LED) 10µF • TSW(µs)• 2 VIN A •µs A 10µF input capacitor is an appropriate selection for a 400kHz buck mode converter with 24V input, 12V output and 1A load. In the buck mode configuration, the input capacitor has large pulsed currents due to the current returned through the Schottky diode when the switch is off. It is important to place the capacitor as close as possible to the Schottky diode and to the GND return of the switch (i.e., the sense resistor). It is also important to consider the ripple current rating of the capacitor. For best reliability, this capacitor should have low ESR and ESL and have an adequate ripple current rating. The RMS input current for a buck mode LED driver is: IIN(RMS) = ILED • √(1–D)D D= V LED VIN where D is the switch duty cycle. MANUFACTURER WEB TDK www.tdk.com Kemet www.kemet.com Murata www.murata.com Taiyo Yuden www.t-yuden.com AVX www.avx.com Output Capacitor Selection The selection of the output capacitor depends on the load and converter configuration, i.e., step-up or step-down and the operating frequency. For LED applications, the equivalent resistance of the LED is typically low and the output filter capacitor should be sized to attenuate the current ripple. Use of an X7R type ceramic capacitor is recommended. To achieve the same LED ripple current, the required filter capacitor is larger in the boost and buck-boost mode applications than that in the buck mode applications. Lower operating frequencies will require proportionately higher capacitor values. The component values shown in the data sheet applications are appropriate to drive the specified LED string. The product of the output capacitor and LED string impedance decides the second dominant pole in the LED current regulation loop. It is prudent to validate the power supply with the actual load (or loads). Power MOSFET Selection For applications operating at high input or output voltages, the power N-channel MOSFET switch is typically chosen for drain voltage VDS rating and low gate charge QG. Consideration of switch on-resistance, RDS(ON), is usually secondary because switching losses dominate power loss. The INTVCC regulator on the LT3795 has a fixed current limit to protect the IC from excessive power dissipation at high VIN, so the MOSFET should be chosen so that the product of QG at 7.7V and switching frequency does not exceed the INTVCC current limit. For driving LEDs, be careful to choose a switch with a VDS rating that exceeds the threshold set by the FB pin in case of an open load fault. Several MOSFET vendors are listed in Table 4. The MOSFETs used in the application circuits in this data sheet have been found to work well with the LT3795. Consult factory applications for other recommended MOSFETs. 3795f For more information www.linear.com/LT3795 19 LT3795 Applications Information greater than the required LED current. For buck mode, select a resistor according to: Table 4. MOSFET Manufacturers VENDOR WEB Vishay Siliconix Fairchild International Rectifier Infineon www.vishay.com www.fairchildsemi.com www.irf.com www.infineon.com High Side PMOS Disconnect Switch Selection A high side PMOS disconnect switch with a minimum VTH of –1V to –2V is recommended in most LT3795 applications to optimize or maximize the PWM dimming ratio and protect the LED string from excessive heating during fault conditions as well. The PMOS disconnect switch is typically selected for drain-source voltage VDS, and continuous drain current ID. For proper operations, VDS rating must exceed the open LED regulation voltage set by the FB pin, and ID rating should be above ILED. Schottky Rectifier Selection The power Schottky diode conducts current during the interval when the switch is turned off. Select a diode rated for the maximum SW voltage. It is important to choose a Schottky diode with sufficiently low leakage current when using the PWM feature for dimming, because leakage increases with temperature and occurs from the output during the PWM low interval. Table 5 has some recommended component vendors. Table 5. Schottky Rectifier Manufacturers VENDOR WEB On Semiconductor www.onsemi.com Diodes, Inc www.diodes.com Central Semiconductor www.centralsemi.com Rohm Semiconductor www.rohm.com RSENSE(BUCK) ≤ For buck-boost mode, select a resistor according to: RSENSE(BUCK − BOOST) ≤ The resistor, RSENSE, between the source of the external N-channel MOSFET and GND should be selected to provide adequate switch current to drive the application without exceeding the 113mV (typical) current limit threshold on the SENSE pin of the LT3795. For buck mode applications, select a resistor that gives a switch current at least 30% VIN • 0.07V (VIN +VLED)ILED For boost, select a resistor according to: RSENSE(BOOST) ≤ VIN • 0.07V VLED • ILED The placement of RSENSE should be close to the source of the N-channel MOSFET and GND of the LT3795. The SENSE input to LT3795 should be a Kelvin connection to the positive terminal of RSENSE. 70mV is used in the equations above to give some margin below the 113mV (typical) sense current limit threshold. Inductor Selection The inductor used with the LT3795 should have a saturation current rating appropriate to the maximum switch current selected with the RSENSE resistor. Choose an inductor value based on operating frequency and input and output voltage to provide a current mode signal on SENSE of approximately 20mV magnitude. The following equations are useful to estimate the inductor value (TSW = 1/fOSC): LBUCK = Sense Resistor Selection 0.07V ILED TSW •RSENSE •VLED(V IN – VLED ) V IN • 0.02V LBUCK−BOOST = LBOOST = TSW •RSENSE •VLED •V IN (VLED +V IN)• 0.02V TSW •RSENSE •VIN (VLED – V IN) VLED • 0.02V Table 6 provides some recommended inductor vendors. 20 3795f For more information www.linear.com/LT3795 LT3795 Applications Information Table 6. Inductor Manufacturers VENDOR WEB Sumida www.sumida.com Würth Elektronik www.we-online.com Coiltronics www.cooperet.com Vishay www.vishay.com Coilcraft www.coilcraft.com Loop Compensation The LT3795 uses an internal transconductance error amplifier whose VC output compensates the control loop. The external inductor, output capacitor and the compensation resistor and capacitor determine the loop stability. The inductor and output capacitor are chosen based on performance, size and cost. The compensation resistor and capacitor at VC are selected to optimize control loop response and stability. For typical LED applications, a 10nF compensation capacitor at VC is adequate, and a series resistor should always be used to increase the slew rate on the VC pin to maintain tighter regulation of LED current during fast transients on the input supply to the converter. Soft-Start Capacitor Selection For many applications, it is important to minimize the inrush current at start-up. The built-in soft-start circuit significantly reduces the start-up current spike and output voltage overshoot. The soft-start interval is set by the softstart capacitor selection according to the equation: TSS = CSS • 2V 28µA A typical value for the soft-start capacitor is 0.1µF. The soft-start pin reduces the oscillator frequency and the maximum current in the switch. Soft-start also operates as fault protection, which forces the converter into hiccup or latchoff mode. Detailed information is provided in the Fault Protection: Hiccup Mode and Latchoff Mode section. Fault Protection: Hiccup Mode and Latchoff Mode If an LED overcurrent condition, INTVCC undervoltage, output short (FB ≤ 0.3V), or thermal limit happens, the TG pin is pulled high to disconnect the LED array from the power path, and the GATE pin is driven low. If the soft-start pin is charging and still below 1.7V, then it will continue to do so with a 28µA source. Once above 1.7V, the pull-up source is disabled and a 2.8µA pull-down is activated. While the SS pin is discharging, the GATE is forced low. When the SS pin is discharged below 0.2V, a new cycle is initiated. This is referred as hiccup mode operation. If the fault still exists when SS crosses below 0.2V, then a full SS charge/ discharge cycle has to complete before switching is enabled. If a resistor is placed between the VREF pin and SS pin to hold SS pin higher than 0.2V during a fault, then the LT3795 will enter latchoff mode with GATE pin low, and TG pin high. To exit latchoff mode, the EN/UVLO pin must be toggled low to high. Board Layout The high speed operation of the LT3795 demands careful attention to board layout and component placement. The exposed pad of the package is the GND terminal of the IC and is also important for thermal management of the IC. It is crucial to achieve a good electrical and thermal contact between the exposed pad and the ground plane of the board. To reduce electromagnetic interference (EMI), it is important to minimize the area of the high dV/dt switching node between the inductor, switch drain and anode of the Schottky rectifier. Use a ground plane under the switching node to eliminate interplane coupling to sensitive signals. The lengths of the high dI/dt traces: 1) from the switch node through the switch and sense resistor to GND, and 2) from the switch node through the Schottky rectifier and filter capacitor to GND should be minimized. The ground points of these two switching current traces should come to a common point then connect to the ground plane under the LT3795. Likewise, the ground terminal of the bypass capacitor for the INTVCC regulator should be placed near the GND of the switching path. Typically, this requirement results in the external switch being closest to the IC, along with the INTVCC bypass capacitor. The ground for the compensation network and other DC control signals should be star connected to the underside of the IC. Do not extensively route high impedance signals such as FB, RT and VC, as they may pick up switching noise. Since there is a small variable DC input bias current to the ISN and ISP inputs, resistance in series with these pins should be 3795f For more information www.linear.com/LT3795 21 LT3795 Applications Information minimized to avoid creating an offset in the current sense threshold. Likewise, minimize resistance in series with the SENSE input to avoid changes (most likely reduction) to the switch current limit threshold. Figure 13 is a suggested two sided layout for a boost converter. Note that the 4-layer layout is recommended for best performance. Please contact the factory for the reference layout design. X VIA FROM ISP X 1 C4 VIA FROM TG X 3 26 4 25 5 24 6 23 7 22 8 21 9 20 10 19 C2 RC R4 VIA FROM X VREF VREF VIA X 11 C1 29 12 RT CIN L1 R3 R2 R1 X C5 VIA FROM VIN INTVCC VIA X 2 17 16 14 15 X GATE VIA 1 18 13 C3 CIN 28 27 R10 R5 LT3795 VIA FROM ISN R9 CC CIN X 2 X VIN RINSNS VIAS TO GROUND PLANE X ROUTING ON THE 2nd LAYER X VIA FROM ISP VIN VIA 3 R8 R7 RSNS 8 6 X 4 X VIA FROM INTVCC PWM 7 M1 5 D1 COUT COUT COUT COUT D2 LED+ 5 4 6 3 7 8 M2 X TG VIA 2 RLED 1 3795 F13 ISN VIA X X ISP VIA COMPONENT DESIGNATIONS REFER TO PAGE 23 CIRCUIT Figure 13. Boost Converter Suggested Layout 22 3795f For more information www.linear.com/LT3795 LT3795 Typical Applications Short-Circuit Robust Boost LED Driver with Input Current Limit and Spread Spectrum Frequency Modulation RINSNS 15mΩ 4A MAXIMUM VIN 8V TO 60V 110V (TRANSIENT) 63V OVLO CIN 2.2µF ×3 100V R2 115k VIN EN/UVLO IVINP OVLO RSNS 15mΩ LT3795 GND CTRL1 FB ISP PWM RLED 620mΩ ISN PWM C2 0.1µF INTVCC M2 TG ISMON D2 R8 100k OPENLED OPENLED SHORTLED M1: INFINEON BSC190N12NS3G M2: VISHAY SILICONIX Si7115DN L1: COOPER HC9-220 D1: DIODES INC PDS5100 D2: VISHAY ES1C LED: CREE XLAMP XR-E RT RC 10k INTVCC INTVCC SHORTLED VC RAMP RT 31.6k 250kHz IVINCOMP C3 47nF 87V LED 400mA C5 4.7µF 3795 TA02a C4 10nF CC 6.8nF Short LED Protection without R6: Hiccup Mode Short LED Protection with R11: Latchoff Mode SS 2V/DIV SS 2V/DIV LED+ 50V/DIV LED+ 50V/DIV SHORTLED 10V/DIV IM2 1A/DIV R10 13.3k M1 SENSE SS LED CURRENT REPORTING IVINN GATE CTRL2 VREF R4 12.4k C1 0.1µF COUT 2.2µF ×4 100V R9 1M R5 NTC 10k R7 100k D1 R1 499k R3 12.4k R6(OPT) 402k L1 22µH SHORTLED 10V/DIV 50ms/DIV IM2 5A/DIV 3796 TA02b 50ms/DIV 3796 TA02c 3795f For more information www.linear.com/LT3795 23 LT3795 Typical Applications SEPIC LED Driver with Input Current Limit VIN 8V TO 60V CIN 2.2µF ×3 100V D1 R1 499k R2 115k R3 12.4k L1B EN/UVLO VIN IVINP GATE 1A RSNS 15mΩ LT3795 GND FB SS PWM ISP RLED 250mΩ PWM ISN LED CURRENT REPORTING C2 0.1µF ISMON M2 TG INTVCC R5 100k COUT 10µF ×3 35V R7 1M R8 40.2k M1 SENSE CTRL2 VREF R4(OPT) 402k IVINN OVLO CTRL1 C1 0.1µF L1A 22µH RINSNS 20mΩ 3A MAXIMUM C5 2.2µF 100V ×2 R6 100k INTVCC OPENLED OPENLED SHORTLED SHORTLED VC M1: INFINEON BSC160N10NS3G M2: VISHAY SILICONIX Si7415DN L1: COILTRONICS DRQ127-220 D1: DIODES INC PDS5100 LED: CREE XLAMP XR-E 25V LED INTVCC C4 4.7µF RT RAMP RT 19.6k 400kHz RC 4.99k CC 10nF IVINCOMP R9 0Ω OPTION FOR DISABLING SSFM C6 47nF 3795 TA03a C3 10nF Efficiency 100 EFFICIENCY (%) 95 90 85 80 75 70 0 10 20 30 40 50 VIN (V) 3796 TA03b 24 3795f For more information www.linear.com/LT3795 LT3795 Typical Applications Buck Mode LED Driver VIN 24V TO 80V RINSNS 50mΩ 1.2A MAXIMUM R1 499k R2 21.5k R3 8.06k VIN EN/UVLO IVINP R5 178k IVINN ISP OVLO ISN VREF R9 10k ISMON C2 10nF D1 M1 GATE IVINCOMP SENSE GND R5 100k OPENLED OPENLED SHORTLED SHORTLED INTVCC VC RT RC 10k RAMP RT 19.6k 400kHz C4 4.7µF SS R10 0Ω OPTION FOR DISABLING SSFM C7 47nF CC 4.7nF INTVCC 3795 TA04a C3 0.1µF Efficiency 100 95 EFFICIENCY (%) M1: VISHAY SILICONIX Si7454DP M2: VISHAY SILICONIX Si7113DN D1: DIODES INC PDS3100 L1: COILTRONICS HC9-220 LED: CREE XLAMP XM-L Q1: ZETEX FMMT593 C6 2.2µF ×3 100V RSNS 15mΩ INTVCC R4 100k 18V LED 2.5A C5 10µF ×2 25V L1 22µH FB LT3795 PWM C1 0.1µF R7 1M R8 100k Q1 TG CTRL2 PWM M2 CIN 0.47µF 100V CTRL1 LED CURRENT REPORTING RLED 100mΩ 90 85 80 75 70 20 30 40 50 60 70 80 VIN (V) 3796 TA04b 3795f For more information www.linear.com/LT3795 25 LT3795 Typical Applications Buck Mode LED Driver with 3000:1 PWM Dimming VIN 16V TO 25V R1 499k R3 17.4k VIN IVINP EN/UVLO OVLO R8 100k 8V LED 1A C5 10µF ×2 25V TG FB LT3795 R9 10k PWM ISMON C2 10nF R10 1M R6 100k IVINN ISP VREF CTRL2 PWM M2 ISN CTRL1 C1 0.1µF RLED 250mΩ CIN 0.47µF R2 26.1k LED CURRENT REPORTING RINSNS 50mΩ 1.2A MAXIMUM D1 M1 GATE IVINCOMP L1 10µH SENSE C6 2.2µF ×3 50V RSNS 33mΩ GND INTVCC R4 100k R5 100k OPENLED SHORTLED M1: VISHAY SILICONIX Si4840BDY M2: VISHAY SILICONIX Si7415DN D1: ZETEX ZLLS2000TA Q1: WÜRTH 744066100 LED: CREE XLAMP XR-E Q1: ZETEX FMMT591 OPENLED INTVCC SHORTLED RT VC RAMP RT 6.65k 1MHz RC 10k SS C4 4.7µF INTVCC 3795 TA05a C3 0.1µF CC 4.7nF 3000:1 PWM Dimming at 100Hz and VIN = 24V PWM 2V/DIV IL 1A/DIV ILED 1A/DIV 1µs/DIV 26 3796 TA05b 3795f For more information www.linear.com/LT3795 LT3795 Package Description Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. FE Package 28-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663 Rev J) Exposed Pad Variation EB 9.60 – 9.80* (.378 – .386) 4.75 (.187) 4.75 (.187) 28 27 26 2524 23 22 21 20 1918 17 16 15 6.60 ±0.10 4.50 ±0.10 2.74 (.108) SEE NOTE 4 0.45 ±0.05 EXPOSED PAD HEAT SINK ON BOTTOM OF PACKAGE 6.40 2.74 (.252) (.108) BSC 1.05 ±0.10 0.65 BSC RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS 2. DIMENSIONS ARE IN MILLIMETERS (INCHES) 3. DRAWING NOT TO SCALE 1 2 3 4 5 6 7 8 9 10 11 12 13 14 0.25 REF 1.20 (.047) MAX 0° – 8° 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE28 (EB) TSSOP REV J 1012 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3795f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representaFor more information www.linear.com/LT3795 tion that the interconnection of its circuits as described herein will not infringe on existing patent rights. 27 LT3795 Typical Application Buck-Boost Mode LED Driver 3A MAXIMUM CIN 4.7µF 50V D1 R1 499k R2 90.9k R3 14.3k PWM LED CURRENT REPORTING L1 22µH C1 0.1µF C2 22nF EN/UVLO VIN IVINP IVINN OVLO Q1 M1 GATE R4 100k OPENLED R5 100k SHORTLED Efficiency vs VIN VIN 100 RSNS 15mΩ CTRL2 VREF CTRL1 PWM ISMON IVINCOMP R7 10k PWM = VREF 95 GND LT3795 FB ISP RLED 250mΩ ISN 90 85 80 M2 75 OPENLED VC RAMP RT INTVCC C4 4.7µF INTVCC SHORTLED M1: VISHAY SILICONIX Si7454DP RC M2: VISHAY SILICONIX Si7113DN 4.99k L1: COOPER HC9-220 D1: DIODES INC PDS5100 CC Q1: ZETEX FMMT593 10nF LED: CREE XLAMP XR-E COUT 2.2µF ×4 50V LED– SENSE TG INTVCC C5 1µF 100V R6 357k EFFICIENCY (%) VIN 8V TO 50V LED- RSNS1 20mΩ RT 19.6k 400kHz SS C6 47nF R8 0Ω OPTION FOR DISABLING SSFM C3 0.1µF 36V LED 1A 70 0 10 20 30 40 50 VIN (V) 3796 TA07b LED- 3795 TA07a Related Parts PART NUMBER DESCRIPTION COMMENTS LT3791 60V, Synchronous Buck-Boost 1MHz LED Controller VIN: 4.7V to 60V, VOUT Range: 0V to 60V, True Color PWM, Analog = 100:1, ISD < 1µA, TSSOP-38E Package LT3796/LT3796-1 100V Constant Current and Constant VIN: 6V to 100V, VOUT(MAX) = 100V, True Color PWM Dimming = 3000:1, ISD < 1µA, Voltage Controller with Dual Current Sense 28-Lead TSSOP Package LT3755/LT3755-1/ High Side 60V, 1MHz LED Controller with LT3755-2 True Color 3,000:1 PWM Dimming VIN: 4.5V to 40V, VOUT Range: 5V to 60V, True Color PWM, Analog = 3000:1, ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages LT3756/LT3756-1/ High Side 100V, 1MHz LED Controller with LT3756-2 True Color 3,000:1 PWM Dimming VIN: 6V to 100V, VOUT Range: 5V to 100V, True Color PWM, Analog = 3000:1, ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages LT3743 Synchronous Step-Down 20A LED Driver with Three-State LED Current Control VIN: 5.5V to 36V, VOUT Range: 5.5V to 35V, True Color PWM, Analog = 3000:1, ISD < 1µA, 4mm × 5mm QFN-28, TSSOP-28E Packages LT3517 1.3A, 2.5MHz High Current LED Driver with 3,000:1 Dimming VIN: 3V to 30V, True Color PWM, Analog = 3000:1, ISD < 1µA, 4mm × 4mm QFN-16 Package LT3518 2.3A, 2.5MHz High Current LED Driver with 3,000:1 Dimming VIN: 3V to 30V, True Color PWM, Analog = 3000:1, ISD < 1µA, 4mm × 4mm QFN-16 Package LT3474/ LT3474-1 36V, 1A (ILED), 2MHz, Step-Down LED Driver VIN: 4V to 36V, VOUT Range = 13.5V, True Color PWM = 400:1, ISD < 1µA, TSSOP-16E Package LT3475/ LT3475-1 Dual 1.5A(ILED), 36V, 2MHz, Step-Down LED Driver VIN: 4V to 36V, VOUT Range = 13.5V, True Color PWM, Analog = 3000:1, ISD < 1µA, TSSOP-20E Package 28 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/LT3795 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT3795 3795f LT 0813 • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2013