Fairchild FCP20N60S Pfcpwm combination controller Datasheet

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AN-8027
FAN480X PFC+PWM Combination Controller Application
FAN4800A / FAN4800C / FAN4801 / FAN4802 / FAN4802L
Introduction
The synchronization of the PWM with the PFC simplifies
the PWM compensation due to the controlled ripple on the
PFC output capacitor (the PWM input capacitor). In
addition to power factor correction, a number of protection
features have been built in to the FAN480X. These include
programmable soft-start, PFC over-voltage protection,
pulse-by-pulse current limiting, brownout protection, and
under-voltage lockout.
This application note describes step-by-step design
considerations for a power supply using the FAN480X
controller. The FAN480X combines a PFC controller and
a PWM controller. The PFC controller employs average
current mode control for Continuous Conduction Mode
(CCM) boost converter in the front end. The PWM
controller can be used in either current mode or voltage
mode for the downstream converter. In voltage mode,
feed-forward from the PFC output bus can be used to
improve the line transient response of PWM stage. In
either mode, the PWM stage uses conventional trailingedge duty cycle modulation, while the PFC uses leadingedge modulation. This proprietary leading/trailing-edge
modulation technique can significantly reduce the ripple
current of the PFC output capacitor.
F1
AC
Input
DBOOST
L BOOST
CIF1
Drv
CBOOST
Q1
FAN4801/2/2L feature programmable two-level PFC
output to improve efficiency at light-load and low-line
conditions.
FAN480X is pin-to-pin compatible with FAN4800 and
ML4800, only requiring adjustment of some peripheral
components. The FAN480X series comparison is
summarized in the Appendix A.
VBOUT
L1
L 11
R FB1
DR1
Q2
Drv
2
Vo1
DR1
DF1
RCS1
CO12
CO11
R RAMP
D1
L22
D2
DR2
Q3
Drv
RLF2
CIC2
CRMS1
CRMS2 R
RMS3
DF2
DR2
Vo2
CO21
CO22
RCS2
R IAC
RLF1
RRMS1
L
2
1
RB
RRMS2
CIC1
RIC
RT
IEA
VEA
IAC
FBPFC
ISENSE
SS
VD
D
OPFC
FBPWM
OPWM
VRMS
CLF1 CSS
CT
VREF
RT/CT
GND
RAMP
ILIMIT
RD
RBIAS
VD
D
CF
RVC
CVC2
CVC1
FAN480X
CRAMP
CB
Vo
1
RF
ROS1
ROS2
Vo2
ROS3
RFB2
CFB
CLF2 CDD CREF
Figure 1. Typical Application Circuit of FAN480X
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
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AN-8027
Functional Description
However, once PFC stops switching operation, the junction
capacitance of bridge diode is not discharged and VIN of
Figure 2 is clamped at the peak of the line voltage. Then,
the voltage of VRMS pin is given by:
Gain Modulator
The gain modulator is the key block for PFC stage because
it provides the reference to the current control error
amplifier for the input current shaping, as shown in Figure
2. The output current of gain modulator is a function of VEA,
IAC , and VRMS. The gain of the gain modulator is given in
the datasheet as a ratio between IMO and IAC with a given
VRMS when VEA is saturated to HIGH. The gain is inversely
proportional to VRMS2, as shown in Figure 3, to implement
line feed-forward. This automatically adjusts the reference
of current control error amplifier according to the line
voltage such that the input power of PFC converter is not
changed with line voltage.
VRMS NS = VLINE
2 RRMS 3
RRMS 1 + RRMS 2 + RRMS 3
(2)
Therefore, the voltage divider for VRMS should be
designed considering the brownout protection trip point
and minimum operation line voltage.
PFC runs
VIN
PFC stops
VIN
IL
IEA
R
ISENS
E
M
RRMS1
M
RIAC
CRMS1
RRMS2
CRMS2
IAC
= I AC ⋅
IA
C
VRMS
VRMS
IMO = G ⋅ I AC
R
K ⋅ (VEA − 0.7)
VRMS 2 (VEA MAX − 0.7)
k
x
2
RRMS3
VEA
Figure 4. VRMS According to the PFC Operation
Gain
Modulator
The rectified sinusoidal signal is obtained by the current
flowing into the IAC pin. The resistor RIAC should be large
enough to prevent saturation of the gain modulator as:
Figure 2. Gain Modulator Block
G∝
1
2VLINE . BO
⋅ G MAX < 159μ A
(3)
RIAC
where VLINE.BO is the line voltage that trips brownout
protection, GMAX is the maximum modulator gain when VRMS
is 1.08V (which can be found in the datasheet), and 159µA is
the maximum output current of the gain modulator.
VRMS 2
Current and Voltage Control of Boost Stage
As shown in Figure 5, the FAN480X employs two control
loops for power factor correction: a current control loop
and a voltage control loop. The current control loop shapes
inductor current, as shown in Figure 6, based on the
reference signal obtained at the IAC pin as:
VRMS
VRMS-UVP
I L ⋅ RCS1 = I MO ⋅ RM = I AC ⋅ G ⋅ RM
Figure 3. Modulation Gain Characteristics
(4)
To sense the RMS value of the line voltage, an averaging
circuit with two poles is typically employed, as shown in
Figure 2. The voltage of VRMS pin in normal PFC
operation is given as:
VRMS = VLINE
2 RRMS 3
2
⋅
RRMS 1 + RRMS 2 + RRMS 3 π
(1)
where VLINE is RMS value of line voltage.
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
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2
AN-8027
V IN
It is typical to set the second boost output voltage as
340V~300V.
VO
IL
RCS1
RF1
ISENSE
RRMS1
CF1
RIAC
CRMS1
RRMS2
CRMS2
IAC
RRMS3
RM
IEA
RM
RIC
IMO
CIC2
IAC
VRMS
+
VEA
-
Drive logic
CIC1
VREF
OPFC
RVC2
Figure 7. Block of Two-Level PFC Output
RFB1
RVC
FBPFC
RVC1
2.5V
Oscillator
RFB2
The internal oscillator frequency of FAN480X is
determined by the timing resistor and capacitor on RT/CT
pin. The frequency of the internal oscillator is given by:
Figure 5. Gain Modulation Block
IAC
fOSC =
I MO
RM
RCS1
1
0.56 ⋅ RT ⋅ CT + 360CT
Because the PWM stage of FAN480X generally uses a
forward converter, it is required to limit the maximum duty
cycle at 50%. To have a small tolerance of the maximum
duty cycle, a frequency divider with toggle flip-flops is
used, as illustrated in Figure 8. The operation frequency of
PFC and PWM stage is one quarter (1/4) of the oscillator
frequency. (For FAN4800C and FAN4802/2L, the
operation frequencies for PFC and PWM stages are one
quarter (1/4) and one half (1/2) of the oscillator frequency,
respectively).
IL
Figure 6. Inductor Current Shaping
The voltage control loop regulates PFC output voltage
using internal error amplifier such that the FBPFC voltage
is same as internal reference of 2.5V.
Brownout Protection
The dead time for the PFC gate drive signal is determined
by the equation:
FAN480X has a built-in internal brownout protection
comparator monitoring the voltage of the VRMS pin. Once
the VRMS pin voltage is lower than 1.05V (0.9V for
FAN4802L), the PFC stage is shutdown to protect the
system from over current. The FAN480X starts up the
boost stage once the VRMS voltage increases above 1.9V
(1.65V for FAN4802L).
tDEAD = 360CT
(7)
The dead time should be smaller than 2% of switching
period to minimize line current distortion around line zero
crossing.
Two-Level PFC Output
VREF
To improve system efficiency at low AC line voltage and
light load condition, FAN480X provides two-level PFC
output voltage. As shown in Figure 7, FAN480X monitors
VEA and VRMS voltages to adjust the PFC output voltage.
When VEA and VRMS are lower than the thresholds, an
internal current source of 20µA is enabled that flows
through RFB2, increasing the voltage of the FBPFC pin.
This causes the PFC output voltage to reduce when 20µA
is enabled, calculated as:
VOPFC 2 =
RFB1 + RFB 2
× (2.5 - 20 μA × RFB 2 )
RFB 2
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
(6)
RT/
CT
OSC
T-FF
T-FF
T Q
T
Q
OPFC, OPWM
OPWM (FAN4800C, FAN4802/2L)
Figure 8. Oscillator Configuration
(5)
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AN-8027
RT/
CT
VBOUT
RRAMP
PFC dead time
REF
1.5V
PWM
-
CRAMP
OPFC
RAMP
+
OPWM
FBPWM
OPWM (FAN4800C, FAN4802/2L)
Figure 10. PWM Ramp Generation Circuit
Figure 9. FAN480X Timing Diagram
PWM Stage
PWM Current Limit
The PWM stage is capable of current-mode or voltagemode operation. In current-mode applications, the PWM
ramp (RAMP) is usually derived directly from a current
sensing resistor or current transformer in the primary of the
output stage and is thereby representative of the current
flowing in the converter’s output stage. ILIMIT, which
provides cycle-by-cycle current limiting, is typically
connected to RAMP in such applications.
The ILIMIT pin is a direct input to the cycle-by-cycle
current limiter for the PWM section. If the input voltage at
this pin exceeds 1V, the output of the PWM is disabled
until the start of the next PWM clock cycle.
VIN OK Comparator
The VIN OK comparator monitors the output of the PFC
stage and inhibits the PWM stage if this voltage is less than
2.4V (96% of its nominal value). Once this voltage goes
above 2.4V, the PWM stage begins to soft-start.
For voltage-mode operation, RAMP can be connected to a
separate RC timing network to generate a voltage ramp
against which FBPWM voltage is compared. Under these
conditions, the use of voltage feed-forward from the PFC
bus can be used for better line transient response.
PWM Soft-Start (SS)
PWM startup is controlled by the soft-start capacitor. A
10µA current source supplies the charging current for the
soft-start capacitor. Startup of the PWM is prohibited until
the soft-start capacitor voltage reaches 1.5V.
No voltage error amplifier is included in the PWM stage,
as this function is generally performed by a programmable
shunt regulator, such as KA431, in the secondary-side. To
facilitate the design of opto-coupler feedback circuitry, an
offset voltage is built into the inverting input of PWM
comparator that allows FBPWM to command a zero
percent duty cycle when its pin voltage is below 1.5V.
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
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AN-8027
Design Considerations
a design example. The design specifications are
summarized in 0. The two-switch forward converter is used
for DC/DC converter stage.
In this section, a design procedure is presented using the
schematic in Figure 11 as reference. A 300W PC power
supply application with universal input range is selected as
Design Specifications
Rated Voltage of Output 1
VOUT1 = 5V
PWM Stage Efficiency
ηPWM = 0.86
Rated Current of Output 1
IOUT1 = 9A
Hold-up Time
tHLD = 20ms
Rated Voltage of Output 2
Vout2 = 12V
Minimum PFC Output Voltage
310V
Rated Current of Output 2
IOUT2 = 16.5A
Nominal PFC output voltage
VO_PFC = 387V
Rated Voltage of Output 3
VOUT3 = -12V
PFC Output Voltage Ripple
12VPP
Rated Current of Output 3
IOUT3 = 0.8A
PFC Inductor Ripple Current
dI = 40%
Rated Voltage of Output 4
VOUT4 = 3.3V
AC Input Voltage Frequency
fline = 50 ~ 60Hz
Rated Current of Output 4
IOUT4 = 13.5A
Switching Frequency
fS = 65KHz
Rated Output Power
PO = 300W
Total Harmonic Distortion
α = 4%
Line Voltage Range
85~264VAC
Magnetic Flux Density
ΔB = 0.27T
Line Frequency
50Hz
Current Density
Dcma = 400C-m/A
Brownout Protection Line Voltage
72VAC
PWM Maximum Duty Cycle
Dmax = 0.35
Overall Stage Efficiency
η = 0.82
5V Output Current Ripple
ILo1 = 44%
12V Output Current Ripple
F1
AC
Input
DBOOST
L BOOST
CIF1
CBOOST
Q1
Drv
ILo2 = 10%
VBOUT
L1
L 11
R FB1
DR1
Q2
Drv
2
Vo1
DR1
DF1
RCS1
CO12
CO11
R RAMP
D1
L22
D2
DR2
Q3
Drv
RLF2
CIC2
CRMS1
CRMS2 R
RMS3
DF2
DR2
Vo2
CO21
CO22
Vo3
RCS2
Vo4
R IAC
RLF1
RRMS1
L
2
1
RB
RRMS2
CIC1
RIC
RT
IEA
VEA
IAC
FBPFC
ISENSE
CLF1 CSS
SS
FBPWM
CT
OPWM
RT/CT
GND
RAMP
ILIMIT
Vo1
RBIAS
VREF
VD
D
OPFC
VRMS
RD
VD
D
CF
RVC
CVC2
CVC1
FAN480X
CRAMP
RF
ROS1
ROS2
Vo2
ROS3
RFB2
CFB
CB
CLF2 CDD CREF
Figure 11. Reference Circuit for Design Example
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
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5
AN-8027
[STEP-1] Define System Specifications
(Design Example) Since the switching frequency is
Since the overall system is comprised of two stages (PFC
and DC/DC), as shown in Figure 12, the input power and
output power of the boost stage are given as:
DMAX . PFC = 1 − 360 ⋅ CT ⋅ f SW = 0.98
PIN =
POUT
The timing resistor is determined as:
1
1
RT = ⋅
= 6.9k Ω
4 0.56 f SW CT
(8)
η
PBOUT =
65kHz, CT is selected as 1nF. Then the maximum duty
cycle of PFC gate drive signal is obtained as:
POUT
(9)
η PWM
where η is the overall efficiency and ηPWM is the forward
converter efficiency.
[STEP-3] Line Sensing Circuit Design
FAN480X senses the RMS value and instantaneous value of
line voltage using the VRMS and IAC pins, respectively, as
shown in Figure 13. The RMS value of the line voltage is
obtained by an averaging circuit using low pass filter with
two poles. Meanwhile, the instantaneous line voltage
information is obtained by sensing the current flowing into
IAC pin through RIAC.
The nominal output current of boost PFC stage is given as:
I BOUT =
POUT
(10)
η PWM VBOUT
PBOUT
PIN
POUT
IBOUT
Boost
PFC
Forward
DC/DC
VBOUT
VIN
IL
VOUT
Figure 12. Two Stage Configuration
(Design Example)
PIN =
POUT
η
PBOUT =
300
=
0.82
POUT
η PWM
I BOUT =
=
RIAC
= 366W
0.86
POUT
IA
C
RRMS1
300
η PWM VBOUT
IAC
= 349W
=
CRMS1
300
120/100Hz
VIN
RRMS2
VRMS
CRMS2
0.86 ⋅ 387
VRMS
fp1
= 0.9 A
fp2
RRMS3
Figure 13. Line Sensing Circuits
[STEP-2] Frequency Setting
The switching frequency is determined by the timing resistor
and capacitor (RT and CT) as:
1
1
f SW ≅ ⋅
(11)
4 0.56 ⋅ RT ⋅ CT
RMS sensing circuit should be designed considering the
nominal operation range of line voltage and brownout
protection trip point as:
It is typical to use a 470pF~1nF capacitor for 50~75kHz
switching frequency operation since the timing capacitor
value determines the maximum duty cycle of PFC gate drive
signal as:
VRMS −UVL = VLINE .BO
DMAX .PFC = 1 −
TOFF .MIN
= 1 − 360 ⋅ CT ⋅ f SW
TSW
2 RRMS 3
2
⋅
RRMS 1 + RRMS 2 + RRMS 3 π
(13)
2 RRMS 3
RRMS 1 + RRMS 2 + RRMS 3
(14)
VRMS −UVH < VLINE .MIN
where VRMS-UVL and VRMS-UVH are the brown OUT/IN
thresholds of VRMS.
(12)
It is typical to set RRMS2 as 10% of RRMS1. The poles of the
low pass filter are given as:
f P1 ≅
fP2 ≅
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
1
2π ⋅ CRMS 1 ⋅ RRMS 2
1
2π ⋅ CRMS 2 ⋅ RRMS 3
(15)
(16)
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AN-8027
To properly attenuate the twice line frequency ripple in
VRMS, it is typical to set the poles around 10~20Hz.
The average of boost inductor current over one switching
cycle at the peak of the line voltage for low line is given as:
The resistor RIAC should be large enough to prevent
saturation of the gain modulator as:
I L. AVG =
I L PK = I L. AVG ⋅ (1 +
LBOOST =
Then the startup of the PFC stage at the minimum line
voltage is checked as:
=
The resistors of the voltage divider network are selected
as RRMS1=2MΩ, RRMS1=200kΩ, and RRMS1=36kΩ.
I L. AVG =
To place the poles of the low pass filter at 15Hz and
22Hz, the capacitors are obtained as:
1
1
CRMS1 =
=
= 53nF
2π ⋅ f P1 ⋅ RRMS 2 2π ⋅15 ⋅ 200 × 103
I L PK =
=
The condition for Resistor RIAC is:
2VLINE .BO MAX
2 ⋅ 72 ⋅ 9
RIAC >
⋅G
=
= 5.8M Ω
159 × 10−6
159 × 10−6
852 ⋅ 0.82 387 − 2 ⋅ 85 10−3
⋅
⋅
= 524 μ H
0.4 ⋅ 300
387
65
2 POUT
2 ⋅ 300
=
= 6.09 A
VLINE .MIN ⋅η 85 ⋅ 0.82
2 POUT
K
⋅ (1 + RB )
VLINE .MIN ⋅η
2
2 ⋅ 300
0.4
⋅ (1 +
) = 7.31A
85 ⋅ 0.82
2
[STEP-5] PFC Output Capacitor Selection
The output voltage ripple should be considered when
selecting the PFC output capacitor. Figure 14 shows the
twice line frequency ripple on the output voltage. With a
given specification of output ripple, the condition for the
output capacitor is obtained as:
Therefore, 6MΩ resistor is selected for RIAC.
[STEP-4] PFC Inductor Design
CBOUT >
The duty cycle of boost switch at the peak of line voltage is
given as:
I BOUT
2π ⋅ f LINE ⋅ VBOUT , RIPPLE
(23)
where IBOUT is nominal output current of boost PFC stage
and VBOUT,RIPPLE is the peak-to-peak output voltage ripple
specification.
(18)
The hold-up time also should be considered when
determining the output capacitor as:
Then, the maximum current ripple of the boost inductor at
the peak of line voltage for low line is given as:
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
VLINE .MIN 2 ⋅η VBOUT − 2VLINE 1
⋅
⋅
K RB ⋅ POUT
VBOUT
f SW
The maximum current of the boost inductor is given as:
1
1
=
= 200nF
2π ⋅ f P 2 ⋅ RRMS 3 2π ⋅ 22 ⋅ 36 × 103
2VLINE .MIN VBOUT − 2VLINE 1
⋅
⋅
LBOOST
VBOUT
f SW
(22)
The average of boost inductor current over one
switching cycle at the peak of the line voltage for low
line is obtained as:
VLINE .MIN ⋅ 2 RRMS 3
= 85 ⋅ 2 ⋅ 0.0162 = 1.95 > 1.9V
RRMS 1 + RRMS 2 + RRMS 3
ΔI L =
2 POUT
K RB
K
)=
⋅ (1 + RB )
2
VLINE .MIN ⋅η
2
(Design Example) With the ripple current
specification (40%), the boost inductor is obtained as:
1.05 π
⋅
= 0.0162
72 2 2
VBOUT − 2VLINE
VBOUT
(21)
The maximum current of boost inductor is given as:
1.05V (VRMS-UVL) and 1.9V (VRMS-UVH), respectively.
Then, the scaling down factor of the voltage divider is:
RRMS 3
V
π
= RMS −UVL ⋅
RRMS 1 + RRMS 2 + RRMS 3 VLINE . BO 2 2
DLP =
VLINE .MIN 2 ⋅η VBOUT − 2VLINE 1
⋅
⋅
K RB ⋅ POUT
VBOUT
f SW
LBOOST =
(Design Example) The brownout protection threshold is
CRMS 2 ≅
(20)
Therefore, with a given current ripple factor
(KRB=ΔIL/ILAVG), the boost inductor value is obtained as:
2VLINE . BO
⋅ G MAX < 159μ A
(17)
RIAC
where VLINE.BO is the brownout protection line voltage,
GMAX is the maximum modulator gain when VRMS is 1.08V
(which can be found in the datasheet), and 159µA is the
maximum output current of the gain modulator.
=
2 POUT
VLINE .MIN ⋅η
CBOUT >
(19)
PBOUT ⋅ tHOLD
VBOUT 2 − VBOUT , MIN 2
(24)
where PBOUT is nominal output power of boost PFC stage,
tHOLD is the required holdup time, and VBOUT,MIN is the
allowable minimum PFC output voltage during hold-up time.
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ID
I D , AVG
I D , AVG = I BOUT (1 − cos(4π ⋅ f LINE ⋅ t ))
I BOUT
VBOUT , RIPPLE =
I BOUT
2π f LINE CBOUT
Figure 15. Two-Level PFC Output Block
The voltage divider network for the PFC output voltage
sensing should be designed such that FBPFC voltage is
2.5V at nominal PFC output voltage:
VBOUT
VBOUT ×
Figure 14. PFC Output Voltage Ripple
RFB 2 = (1 −
I BOUT
0.9
>
=
= 239 μ F
2π ⋅ f LINE ⋅ VBOUT , RIPPLE 2π ⋅ 50 ⋅12
= (1 −
Since minimum allowable output voltage during one
cycle line (20ms) drop-outs is 310V, the capacitor
should be:
CBOUT >
PBOUT ⋅ t HOLD
2
VOUT − VOUT , MIN
2
=
2 ⋅ 349 ⋅ 20 × 10−3
2
387 − 310
2
347
2.5
= 12.9k Ω
)⋅
387 20 × 10−6
RFB1 = (
= 260 μ F
=(
VBOUT
− 1) ⋅ RFB 2
2.5
387
− 1) ⋅ 13 × 103 = 1999k Ω
2.5
2MΩ is selected for RFB1.
[STEP-6] PFC Output Sensing Circuit
[STEP-7] PFC Current-Sensing Circuit Design
To improve system efficiency at low line and light load
condition, FAN480X provides two-level PFC output
voltage. As shown in Figure 15, FAN480X monitors VEA
and VRMS voltages to adjust the PFC output voltage.
Figure 16 shows the PFC compensation circuits. The first
step in compensation network design is to select the currentsensing resistor of PFC converter considering the control
window of voltage loop. Since line feed-forward is used in
FAN480X, the output power is proportional to the voltage
control error amplifier voltage as:
The PFC output voltage when 20µA is enabled is given as:
20 μA × RFB 2
)
2.5
VBOUT 2
2.5
)⋅
VBOUT 20 × 10−6
13kΩ is selected for RFB2.
Thus, 270μF capacitor is selected for the PFC output
capacitor.
VBOUT 2 = VBOUT × (1 -
(26)
(Design Example) Assuming the second level of
PFC output voltage is 347V:
(Design Example) With the ripple specification of
12VPP, the capacitor should be:
CBOUT
RFB 2
= 2.5V
RFB1 + RFB 2
(25)
PBOUT (VEA ) = PBOUT MAX ⋅
VEA − 0.6
VEA SAT − 0.6
(27)
It is typical second boost output voltage as 340V~300V.
where VEASAT is 5.6V and the maximum power limit of PFC
is:
PBOUT MAX =
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
VLINE . BO 2 ⋅ G MAX ⋅ RM
RIAC RCS 1
(28)
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It is typical to set the maximum power limit of PFC stage
around 1.2~1.5 of its nominal power such that the VEA is
around 4~4.5V at nominal output power. By adjusting the
current-sensing resistor for PFC stage, the maximum power
limit of PFC stage can be programmed.
where VRAMP is the peak to peak voltage of ramp signal for
current control PWM comparator, which is 2.55V.
The transfer function of the compensation circuit is given as:
s
1+
)
2π f IC
vIEA 2π f II
⋅
) =
s
vCS 1
s
1+
2π f IP
where:
To filter out the current ripple of switching frequency, an
RC filter is typically used for ISENSE pin. RLF1 should not
be larger than 100Ω and the cut-off frequency of filter
should be 1/2~1/6 of the switching frequency.
Diodes D1 and D2 are required to prevent over-voltage on
ISENSE pin due to the inrush current that might damage
the IC. A fast recovery diode or ultra fast recovery diode is
recommended.
f II =
GMI
1
and
, f IZ =
2π ⋅ CIC1
2π ⋅ RIC ⋅ CIC1
1
f IP =
(31)
(32)
2π ⋅ RIC ⋅ CIC 2
The procedure to design the feedback loop is as follows:
(a) Determine the crossover frequency (fIC) around
1/10~1/6 of the switching frequency. Then calculate
the gain of the transfer function of Equation (30) at
crossover frequency as:
)
vCS1
)
vIEA
=
@ f = f IC
RCS1 ⋅ VBOUT
VRAMP ⋅ 2π f IC ⋅ LBOOST
(33)
(b) Calculate RIC that makes the closed loop gain unity at
crossover frequency:
RIC =
Figure 16. Gain Modulation Block
(34)
@ f = f IC
(c) Since the control-to-output transfer function of power
stage has -20dB/dec slope and -90o phase at the
crossover frequency is 0dB, as shown in Figure 17; it
is necessary to place the zero of the compensation
network (fIZ) around 1/3 of the crossover frequency so
that more than 45° phase margin is obtained. Then the
capacitor CIC1 is determined as:
(Design Example) Setting the maximum power limit
of PFC stage as 450W, the current sensing resistor is
obtained as:
RCS 1 =
1
)
v
GMI ⋅ )CS1
vIEA
VLINE .BO 2 ⋅ G MAX ⋅ RM 722 ⋅ 9 ⋅ 5.7 × 103
=
= 0.098Ω
6 × 106 ⋅ 450
RIAC PBOUT MAX
CIC1 =
Thus, 0.1Ω resistor is selected.
1
RIC ⋅ 2π fC / 3
(35)
(d) Place compensator high-frequency pole (fCP) at least a
decade higher than fIC to ensure that it does not
interfere with the phase margin of the current loop at
its crossover frequency.
[STEP-8] PFC Current Loop Design
The transfer function from duty cycle to the inductor
current of boost power stage is given as:
)
iL
V
) = BOUT
(29)
d sLBOOST
CIC 2 =
1
2π ⋅ f IP ⋅ RIC
(36)
The transfer function from the output of the current control
error amplifier to the inductor current-sensing voltage is
obtained as:
)
vCS 1
RCS 1 ⋅ VBOUT
(30)
) =
vIEA VRAMP ⋅ sLBOOST
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
www.fairchildsemi.com
9
AN-8027
60dB
Closed Loop Gain
Control-to-output
40dB
Compensation
20dB
fIP
0dB
fIZ
fIC
-20dB
-40dB
10Hz
100Hz
1kHz
10kHz
100kHz
1MHz
Figure 17. Current Loop Compensation
Figure 18. Voltage Loop Compensation
(Design Example) Setting the crossover frequency
as 7kHz:
)
vCS 1
RCS 1 ⋅ VBOUT
=
)
vIEA @ f = f
VRAMP ⋅ 2π f IC ⋅ LBOOST
The transfer function of the compensation network is
obtained as:
s
1+
2π fVZ
vˆCOMP 2π fVI
=
⋅
s
vˆOUT
s
1+
2π fVP
IC
0.1 ⋅ 387
=
= 0.66
2.55 ⋅ 2π ⋅ 7 × 103 ⋅ 524 × 10 −6
RIC =
1
)
v
GMI ⋅ )CS 1
vIEA
C IC1 =
=
where:
1
= 17 k Ω
88 × 10−6 ⋅ 0.66
fVI =
@ f = f IC
fVP =
1
1
=
= 4nF
RIC ⋅ 2π fC / 3 17 × 103 ⋅ 2π ⋅ 7 × 103 / 3
1
=
2π ⋅ f IP ⋅ RIC
1
= 0.13nF
2π ⋅ 70 × 103 ⋅ 17 × 103
[STEP-9] PFC Voltage Loop Design
⋅
⋅
1
sCBOUT
1
sCBOUT
(40)
2π ⋅ RVC ⋅ CVC 2
GMV ⋅ I BOUT ⋅ K MAX
2.5
⋅
(41)
5 ⋅ C BOUT ⋅ (2π fVC ) 2 VBOUT
To place the compensation zero at the crossover
frequency, the compensation resistor is obtained as:
CVC1 =
(37)
(38)
RVC =
(b)
Proportional and integration (PI) control with highfrequency pole is typically used for compensation. The
compensation zero (fVZ) introduces phase boost, while the
high-frequency compensation pole (fVP) attenuates the
switching ripple, as shown in Figure 18.
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
1
and
(a) Determine the crossover frequency (fVC) around
1/10~1/5 of the line frequency. Since the control-tooutput transfer function of power stage has -20dB/dec
slope and -90o phase at the crossover frequency, as
shown in Figure 18 as 0dB; it is necessary to place the
zero of the compensation network (fVZ) around the
crossover frequency so that 45° phase margin is
obtained. Then, the capacitor CVC1 is determined as:
Since FAN480X employs line feed-forward, the power
stage transfer function becomes independent of the line
voltage. Then, the low-frequency, small-signal, control-tooutput transfer function is obtained as:
vˆBOUT I BOUT ⋅ K MAX
≅
5
vˆEA
where:
vˆBOUT I BOUT ⋅ K MAX
≅
5
vˆEA
GMV
2.5
1
, fVZ =
⋅
2π ⋅ RVC ⋅ CVC1
VBOUT 2π ⋅ CVC1
The procedure to design the feedback loop is as follows:
Setting the pole of the compensator at 70kHz,
CIC 2 =
(39)
1
2π ⋅ fVC ⋅ CVC1
(42)
Place compensator high-frequency pole (fVP) at least
a decade higher than fC to ensure that it does not
interfere with the phase margin of the voltage
regulation loop at its crossover frequency. It should
also be sufficiently lower than the switching
frequency of the converter so noise can be effectively
attenuated. Then, the capacitor CVC2 is determined as:
CVC 2 =
1
2π ⋅ fVP ⋅ RVC
(43)
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10
AN-8027
(Design Example) Setting the crossover frequency
as 22Hz:
CVC1 =
=
RVC =
GMV ⋅ I BOUT ⋅ K MAX
2.5
⋅
5 ⋅ C BOUT ⋅ (2π fVC ) 2 VBOUT
70 × 10 −6 ⋅ 0.9 ⋅1.27
2.5
⋅
= 20nF
−6
2
5 ⋅ 270 × 10 ⋅ (2π ⋅ 22) 387
1
1
=
= 362k Ω
2π ⋅ fVC ⋅ CVC1 2π ⋅ 22 ⋅ 20 × 10−9
Setting the pole of the compensator at 120Hz:
CVC 2 =
1
2π ⋅ fVP ⋅ RVC
=
1
= 3.7 nF
2π ⋅ 120 ⋅ 362 × 103
[STEP-10] Transformer Design for PWM
Stage
Figure 19 shows the typical secondary-side circuit of
forward converter for multi-output of PC power application.
A common technique for winding multiple outputs with the
same polarity sharing a common ground is to stack the
secondary windings instead of winding each output
winding separately. This approach improves the load
regulation of the stacked outputs. The winding NS1 in
Figure 19 must be sized to accommodate its output current,
plus the current of the output (+12V) stacked on top of it.
To get tight regulation of 3.3V output, magnetic amplifier
(MAG-AMP) is used. The saturable core of MAG-AMP
prevents the diode DREC from fully conducting by
introducing high impedance until it is saturated. This
allows the effective duty cycle of VREC to be controlled to
be regulated the output voltage.
Once the core for the transformer is determined, the
minimum number of turns for the transformer primary-side
to avoid saturation is given by:
MIN
V
DMAX
N P MIN = BOUT
(44)
Ae f SW ΔB
where Ae is the cross sectional area of the core in m2, fSW is
the switching frequency, and ΔB is the maximum flux
density swing in Tesla for normal operation. ΔB is typically
0.2-0.3 T for most power ferrite cores in the case of a
forward converter.
The turn ratio between the primary-side and secondary-side
winding for the first output is determined by:
MIN
N
V
DMAX
n = P = BOUT
(45)
(VO1 + VF 1 )
N S1
where VF is the diode forward-voltage drop.
Next, determine the proper integer for NS1 resulting in Np
larger than Npmin. Once the number of turns of the first
output is determined, the number of turns of other output
(n-th output) can be determined by:
VO ( n ) + VF ( n )
N S (n) =
⋅ N S1
(46)
VO1 + VF 1
The golden ratio between the secondary-side windings for
the best regulation of 3.3V, 5V, and 12V is known as
2:3:7.
(Design Example) The minimum PFC output voltage
is 310V and the maximum duty cycle of PWM
controller is 50%. By adding 5% margin to the
maximum duty cycle, DMAX=0.45 is used for
transformer design. Assuming ERL35 (Ae=107mm2)
core is used and ΔB=0.28, the minimum turns for the
transformer primary side is obtained as:
Additiona
l LC filter
N P MIN =
+12V
NS
The turns ratio for 5V output is obtained as:
2
Np
VBOUT MIN DMAX
310 ⋅ 0.45
=
= 72
Ae f SW ΔB
107 × 10 −6 ⋅ 65 × 103 ⋅ 0.28
n=
Additiona
l LC filter
NS
+5V
N P VBOUT MIN DMAX
310 ⋅ 0.45
=
=
= 25.6
(VO + VF )
(5 + 0.45)
NS
The number of turns for the primary-side winding is
determined as:
1
N p = n ⋅ N S 1 = 2 × 25.6 = 51.2 < N P MIN
MAG AMP
Control
MAG
AMP
+3.3
V
DREC
+
VREC
N p = n ⋅ N S 1 = 3 × 25.6 = 76.8 > N P MIN ∴ N S 1 = 3
Then, the turns ratio for 12V output is obtained as:
Additiona
l LC filter
NS 2 =
-
NS
Additiona
l LC filter
VO 2 + VF 2
12 + 0.7
⋅ N S1 =
⋅ 3 = 6.99 ≅ 7
5 + 0.45
VO1 + VF 1
Therefore, the number of turns for each winding is
obtained as:
-12V
Np=78, NS1=3, NS2=7 (3+4 stack) and NS3=7.
3
Figure 19. Typical Secondary-Side Circuit
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
www.fairchildsemi.com
11
AN-8027
[STEP-11] Coupled Inductor Design for the
PWM Stage
When the forward converter has more than one output, as
shown in Figure 20, coupled inductors are usually employed
to improve the cross regulation and to reduce the ripple.
They are implemented by winding their separate coils on a
single, common core. The turns ratio should be the same as
the transformer turns ratio of the two outputs as:
NS 2 NL2
=
(47)
N S 1 N L1
L2
N
p
NL2
N S2
VO2
NL1
One way to understand the operation of coupled inductor is
to normalize the outputs to one output. Figure 21 shows
how to normalize the second output (VO2) to the first
output (VO1). The transformer and inductor turns are
divided by NS2/NS1, the voltage and current are adjusted by
NS2/NS1. It is assumed that the leakage inductances of the
coupled inductor are much smaller than the magnetizing
inductance and evenly distributed for each winding.
The inductor value of the first output can be obtained by:
VO1 (VO1 + VF 1 )
L1 =
⋅ (1 − DMIN )
ΔI
(48)
f SW ( PO1 + PO 2 ) SUM
I SUM
where:
MIN
V
DMIN = DMAX BOUT
VBOUT
(49)
PO1 + PO 2
I SUM =
VO1
Then, the ripple current for each output is given as:
ΔI O1 ΔI SUM 1
=
⋅
2
I O1
I O1
L1
N S1
ΔI O 2 ΔI SUM N S 1 1
=
⋅
⋅
2
IO 2
N S 2 IO 2
VO1
(50)
(51)
Figure 20. Coupled Inductor
D1
L1
(Design Example) The minimum duty cycle of
PWM stage at nominal input (PFC output) voltage is:
VO1
DMIN = DMAX
IO1
VPOUT ⋅ N S1
NP
0
The sum of two normalize output current is:
D2
VPOUT ⋅ N S 2
NP
VO2
I SUM =
IO2
L2
VO 2 N =
N S1
VO 2 = VO1
NS 2
IO 2 N =
NS 2
IO 2
N S1
LM
PO1 + PO 2 243
=
= 48.6 A
VO1
5
Assuming 16% p-p ripple current in LSUM, the inductor
for the first output is obtained as:
0
ISUM
VBOUT MIN
310
= 0.45
= 0.36
389
VBOUT
L1 =
Normalized
=
D1
LLK
VO1
IO2
VPOUT ⋅ N S1
NP
LLK
5(5 + 0.45)
⋅ (1 − 0.36) = 6.9uH
65 × 103 (5 × 9 + 12 × 16.5) ⋅ 0.16
Then, the ripple current for each output is given as:
ΔI O1 ΔI SUM 1
48.6 × 0.16 1
=
⋅
=
⋅ = 43%
2
2
9
I O1
I O1
0
D2N
VO1 (VO1 + VF 1 )
⋅ (1 − DMIN )
ΔI
f SW ( PO1 + PO 2 ) SUM
I SUM
VO2N
ΔI O 2 ΔI SUM N S 1 1
48.6 × 0.16 3 1
=
⋅
⋅
=
⋅ ⋅
= 10%
IO 2
N S 2 IO 2
2
2
7 16.5
IO2N
Figure 21. Normalized Coupled Inductor Circuit
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
www.fairchildsemi.com
12
AN-8027
[STEP-12] PWM Ramp Circuit Design
For voltage-mode operation, the RAMP pin can be
connected to a DC voltage through a resistor. When it is
connected to the input of forward converter, ramp signal
slope is automatically adjusted according to the input
voltage providing line feed-forward operation. However, it
can cause more power dissipation in the resistor. For better
efficiency and lower standby power consumption, it is
recommended to connect the RAMP pin to the VREF pin.
[STEP-13] Feedback Compensation Design
for PWM Stage
Figure 21 shows the typical cross regulation compensation
circuit configuration for multi-output converters. The small
signal characteristics of the compensation network is given as:
)
vFBPWM
=−
1 + s / ωCZ 1 )
1 + s / ωCZ 2 )
RB
⋅(
vO1 +
vO 2 )
1 + s / ωCP ROS 1 RD CF s
ROS 2 RD CF s
(53)
where:
ωCP =
1
( RB1 // RB 2 )CB
ωCZ 1 =
1
RF C F
ωCZ 2 =
1
( RF + ROS 2 )CF
(54)
VO2
VO1
VREF
RB1
RD
Figure 22. Ramp Generation Circuit for PWM
ROS1
ROS2
FBPWM
It is typical to use 470pF~1nF capacitor on the RAMP pin
and to have the peak of the ramp signal around 2~3V.
RF
RB
The peak of the ram voltage is given as:
VRAMP PK =
1
CRAMP
⋅
VREF
1
⋅
RRAMP 2 f SW
(52)
=
1
CRAMP
⋅
VREF
1
⋅
RRAMP 2 f SW
1
7.5
1
⋅
⋅
= 2.6V
−9
3
1× 10 22 × 10 2 ⋅ 65 × 103
KA431
CB
(Design Example) Selecting CRAMP and RRAMP as
1nF and 22k, the PWM ramp voltage is obtained as:
VRAMP PK =
CF
ROS3
Figure 23. Feedback Compensation Circuit for
PWM Stage
The small signal equivalent circuit for control-to-output
transfer function of the PWM power stage can be simplified
as shown in Figure 24. The transfer function is fourth-order
system because additional LC filters are used to meet the
output voltage ripple specification. Therefore, it is
recommended to use engineering software, such as PSPICE
or Mathlab, to design the feedback loop.
VBOUT ⋅
N S1
NS 2
LM
LLK
LL12
CO11
VO1
RL1
CO12
1
VO2N
VRAMP
VFBPWM
LLK
CO21N
VO2
L22N
CO22N
RL2N
NS1:NS2
Figure 24. Simplified Small Signal Equivalent Circuit
for Control-to-Output Transfer Function
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
www.fairchildsemi.com
13
AN-8027
Design Summary
Application
Output Power
Input Voltage
Output Voltage / Output Current
ATX Power
300W
85~264VAC
12V/16.5A;5V/9A;-12V/0.8A ;3.3V/13.5A
Features
„
Meets 80+ specification
„
FAN4800A fully pin-to-pin compatible with ML4800 and FAN4800 (needs few parts modify)
„
Switch-charge technique of gain modulator can provide better PF and lower THD
„
„
Leading and trailing modulation technique for reduce output ripple
Protections: OVP (Over-Voltage Protection), UVP (Under-Voltage Protection), OLP (Open-Loop Protection), and
maximum current limit
F1
AC
Input
FDA18N50
CIF1
VBOUT
DBOOST
L BOOST
BYC10600
CBOOST
RCS1
100 nF
1000uF
L22 2uH
DF2
FCP11N60
10 k
R RAMP
2200uF
DR2
L2
D2
CO12
FR157
10 k
Q3
FR157
DR2
2200uF
SF34DG
RLF2
0.13nF CIC2
53nF
CRMS1 200 K
200n
36 k
F
CRMS2 R
RMS3
R IAC
6M
RLF1
RT
6.9 k
RB 7.5k
RRMS2
CT
1nF
0.47nF
17k
4nF
RIC
CIC1
IEA
VEA
IAC
FBPFC
ISENSE
SS
VREF
VD
D
OPFC
FBPWM
OPWM
VRMS
CLF1 CSS
RT/CT
GND
RAMP
ILIMIT
1nF
CRAMP
CB
1000uF
Vo3 -12V
L3
1
2M
Vo2 5V
CO22
CO21
1
STPS60L45CW
22k
RRMS1
Vo1 12V
CO11
DF1
DR1
Q2
0.1
D1
2
DR1
FCP11N60
R FB1
2M
Q1
270uF
5
1.8uH
L1
L 11
FYPF2006DN
220uF
CO31
RCS2
1
L
DF2
4
2
CO21
CO22
FR157
10
362k
VD
D
RVC
10
20nF
1k
FR157
3.7nF
CVC2
CVC1
FAN480X
3.3V
Vo4
L43
L4
12.9k
RFB2
CFB
1.2 k
Vo1
RD
CF
300
12V
RF
100
nF
32.4 k
ROS1
11 k
ROS2
CLF2 CDD CREF
4.64 k
3.4 k
1uF 10 k
5.45 k
5V
Vo2
ROS3
Figure 25. Final Schematic of Design Example
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
www.fairchildsemi.com
14
AN-8027
Margin Tape
3mm
Margin Tape
3mm
Mylar Tape 3T
N5
Mylar Tape 1T
N4
Mylar Tape 1T
N3
Mylar Tape 1T
N2
N1
BOBBIN-ERL35
Mylar Tape 3T
Figure 26. Forward Converter Transformer Structure
Winding Specification
No
Pin(s-f)
Wire
N1
3-2
0.6Ф
Insulation: Mylar Tape t = 0.03mm, 3 Layers
N2
8,9-10,11,12
Copper-Foil 10mil
Insulation: Mylar Tape t = 0.03mm, 1 Layers
N3
13-8,9
1.0Ф*4
Insulation: Mylar Tape t = 0.03mm, 1 Layers
N4
10,11,12-14
0.4Ф
Insulation: Mylar Tape t = 0.03mm, 1 Layers
N5
2-6,7
0.6Ф
Insulation: Mylar Tape t = 0.03mm, 3 Layers
Core-ERL35
Insulation: Mylar Tape t = 0.03mm, 3 Layers
Insulation: Copper-Foil Tape t = 0.05mm-pin1 Open Loop
Insulation: Mylar Tape t = 0.03mm, 3 Layers
Turns
37Ts
Winding Method
Solenoid Winding
3Ts
Copper-Foil Width 18mm
4Ts
Solenoid Winding
6Ts
Solenoid Winding
37Ts
Solenoid Winding
Core: ERL35 (Ae=107 mm2)
Bobbin: ERL35
Inductance: 13mH
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
www.fairchildsemi.com
15
AN-8027
Appendix A
FAN480X Series Comparison Table of Relevant Parameters
FAN4800
VDD Maximum Rating
VDD OVP
VCC UVLO
Two-Level PFC
Output
PFC Soft-Start
Brownout
PFC:PWM
Frequency
Frequency Range
Gate Clamp
PFC Multiplier
VINOK
PWM Maximum Duty
Startup Current
Soft-Start Current
PWM Comparator
Level Shift
RAC
New
Generation
New
Generation
New
Generation
New
Generation
FAN4800A
FAN4800C
FAN4801
FAN4802/2L
20V
17.9V/Clamp
10V/13V
30V
28/Auto-Recover
9.3/11V
NO
NO
YES
NO
NO
1:1
YES
YES
1:1
1:2
1:1
68kHz~81kHz
NO
Traditional
2.25V/1.1V
42%~49%
100μA
20μA
50kHz~75kHz
16V
Switching Charge
2.40V/1.15V
49.5%~50%
30μA
10μA
1.0V
1.5V
1~2MΩ
5~8 MΩ
1:2
MOSFET and Diode Reference Specification
PFC MOSFETs
Voltage Rating
500V
600V
Part Number
FQP13N50C, FQPF13N50C, FDP18N50, FDPF18N50, FDA18N50, FDP20N50(T),
FDPF20N50(T)
FCP11N60, FCPF11N60, FCP16N60, FCPF16N60, FCP20N60S, FCPF20N60S, FCA20N60S,
FCP20N60, FCPF20N60
Boost Diodes
600V
FFP08H60S, FFPF10H60S, FFP08S60S, FPF08S60SN, BYC10600
PWM MOSFETs
500V
600V
FQP/PF9N50C, FQPF9N50C, FQP13N50C, FQPF13N50C, FQA13N50C, FDP18N50,
FDPF18N50, FDP20N50(T), FDPF20N50(T)
FCP11N60, FCPF11N60, FCP16N60, FCPF16N60, FCA16N60, FCP20N60S, FCPF20N60S,
FCA20N60S, FCP20N60, FCPF20N60, FCA20N60
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
www.fairchildsemi.com
16
AN-8027
References
FAN480X — PFC/Forward PWM Controller Combo (FAN4800, FAN4801, FAN4802)
AN-6078SC — FAN480X PFC+PWM Combo Controller Application
AN-6004 — 500W Power Factor Corrected (PFC) Design with FAN4810
AN-6032 — FAN4800 Combo Controller Applications
AN-42030 — Theory and Application of the ML4821 Average Current Mode PFC Controller
AN-42009 — ML4824 Combo Controller Applications
ATX 300W 80+ Evaluation Board of FAN4800A+SG6520+FSQ0170
DISCLAIMER
FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY PRODUCTS
HEREIN TO IMPROVE RELIABILITY, FUNCTION, OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY LIABILITY ARISING OUT OF
THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER DOES IT CONVEY ANY LICENSE
UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS.
LIFE SUPPORT POLICY
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR
SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION.
As used herein:
1. Life support devices or systems are devices or systems
2. A critical component is any component of a life support
which, (a) are intended for surgical implant into the body, or
(b) support or sustain life, or (c) whose failure to perform
when properly used in accordance with instructions for use
provided in the labeling, can be reasonably expected to
result in significant injury to the user.
© 2009 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 8/26/09
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
www.fairchildsemi.com
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