ON NCP4306 Secondary side synchronous rectification driver Datasheet

NCP4306
Secondary Side
Synchronous Rectification
Driver for High Efficiency
SMPS Topologies
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The NCP4306 is high performance driver tailored to control a
synchronous rectification MOSFET in switch mode power supplies.
Thanks to its high performance drivers and versatility, it can be used in
various topologies such as DCM or CCM flyback, quasi resonant
flyback, forward and half bridge resonant LLC.
The combination of externally or fixed adjustable minimum
off-time and on-time blanking periods helps to fight the ringing
induced by the PCB layout and other parasitic elements. A reliable and
noise less operation of the SR system is insured due to the Self
Synchronization feature. The NCP4306 also utilizes Kelvin
connection of the driver to the MOSFET to achieve high efficiency
operation at full load and utilizes a light load detection architecture to
achieve high efficiency at light load.
The precise turn−off threshold, extremely low turn−off delay time
and high sink current capability of the driver allow the maximum
synchronous rectification MOSFET conduction time and enables
maximum SMPS efficiency. The high accuracy driver and 5 V gate
clamp enables the use of GaN MOSFETs.
8
1
SOIC−8 NB
CASE 751−07
MARKING DIAGRAMS
8
1
• Self−Contained Control of Synchronous Rectifier in CCM, DCM and
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
© Semiconductor Components Industries, LLC, 2017
December, 2017 − Rev. 1
1
XXXXX
ALYWX
G
XXXAYWG
G
1
SOIC−8 NB
TSOP−6
IC(Pb−Free)
IC (Pb−Free)
XXXXX
A
L
Y
W
G
Features
QR for Flyback or LLC Applications
Precise True Secondary Zero Current Detection
Typically 15 ns Turn off Delay from Current Sense Input to Driver
Rugged Current Sense Pin (up to 200 V)
Ultrafast Turn−off Trigger Interface / Disable Input (10.5 ns)
Adjustable or Fixed Minimum ON−Time
Adjustable or Fixed Minimum OFF-Time with Ringing Detection
Improved Robust Self Synchronization Capability
7 A / 2 A Peak Current Sink / Source Drive Capability
Operating Voltage Range up to VCC = 35 V
Automatic Light−load Disable Mode
GaN Transistor Driving Capability
Low Startup and Disable Current Consumption
Maximum Operation Frequency up to 1 MHz
TSOP6 and SOIC-8 Packages
This is a Pb−Free Device
1
TSOP−6
CASE 318G−02
= Specific Device Code
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
(Note: Microdot may be in either location)
ORDERING INFORMATION
See detailed ordering and shipping information on page 2 of
this data sheet.
Typical Applications
• Notebook Adapters
• High Power Density AC / DC Power
•
•
Supplies (Cell Phone Chargers)
LCD TVs
All SMPS with High Efficiency
Requirements
Publication Order Number:
NCP4306/D
NCP4306
ORDERING INFORMATION TABLE
Table 1. AVAILABLE DEVICES
Device
Package
Package Marking
SOIC8
NCP4306AAAZZZADR2G
6AAAZZZA
NCP4306AADZZZADR2G
6AADZZZA
NCP4306AAHZZZADR2G
6AAHZZZA
TSOP6
NCP4306DADZZDASNT1G
6AC
NCP4306DAHZZAASNT1G
Shipping †
Packing
6AD
SOIC−8
(Pb−Free)
2500 / Tape and Reel
TSOP−6
(Pb−Free)
3000 / Tape and Reel
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specification Brochure, BRD8011 / D.
See the ON Semiconductor Device Nomenclature document (TND310 / D) for a full description of the naming convention
used for image sensors. For reference documentation, including information on evaluation kits, please visit our web site at
www.onsemi.com.
C3
R1
Tr1
M1
RLLD
+VBULK
MIN_TON
LLD
RMIN_TON
RMIN_TOFF
VCC
MIN_TOFF
NCP4306
+VOUT
M3
N2
LLC
STAGE
CONTROL
DRV
GND
CS
TRIG
C2
RTN
M2
N1
D1
N3
M4
C1
OK1
DRV
VCC
MIN_TOFF GND
CS
MIN_TON
LLD
RLLD
RMIN_TON
RMIN_TOFF
R2
C4
TRIG
NCP4306
Figure 1. Typical Application Example – LLC Converter with optional LLD and Trigger Utilization
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2
NCP4306
+VOUT
VBULK
TR1
R1
C1
C2
C5
R3
D3
VCC
FLYBACK
CONTROL
CIRCUITRY
M2
D4
C3
DRV
M1
VCC
FB
GND
C4
CS
DRV
MIN_TOFF GND
OK1
CS
TRIG
LLD
D5
NCP4306
RLLD
R2
RMIN_TON
RMIN_TOFF
MIN_TON
Figure 2. Typical Application Example – DCM, CCM or QR Flyback Converter with
optional LLD and disabled TRIG
+VOUT
VBULK
C1
TR1
R1
C2
C5
D3
R3
VCC
M2
D4
FLYBACK
CONTROL
CIRCUITRY
C3
M1
DRV
CS
VCC
CS
LLD
NCP4306
RMIN_TOFF
GND MIN_TOFF
RLLD
DRV
FB
GND
C4
D5
OK1
Figure 3. Typical Application Example – DCM, CCM or QR Flyback Converter with
NCP4306 in TSOP6 (v Cxxxxxx)
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3
NCP4306
+VOUT
VBULK
TR1
R1
C1
C2
R3
C8
R5
D3
VCC
C4
C3
DRV
C7
M1
GND
DRV
VCC
GND MIN_TOFF
COMP CS
CS
R2
R4
MIN_TON
NCP4306
RMIN_TOFF
R3
M2
D4
PRIMARY
SIDE
FLYBACK
CONTROLLER
RMIN_TON
ZCD
C5
C6
Figure 4. Typical Application Example – Primary Side Flyback Converter and NCP4306 in TSOP6
PIN FUNCTION DESCRIPTION
Table 2. PIN FUNCTION DESCRIPTION
TSOP6
Bxxxxxx
TSOP6
Cxxxxxx
TSOP6
Dxxxxxx
TSOP6
Exxxxxx
TSOP6
Fxxxxxx
TSOP6
Gxxxxxx
SOIC8
Axxxxxx
6
6
6
6
6
6
1
VCC
−
5
5
5
−
2
MIN_TOFF
Adjust the minimum off time
period by connecting resistor to
ground
5
−
4
−
5
−
3
MIN_TON
Adjust the minimum on time
period by connecting resistor to
ground
4
4
−
−
−
4
4
LLD
This input modulates the driver
clamp level and / or turns the driver off during light load conditions
−
−
−
4
4
5
5
TRIG / DIS
Ultrafast turn−off input that can be
used to turn off the SR MOSFET
in CCM applications in order to
improve efficiency. Activates
disable mode if pulled−up for
more than 100 μs
3
3
3
3
3
3
6
CS
Current sense pin detects if the
current flows through the SR
MOSFET and / or its body diode
2
2
2
2
2
2
7
GND
Ground connection for the SR
MOSFET driver and VCC
decoupling capacitor. Ground
connection for minimum tON and
tOFF adjust resistors, LLD and
trigger inputs
GND pin should be wired directly
to the SR MOSFET source
terminal / soldering point using
Kelvin connection
1
1
1
1
1
1
8
DRV
Driver output for the SR MOSFET
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4
Pin Name
Description
Supply voltage pin
NCP4306
Exception time
generator
MIN_TON
EN
ELAPSED
DISABLE
Disable detection
LLD
EXT_ADJ
INT_ADJ
CS
Minimum ON time
generator
CS
detection
ELAPSED
INT_ADJ
EN
DRIVER
dV/dt
CS_ON
CS_OFF
CS_RESET
DRVOUT
DRV
Control logic
VDD
RESET
MIN_TOFF
EXT_ADJ
INT_ADJ
Minimum OFF
time generator
ELAPSED
EN
TRIG
DISABLE
VCC managment
UVLO
VCC
DISABLE
TRIG/DIS
Disable detection
10 μA
VTRIG
Figure 5. Internal Circuit Architecture – NCP4306
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5
GND
NCP4306
ABSOLUTE MAXIMUM RATINGS
Table 3. ABSOLUTE MAXIMUM RATINGS
Rating
Symbol
Value
Unit
VCC
−0.3 to 37.0
V
TRIG / DIS, MIN_TON, MIN_TOFF, LLD Input Voltage (Note 3)
VTRIG / DIS, VMIN_TON, VMIN_TOFF, VLLD
−0.3 to VCC
V
VDRV
−0.3 to 17.0
V
Supply Voltage
Driver Output Voltage
Current Sense Input Voltage
Current Sense Dynamic Input Voltage (tPW = 200 ns)
MIN_TON, MIN_TOFF, LLD, TRIG Input Current
VCS
−4 to 200
V
VCS_DYN
−10 to 200
V
IMIN_TON, IMIN_TOFF, ILLD, ITRIG
−10 to 10
mA
DRV Pin Current (tPW = 10 μs)
IDRV_DYN
−3 to 12
A
VCC Pin Current (tPW = 10 μs)
IVCC_DYN
3
A
Junction to Air Thermal Resistance, 1 oz 1 in2 Copper Area,
SOIC8
RθJ−A_SOIC8
200
°C / W
Junction to Air Thermal Resistance, 1 oz 1 in2 Copper Area
TSOP6
RθJ−A_TSOP6
250
°C / W
TJMAX
150
°C
Maximum Junction Temperature
Storage Temperature
TSTG
−60 to 150
°C
ESD Capability, Human Body Model (except pin CS) (Note 1)
ESDHBM
2000
V
ESD Capability, Human Body Model Pin CS
ESDHBM
600
V
ESD Capability, Machine Model (Note 1)
ESDMM
200
V
ESD Capability, Charged Device Model (Note 1)
ESDCDM
Class C1
-
Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality
should not be assumed, damage may occur and reliability may be affected.
1. This device series contains ESD protection and exceeds the following tests:
Except pin CS: Human Body Model 2000 V per JEDEC Standard JESD22−A114E.
All pins: Machine Model Method 200 V per JEDEC Standard JESD22−A115−A
Charged Machine Model per JEDEC Standard JESD22−C101F
2. This device meets latchup tests defined by JEDEC Standard JESD78D.
3. If voltage higher than 22 V is connected to pin, pin input current increases. Internal ESD clamp contains 24 V Zener diode with 3 kΩ in series.
It is recommended to add serial resistance in case of higher input voltage to limit input pin current.
Table 4. RECOMMENDED OPERATING CONDITION
Parameter
Maximum Operating Voltage
Operating Junction Temperature
Symbol
Min
Max
35
V
−40
125
°C
VCC
TJ
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6
Unit
NCP4306
ELECTRICAL CHARACTERISTICS
Table 5. ELECTRICAL CHARACTERISTICS
−40 ºC ≤ TJ ≤ 125 ºC; VCC = 12 V; CDRV = 0 nF; RMIN_TON = RMIN_TOFF = 10 kΩ or internally set values; VLLD = 3.0 V or LLD internally
disabled; VTRIG / DIS = 0 V; VCS = 4 V, unless otherwise noted. Typical values are at TJ = +25 ºC
Test Conditions
Parameter
Symbol
Min
Typ
Max
Unit
VCCON
3.7
4.0
4.2
V
VCCOFF
3.2
3.5
3.7
SUPPLY SECTION
VCC UVLO
VCC rising
VCC falling
VCC UVLO Hysteresis
Start−up Delay
Current Consumption,
tMIN_TON = tMIN_TOFF = 1 μs,
tLLD = 130 μs
VCC rising from 0 to VCCON + 1 V @ tr
= 10 μs
VCCHYS
0.5
V
tSTART_DEL
50
80
μs
ICC
1.8
2.5
mA
CDRV = 0 nF,
fCS = 100 kHz
xAxxxxx
xBxxxxx
1.7
2.4
CDRV = 1 nF,
fCS = 100 kHz
xAxxxxx
2.8
4.0
xBxxxxx
2.1
3.4
CDRV = 10 nF,
fCS = 100 kHz
xAxxxxx
12
15
xBxxxxx
6.7
9.0
Current Consumption
ICC
1.4
2.2
mA
Current Consumption below UVLO
VCC = VCCOFF – 0.1 V
ICC_UVLO
35
60
μA
Current Consumption in Disable Mode
t > tLLD , VLLD = 0.55 V
ICC_DIS
60
100
μA
VTRIG / DIS = 5 V; VLLD = 0.55 V
60
100
t > tLLD, LLD set internally
37
80
VTRIG / DIS = 5 V, LLD set internally
37
80
DRIVER OUTPUT
Output Voltage Rise−Time
CDRV = 10 nF, 10 % to 90 % VDRVMAX,
VCS = 4 to −1 V
tr
60
100
ns
Output Voltage Fall−Time
CDRV = 10 nF, 90 % to 10 % VDRVMAX,
VCS = −1 to 4 V
tf
25
45
ns
Driver Source Resistance
Driver Sink Resistance
Output Peak Source Current
Output Peak Sink Current
Maximum Driver Pulse Length
Maximum Driver Output Voltage
RDRV_SOURCE
2
Ω
RDRV_SINK
0.5
Ω
IDRV_SOURCE
2
A
IDRV_SINK
7
A
tDRV_ON_MAX
4
ms
VDRVMAX
VCC = 35 V, CDRV > 1 nF,
(ver. xAxxxxx)
VCC = 35 V, CDRV > 1 nF,
(ver. xBxxxxx)
Minimum Driver Output Voltage
VDRVMIN
VCC = VCCOFF + 200 mV,
(ver. xAxxxxxx)
VCC = VCCOFF + 200 mV,
(ver. xBxxxxxx)
9
10
11
4.5
5.0
5.5
3.4
3.7
3.9
3.4
3.7
3.9
V
V
CS INPUT
Total Propagation Delay From CS to
DRV Output On
VCS goes down from 4 to −1 V,
tf_CS <= 5 ns
tPD_ON
30
60
ns
Total Propagation Delay From CS to
DRV Output Off
VCS goes up from −1 to 4 V,
tr_CS <= 5 ns
tPD_OFF
13
23
ns
−75
−40
mV
0
mV
0.6
V
500
nA
Turn On CS Threshold Voltage
Turn Off CS Threshold Voltage
Guaranteed by Design
Turn Off Timer Reset Threshold Voltage
CS Leakage Current
VCS = 200 V
VTH_CS_ON
−120
VTH_CS_OFF
−1
VTH_CS_RESET
0.4
0.5
ICS_LEAKAGE
dV / dt Detector High Threshold
VCS_DVDT_H
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7
3.0
V
NCP4306
Table 5. ELECTRICAL CHARACTERISTICS (continued)
−40 ºC ≤ TJ ≤ 125 ºC; VCC = 12 V; CDRV = 0 nF; RMIN_TON = RMIN_TOFF = 10 kΩ or internally set values; VLLD = 3.0 V or LLD internally
disabled; VTRIG / DIS = 0 V; VCS = 4 V, unless otherwise noted. Typical values are at TJ = +25 ºC
Parameter
Test Conditions
Symbol
Min
Typ
Max
Unit
37
ns
10
ns
CS INPUT
dV / dt Detector Low Threshold
dV / dt Detector Threshold
VCS_DVDT_L
(Note 4)
ver. xxDxxxx
tdV / dt
0.5
13
25
V
TRIGGER DISABLE INPUT
Minimum Trigger Pulse Duration
VTRIG / DIS = 5 V; Shorter pulses may
not be proceeded
Trigger Threshold Voltage
Trigger to DRV Propagation Delay
tTRIG_PW_MIN
VTRIG_TH
VTRIG / DIS goes from 0 to 5 V,
tr_TRIG / DIS <= 5 ns
Trigger Blank Time After DRV Turn−on VCS drops below VTH_CS_ON
Event
1.6
tPD_TRIG
2.0
2.2
V
10.0
16.5
ns
tTRIG_BLANK
30
55
80
ns
75
100
125
μs
1.5
3.0
μs
200
ns
15
μA
10
μs
Delay to Disable Mode
VTRIG / DIS goes from 0 to 5 V
tDIS_TIM
Disable Recovery Timer
VTRIG / DIS goes down from 5 to 0 V;
tMIN_TOFF = 130 ns
tDIS_REC
Minimum Pulse Duration to Disable
Mode End
VTRIG / DIS = 0 V; Shorter pulses may
not be proceeded
tDIS_END
Pull Down Current
VTRIG / DIS = 5 V
Maximum Transition Time
VTRIG / DIS goes from 1 to 3 V or from
3 to 1 V
ITRIG / DIS
7
11
tTRIG_TRAN
MINIMUM TON AND TOFF ADJUST
Minimum tON time
RMIN_TON = 0 Ω (ver. xxxZxxx)
tON_MIN
55
Minimum tOFF time
RMIN_TOFF = 0 Ω (ver. xxxxZxx)
tOFF_MIN
70
Minimum tON time
RMIN_TON = 10 kΩ (ver. xxxZxxx)
tON_MIN
0.90
1.00
1.10
μs
Minimum tOFF time
RMIN_TOFF = 10 kΩ (ver. xxxxZxx)
tOFF_MIN
0.90
1.00
1.10
μs
Minimum tON time
RMIN_TON = 50 kΩ (ver. xxxZxxx)
tON_MIN
4.50
5.00
5.50
μs
Minimum tOFF time
RMIN_TOFF = 50 kΩ (ver. xxxxZxx)
tOFF_MIN
4.40
4.90
5.40
μs
Internal minimum tON time
tON_MIN = 130 ns, (ver. xxxAxxx)
tON_MIN
−20%
tON_MIN
+20%
ns
tON_MIN = 220, 310, 400 ns (ver.
xxx[B−D]xxx)
tON_MIN
−15%
tON_MIN
+15%
ns
tON_MIN = 500, 600, 800, 1000, 1200,
1400, 1700, 2000 ns
(ver. xxx[E−L]xxx)
tON_MIN
−10%
tON_MIN
+10%
ns
tOFF_MIN = 0.9, 1.0, 1.1, 1.2, 1.4, 1.6,
1.8, 2.0, 2.2, 2.4, 2.6, 2.9, 3.2, 3.5,
3.9 μs (ver. xxxx[A−O]xx)
tOFF_MIN
−10%
tOFF_MI
+10%
μs
−19
μA
0.3
V
Internal minimum tOFF time
ns
ns
N
LLD ADJUST
LLD Pull Up Current
LLD Time Selection
(ver. xxxxxZx)
ILLD
IC disabled
VLLD
−20
tLLD = 68 μs
0.40
0.51
0.63
tLLD = 130 μs
0.75
0.89
1.03
tLLD = 280 μs
1.15
1.32
1.50
tLLD = 540 μs
1.68
1.82
1.97
tLLD = 1075 μs
2.20
2.50
2.70
LLD function disabled
LLD Main Time
−21
3.10
VLLD = 0.51 V or ver. xxxxxAx
tLLD
53
68
83
VLLD = 0.89 V or ver. xxxxxBx
100
130
160
VLLD = 1.32 V or ver. xxxxxCx
220
280
340
VLLD = 1.82 V or ver. xxxxxDx
420
540
660
VLLD = 2.45 V or ver. xxxxxEx
840
1075
1310
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8
μs
NCP4306
Table 5. ELECTRICAL CHARACTERISTICS (continued)
−40 ºC ≤ TJ ≤ 125 ºC; VCC = 12 V; CDRV = 0 nF; RMIN_TON = RMIN_TOFF = 10 kΩ or internally set values; VLLD = 3.0 V or LLD internally
disabled; VTRIG / DIS = 0 V; VCS = 4 V, unless otherwise noted. Typical values are at TJ = +25 ºC
Parameter
Test Conditions
Symbol
Min
Typ
Max
Unit
LLD ADJUST
LLD Reduced Time
Disable mode activated
LLD Blanking Time
Disable Recovery Time
tMIN_TOFF = 130 ns
tLLD_RED
0.5 ×
tLLD
μs
tLLD_BLK
0.25 ×
tLLD
μs
tLLD_DIS_REC
1.5
tEXC
4×
tMIN_TON
3.0
μs
EXCEPTION TIMER
Exception Time
(ver. xxHxxxx)
Exception Timer Ratio Accuracy
RatioEXC
4. Test signal:
VCS [V]
4.0
VCS_DVDT_H
t dV/dt
1.5
VCS_DVDT_L
t [ns]
−1.0
Figure 6. Test Signal
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9
−15
μs
+15
%
NCP4306
TYPICAL CHARACTERISTICS
4,2
1,6
4,1
1,4
4,0
1,2
3,8
ICC[mA]
VCC[V]
3,9
VCC on
3,7
VCC off
3,6
1,0
0,6
3,5
0,4
3,4
0,2
3,3
−40
−20
0
20
40
TJ[°C]
60
80
100
0,0
120
TJ = 125 °C
TJ = 85 °C
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
0,8
0
14
3,0
12
2,0
1,5
1,0
0,5
10
15
20
25
30
35
10
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
20
25
30
ICC[mA]
ICC[mA]
2,5
5
15
Figure 8. Current Consumption VCS = 4 V
3,5
0
10
VCC[V]
Figure 7. VCCON and VCCOFF Levels
0,0
5
CDRV = 0 nF
8
CDRV = 1 nF
6
CDRV = 10 nF
4
2
0
−40
35
−20
0
20
VCC[V]
40
60
80
100
120
TJ[°C]
Figure 9. Current Consumption,
fCS = 100 kHz, CDRV = 1 nF, Ver. xAxxxxx
Figure 10. Current Consumption,
fCS = 100 kHz, Ver. xAxxxxx
10
2,5
9
8
1,5
1,0
0,5
0,0
7
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
0
5
10
15
20
25
30
ICC[mA]
ICC[mA]
2,0
6
5
CDRV = 0 nF
4
CDRV = 1 nF
3
CDRV = 10 nF
2
1
0
−40
35
−20
0
20
40
60
80
100
VCC[V]
TJ[°C]
Figure 12. Current Consumption,
fCS = 100 kHz, CDRV = 1 nF, Ver. xBxxxx
Figure 11. Current Consumption,
fCS = 100 kHz, Ver. xBxxxxx
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10
120
NCP4306
60
100
90
80
40
ICC_DIS[μA]
ICC_UVLO[μA]
50
30
20
10
70
60
50
40
30
0
−40
−20
0
20
40
60
80
100
20
−40
120
−20
0
20
TJ[°C]
100
120
120
90
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
TJ = 55 °C
100
80
70
ICC_DIS[μA]
ICC_DIS[μA]
80
Figure 14. Current Consumption in Disable Mode
VCS = 4 V, t > tLLD
100
60
50
40
80
60
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
40
20
30
20
−40
−20
0
20
40
TJ[°C]
60
80
100
0
120
0
0,0
90
−0,1
TJ = 85 °C
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
60
50
40
30
15
20
−0,3
ICS[mA]
70
10
15
25
30
35
−0,2
TJ = 125 °C
TJ = 105 °C
80
5
10
Figure 16. Current Consumption in Disable Mode,
VCS = 4 V, t > tLLD
100
0
5
VCC[V]
Figure 15. Current Consumption in Disable Mode,
VTRIG/DIS = 5 V
ICC_DIS[μA]
60
TJ[°C]
Figure 13. Current Consumption below UVLO,
VCC = VCCOFF − 0.1 V
20
40
20
25
30
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
−0,4
−0,5
−0,6
−0,7
−0,8
−0,9
35
−1,0
−1 −0,8 −0,6 −0,4 −0,2
VCC[V]
0
0,2
0,4
VCS[V]
Figure 17. Current Consumption in Disable Mode,
VTRIG/DIS = 5 V
Figure 18. CS Input Current
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0,6
0,8
1
2,2
2,0
1,8
1,6
1,4
1,2
1,0
0,8
0,6
0,4
0,2
0,0
−1,0 −0,8
−40
−50
VTH_CS_ON[mV]
ICC[mA]
NCP4306
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
−60
−70
−80
−90
−100
−110
−0,6 −0,4 −0,2 0,0
VCS[V]
0,2
0,4
0,6
−120
0,8 1,0
−40
Figure 19. Supply Current vs. CS Voltage
0
20
40
TJ[°C]
60
80
100
120
100
120
100
120
Figure 20. CS Turn−on Threshold
0,60
1,0
VTH_CS_RESET[V]
0,5
VTH_CS_OFF[mV]
−20
0,0
−0,5
−1,0
0,55
0,50
0,45
−1,5
−2,0
−40
−20
0
20
40
60
80
100
0,40
−40
120
−20
0
20
TJ[°C]
tPD_ON[ns]
ICS_LEAKAGE[nA]
400
300
200
100
0
20
40
80
Figure 22. CS Reset Threshold
500
−20
60
TJ[°C]
Figure 21. CS turn−off Threshold
0
−40
40
60
80
100
120
60
55
50
45
40
35
30
25
20
15
10
−40
TJ[°C]
−20
0
20
40
60
80
TJ[°C]
Figure 23. CS Input Leakage VCS = 200 V
Figure 24. Propagation Delay from CS
to DRV Output On
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24
22
20
18
16
14
12
10
8
6
4
−40
2,2
2,0
VTRIG_TH[V]
tPD_OFF[ns]
NCP4306
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
1,8
1,6
1,4
1,2
−20
0
20
40
60
80
100
1,0
120
0
5
10
15
TJ[°C]
20
30
25
35
VCC[V]
Figure 25. Propagation Delay from CS
to DRV Output Off
Figure 26. Trigger Pin Threshold
15
2,3
14
2,2
ITRIG/DIS[μA]
VTRIG_TH[V]
13
2,1
2,0
1,9
12
11
10
9
1,8
1,7
−40
8
−20
0
20
40
60
80
100
7
−40
120
−20
0
20
TJ[°C]
Figure 27. Trigger Pin Threshold
60
80
100
120
Figure 28. Trigger Pin Pull Down Current
120
17
115
15
110
13
11
tDIS_TIM[μs]
tPD_TRIG[ns]
40
TJ[°C]
9
7
5
105
100
95
90
85
3
−40
−20
0
20
40
60
80
100
80
−40
120
TJ[°C]
−20
0
20
40
60
80
100
TJ[°C]
Figure 29. Propagation Delay from TRIG
to DRV Output Off
Figure 30. Delay to Disable Mode,
VTRIG/DIS = 5 V
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120
1,10
1,08
1,06
1,04
1,02
1,00
0,98
0,96
0,94
0,92
0,90
−40
tMIN_TON[μs]
tMIN_TON[μs]
NCP4306
−20
0
20
40
60
80
100
120
5,5
5,4
5,3
5,2
5,1
5,0
4,9
4,8
4,7
4,6
4,5
−40
−20
0
20
TJ[°C]
0
20
40
60
80
100
120
tMIN_TOFF[μs]
5,4
5,3
5,2
5,1
5,0
4,9
4,8
4,7
4,6
4,5
4,4
−40
−20
0
20
TJ[°C]
100
120
40
60
80
100
120
TJ[°C]
Figure 33. Minimum on Time RMIN_TOFF = 10 kW
Figure 34. Minimum on Time RMIN_TOFF = 50 kW
−18,0
0,0
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
−10,0
−15,0
−18,5
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
TJ = 55 °C
−19,0
ILLD[μA]
−5,0
ILLD[μA]
80
Figure 32. Minimum on Time RMIN_TON = 50 kW
tMIN_TOFF[μs]
−20
60
TJ[°C]
Figure 31. Minimum on Time RMIN_TON = 10 kW
1,10
1,08
1,06
1,04
1,02
1,00
0,98
0,96
0,94
0,92
0,90
−40
40
−19,5
−20,0
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
−20,5
−21,0
−20,0
−21,5
−25,0
0
5
10
15
20
VCC[V]
25
30
35
−22,0
0
5
10
15
20
25
VCC[V]
Figure 35. LLD Current, VLLD = 3.0 V
Figure 36. LLD current, VLLD = 2.5 V
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30
35
−15.0
640
−16.0
620
−17.0
−18.0
−19.0
600
TJ = 125 °C
TJ = 105 °C
TJ = 85 °C
−20.0
TJ = 55 °C
TJ = 25 °C
TJ = 0 °C
TJ = −20 °C
TJ = −40 °C
−21.0
−22.0
−23.0
−24.0
−25.0
580
tLLD[μs]
ILLD[μA]
NCP4306
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
560
540
520
500
480
460
440
−40
4.0
−20
0
20
VLLD[V]
Figure 37. LLD Current
10,0
5,1
9,8
VDRV[V]
VDRV[V]
5,3
VCC = 12 V, CDRV = 0 nF
VCC = 12 V, CDRV = 1 nF
VCC = 12 V, CDRV = 10 nF
VCC = 35 V, CDRV = 0 nF
VCC = 35 V, CDRV = 1 nF
VCC = 35 V, CDRV = 10 nF
9,6
9,4
9,2
100
120
−20
0
20
40
TJ[°C]
60
80
100
VCC = 12 V, CDRV = 0 nF
VCC = 12 V, CDRV = 1 nF
VCC = 12 V, CDRV = 10 nF
VCC = 35 V, CDRV = 0 nF
VCC = 35 V, CDRV = 1 nF
VCC = 35 V, CDRV = 10 nF
4,9
4,7
4,5
4,3
−40
120
Figure 39. Driver Output Voltage, Ver. xAxxxxx
−20
0
20
40
TJ[°C]
60
80
100
120
Figure 40. Driver Output Voltage, Ver. xBxxxxx
38
4,6
33
4,4
RatioEXC[−]
tdV/dt[ns]
80
5,5
10,2
28
23
18
13
−40
60
Figure 38. LLD Time, VLLD = 1.82 V (or Internal Option)
10,4
9,0
−40
40
TJ[°C]
4,2
4,0
3,8
3,6
−20
0
20
40
60
80
100
3,4
−40
120
TJ[°C]
−20
0
20
40
60
80
100
TJ[°C]
Figure 41. dV/dt Detector Time Threshold,
Ver. xxDxxxx
Figure 42. Exception Timer Ratio to tMIN_TON,
Ver. xxHxxxx
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120
NCP4306
GENERAL DESCRIPTION
The NCP4306 is designed to operate either as a standalone
IC or as a companion IC to a primary side controller to help
achieve efficient synchronous rectification in switch mode
power supplies. This controller features a high current gate
driver along with high−speed logic circuitry to provide
appropriately timed drive signals to a synchronous
rectification MOSFET. With its novel architecture, the
NCP4306 has enough versatility to keep the synchronous
rectification system efficient under any operating mode.
The NCP4306 works from an available voltage with range
from 4.0 / 3.5 V to 35 V (typical). The wide VCC range
allows direct connection to the SMPS output voltage of most
adapters such as notebooks, cell phone chargers and LCD
TV adapters.
Precise turn−off threshold of the current sense comparator
together with an accurate offset current source allows the
user to adjust for any required turn−off current threshold of
the SR MOSFET switch using a single resistor. Compared
to other SR controllers that provide turn−off thresholds in
the range of −10 mV to −5 mV, the NCP4306 offers a
turn−off threshold of 0 mV. When using a low RDS_ON SR
(1 mΩ) MOSFET our competition, with a −10 mV turn off,
will turn off with 10 A still flowing through the SR FET,
while our 0 mV turn off turns off the FET at 0 A;
significantly reducing the turn−off current threshold and
improving efficiency. Many of the competitor parts
maintain a drain source voltage across the MOSFET causing
the SR MOSFET to operate in the linear region to reduce
turn−off time. Thanks to the 6 A sink current of the
NCP4306 significantly reduces turn off time allowing for a
minimal drain source voltage to be utilized and efficiency
maximized.
To overcome false triggering issues after turn−on and
turn−off events, the NCP4306 provides adjustable minimum
on−time and off−time blanking periods. Blanking times can
be set internally during production or adjusted
independently of IC VCC using external resistors connected
to GND (internal or external option depends on IC variant).
If needed, externally set blanking periods can be modulated
using additional components.
An extremely fast turn−off comparator, implemented on
the current sense pin, allows for NCP4306 implementation
in CCM applications without any additional components or
external triggering.
An ultrafast trigger input offers the possibility to further
increase efficiency of synchronous rectification systems
operated in CCM mode (for example, CCM flyback or
forward). The time delay from trigger input to driver turn off
event is tPD_TRIG. Additionally, the trigger input can be used
to disable the IC and activate a low consumption standby
mode. This feature can be used to decrease standby
consumption of an SMPS. If the trigger input is not wanted
than the trigger pin can be tied to GND.
An output driver features capability to keep SR transistor
turned−off even when there is no supply voltage for the
NCP4306. SR transistor drain voltage goes up and down
during SMPS operation and this is transferred through drain
gate capacitance to gate and may open transistor. The
NCP4306 keeps DRV pin pulled low even without any
supply voltage and thanks to this the risk of turned−on SR
transistor before enough VCC is applied to the NCP4306 is
eliminated.
Finally, the NCP4306 features a Light Load Detection
function that can be set internally or externally at LLD pin
by resistor connected to ground. This function detects light
load or no load conditions and during them between
conduction phases it decreases current consumption. This
helps to improve SMPS efficiency. If LLD function is not
needed pin can be left open.
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NCP4306
SUPPLY SECTION
Supply voltage should be connected to VCC pin.
Minimum voltage for proper operation is 4.0 / 3.5 V
typically and maximum level is 35 V. Decoupling capacitor
between VCC and GND pin is needed for proper operation
and its recommended value is 1 μF. If IC is supplied from
SMPS output voltage, few ohm resistor is recommended
between SMPS output voltage and VCC pin. Resistor task
is to divide decoupling cap from output to avoid closing HF
currents through NCP4306 decoupling cap, because these
currents may causes drops at GND connection that affects
SR transistor sensing and incorrect SR transistor turn−off.
SR transistor is usually used in low side configuration
(placed in return path), but it may be also used in high side
configuration (placed in positive line). It is not possible to
use SMPS VOUT for SR supply in high side configuration so
it is needed to provide supply differently. One possibility is
to use auxiliary winding as shown in Figure 43. Voltage from
auxiliary winding is rectified, filtered and use as supply
voltage.
C5
C4
R6
C1
R2
C2
R7
R8
R9
D4
TR1
NCP4306
D3
M2
VCC
FLYBACK
CONTROL
CIRCUITRY
FB
D1
C3
DRV
CS
D2
+VOUT
M1
OK1
R1
D5
C1
R3
GND
Figure 43. High Side Configuration Supplied from Auxiliary Winding
If auxiliary winding is not acceptable, transformer
forward voltage can be used as supply source (Figure 44).
Forward voltage is regulated by simple voltage regulator to
fit NCP4306 VCC restriction. Penalty for this solution is
slightly lower efficiency.
C4
R6
C5
R7
R8
R9
D4
VBULK
R2
C1
D5
C2
NCP4306
M2
D3
FLYBACK
CONTROLL
FB
CS
C3
DRV
+VOUT
D1
D2
C6
M1
OK1
R1
D6
R3
Figure 44. High Side Configuration Supplied from Transformer Forward Voltage
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17
GND
NCP4306
Auxiliary winding or forward voltage can be used as
supply source also for low side configuration if VOUT is not
high enough (Figure 45). Do not focus just on SR controller
UVLO, but also on SR transistor characteristics. Some
transistors may be not turned−on enough even at 5 V so in
these case SR controller supply voltage should be increased.
+VOUT < 5V
VBULK
R1
C1
R3
D4
C2
C8
R6
R7
D3
C7
VCC
ZCD
C4
R4
D4
PRIMARY
SIDE
FLYBACK
CONTROLLER
C3
DRV
M2
M1
R8
D5
GND
C9
COMP CS
R2
R5
NCP4306
C5
C6
Figure 45. Low Side Configuration Supplied from Transformer Forward Voltage for Low VOUT SMPS
Current Sense Input
Because of parasitic impedances, significant ringing can
occur in the application. To overcome false sudden turn−off
due to mentioned ringing, the minimum conduction time of
the SR MOSFET is activated. Minimum conduction time
can be adjusted using the RMIN_TON resistor or can be
chosen from internal fixed values.
Figure 46 shows the internal connection of the CS
circuitry on the current sense input. When the voltage on the
secondary winding of the SMPS reverses, the body diode of
M1 starts to conduct current and the voltage of M1’s drain
drops approximately to −1 V. Once the voltage on the CS pin
is lower than VTH_CS_ON threshold, M1 is turned−on.
+ VOUT
+
SR MOSFET
M1
VTH_CS_ON
CS_ON
CS_OFF
To Internal logic
CS_RESET
VTH_CS_RESET
dV / dt
Detector
Figure 46. Current Sensing Circuitry Functionality
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High dV / dt
NCP4306
The SR MOSFET is turned−off as soon as the voltage on
the CS pin is higher than VTH_CS_OFF (typically −0.5 mV).
For the same ringing reason, a minimum off−time timer is
asserted once the VCS goes above VTH_CS_RESET. The
minimum off−time can be externally adjusted using
RMIN_TOFF resistor or can be chosen from internally fixed
values (depends on version). The minimum off−time
generator can be re−triggered by MIN_TOFF reset
comparator if some spurious ringing occurs on the CS input
after SR MOSFET turn−off event. This feature significantly
simplifies SR system implementation in flyback converters.
In an LLC converter the SR MOSFET M1 channel
conducts while secondary side current is decreasing (refer to
Figure 47). Therefore the turn−off current depends on
MOSFET RDSON. The −0.5 mV threshold provides an
optimum switching period usage while keeping enough time
margin for the gate turn−off. To ensure proper switching, the
min_tOFF timer is reset, when the VDS of the MOSFET rings
and falls down past the VTH_CS_RESET. The minimum
off−time needs to expire before another drive pulse can be
initiated. Minimum off−time timer is started again when
VDS rises above VTH_CS_RESET.
VDS = VCS
ISEC
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VDRV
Turn on delay
Min ON−time
Turn off delay
Min tOFF timer was
stopped here because of
VCS<VTH_CS_RESET
tMIN_TON
Min OFF−time
tMIN_TOFF
Figure 47. CS Input Comparators Thresholds and Blanking Periods Timing in LLC
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19
NCP4306
VDS = VCS
ISEC
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VDRV
Turn off delay
Turn on delay
Min ON−time
Min tOFF timer was
stopped here because of
VCS<VTH_CS_RESET
tMIN_TON
Min OFF−time
tMIN_TOFF
Figure 48. CS Input Comparators Thresholds and Blanking Periods Timing in Flyback
SR Transistor Selection
which was selected with minimum available RDS(ON)
resistance requirement only.
Sensing VDS drop across the SR transistor, which is
ideally product of transistor’s RDS(ON) and secondary side
current, is affected by voltage drop at parasitic inductance of
package (bonding, leads, …) and PCB layout−(refer to
Figure 49). The current that flows through the SR MOSFET
experiences a high Δi(t) / Δt that induces an error voltage on
the SR MOSFET bonds and leads due to their parasitic
inductance. This error voltage is proportional to the
derivative of the SR MOSFET current; and shifts the CS
input voltage to zero when significant current still flows
through the MOSFET channel. As a result, the SR MOSFET
is turned−off prematurely and the efficiency of the SMPS is
not optimized – refer to Figure 50 for a better understanding.
An SMPS designer should consider all important SR
MOSFET parameters for its optimum selection in given
application and not stick only to the lowest RDS(ON)
requirement. The lower RDS(ON) device is selected the more
significant role the lead parasitic inductances play in turn−of
threshold sensing i.e. the more premature turn−off will
happen (refer to section below for parasitic inductance
impact to VDS sensing). The lower RDS(ON) switch also
usually features higher input capacitance that increases
driving losses. The higher output capacitance and higher
reverse recovery charge of body diode then results in higher
drain−to source voltage peaks in CCM applications. Thus
the higher RDS(ON) MOSFET can usually provide better or
at least same efficiency result when compare to a switch
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20
NCP4306
VDS
a)
VL_LAYOUT
LLAYOUT
VL_DRAIN
LDRAIN
ISEC
VRDS_ON
VL_SOURCE
VL_LAYOUT
LLAYOUT
LSOURCE
RDS_ON
MOSFET equivalent circuit
To VCC
1 μF
100 nF
Decoupling
Capacitors
NCP4306
VDS
ISEC
b)
VL_LAYOUT
LLAYOUT
VL_DRAIN
LDRAIN
VRDS_ON
RDS_ON
VL_SOURCE
LSOURCE
VL_LAYOUT
LLAYOUT
MOSFET equivalent circuit
To VCC
1 μF
100 nF
Decoupling
Capacitors
NCP4306
Figure 49. SR System Connection Including MOSFET and Layout Parasitic Inductances in a) LLC and b) Flyback
Application
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21
NCP4306
a)
b)
Figure 50. Waveforms from SR System Implemented in a) LLC and b) Flyback Application and Using MOSFET in
TO220 Package With Long Leads – SR MOSFET channel Conduction Time is Reduced
current Δi / Δt and high operating frequency is to use
lead−less SR MOSFET i.e. SR MOSFET in SMT package.
The parasitic inductance of a SMT package is negligible
causing insignificant CS turn−off threshold shift and thus
minimum impact to efficiency (refer to Figure 51).
Note that the efficiency impact caused by the error voltage
due to the parasitic inductance increases with lower
MOSFETs RDS_ON and / or higher operating frequency.
It is thus beneficial to minimize SR MOSFET package
leads length in order to maximize application efficiency. The
optimum solution for applications with high secondary
a)
b)
Figure 51. Waveforms from SR System Implemented in a) LLC b) Flyback Application and Using MOSFET in SMT
Package with Minimized Parasitic Inductance – SR MOSFET Channel Conduction Time is Optimized
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NCP4306
resistance. In reality there will be small parasitic impedance
on the CS path due to the bonding wires, leads and soldering.
To assure the best efficiency results, a Kelvin connection of
the SR controller to the power circuitry should be
implemented. The GND pin should be connected to the SR
MOSFET source soldering point and current sense pin
should be connected to the SR MOSFET drain soldering
point − refer to Figure 49. Using a Kelvin connection will
avoid any impact of PCB layout parasitic elements on the SR
controller functionality; SR MOSFET parasitic elements
will still play a role in attaining an error voltage. Figure 52
and Figure 53 show examples of SR system layouts using
MOSFETs in D2PAK and SO8FL packages.
It can be deduced from the above paragraphs on the
induced error voltage and parameter tables that turn−off
threshold precision is quite critical. If we consider a SR
MOSFET with RDS_ON of 1 mΩ, the 1 mV error voltage on
the CS pin results in a 1 A turn−off current threshold
difference; thus the PCB layout is very critical when
implementing the SR system. Note that the CS turn−off
comparator is referred to the GND pin. Any parasitic
impedance (resistive or inductive – even on the magnitude
of mΩ and nH values) can cause a high error voltage that is
then evaluated by the CS comparator. Ideally the CS
turn–off comparator should detect voltage that is caused by
secondary current directly on the SR MOSFET channel
Figure 52. Recommended Layout When Using SR
MOSFET in D2PAK Package
Figure 53. Recommended Layout When Using SR
MOSFET in SMT Package SO8 FL
Trigger / Disable input
advantageous than leaving end of conduction phase on body
diode. Reason is body diode has usually longer recovery
time and resulting overlap time (simultaneously conduction
primary and secondary side switches) is longer. There are
several possibilities for transferring the trigger signal from
the primary to the secondary side – refer to Figure 68 and
Figure 69.
The trigger signal is blanked for tTRIG_BLANK after the
DRV turn−on process has begun. The blanking technique is
used to increase trigger input noise immunity against the
parasitic ringing that is present during the turn on process
due to the SMPS layout. The trigger input is supersedes the
CS input except trigger blanking period. TRIG / DIS signal
turns the SR MOSFET off or prohibits its turn−on when the
TRIG / DIS pin is pulled above VTRIG_TH.
The SR controller enters disable mode when the trigger
pin is pulled−up for more than tDIS_TIM. In disable mode the
IC consumption is significantly reduced. To recover from
disable mode and enter normal operation, the TRIG / DIS
pin has to be pulled low at least for tDIS_END.
The NCP4306 features an ultrafast trigger input that
exhibits a maximum of tPD_TRIG delay from its activation to
the start of SR MOSFET turn−off of process. This input can
be used in applications operated in deep Continues
Conduction Mode (CCM) to further increase efficiency and
/ or to activate disable mode of the SR driver in which the
consumption of the NCP4306 is reduced to maximum of
ICC_DIS.
NCP4306 is capable to turn−off the SR MOSFET reliably
in CCM applications just based on CS pin information only,
without using the trigger input. However, natural delay of
the ZCD comparator and DRV turn−off delay increase
overlap between primary and secondary MOSFETs
switching (also known as cross conduction). If one wants to
achieve absolutely maximum efficiency with deep CCM
applications, then the trigger signal coming from the
primary side should be applied to the trigger pin. It is good
to set trigger pulse in way there is just short overlap between
primary and secondary switches. Short overlap is usually
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NCP4306
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
VDRV
t1 t2 t3
t4
t5
t6
t7
t8
t
t9
Figure 54. Trigger Input Functionality Waveforms Using the Trigger to Turn−off and Block the DRV Signal
/ DIS pin almost immediately turns off the drive to the SR
MOSFET, turning off the MOSFET. The DRV is not
turned−on in other case (t6) because the trigger pin is high
in the time when CS pin signal crosses turn−on threshold.
This figure clearly shows that the DRV can be asserted only
on falling edge of the CS pin signal in case the trigger input
is at low level (t2).
Figure 54 shows basic TRIG / DIS input functionality. At
t1 the TRIG / DIS pin is pulled low to enter into normal
operation. At t2 the CS pin is dropped below the
VTH_CS_ON, signaling to the NCP4306 to start to turn the SR
MOSFET on. At t3 the NCP4306 begins to drive the
MOSFET. At t4, the SR MOSFET is conducting and the
TRIG / DIS pin is pulled high. This high signal on the TRIG
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
TRIG / DIS blank
tTRIGBLANK
Min ON−time
VDRV
t1 t2
t
t3
Figure 55. Trigger Input Functionality Waveforms – Trigger Blanking
shortly after time t2. Due to the Trigger blanking clock
(tTRIG_BLANK) the Trigger’s high signal does not affect the
DRV signal until the tTRIG_BLANK timer has expired. At
time t3 the TRIG / DIS signal is reevaluated and the DRV
In Figure 55 above, at time t1 the CS pin falls below the
VTH_CS_ON while the Trigger is low setting in motion the
DRV signal that appears at t2. At time t2 the DRV signal and
Trigger blanking clock begin. TRIG / DIS signal goes high
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NCP4306
signal is turned off. The TRIG / DIS input is blanked for
tTRIG_BLANK after DRV set signal to avoid undesirable
behavior during SR MOSFET turn−on event. The blanking
time in combination with high threshold voltage
(VTRIG_TH) prevent triggering on ringing and spikes that are
present on the TRIG / DIS input pin during the SR MOSFET
turn−on process. Controller’s response to the narrow pulse
on the TRIG / DIS pin is depicted in Figure 55 – this short
trigger pulse enables to turn the DRV on for tTRIG_BLANK.
Note that this case is valid only if device not entered disable
mode before.
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
VTRIG_BLANK
TRIG / DIS blank
MIN ON−TIME
VDRV
t0
t1 t2 t3
t4
t
t5 t6
Figure 56. Trigger Input Functionality Waveforms – Trigger Blanking Acts Like a Filter
Figure 56 above shows almost the same situation as in
Figure 55 with one main exception; the TRIG / DIS signal
was not high after trigger blanking timer expired so the DRV
signal remains high. The advantage of the trigger blanking
time during DRV turn−on is evident from Figure 56 since it
acts like a filter on the TRIG / DIS pin. Rising edge of the
DRV signal may cause spikes on the trigger input. If it wasn’t
for the TRIG / DIS blanking these spikes, in combination
with ultra−fast performance of the trigger logic, could turn
the SR MOSFET off in an inappropriate time.
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
Min ON−time
VDRV
t0
t1
t2 t3
t4
t5
t6
t7
t8
t
Figure 57. Trigger Input Functionality Waveforms – Trigger over Ride, CS Turn Off and Min On−time
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NCP4306
Figure 57 depicts all possible driver turn−off events in
details when correct VCC is applied. Controller driver is
disabled based on TRIG / DIS input signal in time t2; the
TRIG / DIS input overrides the minimum on−time period.
Driver is turned−off according to the CS (VDS) signal (t5
marker) and when minimum on−time period elapsed
already. TRIG / DIS signal needs to be low during this event.
If the CS (VDS) voltage reaches VTH_CS_OFF threshold
before minimum on−time period ends (t7) and the TRIG /
DIS pin is low the DRV is turned−off on the falling edge of
the minimum on−time period (t8 time marker in Figure 57).
This demonstrates the fact that the Trigger over rides the
minimum on−time. Minimum on−time has higher priority
than the CS signal.
In Figure 58 the TRIG / DIS input is low the whole time
and the DRV pulses are purely a function of the CS signal
and the minimum on−time. The first DRV pulse terminated
based on the CS signal and another two DRV pulses are
prolonged till the minimum on−time period end despite the
CS signal crosses the VTH_CS_OFF threshold earlier.
If a minimum on−time is too long the situation that occurs
after time marker t6 Figure 58 can occur, is not correct and
should be avoided. The minimum tON period should be
selected shorter to overcome situation that the SR MOSFET
is turned−on for too long time. The secondary current then
changes direction and energy flows back to the transformer
that result in reduced application efficiency and also in
excessive ringing on the primary and secondary MOSFETs.
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
Min ON−time
VDRV
t0
t1
t2
t3
t4 t5
t6
t7 t8 t9
t
Figure 58. Minimum On−Time Priority
pin voltage goes above VTH_CS_RESET threshold. Next cycle
starts in time t6. The TRIG / DIS goes low and enables the
DRV before VDS drops below VTH_CS_ON threshold voltage
thus the DRV turns−on in time t6. The TRIG / DIS signal
rises up to HIGH level at time t7, consequently DRV
turns−off and IC waits for high CS voltage to start minimum
off−time execution.
Figure 59 shows IC behavior in case the trigger signal
features two pulses during one cycle of the VDS (CS) signal.
The TRIG / DIS goes low enables the DRV just before time
t1 and DRV turns−on because the VDS voltage drops under
VTH_CS_ON threshold voltage. The TRIG / DIS signal
disables driver at time t2. The TRIG / DIS drops down to
LOW level in time t3, but IC waits for complete minimum
off−time. Minimum off−time execution is blocked until CS
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NCP4306
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
Min ON−time
Min OFF−time
VDRV
t0
t1
t2
t3 t4
t5
t6
t7 t8 t9
t10
t
Figure 59. TRIG / DIS Input Functionality Waveforms – Two Pulses at One Cycle
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
tDIS_TIM
Min ON−time
VDRV
Power
consumption
t0
t1
t2 t3
t4
t
Figure 60. Trigger Input Functionality Waveforms – Disable Mode Activation
transition to disable mode. Figure 61 shows disable mode
transition 2nd case – i.e. when trigger rising edge comes
during the trigger blank period. Figure 62 shows entering
into disable mode and back to normal sequences.
In Figure 60 above, at t2 the CS pin rises to VTH_CS_OFF
and the SR MOSFET is turned−off. At t3 the TRIG / DIS
signal is held high for more than tDIS_TIM. NCP4306 enters
disable mode after tDIS_TIM. Driver output is disabled in
disable mode. The DRV stays low (disabled) during
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NCP4306
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
tDIS_TIM
Min ON−time
tTRIGBLANK
VDRV
Power
consumption
t0
t1 t2
t3
t
Figure 61. Trigger Input Functionality Waveforms – Disable Mode Clock Initiation
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
tDIS_TIM
tDIS_REC
VDRV
Min OFF−time
Power
consumption
Disable Mode
t0
t1
t2
t3 t4
Figure 62. Trigger Input Functionality Waveforms – Disable and Normal Modes
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t
NCP4306
Figure 63 and Figure 64 shows exit from disable mode in
detail. NCP4306 requires time up to tDIS_REC to recover all
internal circuitry to normal operation mode when
recovering from disable mode. The driver is then enabled
after complete tMIN_TOFF period when CS (VDS) voltage is
over VTH_CS_RESET threshold. Driver turns−on in the next
cycle on CS (VDS) falling edge signal only (t5 − Figure 63).
The DRV stays low during recovery time period. TRIG / DIS
input has to be low at least for tDIS_END time to end disable
mode and start with recovery. Trigger can go back high after
tDIS_END without recovery interruption.
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
Min ON−time
Power
consumption
Disable Mode
VDRV
Rec
time
Waits for
complete Normal Mode
tMIN_TOFF
t
t0
t1
t2
t3
t4
t5
t6
t7
t8
Note: Rec Time = Recovery Time
Figure 63. Trigger Input Functionality Waveforms – Exit from Disable Mode before the Falling
Edge of the CS Signal
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NCP4306
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
tDIS_END
Min OFF−time
VDRV
Power
consumption
Waits for
complete
time
tMIN_TOFF
Rec
Disable Mode
t0
t1 t2
t4
t3
Normal Mode
t
t5
Note: Rec Time = Recovery Time
Figure 64. Trigger Input Functionality Waveforms
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
Min OFF−time
tMIN_TOFF
tDIS_REC
VDRV
Power
consumption
Disable Mode
Waits for
complete
tMIN_TOFF
Recovery
t0
t1
t2 t3
t4
Normal Mode
t5
t6
t7 t8
Figure 65. Trigger Input Functionality Waveforms
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30
tt
NCP4306
normal mode doesn’t start. VDS voltage goes high again at
time t4 and this event starts new minimum off−time timer
execution. Next VDS falling edge below VTH_CS_ON level
activates driver.
Figure 65 shows detail IC behavior after disable mode is
ended. The trigger pin voltage goes low at t1 and after
tDIS_REC IC leaves disable mode (t2). Time interval between
t2 and t3 is too short for complete minimum off−time so
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG / DIS
tMIN_TOFF
Min OFF−time
Disable Mode
VDRV
Power
consumption
t0
tDIS_REC
Waits for
complete
tMIN_TOFF
Recovery
t1
t4
t2 t3
Normal Mode
t5
t6
t
t7 t8
Figure 66. Trigger Input Functionality Waveforms
the IC waits to another time when VDS voltage is positive
and then is again started the minimum off−time timer. The
IC returns into normal mode after whole minimum off−time
elapses.
Different situation of leaving from disable mode is shown
at Figure 66. Minimum off−time execution starts at time t2,
but before time elapses VDS voltage falls to negative
voltage. This interrupts minimum off−time execution and
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
Min OFF−time
Not Complete
tMIN_TOFF −
IC is not
activated
tMIN_TOFF
Complete
tMIN_TOFF −
activates IC
tMIN_TOFF
tMIN_TOFF −
is stopped due to
VDS drops below
VTH_CS_RESET
tMIN_TON
Min ON−time
VDRV
VCC
VCCON
t
t1 t3
t2 t4
t5 t7
t6
t8
t9 t10
t11 t13 t14
t12
Figure 67. NCP4306 Operation after Start−Up Event
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NCP4306
Start−up event waveforms are shown at Figure 67. A
start−up event is very similar to an exit from disable mode
event. The IC waits for a complete minimum off−time event
(CS pin voltage is higher than VTH_CS_RESET) until drive
pulses can continue. Figure 67 shows how the minimum
off−time timer is reset when CS voltage is oscillating
through VTH_CS_RESET level. The NCP4306 starts
operation at time t1 (time t1 can be seen as a wake−up event
from the disable mode through TRIG / DIS or LLD pin).
Internal logic waits for one complete minimum off−time
period to expire before the NCP4306 can activate the driver
after a start−up or wake−up event. The minimum off−time
timer starts to run at time t1, because VCS is higher than
VTH_CS_RESET. The timer is then reset, before its set
minimum off−time period expires, at time t2 thanks to CS
voltage lower than VTH_CS_RESET threshold. The
aforementioned reset situation can be seen again at time t3,
t4, t5 and t6. A complete minimum off−time period elapses
between times t7 and t8 allowing the IC to activate a driver
output after time t8.
Optional primary triggering techniques for CCM flyback
application are shown in Figure 68 and Figure 69. NCP4306
can operate properly without triggering in CCM, but use of
triggering can reduce the commutation losses and the SR
MOSFET drain voltage spike, which results in improved
efficiency in CCM.
VBULK
C1
R3
+ VOUT
TR1
R4
C3
C6
D2
M2
VCC
FLYBACK
CONTROL
CIRCUITRY
FB
D3
DRV
D1
C2
M1
CS
GND
C4
R5
R6
R7
R1
D4
NCP4306
R2
C5
OK1
D5
R7
Figure 68. Optional Primary Triggering in Deep CCM Application Using Auxiliary Winding
The application shown in Figure 68 is simplest and the
most cost effective solution for primary SR triggering. This
method uses auxiliary winding made of triple insulated wire
placed close to the primary winding section. This auxiliary
winding provides information about primary turn−on event
to the SR controller before the secondary winding reverses.
This is possible thanks to the leakage between primary and
secondary windings that creates natural delay in energy
transfer. This technique provides approximately 0.5%
efficiency improvement when the application is operated in
deep CCM and a transformer that has a leakage of 1% of
primary inductance is used.
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NCP4306
VBULK
C1
TR1
R2
+ VOUT
R4
C4
D2
VCC
FLYBACK
CONTROL
CIRCUITRY
FB
CS
C7
M2
D3
C3
DRV
GND
C6
D1
M1
R5
R6
R7
R1
R3
D4
NCP4306
OK1
D5
TR2
R8
C5
Figure 69. Optional Primary Triggering in Deep CCM Application Using Trigger Transformer
to assure LPRIMARY = LSECONDARY > 10 μH. Proper safety
insulation between primary and secondary sides can be
easily assured by using triple insulated wire for one or,
better, both windings.
This
primary
triggering
technique
provides
approximately 0.5% efficiency improvement when the
application is operated in deep CCM and transformer with
leakage of 1% of primary inductance is used.
It is also possible to use capacitive coupling (use
additional capacitor with safety insulation) between the
primary and secondary to transmit the trigger signal. We do
not recommend this technique as the parasitic capacitive
currents between primary and secondary may affect the
trigger signal and thus overall system functionality.
Application from Figure 69 uses an ultra−small trigger
transformer to transfer primary turn−on information directly
from the primary controller driver pin to the SR controller
trigger input. Because the trigger input is rising edge
sensitive, it is not necessary to transmit the entire primary
driver pulse to the secondary. The coupling capacitor C5 is
used to allow the trigger transformer’s core to reset and also
to prepare a needle pulse (a pulse with width shorter than
100 ns) to be transmitted to the NCP4306 TRIG / DIS input.
The advantage of needle trigger pulse usage is that the
required volt−second product of the pulse transformer is
very low and that allows the designer to use very small and
cheap magnetic. The trigger transformer can even be
prepared on a small toroidal ferrite core with outer diameter
of 4 mm and four turns for primary and secondary windings
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NCP4306
Minimum tON and tOFF adjustment
Fixed versions are defined internally and can’t be
modified later or changed during operation.
The adjustment of minimum tON and tOFF periods are
done based on an internal timing capacitance and external
resistors connected to the GND pin – refer to Figure 70 for
a better understanding.
The NCP4306 offers fixed or an adjustable minimum
on−time and off−time blanking periods (depends on IC
version) that ease the implementation of a synchronous
rectification system in any SMPS topology. These timers
avoid false triggering on the CS input after the MOSFET is
turned on or off.
VDD
To Internal Logic
VREF
IR_MIN_TON
tMIN_TON
MIN_TON
Discharge
Switch
Ct
IR_MIN_TON
RMIN_TON NCP4306
GND
Figure 70. Internal Connection of the MIN_TON Generator (the MIN_TOFF Works in the Same Way)
Current through the MIN_TON adjust resistor can be
calculated as:
I R_MIN_TON +
V ref
R MIN_TON
If the internal current mirror creates the same current
through RMIN_TON as used the internal timing capacitor (Ct)
charging, then the minimum on−time duration can be
calculated using this equation.
(eq. 1)
t MIN_ON + V t
V ref
I R_MIN_TON
V ref
+ Ct
V ref
+ Ct
R MIN_TON
(eq. 2)
R MIN_TON
the accuracy of equations 7 and 8 when MIN_TON or
MIN_TOFF times are selected near to their minimum
possible values. Please refer to Figure 71 and Figure 72 for
measured minimum on and off time charts.
5,0
5,0
4,5
4,5
4,0
4,0
3,5
3,5
tMIN_TOFF [μs]
tMIN_TON [μs]
The internal capacitor size would be too large if
IR_MIN_TON was used. The internal current mirror uses a
proportional current, given by the internal current mirror
ratio. Note that the internal timing comparator delay affects
3,0
2,5
2,0
1,5
1,0
2,5
2,0
1,5
1,0
0,5
0,0
3,0
0,5
0
5
10
15
20 25 30
RMIN_TON [kΩ]
35
40
45
0,0
50
Figure 71. MIN_TON Adjust Characteristic
0
5
10
15
20 25 30 35
RMIN_TOFF [kΩ]
40
45
Figure 72. MIN_TOFF Adjust Characteristic
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50
NCP4306
The absolute minimum tON duration is internally clamped
to 55 ns and minimum tOFF duration to 70 ns in order to
prevent any potential issues with the minimum tON and / or
tOFF input being shorted to GND.
The NCP4306 features dedicated anti−ringing protection
system that is implemented with a minimum tOFF blank
generator. The minimum off−time one−shoot generator is
restarted in the case when the CS pin voltage crosses
VTH_CS_RESET threshold and MIN_TOFF period is active.
The total off−time blanking period is prolonged due to the
ringing in the application (refer to Figure 47).
Some applications may require adaptive minimum on and
off time blanking periods. It is possible to modulate blanking
periods by using an external NPN transistor – refer to Figure
73. The modulation signal can be derived based on the load
current, feedback regulator voltage or other application
parameter.
VDD
To Internal Logic
VREF
IR_MIN_TON
tMIN_TON
MIN_TON
RMIN_TON
MIN_TON modulation Input
RMIN_TON_2
Modulation
Current
Discharge
Switch
IR_MIN_TON
NCP4306
GND
Figure 73. Possible Connection for MIN_TON and MIN_TOFF Modulation
dV / dt Detection – Flyback feature
primary side switch is turned on and off again so SR
controller doesn’t turn on SR mosfet. Whole secondary side
current flows through body diode that makes power loss.
Figure 74 shows situation without dV / dt detection. Here
can be seen that without detection next conduction cycle
may be not taken through activated SR transistor. Reason is
not elapsed minimum off−time blanking interval.
The NCP4306 includes optional feature for flyback type
converters, which operates with shorter primary on−time
than ringing period after demagnetization phase during
medium / high loads. These applications are for example
USB−PD or Quick Charge adapters. Difficulty with this
situation is that minimum off−time doesn’t elapse before
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NCP4306
tMIN_TOFF has to be set to
longer time length of ringing
period
tMIN_TOFF
VDS = VCS
ISEC
Primary on−time is very short
(shorter than ringing period) for low VOUT
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VDRV
tMIN_TOFF timer is stopped here
because of VCS< VTH_CS_RESET
tMIN_TON
tMIN_TOFF
Min ON−time
Driver is not turned−on because
tMIN_TOFF doesn’t elapse
Turn−on delay Turn−off delay
Min OFF−time
tMIN_TOFF
Figure 74. Situation without dV / dt Detection Feature
Figure 75 shows how system with activated dV / dt
detection behaves. Min_toff blanking interval is also reset
during voltage drops at CS pin, but if high negative dV / dt
occurs at CS pin, min_toff interval is shorted and SR
controller is ready to detect CS voltage lower than
VCS_TH_ON and turn SR transistor on. Negative dV / dt at CS
pin after primary switch is turned off is high in compare to
slope that comes during ringing after demagnetization.
Thanks to this we can safely detect end of primary on−time
from ringing.
VDS = VCS
tMIN_TOFF has to be set
to longer time than length
of ringing period tMIN_TOFF
ISEC
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VDRV
Min ON−time
Turn−on delay
tMIN_TON
Min OFF−time
Turn−off delay
CS voltage drops below
VTH_CS_ON but at time
when min tOFF doesn’t
elapse and just low dV / dt
is detected so DRV is not
activated
tMIN_TOFF timer is stopped here
because of VCS< VTH_CS_RESET
tMIN_TOFF
Negative dV / dt
detector at CS pin
tMIN_TOFF
tMIN_TOFF timer doesn’t
elapse but high dV / dt is
detected so DRV is enable
Figure 75. Situation with Enabled dV / dt Detection
Exception Timer – LLC feature
part followed by distorted sine. Examples of current shape
is shown in Figure 76. This figure shows different current
shapes at different loading. Lower loading makes shape
more distorted from ideal sine.
Exception timer is special feature for LLC type SMPS. It
is mainly targeted to operation under light / medium load,
where secondary side SMPS current shape is not sine, but it
contains part of capacitive peak optionally with no current
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NCP4306
ISEC
t
Figure 76. Current Shapes in LLC Examples
above VTH_CS_OFF threshold. Turn−off process can be
masked by min_ton blanking interval, but in this case is
needed to set it at least to 1.5 μs that can make issue during
very light load where current flows just short time and long
min_ton may cause reverse current from output capacitors
back to transformer and may change soft switching
condition to hard switching at primary side.
3,00
2
2,50
0
2,00
-2
VDS[mV]
ISEC[A]
Problematic shapes may cause prematurely SR transistor
turn−off, because CS voltage may get to zero or to positive
voltage (due to low current or high dI / dt and parasitic
inductance). Sensed voltage drop can be seen in Figure 77.
This situation is valid for SR mosfet with RDSON = 1 mΩ and
with package (SMT) parasitic inductance LPACPAR =
0.5 nH. There can be seen that SR transistor should be turned
off in time between 0.4 to 1.5 μs, because CS voltage is
1,50
-4
1,00
-6
0,50
-8
-10
0,00
0
1
2
3
4
5
6
7
VTH_CS_OFF
Drop just at
RDSon
0
t[μs]
1
2
Drop at RDSon
and parasitic
inductance
3
4
5
6
7
t[μs]
Figure 77. Sensed Voltage Drop at SR Transistor in LLC during Light / Medium Load
To early SR transistor turn−off is not issue just from
efficiency point of view, but also from system stability point
of view. When load is decreased, feedback loop asks primary
side for lower power that changes secondary side current
shape and SR driver can be turned off shortly after min_toff
elapses. This causes lower efficiency transfer to secondary
side and output voltage starts to decrease. Feedback loop
asks for more power, secondary current shape changes and
SR driver starts to conduct whole period again that improves
energy transfer efficiency and output voltage starts to
increase. This has to be again regulated by feedback loop and
everything starts from begin and make SMPS oscillations
that can be accompany with audible noise.
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NCP4306
Regulation loop decreases transferred power because thanks to SR
Regulation loop increases transferred power because thanks to low
it is too much voltage at the output
SR conduction angle lot of power is lose at body diode
ISEC
VDS1
VDRV1
MIN_TON
VDS2
Bodydiode conducts
Bodydiode conducts
CS voltage goes above 0 V after
min_ton = SR is turned off
VDRV2
MIN_TON
Bodydiode conducts
Bodydiode conducts
Figure 78. LLC System Oscillation due to Short SR Transistor Conduction
Operation of new feature is shown in Figure 79. Current
shape makes drop at SR transistor with 1 mΩ and 0.5 nH
shown as VDS that is sensed at CS pin and on and off
comparators decide about SR operation based on this
voltage. Driver is turned on and exception timer is started
when VCS drops below VCS_TH_ON. During minimum
on−time blanking interval off comparator is not active. CS
pin voltage is above VTH_CS_OFF after minimum on−time
elapses so driver is turned−off and because exception timer
doesn’t elapse, min_ton blanking interval is started. During
this time on comparator output is blanked. Reason is to avoid
quick driver turning on and off that would just increase
consumption. When min_ton blanking interval elapses CS
voltage is again below VTH_CS_ON and exception timer is
not elapsed, driver can be turned on again simultaneously
with minimum on−time interval. Driver is turned off again
almost at the end of conduction phase, but this is correct turn
off. Min−ton blanking interval doesn’t start, because
exception timer elapsed before so SR controller waits for
VCS > VCS_TH_RESET to start minimum off time blanking
timer.
Exception timer length is given as multiple of minimum
on time interval. It should be not set to longer time than
t EXC t
3
1
f SWMAX
(eq. 3)
where fSWMAX is maximum LLC switching frequency.
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NCP4306
ISEC
∼2.7 A
∼0.25 A
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
Body diode drop
(not in scale)
Min ON−time
Min ON−time_2
Driver can be
turned on again up
to this point
Exception timer
DRV
Min OFF−time
Figure 79. Exception Timer Operation
Light Load Detection
informs about CS voltage lower than zero (body diode or SR
transistor conducting), LLD timer with set able nominal
time and possibility to reduce it to one half and finally D flip
flop with Disable signal output. Nominal time can be set by
resistor at LLD pin connected to ground or internally during
production. Recommended resistor values are shown in
Table 6. In case of very noisy system, capacitor in parallel
to LLD resistor may be used. Capacitor value impacts
start−up time, because capacitor has to be charged above
disable threshold by internal LLD current source.
Light load detection feature is feature which task is to
decrease SR controller consumption during time when SR
transistor switching is not needed. This is usually during no
load and light load condition when static SR controller
consumption starts to play role. Goal is to disable controller
during no switching time to eliminate static consumption
and turn−on SR transistor as soon as possible when
switching comes.
Internal simplified block diagram is shown in Figure 80.
Main parts of this system are comparator at CS input that
VCSLLD
CS
Set max
Set max
to tLLD / 2
to tLLD
LLD
Timer
RESET
ELAPSED
tim > 1/4 tLLD
Figure 80. LLD Internal Block Diagram
www.onsemi.com
39
S
R
Q
Disable
NCP4306
Table 6. PIN FUNCTION DESCRIPTION
LLD setting tLLD [ms]
IC disabled
70
130
280
540
1075
LLD disabled
RLLD [kW]
<12
27
43
68
91
120
>470*
*floating pin allowed, small cap for noise robustness improvement recommended
and IC starts to wake up (takes tLLD_DIS_REC, system wake
up is controlled same as exit from disable mode by TRIG /
DIS pin). End of conduction phase (CS voltage goes
positive) starts LLD timer. If next conduction phase comes
shortly after first (pulses in skip burst) so shortly than tLLD
/ 4 just LLD timer is reset. LLD timer length is set back to
tLLD only when new conduction phase comes after previous
in time between tLLD / 4 to tLLD / 2. This situation happens
when load is slowly increased and skip bursts come more
often.
Logic function is also described by bubble diagram in
Figure 81. LLD timer is running every time when CS pin
voltage is positive (body diode and or transistor not
conducting). If conduction doesn’t come sooner than LLD
timer elapses, DISABLE flag is set (IC is sent into low
consumption mode), LLD timer length is changed to tLLD /
2 (this adds some hysteresis in system and helps keeping
overall system stable) and timer is also reset. SR controller
waits for falling edge at CS pin (begin of new conduction
cycle). When CS goes negative, disable mode is deactivated
Start
LLD TIM is
RUNNING
LLD_CMP &
TIM CNT < 1/4 tLLD
Reset TIM
DISABLE = 0
LLD_CMP
DISABLE = 1
tTIM = 1/2 tLLD
Reset TIM
Reset TIM
DISABLE = 0
tTIM = tLLD
Figure 81. LLD Operation Bubble Diagram
Example of LLD operation with flyback convertor can be
seen in Figure 82. SMPS works under heavy load from point
0 to 1 where switching pulses comes regularly at high
frequency that resets LLD timer soon after begin of
counting. Load is significantly decreased to light load at
point 1 so primary controller turns to skip mode. LLD timer
elapses during skip so controller enters disable mode with
very low consumption and change LLD timer maximum to
tLLD / 2. Switching pulse in skip comes at time 3, this resets
LLD timer and starts IC wake−up. Controller is waked up
fully before point 4 and turns−on SR transistor. There is
again no switching from 4 to 6 and thanks to it, LLD timer
elapses at point 5 and controller enters disable mode again.
Disable mode is ended at time 6, because new cycle comes.
SR controller wakes−up and next pulse in skip burst is
conducted via SR transistor. Time between 7 and 8 is delay
between skip burst. Time is still less than tLLD / 4, LLD timer
interval is not changed. Pulse at time 8 is fully conducted via
SR transistor, because controller was not in disable mode
before pulse came. No switching period between 9 and 11 is
longer than tLLD / 2 that changes LLD timer setting to tLLD.
This is because shorter delay between skip burst means
higher load. Pulses are transferred via SR transistor at time
11 and 12, because disable mode was not activated. Load is
being decreased again between time 12 to 15 so at time 15
SR controller enters disable mode and LLD timer time is
reduced again to tLLD / 2. Second pulse in skip burst is again
transferred via turned on SR transistor. Disable mode is
activated after tLLD / 2 at time 18. Load is sharply changed
at time 19 that means LLD timer is reset each pulse and
timers time is kept at tLLD / 2. Load is removed at time 20 and
disable is activated at time 21. Suitable LLD timer setting for
flyback type of SMPS is 540 or 1075 μs (for special type
280 μs).
www.onsemi.com
40
NCP4306
VDS = VCS
t1
t2
t4
t2
t1
t2
DRV
DIS
ICC
0
t4
t1
t2
LLD tim
tLLD
1
2 3 4
5 6
7 8 9 101112
13 14
15
16 17 18 19
20
21 22
Figure 82. LLD Operation with Flyback SMPS
at time 8 respectively 9. Skip burst ends at time 12, LLD
timers elapse at time 13 and 14 (reached tLLD / 2) and SR
controllers enter disable mode. Controllers wake up at time
15 and 16 same as was in time 6 and 7. SMPS goes into skip
in time 21, but load is connected soon and SMPS starts to
operate under higher load from time 22. LLD timers reach
time higher than tLLD / 4 but lower than tLLD / 2 so LLD
timers maximum is set to tLLD. LLD timer setting for LLC
may be set to lower times.
Example of LLD operation with LLC convertor can be
seen in Figure 83. SMPS works under heavy load from point
0 to 3. Both LLD timers are reset each cycle before LLD
timer reaches tLLD / 4 and disable mode is not activated.
SMPS load decreases at point 3 and goes into skip. LLD
timers elapse during no switching time and change LLD
timer time to tLLD / 2. When skip burst comes at time 6
channel 2 starts to wake up, channel 1 starts to wake up at
time 7. Both channels are ready to conduct via SR transistor
VDS1 = VCS1
VDS2 = VCS2
DRV1
DRV2
DIS1
DIS2
t2
0
1
2 3
t4
t1
LLD tim1
tLLD1
LLD tim2
tLLD1
4 5 6 7 8 9 10 11 12
13 14
15 16 17
Figure 83. LLD Operation with LLC SMPS
www.onsemi.com
41
19 20 21 22 23 24 25 26 27 28
NCP4306
Operation flow
and dV / dt features are never activated both at same time.
Operation starts in bubble start where system comes when
VCC is higher than UVLO level and / or disable mode is
activated (by LLD or TRIG / DIS pin).
Followed bubble diagram at Figure 84 shows overall
operation flow. Black bubbles are fundamental parts of
system. States for dV / dt feature are colored by blue color
and states for LLC feature (exception timer) are in red. LLC
Figure 84. Overall Operation Bubble Diagram
Power dissipation calculation
significantly. Therefore, the MOSFET switch always
operates under Zero Voltage Switching (ZVS) conditions
when in a synchronous rectification system.
The following steps show how to approximately calculate
the power dissipation and DIE temperature of the NCP4306
controller. Note that real results can vary due to the effects
of the PCB layout on the thermal resistance.
It is important to consider the power dissipation in the
MOSFET driver of a SR system. If no external gate resistor
is used and the internal gate resistance of the MOSFET is
very low, nearly all energy losses related to gate charge are
dissipated in the driver. Thus it is necessary to check the SR
driver power losses in the target application to avoid over
temperature and to optimize efficiency.
In SR systems the body diode of the SR MOSFET starts
conducting before SR MOSFET is turned−on, because there
is some delay from VTH_CS_ON detect to turn−on the driver.
On the other hand, the SR MOSFET turn off process always
starts before the drain to source voltage rises up
Step 1 – MOSFET gate to source capacitance:
During ZVS operation the gate to drain capacitance does
not have a Miller effect like in hard switching systems
because the drain to source voltage does not change (or its
change is negligible).
Figure 85. Typical MOSFET Capacitances Dependency on VDS and VGS Voltages
www.onsemi.com
42
NCP4306
C iss + C gs ) C gd
(eq. 4)
C rss + C gd
(eq. 5)
C oss + C ds ) C gd
(eq. 6)
NCP4306 offers both a 5 V gate clamp and a 10 V gate
clamp for those MOSFET that require higher gate to source
voltage.
The total driving loss can be calculated using the selected
gate driver clamp voltage and the input capacitance of the
MOSFET:
Therefore, the input capacitance of a MOSFET operating
in ZVS mode is given by the parallel combination of the gate
to source and gate to drain capacitances (i.e. Ciss capacitance
for given gate to source voltage). The total gate charge,
Qg_total, of most MOSFETs on the market is defined for hard
switching conditions. In order to accurately calculate the
driving losses in a SR system, it is necessary to determine the
gate charge of the MOSFET for operation specifically in a
ZVS system. Some manufacturers define this parameter as
Qg_ZVS. Unfortunately, most datasheets do not provide this
data. If the Ciss (or Qg_ZVS) parameter is not available then
it will need to be measured. Please note that the input
capacitance is not linear (as shown Figure 85) and it needs
to be characterized for a given gate voltage clamp level.
P DRV_total + V CC
−
+
RDRV_SINK_EQ
f SW
(eq. 7)
The total driving power loss won’t only be dissipated in
the IC, but also in external resistances like the external gate
resistor (if used) and the MOSFET internal gate resistance
(Figure 86). Because NCP4306 features a clamped driver,
it’s high side portion can be modeled as a regular driver
switch with equivalent resistance and a series voltage
source. The low side driver switch resistance does not drop
immediately at turn−off, thus it is necessary to use an
equivalent value (RDRV_SIN_EQK) for calculations. This
method simplifies power losses calculations and still
provides acceptable accuracy. Internal driver power
dissipation can then be calculated using equation 10:
VCC
RDRV_SOURCE_EQ
C G_ZVS
Where:
VCC is the NCP4306 supply voltage
VCLAMP is the driver clamp voltage
Cg_ZVS is the gate to source capacitance of the
MOSFET in ZVS mode
fsw is the switching frequency of the target application
Step 2 – Gate drive losses calculation:
Gate drive losses are affected by the gate driver clamp
voltage. Gate driver clamp voltage selection depends on the
type of MOSFET used (threshold voltage versus channel
resistance). The total power losses (driving loses and
conduction losses) should be considered when selecting the
gate driver clamp voltage. Most of today’s MOSFETs for SR
systems feature low RDS_ON for 5 V VGS voltage. The
VCC − VCLAMP
V CLAMP
DRV
RG_EXT
SR MOSFET
RG_INT
GND
CG_ZVS
Figure 86. Equivalent Schematic of Gate Drive Circuitry
www.onsemi.com
43
NCP4306
P DRV_IC +
1
2
+)
C g_ZVS
1
2
C g_ZVS
V
CLAMP 2
V
CLAMP 2
f SW
f SW
ǒ
Ǔ
R DRV_SINK_EQ
R DRV_SINK_EQ ) R G_EXT ) R g_int
ǒ
R DRV_SOURCE_EQ
) C g_ZVS
V CLAMP
f SW
(V CC ) V CLAMP) (eq. 8)
Ǔ
(eq. 9)
R DRV_SOURCE_EQ ) R G_EXT ) R g_int
Where:
RDRV_SINK_EQ is the NCP4306 driver low side switch
equivalent resistance (1.6 Ω)
RDRV_SOURCE_EQ is the NCP4306 driver high side
switch equivalent resistance (7 Ω)
RG_EXT is the external gate resistor (if used)
Rg_int is the internal gate resistance of the MOSFET
P CC + V CC
I CC
(eq. 10)
Step 4 – IC die temperature arise calculation:
The die temperature can be calculated now that the total
internal power losses have been determined (driver losses
plus internal IC consumption losses). The SO8 package
thermal resistance is specified in the maximum ratings table
for a 35 μm thin copper layer with no extra copper plates on
any pin (i.e. just 0.5 mm trace to each pin with standard
soldering points are used).
The die temperature is calculated as:
Step 3 – IC consumption calculation:
In this step, power dissipation related to the internal IC
consumption is calculated. This power loss is given by the
ICC current and the IC supply voltage. The ICC current
depends on switching frequency and also on the selected min
tON and tOFF periods because there is current flowing out
from the MIN_TON and MIN_TOFF pins. The most
accurate method for calculating these losses is to measure
the ICC current when CLOAD = 0 nF and the IC is switching
at the target frequency with given min_tON and min_tOFF
adjust resistors. IC consumption losses can be calculated as:
T DIE + (P DRV_IC ) P CC)
R qJ*A ) T A
(eq. 11)
Where:
PDRV_IC is the IC driver internal power dissipation
PCC is the IC control internal power dissipation
R J−A is the thermal resistance from junction to ambient
TA is the ambient temperature
www.onsemi.com
44
NCP4306
OPN coding table
NCP4306 OPN is built from prefix of NCP4306 and
postfix that consist of seven letters. Meaning of these letters
are shown in table 7.
Table 7. OPN CODING TABLE
NCP4306xxxxxxx
Postfix Index
Parameter
Postfix
1
Pinout
A
MIN_TON, MIN_TOFF, LLD, TRIG / DIS − 8 pins
2
3
4
5
DRV
dV / dt + exception
MIN_TON
MIN_TOFF
Parameter
B
MIN_TON, LLD
C
MIN_TOFF, LLD
D
MIN_TON, MIN_TOFF
E
MIN_TOFF, TRIG / DIS
F
MIN_TON, TRIG / DIS
G
TRIG / DIS, LLD
H
None
A
DRV CLMP = 10 V
B
DRV CLMP = 5 V
A
None
D
Flyback (dV / dt) − 100 V / μs
H
LLC exception − multiplier 4
A
130 ns
B
220 ns
C
310 ns
D
400 ns
E
500 ns
F
600 ns
G
800 ns
H
1000 ns
I
1200 ns
J
1400 ns
K
1700 ns
L
2000 ns
Z
External
A
0.9 μs
B
1.0 μs
C
1.1 μs
D
1.2 μs
E
1.4 μs
F
1.6 μs
G
1.8 μs
H
2.0 μs
I
2.2 μs
J
2.4 μs
K
2.6 μs
L
2.9 μs
M
3.2 μs
N
3.5 μs
O
3.9 μs
Z
External
www.onsemi.com
45
NCP4306
Table 7. OPN CODING TABLE (continued)
NCP4306xxxxxxx
Postfix Index
Parameter
Postfix
6
LLD
A
68 μs
B
130 μs
C
280 μs
D
540 μs
E
1075 μs
F
Disabled
Z
External
A
−
7
Reserved
www.onsemi.com
46
Parameter
NCP4306
PACKAGE DIMENSIONS
SOIC−8 NB
CASE 751−07
ISSUE AK
−X−
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEW
STANDARD IS 751−07.
A
8
5
S
B
0.25 (0.010)
M
Y
M
1
4
−Y−
K
G
C
N
DIM
A
B
C
D
G
H
J
K
M
N
S
X 45 _
SEATING
PLANE
−Z−
0.10 (0.004)
H
D
0.25 (0.010)
M
Z Y
S
X
M
J
S
SOLDERING FOOTPRINT*
1.52
0.060
7.0
0.275
4.0
0.155
0.6
0.024
1.270
0.050
SCALE 6:1
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM / D.
www.onsemi.com
47
MILLIMETERS
MIN
MAX
4.80
5.00
3.80
4.00
1.35
1.75
0.33
0.51
1.27 BSC
0.10
0.25
0.19
0.25
0.40
1.27
0_
8_
0.25
0.50
5.80
6.20
INCHES
MIN
MAX
0.189
0.197
0.150
0.157
0.053
0.069
0.013
0.020
0.050 BSC
0.004
0.010
0.007
0.010
0.016
0.050
0 _
8 _
0.010
0.020
0.228
0.244
NCP4306
PACKAGE DIMENSIONS
TSOP−6
CASE 318G−02
ISSUE V
D
H
ÉÉ
ÉÉ
6
E1
1
NOTE 5
5
2
L2
4
GAUGE
PLANE
E
3
L
b
C
DETAIL Z
e
0.05
M
A
A1
NOTES:
1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994.
2. CONTROLLING DIMENSION: MILLIMETERS.
3. MAXIMUM LEAD THICKNESS INCLUDES LEAD FINISH. MINIMUM
LEAD THICKNESS IS THE MINIMUM THICKNESS OF BASE MATERIAL.
4. DIMENSIONS D AND E1 DO NOT INCLUDE MOLD FLASH,
PROTRUSIONS, OR GATE BURRS. MOLD FLASH, PROTRUSIONS, OR
GATE BURRS SHALL NOT EXCEED 0.15 PER SIDE. DIMENSIONS D
AND E1 ARE DETERMINED AT DATUM H.
5. PIN ONE INDICATOR MUST BE LOCATED IN THE INDICATED ZONE.
SEATING
PLANE
c
DETAIL Z
DIM
A
A1
b
c
D
E
E1
e
L
L2
M
RECOMMENDED
SOLDERING FOOTPRINT*
6X
0.60
3.20
6X
0.95
0.95
PITCH
DIMENSIONS: MILLIMETERS
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM / D.
www.onsemi.com
48
MIN
0.90
0.01
0.25
0.10
2.90
2.50
1.30
0.85
0.20
05
MILLIMETERS
NOM
MAX
1.00
1.10
0.06
0.10
0.38
0.50
0.18
0.26
3.00
3.10
2.75
3.00
1.50
1.70
0.95
1.05
0.40
0.60
0.25 BSC
−
105
NCP4306
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NCP4306/D