MPS MP1583DP 3a, 23v, 385khz step-down converter Datasheet

MP1583
3A, 23V, 385KHz
Step-Down Converter
The Future of Analog IC Technology
DESCRIPTION
FEATURES
The MP1583 is a step-down regulator with a
built-in internal Power MOSFET. It achieves 3A
of continuous output current over a wide input
supply range with excellent load and line
regulation.
•
•
•
•
•
•
•
•
•
•
•
•
Current mode operation provides fast transient
response and eases loop stabilization.
Fault
condition
protection
includes
cycle-by-cycle current limiting and thermal
shutdown. An adjustable soft-start reduces the
stress on the input source at startup. In
shutdown mode the regulator draws 20μA of
supply current.
The MP1583 requires a minimum number of
external components, providing a compact
solution.
3A Output Current
Programmable Soft-Start
100mΩ Internal Power MOSFET Switch
Stable with Low ESR Output Ceramic Capacitors
Up to 95% Efficiency
20μA Shutdown Mode
Fixed 385KHz Frequency
Thermal Shutdown
Cycle-by-Cycle Over Current Protection
Wide 4.75V to 23V Operating Input Range
Output Adjustable from 1.22V to 21V
Under-Voltage Lockout
APPLICATIONS
•
•
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Distributed Power Systems
Battery Chargers
Pre-Regulator for Linear Regulators
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
TYPICAL APPLICATION
Efficiency Curve
OPEN =
AUTOMATIC
STARTUP
10μ F
CERAMIC
2
7
8
IN
BS
SW
EN
MP1583
SS
GND
4
10nF
FB
COMP
6
100
10nF
1
3
5
B330A
5.6nF
3.9kΩ
15μ H
10.5kΩ
10kΩ
OUTPUT
2.5V
3A
22μ F
CERAMIC
VIN = 10V
VOUT=5.0V
90
EFFICIENCY (%)
INPUT
4.75V to 23V
VOUT=2.5V
80
VOUT=3.3V
70
60
50
0
0.5 1.0 1.5 2.0 2.5 3.0 3.5
LOAD CURRENT (A)
MP1583_EC01
MP1583 Rev. 3.1
6/20/2011
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number*
MP1583DN
MP1583DP
Package
SOIC8E
PDIP8
Top Marking
MP1583DN
MP1583DP
Free Air Temperature (TA)
–40°C to +85°C
–40°C to +85°C
* For Tape & Reel, add suffix –Z (e.g. MP8736DL–Z)
For RoHS compliant packaging, add suffix –LF (e.g. MP8736DL–LF–Z)
PACKAGE REFERENCE
TOP VIEW
SOIC8N/PDIP8
BS
1
8
SS
IN
2
7
EN
SW
3
6
COMP
GND
4
5
FB
EXPOSED PAD
(SOIC8N ONLY)
CONNECT TO PIN 4
MP1583_PD01
ABSOLUTE MAXIMUM RATINGS (1)
Thermal Resistance
Supply Voltage VIN .......................–0.3V to +28V
Switch Voltage VSW ................. –1V to VIN + 0.3V
Bootstrap Voltage VBS ....VSW – 0.3V to VSW + 6V
FB, COMP and SS Pins.................–0.3V to +6V
(2)
Continuous Power Dissipation (TA = +25°C)
SOIC8E...................................................... 2.5W
PDIP8 ........................................................ 1.2W
Junction Temperature ...............................150°C
Lead Temperature ....................................260°C
Storage Temperature ............. –65°C to +150°C
SOIC8E .................................. 50 ...... 10... °C/W
PDIP8 .................................... 104 ..... 45... °C/W
Recommended Operating Conditions
(3)
Input Voltage VIN ............................4.75V to 23V
Operating Junct. Temp (TJ)...... -40°C to +125°C
MP1583 Rev. 3.1
6/20/2011
(4)
θJA
θJC
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ(MAX), the junction-toambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD(MAX)=(TJ(MAX)TA)/θJA. Exceeding the maximum allowable power dissipation
will cause excessive die temperature, and the regulator will go
into thermal shutdown. Internal thermal shutdown circuitry
protects the device from permanent damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7, 4-layer PCB
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameters
Symbol Condition
Shutdown Supply Current
Supply Current
VEN = 0V
VEN = 2.8V, VFB =1.4V
Feedback Voltage
VFB
Error Amplifier Voltage Gain
AVEA
Error Amplifier Transconductance
GEA
High-Side Switch On-Resistance
Low-Side Switch On-Resistance
High-Side Switch Leakage Current
Current Limit
Current Sense to COMP Transconductance
Oscillation Frequency
Short Circuit Oscillation Frequency
Maximum Duty Cycle
Minimum Duty Cycle
EN Shutdown Threshold Voltage
Enable Pull Up Current
EN UVLO Threshold
EN UVLO Threshold Hysteresis
Soft-Start Period
Min
4.75V ≤ VIN ≤ 23V
Units
20
1.0
30
1.2
µA
mA
400
ΔICOMP = ±10μA
500
VEN = 0V, VSW = 0V
4.0
GCS
fS
VFB = 0V
VFB = 1.0V
VFB = 1.5V
VEN = 0V
VEN Rising
335
25
0.9
1.1
2.37
CSS = 0.1µF
Thermal Shutdown
MP1583 Rev. 3.1
6/20/2011
Max
1.194 1.222 1.250
RDS(ON)1
RDS(ON)2
DMAX
Typ
800
0.1
10
0
4.9
3.8
385
40
90
1.2
1.8
2.54
210
10
160
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V
V/V
1120
10
6.0
435
55
0
1.5
2.5
2.71
µA/V
Ω
Ω
µA
A
A/V
KHz
KHz
%
%
V
µA
V
mV
ms
°C
3
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
Soft-Start
Efficiency Curve
100
CSS Open, VIN = 10V, VOUT = 3.3V,
1.5A Resistive Load
VIN = 7V
VOUT=5.0V
EFFICIENCY (%)
90
VOUT=2.5V
80
VOUT=3.3V
70
VOUT
2V/div.
IL
1A/div.
60
50
VEN
5V/div.
0
0.5 1.0 1.5 2.0 2.5 3.0 3.5
LOAD CURRENT (A)
MP1583-TPC02
MP1583-TPC01
VEN
5V/div.
VEN
5V/div.
VOUT
2V/div.
VOUT
2V/div.
IL
1A/div.
IL
1A/div.
1ms/div.
MP1583-TPC03
MP1583-TPC04
PIN FUNCTIONS
Pin #
1
2
3
4
5
6
Name Description
High-Side Gate Drive Bootstrap Input. BS supplies the drive for the high-side N-Channel MOSFET
BS
switch. Connect a 4.7nF or greater capacitor from SW to BS to power the high-side switch.
Power Input. IN supplies the power to the IC. Drive IN with a 4.75V to 23V power source. Bypass
IN
IN to GND with a suitably large capacitor to eliminate noise on the input to the IC. See Input
Capacitor
Power Switching Output. SW is the switching node that supplies power to the output. Connect the
SW
output LC filter from SW to the output load. Note that a capacitor is required from SW to BS to
power the high-side switch.
GND Ground. (Note: For the SOIC8E package, connect the exposed pad on backside to Pin 4).
Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive voltage
FB
divider from the output voltage. The feedback threshold is 1.222V. See Setting the Output Voltage
Compensation Node. COMP is used to compensate the regulation control loop. Connect a series
COMP
RC network from COMP to GND to compensate the regulation control loop. See Compensation
MP1583 Rev. 3.1
6/20/2011
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
PIN FUNCTIONS (continued)
Pin #
Name Description
7
EN
Enable/UVLO. A voltage greater than 2.71V enables operation. For complete low
current shutdown the EN pin voltage needs to be at less than 900mV. When the
voltage on EN exceeds 1.2V, the internal regulator will be enabled and the soft-start
capacitor will begin to charge. The MP1583 will start switching after the EN pin
voltage reaches 2.71V. There is 7V zener connected between EN and GND. If EN is
driven by external signal, the voltage should never exceed 7V.
8
SS
Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND to
set the soft-start period. To disable the soft-start feature, leave SS unconnected.
OPERATION
The voltage at COMP is compared to the
internally measured switch current to control the
output voltage.
The MP1583 is a current-mode step-down
regulator. It regulates input voltages from 4.75V to
23V down to an output voltage as low as 1.222V,
and is able to supply up to 3A of load current.
The converter uses an internal N-Channel
MOSFET switch to step-down the input voltage to
the regulated output voltage. Since the MOSFET
requires a gate voltage greater than the input
voltage, a boost capacitor connected between
SW and BS drives the gate. The capacitor is
internally charged when SW is low.
The MP1583 uses current-mode control to
regulate the output voltage. The output voltage is
measured at FB through a resistive voltage
divider and amplified through the internal error
amplifier.
The
output
current
of
the
transconductance error amplifier is presented at
COMP where a RC network compensates the
regulation control system.
An internal 10Ω switch from SW to GND is used
to insure that SW is pulled to GND when SW is
low in order to fully charge the BS capacitor.
IN 2
CURRENT
SENSE
AMPLIFIER
INTERNAL
REGULATORS
OSCILLATOR
40/385KHz
+
1.2V
--
EN 7
--
2.54V
+
FREQUENCY
FOLDBACK
COMPARATOR
SHUTDOWN
COMPARATOR
SLOPE
COMP
5V
CLK
--
+
-1μ A
LOCKOUT
COMPARATOR
+
Σ
S
Q
R
Q
CURRENT
COMPARATOR
1
BS
3
SW
4
GND
7V
1.8V
--
+
--
0.7V
1.222V
5
FB
+
ERROR
AMPLIFIER
GM = 800μ A/V 6
COMP
8
SS
Figure 1—Functional Block Diagram
MP1583 Rev. 3.1
6/20/2011
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
APPLICATION INFORMATION
COMPONENT SELECTION
Setting the Output Voltage
The output voltage is set using a resistive
voltage divider from the output voltage to the
FB pin. The voltage divider divides the output
voltage down to the feedback voltage by the
ratio:
V FB = VOUT
R2
R1 + R 2
Where VFB is the feedback voltage and VOUT is
the output voltage.
Thus the output voltage is:
VOUT = 1.22V ×
R1 = 8.18 × (VOUT − 1.22V )( kΩ )
For example, for a 3.3V output voltage, R2 is
10kΩ, and R1 is 17kΩ.
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will result in less ripple current and
lower output ripple voltage. However, larger
value inductors have a larger physical size,
higher series resistance, and/or lower
saturation current. A good rule for determining
the inductance to use is to allow the inductor
peak-to-peak ripple current to be approximately
30% of the maximum switch current limit. Also,
make sure that the peak inductor current is
below the maximum switch current limit. The
inductance value can be calculated by:
⎛ V
VOUT
× ⎜⎜ 1 − OUT
VIN
f S × ΔI L ⎝
The peak inductor current can be calculated by:
I LP = I LOAD +
⎛ V
VOUT
× ⎜⎜ 1 − OUT
2 × fS × L ⎝
V IN
⎞
⎟⎟
⎠
Where ILOAD is the load current.
Table 1 lists a number of suitable inductors
from various manufacturers. The choice of
which inductor to use mainly depends on the
price vs. size requirements and any EMI
requirements.
Table 1—Inductor Selection Guide
R1 + R 2
R2
A typical value for R2 can be as high as 100kΩ,
but a typical value is 10kΩ. Using that value, R1
is determined by:
L=
Choose an inductor that will not saturate under
the maximum inductor peak current.
Vendor/
Model
Package
Dimensions
(mm)
Core
Type
Core
Material
W
L
H
CR75
Open
Ferrite
7.0
7.8
5.5
CDH74
Open
Ferrite
7.3
8.0
5.2
CDRH5D28 Shielded
Ferrite
5.5
5.7
5.5
CDRH5D28 Shielded
Ferrite
5.5
5.7
5.5
CDRH6D28 Shielded
Ferrite
6.7
6.7
3.0
CDRH104R Shielded
Ferrite
10.1 10.0
3.0
Sumida
Toko
D53LC
Type A
Shielded
Ferrite
5.0
5.0
3.0
D75C
Shielded
Ferrite
7.6
7.6
5.1
D104C
Shielded
Ferrite
10.0 10.0
4.3
D10FL
Open
Ferrite
9.7
1.5
4.0
DO3308
Open
Ferrite
9.4
13.0
3.0
DO3316
Open
Ferrite
9.4
13.0
5.1
Coilcraft
⎞
⎟⎟
⎠
Where VIN is the input voltage, fS is the 385KHz
switching frequency and ΔIL is the peak-to-peak
inductor ripple current.
MP1583 Rev. 3.1
6/20/2011
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off.
Use a Schottky diode to reduce losses due to
the diode forward voltage and recovery times.
The input capacitor can be electrolytic, tantalum
or ceramic. When using electrolytic or tantalum
capacitors, a small, high quality ceramic
capacitor (i.e. 0.1μF) should be placed as close
to the IC as possible.
Choose a diode whose maximum reverse voltage
rating is greater than the maximum input voltage,
and whose current rating is greater than the
maximum load current. Table 2 lists example
Schottky diodes and manufacturers.
When using ceramic capacitors, make sure that
they have enough capacitance to provide
sufficient charge to prevent excessive voltage
ripple at the input. The input voltage ripple
caused by capacitance can be estimated by:
Table 2—Diode Selection Guide
Diode
SK33
SK34
B330
B340
MBRS330
MBRS340
Voltage/Current Manufacture
Rating
30V, 3A
40V, 3A
30V, 3A
40V, 3A
30V, 3A
40V, 3A
Diodes Inc.
Diodes Inc.
Diodes Inc.
Diodes Inc.
On Semiconductor
On Semiconductor
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
will also suffice.
Since the input capacitor absorbs the input
switching current it requires an adequate ripple
current rating. The RMS current in the input
capacitor can be estimated by:
I C 1 = I LOAD ×
VOUT ⎛⎜ VOUT
× 1−
V IN ⎜⎜⎝ V IN
⎞
⎟
⎟⎟
⎠
The worst-case condition occurs at VIN = 2VOUT,
where:
I C1 =
I LOAD
2
For simplification, choose an input capacitor
whose RMS current rating is greater than half of
the maximum load current.
MP1583 Rev. 3.1
6/20/2011
ΔV IN =
I LOAD VOUT ⎛ VOUT
×
× ⎜1 −
f S × C1 VIN ⎜⎝
V IN
⎞
⎟⎟
⎠
Where C1 is the input capacitance value.
Output Capacitor
The output capacitor is required to maintain the
DC output voltage. Ceramic, tantalum or low
ESR electrolytic capacitors are recommended.
Low ESR capacitors are preferred so as to
keep the output voltage ripple low. The output
voltage ripple can be estimated by:
ΔVOUT =
VOUT ⎛ VOUT
× ⎜1 −
f S × L ⎜⎝
VIN
⎞
⎞ ⎛
1
⎟
⎟⎟ × ⎜⎜ RESR +
8 × f S × C 2 ⎟⎠
⎠ ⎝
Where L is the inductor value, C2 is the output
capacitance value and RESR is the equivalent
series resistance (ESR) value of the output
capacitor.
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance, which is the
main cause for the output voltage ripple. For
simplification, the output voltage ripple can be
estimated by:
ΔVOUT =
⎛ V
× ⎜⎜ 1 − OUT
VIN
× L × C2 ⎝
VOUT
8× fS
2
⎞
⎟⎟
⎠
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the
output ripple can be approximated to:
ΔVOUT =
VOUT ⎛ VOUT ⎞
× ⎜1 −
⎟ × R ESR
fS × L ⎝
VIN ⎠
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
The MP1583 can be optimized for a wide range
of capacitance and ESR values.
Compensation Components
The MP1583 employs current mode control for
easy compensation and fast transient response.
The system stability and transient response are
controlled through the COMP pin. COMP is the
output of the internal transconductance error
amplifier.
A
series
capacitor-resistor
combination sets a pole-zero combination to
control the characteristics of the control system.
The DC gain of the voltage feedback loop is:
AVDC = RLOAD × GCS × AVEA ×
VFB
VOUT
Where AVEA is the error amplifier voltage gain,
GCS is the current sense transconductance and
RLOAD is the load resistor value.
The system has two poles of importance. One
is due to the compensation capacitor (C3) and
the output resistor of error amplifier while the
other is due to the output capacitor and the load
resistor. These poles are located at:
f P1 =
G EA
2π × C 3 × AVEA
f P2 =
1
2π × C 2 × RLOAD
is
Where
GEA
transconductance.
the
error
amplifier
1
=
2π × C 3 × R3
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero
due to the ESR and capacitance of the output
capacitor is located at:
f ESR
MP1583 Rev. 3.1
6/20/2011
1
=
2π × C 2 × RESR
f P3 =
the
the
to
the
1
2π × C 6 × R 3
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
(where the feedback loop has unity gain) is
important.
Lower crossover frequencies result in slower
line and load transient responses, while higher
crossover frequencies could cause system
instability. A good standard is to set the
crossover frequency to approximately one-tenth
of the switching frequency. The switching
frequency for the MP1583 is 385KHz, so the
desired crossover frequency is around 38KHz.
Table 3 lists the typical values of compensation
components for some standard output voltages
with various output capacitors and inductors.
The values of the compensation components
have been optimized for fast transient
responses and good stability at given conditions.
Table 3—Compensation Values for Typical
Output Voltage/Capacitor Combinations
(Please reference Fig. 3 and Fig. 4)
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). This zero is located
at:
f Z1
In this case, a third pole set by
compensation capacitor (C6) and
compensation resistor (R3) is used
compensate the effect of the ESR zero on
loop gain. This pole is located at:
VOUT
C2
R3
C3
C6
2.5V
22μF
Ceramic
3.9kΩ
5.6nF
None
3.3V
22μF
Ceramic
4.7kΩ
4.7nF
None
5V
22μF
Ceramic
7.5kΩ
4.7nF
None
12V
22μF
Ceramic
16.9kΩ
1.5nF
None
2.5V
560μF Al.
30mΩ ESR
91kΩ
1nF
150pF
3.3V
560μF Al
30mΩ ESR
120kΩ
1nF
120pF
5V
470μF Al.
30mΩ ESR
100kΩ
1nF
120pF
12V
220μF Al.
30mΩ ESR
169kΩ
1nF
39pF
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
4)
Route SW away from sensitive analog
areas such as FB.
5)
Connect IN, SW, and especially GND
respectively to a large copper area to cool
the chip to improve thermal performance
and long-term reliability. For single layer,
do not solder exposed pad of the IC
Where R3 is the compensation resistor value.
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3
at the location of the ESR zero. Determine C6
by the equation:
C6 =
C2 × R ESR
R3
FB 5
6
SGND
SGND
C6
C3
R3
4 GND
C5
f
1
< S
2π × C2 × R ESR
2
R1
R2
C4
3. Determine if the second compensation
capacitor (C6) is required. It is required if the
ESR zero of the output capacitor is located at
less than half of the 385KHz switching
frequency, or if the following relationship is valid:
R4
COMP
4
2π × R3 × f C
Ensure all feedback connections are short
and direct. Place the feedback resistors
and compensation components as close to
the chip as possible.
SW
C3 >
3)
3
2. Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, below one forth of
the crossover frequency provides sufficient
phase margin. Determine C3 by the following
equation:
Keep the connection of low-side MOSFET
between SW pin and input power ground
as short and wide as possible.
EN 7
Where fC is the desired crossover frequency
(which typically has a value no higher than
38KHz).
2)
IN
2π × C2 × f C VOUT
×
G EA × G CS
VFB
Keep the path of switching current short
and minimize the loop area formed by Input
cap, high-side and low-side MOSFETs.
2
R3 =
1)
SS/REF 8
1. Choose the compensation resistor (R3) to set
the desired crossover frequency. Determine R3
by the following equation:
PCB Layout Guide
PCB layout is very important to achieve stable
operation. Please follow these guidelines and
take Figure2 and 3 for references.
1 BS
To optimize the compensation components for
conditions not listed in Table 2, the following
procedure can be used.
L1
C1
D1
PGND
C2
Figure 2―PCB Layout (Single Layer)
MP1583 Rev. 3.1
6/20/2011
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
C4
R4
FB 5
SGND
4 GND
6
SW
3
COMP
EN 7
IN
2
SS/REF 8
1 BS
R2
C2
R1
C3
C6
External Bootstrap Diode
An external bootstrap diode may enhance the
efficiency of the regulator, the applicable
conditions of external BST diode are:
R3
L1
z
VOUT=5V or 3.3V; and
z
Duty cycle is high: D=
VOUT
>65%
VIN
In these cases, an external BST diode is
recommended from the output of the voltage
regulator to BST pin, as shown in Fig.4
External BST Diode
IN4148
BST
C1
D1
PGND
MP1583
C5
Top Layer
SW
CBST
L
+
COUT
5V or 3.3V
Figure 4—Add Optional External Bootstrap
Diode to Enhance Efficiency
The recommended external BST diode is
IN4148, and the BST cap is 0.1~1µF.
Bottom Layer
Figure 3―PCB Layout (Double Layer)
MP1583 Rev. 3.1
6/20/2011
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10
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
TYPICAL APPLICATION CIRCUITS
C5
10nF
INPUT
4.75V to 23V
2
OPEN = AUTOMATIC
STARTUP
C1
10μ F/25V
CERAMIC
7
L1
15μ H
1
IN
BS
SW
EN
3
R1
16.9kΩ
MP1583
8
SS
GND
FB
COMP
4
C4
10nF
5
6
C6
NS
C3
4.7nF
R3
4.7kΩ
OUTPUT
3.3V
3A
D1
R2
10kΩ
C2
22μ F/10V
Murata
Figure 5—3.3V output 3A solution with Murata 22µF, 10V Ceramic Output Capacitor
MP1583 Rev. 3.1
6/20/2011
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11
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
PACKAGE INFORMATION
SOIC8E (EXPOSED PAD)
0.189(4.80)
0.197(5.00)
8
0.124(3.15)
0.136(3.45)
5
0.150(3.80)
0.157(4.00)
PIN 1 ID
1
0.228(5.80)
0.244(6.20)
0.089(2.26)
0.101(2.56)
4
TOP VIEW
BOTTOM VIEW
SEE DETAIL "A"
0.013(0.33)
0.020(0.51)
0.051(1.30)
0.067(1.70)
SEATING PLANE
0.000(0.00)
0.006(0.15)
0.0075(0.19)
0.0098(0.25)
SIDE VIEW
0.050(1.27)
BSC
FRONT VIEW
0.010(0.25)
x 45o
0.020(0.50)
GAUGE PLANE
0.010(0.25) BSC
0.050(1.27)
0.024(0.61)
0o-8o
0.016(0.41)
0.050(1.27)
0.063(1.60)
DETAIL "A"
0.103(2.62)
0.138(3.51)
RECOMMENDED LAND PATTERN
MP1583 Rev. 3.1
6/20/2011
0.213(5.40)
NOTE:
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN
BRACKET IS IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,
PROTRUSIONS OR GATE BURRS.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
OR PROTRUSIONS.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.004" INCHES MAX.
5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.
6) DRAWING IS NOT TO SCALE.
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
PDIP8
0.387 (9.830)
0.367 (9.322)
0.325(8.255)
0.300(7.620)
PIN 1 IDENT.
0.260 (6.604)
0.240 (6.096)
0.014 (0.356)
0.008 (0.200)
3°~11°
Lead Bend
0.392(9.957)
0.332(8.433)
0.065 (1.650)
0.050 (1.270)
0.040 (1.016)
0.020 (0.508)
0.145(3.683)
0.134(3.404)
0.035 (0.889)
0.015 (0.381)
0.140(3.556)
0.120(3.048)
0.100 BSC(2.540)
0.021(0.533)
0.015(0.381)
NOTE:
1) Control dimension is in inches
. Dimension in bracket is millimeters.
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS
products into any application. MPS will not assume any legal responsibility for any said applications.
MP1583 Rev. 3.1
6/20/2011
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2011 MPS. All Rights Reserved.
13