ON CS5155HGDR16 Cpu 5−bit synchronous buck controller Datasheet

CS5155H
CPU 5−Bit Synchronous
Buck Controller
The CS5155H is a 5−bit synchronous dual N−Channel buck
controller. It is designed to provide unprecedented transient response
for today’s demanding high−density, high−speed logic. The regulator
operates using a proprietary control method, which allows a 100 ns
response time to load transients. The CS5155H is designed to operate
over a 4.25−20 V range (VCC) using 12 V to power the IC and 5.0 V or
12 V as the main supply for conversion.
The CS5155H is specifically designed to power Pentium® II
processors and other high performance core logic. It includes the
following features: on board, 5−bit DAC, short circuit protection,
1.0% output tolerance, VCC monitor, and programmable Soft Start
capability. The CS5155H is backwards compatible with the 4−bit
CS5150, allowing the mother board designer the capability of using
either the CS5150 or the CS5155H with no change in layout. The
CS5155H is available in 16 pin surface mount packages.
Features
Dual N−Channel Design
Excess of 1.0 MHz Operation
100 ns Transient Response
5−Bit DAC
Backward Compatible with 4−Bit CS5150H/CS5151H
30 ns Gate Rise/Fall Times
1.0% DAC Accuracy
5.0 V & 12 V Operation
Remote Sense
Programmable Soft Start
Lossless Short Circuit Protection
VCC Monitor
25 ns FET Nonoverlap Time
Adaptive Voltage Positioning
V2™ Control Topology
Current Sharing
Overvoltage Protection
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© Semiconductor Components Industries, LLC, 2006
July, 2006 − Rev. 3
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MARKING
DIAGRAM
16
16
1
CS5155H
AWLYWW
SOIC−16
D SUFFIX
CASE 751B
A
WL, L
YY, Y
WW, W
1
= Assembly Location
= Wafer Lot
= Year
= Work Week
PIN CONNECTIONS
VID0
VID1
VID2
VID3
SS
VID4
COFF
VFFB
1
16
VFB
COMP
LGND
VCC1
VGATE(L)
PGND
VGATE(H)
VCC2
ORDERING INFORMATION
Device
1
Package
Shipping
CS5155HGD16
SO−16
48 Units/Rail
CS5155HGDR16
SO−16
2500 Tape & Reel
Publication Order Number:
CS5155H/D
CS5155H
5.0 V
12 V
0.1 μF
VCC1
VID0
VID0
VID1
VID1
VID2
VID2
VID3
VID3
VID4
VID4
COFF
330 pF
VCC2
VGATE(H)
2.0 μH
1.3 V to 3.5 V @ 13 A
IRL3103
VGATE(L)
CS5155H
PGND
SS
0.1 μF
1200 μF/16 V × 3
AIEI
IRL3103
COMP
VFB
3.3 k
0.33 μF
LGND
VFFB
1200 μF/16 V × 5
AIEI
100 pF
Figure 1. Application Diagram, Switching Power Supply for Core Logic − Pentium) II Processor
ABSOLUTE MAXIMUM RATINGS*
Rating
Value
Unit
0 to 150
°C
260 peak
230 peak
°C
−65 to +150
°C
2.0
kV
Operating Junction Temperature, TJ
Lead Temperature Soldering:
Wave Solder (through hole styles only) (Note 1)
Reflow: (SMD styles only) (Note 2)
Storage Temperature Range, TS
ESD Susceptibility (Human Body Model)
1. 10 second maximum.
2. 60 second maximum above 183°C.
*The maximum package power dissipation must be observed.
ABSOLUTE MAXIMUM RATINGS
Pin Name
Max Operating Voltage
Max Current
VCC1
16 V/−0.3 V
25 mA DC/1.5 A peak
VCC2
20 V/−0.3 V
20 mA DC/1.5 A peak
SS
6.0 V/−0.3 V
−100 μA
COMP
6.0 V/−0.3 V
200 μA
VFB
6.0 V/−0.3 V
−0.2 μA
COFF
6.0 V/−0.3 V
−0.2 μA
VFFB
6.0 V/−0.3 V
−0.2 μA
VID0 − VID4
6.0 V/−0.3 V
−50 μA
VGATE(H)
20 V/−0.3 V
100 mA DC/1.5 A peak
VGATE(L)
16 V/−0.3 V
100 mA DC/1.5 A peak
LGND
0V
25 mA
PGND
0V
100 mA DC/1.5 A peak
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CS5155H
ELECTRICAL CHARACTERISTICS (0°C < TA < +70°C; 0°C < TJ < +125°C; 8.0 V < VCC1 < 14 V; 5.0 V < VCC2 < 20 V;DAC
Code: VID4 = VID2 = VID1 = VID0 = 1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1.0 nF; COFF = 330 pF; CSS = 0.1 μF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Error Amplifier
VFB Bias Current
VFB = 0 V
−
0.3
1.0
μA
Open Loop Gain
1.25 V < VCOMP < 4.0 V; Note 3
50
60
−
dB
Unity Gain Bandwidth
Note 3
500
3000
−
kHz
COMP SINK Current
VCOMP = 1.5 V; VFB = 3.0 V; VSS > 2.0 V
0.4
2.5
8.0
mA
COMP SOURCE Current
VCOMP = 1.2 V; VFB = 2.7 V; VSS = 5.0 V
30
50
80
μA
COMP CLAMP Current
VCOMP = 0 V; VFB = 2.7 V
0.4
1.0
1.6
mA
COMP High Voltage
VFB = 2.7 V; VSS = 5.0 V
4.0
4.3
5.0
V
COMP Low Voltage
VFB = 3.0 V
−
160
600
mV
PSRR
8.0 V < VCC1 < 14 V @ 1.0 kHz; Note 3
60
85
−
dB
VCC1 Monitor
Start Threshold
Output switching
3.75
3.90
4.05
V
Stop Threshold
Output not switching
3.70
3.85
4.00
V
Hysteresis
Start−Stop
−
50
−
mV
Out SOURCE Sat at 100 mA
Measure VCC1 − VGATE(L); VCC2 − VGATE(H)
−
1.2
2.0
V
Out SINK Sat at 100 mA
Measure VGATE(H) − VPGND; VGATE(L) − VPGND
−
1.0
1.5
V
Out Rise Time
1.0 V < VGATE(H) < 9.0 V; 1.0 V < VGATE(L) < 9.0 V;
VCC1 = VCC2 = 12 V
−
30
50
ns
Out Fall Time
9.0 V > VGATE(H) > 1.0 V; 9.0 V > VGATE(L) > 1.0 V;
VCC1 = VCC2 = 12 V
−
30
50
ns
Shoot−Through Current
Note 3
−
−
50
mA
Delay VGATE(H) to VGATE(L)
VGATE(H) falling to 2.0 V; VCC1 = VCC2 = 8.0 V;
VGATE(L) rising to 2.0 V
−
25
50
ns
Delay VGATE(L) to VGATE(H)
VGATE(L) falling to 2.0 V; VCC1 = VCC2 = 8.0 V;
VGATE(H) rising to 2.0 V
−
25
50
ns
VGATE(H), VGATE(L) Resistance
Resistor to LGND. Note 3
20
50
100
kΩ
VGATE(H), VGATE(L) Schottky
LGND to VGATE(H) @ 10 mA
LGND to VGATE(L) @ 10 mA
−
600
800
mV
VGATE(H) and VGATE(L)
Soft Start (SS)
Charge Time
−
1.6
3.3
5.0
ms
Pulse Period
−
25
100
200
ms
Duty Cycle
(Charge Time /Pulse Period) × 100
1.0
3.3
6.0
%
COMP Clamp Voltage
VFB = 0 V; VSS = 0
0.50
0.95
1.10
V
VFFB SS Fault Disable
VGATE(H) = Low; VGATE(L) = Low
0.9
1.0
1.1
V
−
2.5
3.0
V
High Threshold
−
PWM Comparator
Transient Response
VFFB = 0 to 5.0 V to VGATE(H) = 9.0 V to 1.0 V;
VCC1 = VCC2 = 12 V
−
100
125
ns
VFFB Bias Current
VFFB = 0 V
−
0.3
−
μA
3. Guaranteed by design, not 100% tested in production.
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CS5155H
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < +70°C; 0°C < TJ < +125°C; 8.0 V < VCC1 < 14 V; 5.0 V < VCC2 < 20 V;DAC
Code: VID4 = VID2 = VID1 = VID0 = 1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1.0 nF; COFF = 330 pF; CSS = 0.1 μF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
DAC
Input Threshold
VID0, VID1, VID2, VID3, VID4
1.00
1.25
2.40
V
Input Pull Up Resistance
VID0, VID1, VID2, VID3, VID4
25
50
100
kΩ
4.85
5.00
5.15
V
−
−
1.0
%
Pull Up Voltage
−
Accuracy (all codes except 11111)
Measure VFB = VCOMP, 25°C ≤ TJ ≤ 125°C
VID4
VID3
VID2
VID1
VID0
0
1
1
1
1
−
1.3266
1.3400
1.3534
V
0
1
1
1
0
−
1.3761
1.3900
1.4039
V
0
1
1
0
1
−
1.4256
1.4400
1.4544
V
0
1
1
0
0
−
1.4751
1.4900
1.5049
V
0
1
0
1
1
−
1.5246
1.5400
1.5554
V
0
1
0
1
0
−
1.5741
1.5900
1.6059
V
0
1
0
0
1
−
1.6236
1.6400
1.6564
V
0
1
0
0
0
−
1.6731
1.6900
1.7069
V
0
0
1
1
1
−
1.7226
1.7400
1.7574
V
0
0
1
1
0
−
1.7721
1.7900
1.8079
V
0
0
1
0
1
−
1.8216
1.8400
1.8584
V
0
0
1
0
0
−
1.8711
1.8900
1.9089
V
0
0
0
1
1
−
1.9206
1.9400
1.9594
V
0
0
0
1
0
−
1.9701
1.9900
2.0099
V
0
0
0
0
1
−
2.0196
2.0400
2.0604
V
0
0
0
0
0
−
2.0691
2.0900
2.1109
V
1
1
1
1
1
−
1.2191
1.2440
1.2689
V
1
1
1
1
0
−
2.1186
2.1400
2.1614
V
1
1
1
0
1
−
2.2176
2.2400
2.2624
V
1
1
1
0
0
−
2.3166
2.3400
2.3634
V
1
1
0
1
1
−
2.4156
2.4400
2.4644
V
1
1
0
1
0
−
2.5146
2.5400
2.5654
V
1
1
0
0
1
−
2.6136
2.6400
2.6664
V
1
1
0
0
0
−
2.7126
2.7400
2.7674
V
1
0
1
1
1
−
2.8116
2.8400
2.8684
V
1
0
1
1
0
−
2.9106
2.9400
2.9694
V
1
0
1
0
1
−
3.0096
3.0400
3.0704
V
1
0
1
0
0
−
3.1086
3.1400
3.1714
V
1
0
0
1
1
−
3.2076
3.2400
3.2724
V
1
0
0
1
0
−
3.3066
3.3400
3.3734
V
1
0
0
0
1
−
3.4056
3.4400
3.4744
V
1
0
0
0
0
−
3.5046
3.5400
3.5754
V
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CS5155H
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < +70°C; 0°C < TJ < +125°C; 8.0 V < VCC1 < 14 V; 5.0 V < VCC2 < 20 V;DAC
Code: VID4 = VID2 = VID1 = VID0 = 1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1.0 nF; COFF = 330 pF; CSS = 0.1 μF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Supply Current
ICC1
No Switching
−
8.5
13.5
mA
ICC2
No Switching
−
1.6
3.0
mA
Operating ICC1
VFB = COMP = VFFB
−
8.0
13
mA
Operating ICC2
VFB = COMP = VFFB
−
2.0
5.0
mA
COFF
Normal Charge Time
VFFB = 1.5 V; VSS = 5.0 V
1.0
1.6
2.2
μs
Extension Charge Time
VSS = VFFB = 0
5.0
8.0
11.0
μs
Discharge Current
COFF to 5.0 V; VFB > 1.0 V
5.0
−
−
mA
Time Out Time
VFB = VCOMP; VFFB = 2.0 V;
Record VGATE(H) Pulse High Duration
10
30
65
μs
Fault Mode Duty Cycle
VFFB = 0V
35
50
70
%
Time Out Timer
PACKAGE PIN DESCRIPTION
PACKAGE PIN #
SO−16
PIN SYMBOL
FUNCTION
1, 2, 3, 4, 6
VID0−VID4
Voltage ID DAC input pins. These pins are internally pulled up to 5.0 V providing logic
ones if left open. VID4 selects the DAC range. When VID4 is High (logic one), the DAC
range is 2.14 V to 3.54 V with 100 mV increments. When VID4 is Low (logic zero), the
DAC range is 1.34 V to 2.09 V with 50 mV increments. VID0 − VID4 select the desired
DAC output voltage. Leaving all 5 DAC input pins open results in a DAC output voltage
of 1.244 V, allowing for adjustable output voltage, using a traditional resistor divider.
5
SS
Soft Start Pin. A capacitor from this pin to LGND in conjunction with internal 60 μA current source provides Soft Start function for the controller. This pin disables fault detect
function during Soft Start. When a fault is detected, the Soft Start capacitor is slowly discharged by internal 2.0 μA current source setting the time out before trying to restart the
IC. Charge/discharge current ratio of 30 sets the duty cycle for the IC when the regulator
output is shorted.
7
COFF
A capacitor from this pin to ground sets the time duration for the on board one shot,
which is used for the constant off time architecture.
8
VFFB
Fast feedback connection to the PWM comparator. This pin is connected to the regulator
output. The inner feedback loop terminates on time.
9
VCC2
Boosted power for the high side gate driver.
10
VGATE(H)
11
PGND
12
VGATE(L)
13
VCC1
Input power for the IC and low side gate driver.
14
LGND
Signal ground for the IC. All control circuits are referenced to this pin.
15
COMP
Error amplifier compensation pin. A capacitor to ground should be provided externally to
compensate the amplifier.
16
VFB
Error amplifier DC feedback input. This is the master voltage feedback which sets the
output voltage. This pin can be connected directly to the output or a remote sense trace.
High FET driver pin capable of 1.5 A peak switching current. Internal circuit prevents
VGATE(H) and VGATE(L) from being in high state simultaneously.
High current ground for the IC. The MOSFET driver is referenced to this pin. Input capacitor ground and the source of lower FET should be tied to this pin.
Low FET driver pin capable of 1.5 A peak switching current.
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CS5155H
VCC1
VCC2
VCC1 Monitor
− Comparator
5.0 V
+
−
3.90 V
3.85V
60 μA
+
SS High
Comparator
FAULT
Q
Q
S
FAULT
Latch
0.7 V
2.0 μA
PGnd
FAULT
VCC1
−
VID0
VID2
R
+
SS
VID1
VGATE(H)
SS Low
Comparator
5 BIT
DAC
VID3
Error
Amplifier
+
−
VGATE(L)
2.5 V
PGnd
PWM
Comparator
VID4
−
VFB
Maximum
On−Time
Timeout
+
Slow Feedback
Fast Feedback
−
+
LGnd
1.0 V
S
Normal
Off−Time
Timeout
Extended
Off−Time
Timeout
COMP
VFFB
R
Q
Q
PMW
Latch
GATE(H) = ON
GATE(H) = OFF
COFF
One Shot
R
Off−Time
Timeout
COFF
Q
S
VFFB Low
Comparator
Time−Out
Timer
(30 μs)
PWM COMP
Edge Triggered
Figure 2. Block Diagram
APPLICATIONS INFORMATION
THEORY OF OPERATION
PWM
Comparator
+
VGATE(H)
C
VGATE(L)
−
V2 Control Method
The V2 method of control uses a ramp signal that is
generated by the ESR of the output capacitors. This ramp is
proportional to the AC current through the main inductor
and is offset by the value of the DC output voltage. This
control scheme inherently compensates for variation in
either line or load conditions, since the ramp signal is
generated from the output voltage itself. This control
scheme inherently compensates for variation in either line or
load conditions, since the ramp signal is generated from the
output voltage itself. This control scheme differs from
traditional techniques such as voltage mode, which
generates an artificial ramp, and current mode, which
generates a ramp from inductor current.
Ramp
Signal
VFFB
Error
Amplifier
COMP
Error
Signal
Output
Voltage
Feedback
VFB
−
E
+
Figure 3. V2 Control Diagram
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Reference
Voltage
CS5155H
The V2 control method is illustrated in Figure 3. The
output voltage is used to generate both the error signal and
the ramp signal. Since the ramp signal is simply the output
voltage, it is affected by any change in the output regardless
of the origin of that change. The ramp signal also contains
the DC portion of the output voltage, which allows the
control circuit to drive the main switch to 0% or 100% duty
cycle as required.
A change in line voltage changes the current ramp in the
inductor, affecting the ramp signal, which causes the V2
control scheme to compensate the duty cycle. Since the
change in inductor current modifies the ramp signal, as in
current mode control, the V2 control scheme has the same
advantages in line transient response.
A change in load current will have an affect on the output
voltage, altering the ramp signal. A load step immediately
changes the state of the comparator output, which controls
the main switch. Load transient response is determined only
by the comparator response time and the transition speed of
the main switch. The reaction time to an output load step has
no relation to the crossover frequency of the error signal
loop, as in traditional control methods.
The error signal loop can have a low crossover frequency,
since transient response is handled by the ramp signal loop.
The main purpose of this ‘slow’ feedback loop is to provide
DC accuracy. Noise immunity is significantly improved,
since the error amplifier bandwidth can be rolled off at a low
frequency. Enhanced noise immunity improves remote
sensing of the output voltage, since the noise associated with
long feedback traces can be effectively filtered.
Line and load regulation are drastically improved because
there are two independent voltage loops. A voltage mode
controller relies on a change in the error signal to
compensate for a deviation in either line or load voltage.
This change in the error signal causes the output voltage to
change corresponding to the gain of the error amplifier,
which is normally specified as line and load regulation. A
current mode controller maintains fixed error signal under
deviation in the line voltage, since the slope of the ramp
signal changes, but still relies on a change in the error signal
for a deviation in load. The V2 method of control maintains
a fixed error signal for both line and load variation, since the
ramp signal is affected by both line and load.
Constant off time provides a number of advantages.
Switch duty cycle can be adjusted from 0 to 100% on a pulse
by pulse basis when responding to transient conditions. Both
0% and 100% duty cycle operation can be maintained for
extended periods of time in response to load or line
transients. PWM slope compensation to avoid
sub−harmonic oscillations at high duty cycles is avoided.
Switch on time is limited by an internal 30 μs timer,
minimizing stress to the power components.
Programmable Output
The CS5155H is designed to provide two methods for
programming the output voltage of the power supply. A five
bit on board digital to analog converter (DAC) is used to
program the output voltage within two different ranges. The
first range is 2.14 V to 3.54 V in 100 mV steps, the second
is 1.34 V to 2.09 V in 50 mV steps, depending on the digital
input code. If all five bits are left open, the CS5155H enters
adjust mode. In adjust mode, the designer can choose any
output voltage by using resistor divider feedback to the VFB
and VFFB pins, as in traditional controllers. The CS5155H
is specifically designed to be backwards compatible with the
CS5150, which uses a four bit DAC code.
Start Up
Until the voltage on the VCC1 supply pin exceeds the 3.9 V
monitor threshold, the Soft Start and gate pins are held low.
The FAULT latch is reset (no Fault condition). The output
of the error amplifier (COMP) is pulled up to 1.0 V by the
comparator clamp. When the VCC1 pin exceeds the monitor
threshold, the GATE(H) output is activated, and the Soft
Start capacitor begins charging. The GATE(H) output will
remain on, enabling the NFET switch, until terminated by
either the PWM comparator, or the maximum on time timer.
If the maximum on time is exceeded before the regulator
output voltage achieves the 1.0 V level, the pulse is
terminated. The GATE(H) pin drives low, and the GATE(L)
pin drives high for the duration of the extended off time. This
time is set by the time out timer and is approximately equal
to the maximum on time, resulting in a 50% duty cycle. The
GATE(L) pin will then drive low, the GATE(H) pin will
drive high, and the cycle repeats.
When regulator output voltage achieves the 1.0 V level
present at the COMP pin, regulation has been achieved and
normal off time will ensue. The PWM comparator
terminates the switch on time, with off time set by the COFF
capacitor. The V2 control loop will adjust switch duty cycle
as required to ensure the regulator output voltage tracks the
output of the error amplifier.
The Soft Start and COMP capacitors will charge to their
final levels, providing a controlled turn on of the regulator
output. Regulator turn on time is determined by the COMP
Constant Off Time
To maximize transient response, the CS5155H uses a
constant off time method to control the rate of output pulses.
During normal operation, the off time of the high side switch
is terminated after a fixed period, set by the COFF capacitor.
To maintain regulation, the V2 control loop varies switch on
time. The PWM comparator monitors the output voltage
ramp, and terminates the switch on time.
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7
CS5155H
capacitor charging to its final value. Its voltage is limited by
the Soft Start COMP clamp and the voltage on the Soft Start
pin (see Figures 4 and 5).
M 10.0 μs
Trace 1− Regulator Output Voltage (5.0 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
M 250 μs
Figure 6. CS5155H Demonstration Board Enable
Startup Waveforms
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 2− Inductor Switching Node (2.0 V/div.)
Trace 3− 12 V Input (VCC1 and VCC2) (5.0 V/div.)
Trace 4− 5.0 V Input (1.0 V/div.)
Normal Operation
Figure 4. CS5155H Demonstration Board Startup in
Response to Increasing 12 V and 5.0 V Input Voltages.
Extended Off Time is Followed by Normal Off Time
Operation when Output Voltage Achieves Regulation to
the Error Amplifier Output.
During normal operation, switch off time is constant and
set by the COFF capacitor. Switch on time is adjusted by the
V2 control loop to maintain regulation. This results in
changes in regulator switching frequency, duty cycle, and
output ripple in response to changes in load and line. Output
voltage ripple will be determined by inductor ripple current
working into the ESR of the output capacitors (see Figures
7 and 8).
M 2.50 ms
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 3− COMP PIn (error amplifier output) (1.0 V/div.)
Trace 4− Soft Start Pin (2.0 V/div.)
Figure 5. CS5155H Demonstration Board Startup
Waveforms
M 1.00 μs
Trace 1− Regulator Output Voltage (10 mV/div.)
If the input voltage rises quickly, or the regulator output
is enabled externally, output voltage will increase to the
level set by the error amplifier output more rapidly, usually
within a couple of cycles (see Figure 6).
Trace 2− Inductor Switching Node (5.0 V/div.)
Figure 7. Peak−to−Peak Ripple on VOUT = 2.8 V,
IOUT = 0.5 A (Light Load)
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CS5155H
level, the output capacitor is pre−positioned −40 mV (see
Figures 9, 10, and 11). For best transient response, a
combination of a number of high frequency and bulk output
capacitors are usually used.
If the maximum on time is exceeded while responding to
a sudden increase in load current, a normal off time occurs
to prevent saturation of the output inductor.
M 1.00 μs
Trace 1− Regulator Output Voltage (10 mV/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Figure 8. Peak−to−Peak Ripple on VOUT = 2.8 V,
IOUT = 13 A (Heavy Load)
Transient Response
The CS5155H V2 control loop’s 100 ns reaction time
provides unprecedented transient response to changes in
input voltage or output current. Pulse by pulse adjustment of
duty cycle is provided to quickly ramp the inductor current
to the required level. Since the inductor current cannot be
changed instantaneously, regulation is maintained by the
output capacitor(s) during the time required to slew the
inductor current.
Overall load transient response is further improved
through a feature called “adaptive voltage positioning”. This
technique pre−positions the output capacitor’s voltage to
reduce total output voltage excursions during changes in
load.
Holding tolerance to 1.0% allows the error amplifier’s
reference voltage to be targeted +40 mV high without
compromising DC accuracy. A “droop resistor”,
implemented through a PC board trace, connects the error
amplifier’s feedback pin (VFB) to the output capacitors and
load and carries the output current. With no load, there is no
DC drop across this resistor, producing an output voltage
tracking the error amplifier’s, including the +40 mV offset.
When the full load current is delivered, an 80 mV drop is
developed across this resistor. This results in output voltage
being offset −40 mV low.
The result of adaptive voltage positioning is that
additional margin is provided for a load transient before
reaching the output voltage specification limits. When load
current suddenly increases from its minimum level, the
output capacitor is pre−positioned +40 mV. Conversely,
when load current suddenly decreases from its maximum
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 2− Regulator Output Voltage (20 V/div.)
Figure 9. CS5155H Demonstration Board Response
to a 0.5 to 13 A Load Pulse (Output Set for 2.8 V)
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Trace 3− Output Current (0.5 to 13 Amps) (20 V/div.)
Figure 10. CS5155H Demonstration Board Response to
13 A Load Turn On (Output Set for 2.8 V). Upon
Completing a Normal Off Time, The V2 Control Loop
Immediately Connects the Inductor to the Input
Voltage, Providing 100% Duty Cycle. Regulation is
Achieved in Less Than 20 ms
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CS5155H
traces than occurs with constant current limit protection (see
Figures 12 and 13).
If the short circuit condition is removed, output voltage
will rise above the 1.0 V level, preventing the FAULT latch
from being set, allowing normal operation to resume.
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Trace 3− Output Current (13 to 0,5 Amps) (20 mV/div.)
Figure 11. CS5155H Demonstration Board Response to
13 A Load Turn Off (Output Set for 2.8 V). V2 Control
Topology Immediately Connects Inductor to Ground,
Providing 0% Duty Cycle. Regulation is Achieved in
Less Than 10 ms
M 25.0 ms
Trace 4− 5.0 V Supply Voltage (2.0 V/div.)
Trace 3− Soft Start Timing Capacitor (1.0 V/div.)
Trace 2− Inductor Switching Node (2.0 V/div.)
Figure 12. CS5155H Demonstration Board Hiccup
Mode Short Circuit Protection. Gate Pulses are
Delivered While the Soft Start Capacitor Charges, and
Cease During Discharge
PROTECTION AND MONITORING FEATURES
VCC1 Monitor
To maintain predictable startup and shutdown
characteristics an internal VCC1 monitor circuit is used to
prevent the part from operating below 3.75 V minimum
startup. The VCC1 monitor comparator provides hysteresis
and guarantees a 3.70 V minimum shutdown threshold.
Short Circuit Protection
A lossless hiccup mode short circuit protection feature is
provided, requiring only the Soft Start capacitor to
implement. If a short circuit condition occurs (VFFB < 1.0 V),
the VFFB low comparator sets the FAULT latch. This causes
the MOSFET to shut off, disconnecting the regulator from
it’s input voltage. The Soft Start capacitor is then slowly
discharged by a 2.0 μA current source until it reaches it’s
lower 0.7 V threshold. The regulator will then attempt to
restart normally, operating in it’s extended off time mode
with a 50% duty cycle, while the Soft Start capacitor is
charged with a 60 μA charge current.
If the short circuit condition persists, the regulator output
will not achieve the 1.0 V low VFFB comparator threshold
before the Soft Start capacitor is charged to it’s upper 2.5 V
threshold. If this happens the cycle will repeat itself until the
short is removed. The Soft Start charge/discharge current
ratio sets the duty cycle for the pulses (2.0 μA/60 μA =
3.3%), while actual duty cycle is half that due to the
extended off time mode (1.65%).
This protection feature results in less stress to the
regulator components, input power supply, and PC board
M 50.0 μs
Trace 4− 5.0 V from PC Power Supply (2.0 V/div.)
Trace 2− Inductor Switching Node (2.0 V/div.)
Figure 13. Startup with Regulator Output Shorted
Overvoltage Protection
Overvoltage protection (OVP) is provided as result of the
normal operation of the V2 control topology and requires no
additional external components. The control loop responds
to an overvoltage condition within 100 ns, causing the top
MOSFET to shut off, disconnecting the regulator from it’s
input voltage. The bottom MOSFET is then activated,
resulting in a “crowbar” action to clamp the output voltage
and prevent damage to the load (see Figures 14 and 15 ). The
regulator will remain in this state until the overvoltage
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CS5155H
5.0 V
condition ceases or the input voltage is pulled low. The
bottom FET and board trace must be properly designed to
implement the OVP function.
MMUN2111T1 (SOT−23)
5 SS
CS5155H
8 V
FFB
IN4148
Shutdown
Input
Figure 16. Implementing Shutdown with the CS5155H
M 10.0 μs
Trace 4− 5.0 V from PC Power Supply (5.0 V/div.)
Trace 1− Regulator Output Voltage (1.0 V/div.)
External Power Good Circuit
Trace 2− Inductor Switching Node 5.0 V/div.)
An optional Power Good signal can be generated through
the use of four additional external components (see Figure
17). The threshold voltage of the Power Good signal can be
adjusted per the following equation:
Figure 14. OVP Response to an Input−to−Output
Short Circuit by Immediately Providing 0% Duty
Cycle, Crow−Barring the Input Voltage to Ground
VPower Good +
(R1 ) R2) 0.65 V
R2
This circuit provides an open collector output that drives
the Power Good output to ground for regulator voltages less
than VPower Good.
5.0 V
R3
10 k
VOUT
CS5155H
M 5.00 ms
R1
10 k
PN3904
Power Good
PN3904
R2
6.2 k
Trace 4− 5.0 V from PC Power Supply (2.0 V/div.)
Trace 1− Regulator Output Voltage (1.0 V/div.)
Figure 17. Implementing Power Good with the CS5155H
Figure 15. OVP Response to an Input−to−Output Short
Circuit by Pulling the Input Voltage to Ground
External Output Enable Circuit
On/off control of the regulator can be implemented
through the addition of two additional discrete components
(see Figure 16). This circuit operates by pulling the Soft
Start pin high, and the VFFB pin low, emulating a short
circuit condition.
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CS5155H
M 2.50 ms
M 1.00 μs
Trace 3 − 12 V Input (VCC1) and (VCC2) (10 V/div.)
Trace 3 = VGATE(H) (10 V/div.)
Trace 4− 5.0 V Input (2.0 V/div.)
Math 1 = VGATE(H) − 5.0 VIN
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 4 = VGATE(L) (10 V/div.)
Trace 2− Power Good Signal (2.0 V/div.)
Trace 2− Inductor Switching Nodes (5.0 V/div.)
Figure 18. CS5155H Demonstration Board During
Power Up. Power Good Signal is Activated when
Output Voltage Reaches 1.70 V.
Figure 19. CS5155H Gate Drive Waveforms Depicting
Rail to Rail Swing
The most important aspect of MOSFET performance is
RDSON, which effects regulator efficiency and MOSFET
thermal management requirements.
The power dissipated by the MOSFETs may be estimated
as follows;
Switching MOSFET:
Selecting External Components
The CS5155H can be used with a wide range of external
power components to optimize the cost and performance of
a particular design. The following information can be used
as general guidelines to assist in their selection.
Power + ILOAD2
NFET Power Transistors
RDSON
duty cycle
Synchronous MOSFET:
Both logic level and standard MOSFETs can be used. The
reference designs derive gate drive from the 12 V supply
which is generally available in most computer systems and
use logic level MOSFETs. A charge pump may be easily
implemented to support 5.0 V or 12 V only systems
(maximum of 20 V). Multiple MOSFETs may be paralleled
to reduce losses and improve efficiency and thermal
management.
Voltage applied to the MOSFET gates depends on the
application circuit used. Both upper and lower gate driver
outputs are specified to drive to within 1.5 V of ground when
in the low state and to within 2.0 V of their respective bias
supplies when in the high state. In practice, the MOSFET
gates will be driven rail to rail due to overshoot caused by the
capacitive load they present to the controller IC. For the
typical application where VCC1 = VCC2 = 12 V and 5.0 V is
used as the source for the regulator output current, the
following gate drive is provided;
Power + ILOAD2
RDSON
(1 * duty cycle)
Duty Cycle =
VOUT ) (ILOAD
ƪ
RDSON OF SYNCH FET)
VIN)(ILOAD RDSON OF SYNCH FET)
* (ILOAD RDSON OF SWITCH FET)
ƫ
Off Time Capacitor (COFF)
The COFF timing capacitor sets the regulator off time:
TOFF + COFF
4848.5
When the VFFB pin is less than 1.0 V, the current charging
the COFF capacitor is reduced. The extended off time can be
calculated as follows:
TOFF + COFF
24, 242.5
Off time will be determined by either the TOFF time, or the
time out timer, whichever is longer.
VGATE(H) + 12 V * 5.0 V + 7.0 V, VGATE(L) + 12 V
(see Figure 19.)
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CS5155H
The preceding equations for duty cycle can also be used
to calculate the regulator switching frequency and select the
COFF timing capacitor:
COFF +
Perioid
regulator output voltage. Key specifications for input
capacitors are their ripple rating, while ESR is important for
output capacitors. For best transient response, a combination
of low value/high frequency and bulk capacitors placed
close to the load will be required.
(1 * duty cycle)
4848.5
where:
Period +
Output Inductor
1
switching frequency
The inductor should be selected based on its inductance,
current capability, and DC resistance. Increasing the
inductor value will decrease output voltage ripple, but
degrade transient response.
Schottky Diode for Synchronous MOSFET
A Schottky diode may be placed in parallel with the
synchronous MOSFET to conduct the inductor current upon
turn off of the switching MOSFET to improve efficiency.
The CS5155H reference circuit does not use this device due
to it’s excellent design. Instead, the body diode of the
synchronous MOSFET is utilized to reduce cost and
conducts the inductor current. For a design operating at
200 kHz or so, the low non−overlap time combined with
Schottky forward recovery time may make the benefits of
this device not worth the additional expense (see Figure 8,
channel 2). The power dissipation in the synchronous
MOSFET due to body diode conduction can be estimated by
the following equation:
Power + VBD
ILOAD
conduction time
THERMAL MANAGEMENT
Thermal Considerations for Power
MOSFETs and Diodes
In order to maintain good reliability, the junction
temperature of the semiconductor components should be
kept to a maximum of 150°C or lower. The thermal
impedance (junction to ambient) required to meet this
requirement can be calculated as follows:
Thermal Impedance +
A heatsink may be added to TO−220 components to
reduce their thermal impedance. A number of PC board
layout techniques such as thermal vias and additional copper
foil area can be used to improve the power handling
capability of surface mount components.
switching frequency
Where VBD = the forward drop of the MOSFET body
diode. For the CS5155H demonstration board as shown in
Figure 8;
Power + 1.6 V
13 A
100 ns
TJUNCTION(MAX) * TAMBIENT
Power
233 kHz + 0.48 W
EMI Management
This is only 1.3% of the 36.4 W being delivered to the
load.
As a consequence of large currents being turned on and off
at high frequency, switching regulators generate noise as a
consequence of their normal operation. When designing for
compliance with EMI/EMC regulations, additional
components may be added to reduce noise emissions. These
components are not required for regulator operation and
experimental results may allow them to be eliminated. The
input filter inductor may not be required because bulk filter
and bypass capacitors, as well as other loads located on the
board will tend to reduce regulator di/dt effects on the circuit
board and input power supply. Placement of the power
component to minimize routing distance will also help to
reduce emissions.
“Droop” Resistor for Adaptive Voltage Positioning
Adaptive voltage positioning is used to reduce output
voltage excursions during abrupt changes in load current.
Regulator output voltage is offset +40 mV when the
regulator is unloaded, and −40 mV at full load. This results
in increased margin before encountering minimum and
maximum transient voltage limits, allowing use of less
capacitance on the regulator output (see Figure 9).
To implement adaptive voltage positioning, a “droop”
resistor must be connected between the output inductor and
output capacitors and load. This is normally implemented by
a PC board trace of the following value:
RDROOP + 80 mV
IMAX
2.0 μH
Adaptive voltage positioning can be disabled for
improved DC regulation by connecting the VFB pin directly
to the load using a separate, non−load current carrying
circuit trace.
33 Ω
1000 pF
Input and Output Capacitors
These components must be selected and placed carefully
to yield optimal results. Capacitors should be chosen to
provide acceptable ripple on the input supply lines and
Figure 20. Filter Components
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13
CS5155H
2.0 μH
carry the full output current. (Typical trace is 1.0 inch
long, 0.17 inch wide). Care should be taken to
minimize any additional losses after the feedback
connection point to maximize regulation.
7. If DC regulation is to be optimized (at the expense of
degraded transient regulation), adaptive voltage
positioning can be disabled by connecting to VFB pin
directly to the load with a separate trace (remote
sense).
8. Place 5.0 V input capacitors close to the switching
MOSFET and synchronous MOSFET.
Route gate drive signals VGATE(H) (pin 10) and
VGATE(L) (pin 12 when used) with a trace that are a
minimum of 0.025 inches wide.
+
1200 pF × 3.0/16 V
Figure 21. Input Filter
Layout Guidelines
1. Place 12 V filter capacitor next to the IC and connect
capacitor ground to pin 11 (PGND).
2. Connect pin 11 (PGND) with a separate trace to the
ground terminals of the 5.0 V input capacitors.
3. Place fast feedback filter capacitor next to pin 8 (VFFB)
and connect it’s ground terminal with a separate, wide
trace directly to pin 14 (LGND).
4. Connect the ground terminals of the Compensation
capacitor directly to the ground of the fast feedback
filter capacitor to prevent common mode noise from
effecting the PWM comparator.
5. Place the output filter capacitor(s) as close to the load
as possible and connect the ground terminal to pin 14
(LGND).
6. To implement adaptive voltage positioning, connect
both slow and fast feedback pins 16 (VFB) and 8
(VFFB) to the regulator output right at the inductor
terminal. Connect inductor to the output capacitors via
a trace with the following resistance:
To the negative terminal
of the input capacitors
VCC
0.1 μF
15
11
1.0 μF
VCOMP
8
5
100 pF
VFFB
SOFT START
RTRACE + 80 mV
IMAX
OFF TIME
This causes the output voltage to be +40 mV with no
load, and −40 mV with a full load, improving regulator
transient response. This trace must be wide enough to
To the negative terminal of the output capacitors
Figure 22. Layout Guidelines
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CS5155H
5.0V
MBRS
120
0.1 μF
MBRS120
1.0 μF
+
1.0 μF
MBRS120
VCC2
VCC1
100 μF/10 V × 3
Tantalum
Si4410DY
VGATE(H)
3.0 μH
3.3 V/10 A
VID0
VID1
VID2
CS5155H
VID4
PGND
COFF
330 pF
SS
0.1 μF
Si9410DY
VGATE(L)
VID3
VFB
COMP
3.3 k
VFFB
LGND
+
0.33 μF
100 pF
100 μF/10 V × 3
Tantalum
Figure 23. Additional Application Diagram, 5.0 V to 3.3 V/10 A Converter
3.3 V
12 V
+
1.0 μF
VCC1
VCC2
5.0 μH
Si9410
VGATE(H)
33 μF/25 V × 3
Tantalum
2.5 V/7.0 A
VID0
VID1
VFB
VID2
VID3
VID4
+
CS5155H
100 μF/10 V × 2
Tantalum
Si9410
VGATE(L)
COFF
330 pF
SS
0.1 μF
PGND
3.3 k
COMP
LGND
0.33 μF
VFFB
100 pF
Figure 24. Additional Application Diagram, 3.3 V to 2.5 V/7.0 A Converter with 12 V Bias
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CS5155H
5.0V
MBRS
120
0.1 μF
MBRS120
1.0 μF
VCC1
+
1.0 μF
MBRS120
VCC2
100 μF/10 V × 3
Tantalum
Remote
Sense
Si4410
VGATE(H)
3.0 μH
3.3 V/10 A
VID0
VID1
VFB
VID2
VID3
10 Ω
100 μF/10 V × 3
+
Tantalum
CS5155H
VID4
Si9410
VGATE(L)
COFF
330 pF
SS
0.1 μF
PGND
COMP
3.3 k
VFFB
LGND
Connect to other
circuits for current
sharing
100 pF
0.33 μF
Figure 25. Additional Application Diagram, 5.0 V to 3.3 V/10 A Converter with Current Sharing
12 V 1N5818
12 V
1N5818
22 Ω
1/4 W
1.0 μF
+
1N4746
18 V 1.0 W
1.0 μF
VCC1
VCC2
VID0
VID2
VID3
VID4
330 pF
FY10AAJ03
CS5155H
3.3 V/5.0 A
FY10AAJ03
+
VGATE(L)
COFF
FY10AAJ03
1200 μF/10 V × 2
Aluminum
Electrolytic
PGND
3.3 k
COMP
LGND
0.33 μF
1.1 μH
VFB
SS
0.1 μF
0.1 μF
VGATE(H)
VID1
820 μF/16 V × 4
Aluminum
Electrolytic
VFFB
100 pF
Figure 26. Additional Application Diagram, 12 V to 3.3 V/5.0 A Converter with Remote Sense
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CS5155H
PACKAGE DIMENSIONS
SO−16
D SUFFIX
CASE 751B−05
ISSUE J
−A−
16
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
9
−B−
1
P
8 PL
0.25 (0.010)
8
M
B
S
G
R
K
DIM
A
B
C
D
F
G
J
K
M
P
R
F
X 45 _
C
−T−
SEATING
PLANE
J
M
D
16 PL
0.25 (0.010)
M
T B
S
A
S
MILLIMETERS
MIN
MAX
9.80
10.00
3.80
4.00
1.35
1.75
0.35
0.49
0.40
1.25
1.27 BSC
0.19
0.25
0.10
0.25
0_
7_
5.80
6.20
0.25
0.50
INCHES
MIN
MAX
0.386
0.393
0.150
0.157
0.054
0.068
0.014
0.019
0.016
0.049
0.050 BSC
0.008
0.009
0.004
0.009
0_
7_
0.229
0.244
0.010
0.019
PACKAGE THERMAL DATA
Parameter
16−SO
Unit
RΘJC
Typical
28
°C/W
RΘJA
Typical
115
°C/W
V2 is a trademark of Switch Power, Inc.
Pentium is a registered trademark of Intel Corporation.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates,
and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death
associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
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For additional information, please contact your local
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CS5155H/D