LINER LTC3838 Dual, fast, accurate step-down dc/dc controller Datasheet

LTC3838
Dual, Fast, Accurate StepDown DC/DC Controller with
Differential Output Sensing
DESCRIPTION
FEATURES
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Wide VIN Range: 4.5V to 38V, VOUT: 0.6V to 5.5V
±0.67% Output Voltage Accuracy Over Temperature,
Differential Output Voltage Sensing, Allowing Up to
±500mV Line Loss at Remote Ground on Channel 1
±1% Output on the Independent 2nd Channel
Controlled On-Time, Valley Current Mode Control
Fast Load Transient Response
Detect Transient (DTR) Reduces VOUT Overshoot
Frequency Programmable from 200kHz to 2MHz,
Synchronizable to External Clock
tON(MIN) = 30ns, tOFF(MIN) = 90ns
RSENSE or Inductor DCR Current Sensing
Overvoltage Protection and Current Limit Foldback
Power Good Output Voltage Monitor
Output Voltage Tracking and Adjustable Soft Start-Up
Thermally Enhanced 38-Pin (5mm × 7mm) QFN and
TSSOP packages
APPLICATIONS
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Distributed Power Systems
Point-of-Load Converters
Computing Systems
Data Communication Systems
The LTC®3838 is a dual, PolyPhase® synchronous step-down
DC/DC switching regulator controller. Two independent
channels drive all N-channel power MOSFETs. The controlled
on-time, valley current mode control architecture allows for
fast transient response and constant frequency switching
in steady-state operation, independent of VIN, VOUT and
load current. Its load-release transient detection feature
significantly reduces overshoot at low output voltages.
Differential output voltage sensing, along with a precision internal reference, offers an accurate ±0.67% output
regulation on Channel 1, even if the remote output ground
deviates from local ground by ±500mV. The second channel
can either provide an independent ±1% output, or together
with the first channel of this controller, serve as one of the
PolyPhase channels for a single-output voltage.
The switching frequency can be programmed from 200kHz
to 2MHz with an external resistor, and can be synchronized to
an external clock. Very low tON and tOFF times allow for near
0% and near 100% duty cycles, respectively. Voltage tracking soft start-up and multiple safety features are provided.
L, LT, LTC, LTM, PolyPhase, OPTI-LOOP, Linear Technology and the Linear logo are registered
trademarks and Hot Swap, No RSENSE and UltraFast are trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners. Protected by U.S.
Patents, including 5481178, 5847554, 6580258, 6304066, 6476589, 6774611.
TYPICAL APPLICATION
1.2V/1.5V, 15A, 350kHz, DCR Sense, Step-Down Converter (Refer to Figure 16 for Full Design)
Efficiency/Power Loss
VIN
4.5V TO 38V
100
SENSE2–
+
+
SENSE1
10k
TG1
TG2
LTC3838
BOOST2
DRVCC1
DRVCC2
0.56μH
0.1μF
0.1μF
w2
BG1
BG2
4.7μF
10k
90
SW2
BOOST1
15k
VOUT2
1.5V
15A
+
10k
330μF
w2
2.0
80
POWER
LOSS
60
VIN = 12V
VOUT = 1.2V
0.1
TRACK/SS1 TRACK/SS2
115k
ITH1
RT
SGND
1.0
0.5
50
40
VFB2
1.5
EFFICIENCY
70
PGND
VOUTSENSE1+
VOUTSENSE1–
2.5
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
POWER LOSS (W)
+
330μF
INTVCC
SW1
0.56μH
VOUT1
1.2V
15A
SENSE2
VIN
EFFICIENCY (%)
SENSE1–
1
LOAD CURRENT (A)
0
10
3838 F16b
ITH2
3838 TA01a
3838fa
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LTC3838
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN Voltage ................................................. –0.3V to 40V
BOOST1, BOOST2 Voltages ....................... –0.3V to 46V
SW1, SW2 Voltages ...................................... –5V to 40V
INTVCC, DRVCC1, DRVCC2, EXTVCC, PGOOD1,
PGOOD2, RUN1, RUN2, (BOOST1-SW1),
(BOOST2-SW2), MODE/PLLIN Voltages ...... –0.3V to 6V
VOUTSENSE1+, VOUTSENSE1–, SENSE1+, SENSE2+,
SENSE1–, SENSE2– Voltages ....................... –0.6V to 6V
TRACK/SS1, TRACK/SS2 Voltages .............. –0.3V to 5V
DTR1, DTR2, PHASMD, RT, VRNG1, VRNG2, VFB2,
ITH1, ITH2 Voltages ................ –0.3V to (INTVCC + 0.3V)
Operating Junction Temperature Range
(Note 2, 3, 4) .......................................... –40°C to 125°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
FE Package ....................................................... 300°C
PIN CONFIGURATION
TOP VIEW
BOOST2
PGOOD2
RUN2
DTR2
SENSE2–
SENSE2+
VFB2
TOP VIEW
38 37 36 35 34 33 32
TRACK/SS2 1
31 TG2
ITH2 2
30 SW2
VRNG2 3
29 BG2
PHASMD 4
28 DRVCC2
27 EXTVCC
MODE/PLLIN 5
CLKOUT 6
26 INTVCC
39
PGND
SGND 7
25 PGND
24 VIN
RT 8
23 DRVCC1
VRNG1 9
DTR2
1
38 RUN2
SENSE2–
2
37 PGOOD2
SENSE2+
3
36 BOOST2
VFB2
4
35 TG2
TRACK/SS2
5
34 SW2
ITH2
6
33 BG2
VRNG2
7
32 DRVCC2
PHASMD
8
31 EXTVCC
MODE/PLLIN
9
CLKOUT 10
SGND 11
RT 12
30 INTVCC
39
PGND
29 PGND
28 VIN
27 DRVCC1
22 BG1
VRNG1 13
26 BG1
TRACK/SS1 11
21 SW1
ITH1 14
25 SW1
VOUTSENSE1+ 12
20 TG1
TRACK/SS1 15
24 TG1
BOOST1
PGOOD1
RUN1
22 PGOOD1
DTR1
23 BOOST1
VOUTSENSE1– 17
SENSE1–
VOUTSENSE1+ 16
SENSE1+
13 14 15 16 17 18 19
VOUTSENSE1–
ITH1 10
UHF PACKAGE
38-LEAD (5mm w 7mm) PLASTIC QFN
TJMAX = 125°C, θJA = 34°C/W
EXPOSED PAD (PIN 39) IS PGND, MUST BE SOLDERED TO PCB
SENSE1+ 18
21 RUN1
SENSE1–
20 DTR1
19
FE PACKAGE
38-LEAD PLASTIC TSSOP
TJMAX = 125°C, θJA = 28°C/W
EXPOSED PAD (PIN 39) IS PGND, MUST BE SOLDERED TO PCB
3838fa
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LTC3838
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3838EUHF#PBF
LTC3838EUHF#TRPBF
3838
38-Lead (5mm × 7mm) Plastic QFN
–40°C to 125°C
LTC3838IUHF#PBF
LTC3838IUHF#TRPBF
3838
38-Lead (5mm × 7mm) Plastic QFN
–40°C to 125°C
LTC3838EFE#PBF
LTC3838EFE#TRPBF
LTC3838FE
38-Lead Plastic TSSOP
–40°C to 125°C
LTC3838IFE#PBF
LTC3838IFE#TRPBF
LTC3838FE
38-Lead Plastic TSSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted (Note 3).
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loops
VIN
Input Voltage Operating Range
4.5
38
V
VOUT1,2(REG)
Regulated Output Voltage Operating Range VOUT1 Regulated Differentially with Respect to
VOUTSENSE1–, VOUT2 Regulated with Respect to
SGND
0.6
5.5
V
IQ
Input DC Supply Current
Both Channels Enabled
Only One Channel Enabled
Shutdown Supply Current
MODE/PLLIN = 0V, No Load
RUN1 or RUN2 (But Not Both) = 0V
RUN1 = RUN2 = 0V
VOUTSENSE1(REG) Regulated Differential Feedback Voltage on ITH1 = 1.2V (Note 5)
Channel 1 (VOUTSENSE1+ – VOUTSENSE1–)
TA = 25°C
TA = 0°C to 85°C
TA = –40°C to 125°C
3
2
15
mA
mA
μA
l
l
0.5985
0.596
0.594
0.6
0.6
0.6
0.6015
0.604
0.606
V
V
V
l
l
0.594
0.591
0.6
0.6
0.606
0.609
V
V
ITH2 = 1.2V (Note 5)
TA = 25°C
TA = 0°C to 85°C
TA = –40°C to 125°C
l
l
0.597
0.594
0.592
0.6
0.6
0.6
0.603
0.606
0.608
V
V
V
Regulated Feedback Voltage on Channel 2
Over Line, Load
VIN = 4.5V to 38V, ITH2 = 0.5V to 1.9V (Note 5)
TA = 0°C to 85°C
TA = –40°C to 125°C
l
l
0.592
0.588
0.6
0.6
0.608
0.612
V
V
IVOUTSENSE1+
VOUTSENSE1+ Input Bias Current
VOUTSENSE1+ – VOUTSENSE1– = 0.6V
±5
±25
nA
IVOUTSENSE1–
VOUTSENSE1– Input Bias Current
VOUTSENSE1+ – VOUTSENSE1– = 0.6V
–25
–50
μA
IVFB2
VFB2 Input Bias Current
VFB2 = 0.6V
–5
±50
nA
gm(EA)1,2
Error Amplifier Transconductance
ITH = 1.2V (Note 5)
1.7
mS
tON(MIN)1,2
Minimum Top Gate On-Time
VIN = 38V, VOUT = 0.6V, RT = 20k (Note 6)
30
ns
tOFF(MIN)1,2
Minimum Top Gate Off-Time
(Note 6)
90
ns
Regulated Differential Feedback Voltage on VIN = 4.5V to 38V, ITH1 = 0.5V to 1.9V (Note 5),
Channel 1 Over Line, Load and Common
–0.5V < VOUTSENSE1– < 0.5V
Mode
TA = 0°C to 85°C
TA = –40°C to 125°C
VFB2(REG)
Regulated Feedback Voltage on Channel 2
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LTC3838
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted (Note 3).
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
VSENSE(MAX)1,2
Maximum Valley Current Sense Threshold
(VSENSE1,2+ – VSENSE1,2–)
VRNG = 2V, VFB = 0.57V, VSENSE– = 2.5V
VRNG = 0V, VFB = 0.57V, VSENSE– = 2.5V
VRNG = INTVCC, VFB = 0.57V, VSENSE– = 2.5V
80
21
39
100
30
50
120
40
61
mV
mV
mV
VSENSE(MIN)1,2
Minimum Valley Current Sense Threshold
(VSENSE1,2+ – VSENSE1,2–)
(Forced Continuous Mode)
VRNG = 2V, VFB = 0.63V, VSENSE– = 2.5V
VRNG = 0V, VFB = 0.63V, VSENSE– = 2.5V
VRNG = INTVCC, VFB = 0.63V, VSENSE– = 2.5V
–50
–15
–25
ISENSE1,2+
SENSE1,2+ Pins Input Bias Current
VSENSE+ = 0.6V
VSENSE+ = 5V
±5
1
ISENSE1,2–
SENSE1,2– Pins Input Bias Current
(Internal 500k Resistor to SGND)
VSENSE– = 0.6V
VSENSE– = 5V
1.2
10
Current Sensing
l
l
l
mV
mV
mV
±50
±2
nA
μA
μA
μA
Start-Up and Shutdown
VRUN1,2
IRUN1,2
UVLO
ITRACK/SS1,2
l
RUN Pin On Threshold
VRUN1,2 Rising
RUN Pin On Hysteresis
VRUN1,2 Falling from On Threshold
100
RUN Pin Pull-Up Current when Off
RUN1,2 = SGND
1.2
μA
RUN Pin Pull-Up Current Hysteresis
IRUN1,2(HYS) = IRUN1,2(ON) – IRUN1,2(OFF)
5
μA
INTVCC Undervoltage Lockout
INTVCC Falling
INTVCC Rising
Soft-Start Pull-Up Current
l
l
1.1
3.3
0V < TRACK/SS1,2 < 0.6V
1.2
3.7
4.2
1.3
V
mV
4.5
1
V
V
μA
Frequency and Clock Synchronization
Clock Output Frequency
(Steady-State Switching Frequency)
RT = 205k
RT = 80.6k
RT = 18.2k
Channel 2 Phase (Relative to Channel 1)
PHASMD = SGND
PHASMD = Floating
PHASMD = INTVCC
180
180
240
Deg
Deg
Deg
CLKOUT Phase (Relative to Channel 1)
PHASMD = SGND
PHASMD = Floating
PHASMD = INTVCC
60
90
120
Deg
Deg
Deg
450
200
500
2000
550
kHz
kHz
kHz
VPLLIN(H)
Clock Input High Level Into MODE/PLLIN
VPLLIN(L)
Clock Input Low Level Into MODE/PLLIN
RMODE/PLLIN
MODE/PLLIN Input DC Resistance
With Respect to SGND
600
kΩ
RTG(UP)1,2
TG Driver Pull-Up On Resistance
TG High
2.5
Ω
RTG(DOWN)1,2
TG Driver Pull-Down On Resistance
TG Low
1.2
Ω
RBG(UP)1,2
BG Driver Pull-Up On Resistance
BG High
2.5
Ω
RBG(DOWN)1,2
BG Driver Pull-Down On Resistance
BG Low
0.8
Ω
tD(TG/BG)1,2
Top Gate Off to Bottom Gate On Delay Time (Note 6)
20
ns
tD(BG/TG)1,2
Bottom Gate Off to Top Gate On Delay Time (Note 6)
15
ns
2
V
0.5
V
Gate Drivers
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LTC3838
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted (Note 3).
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Internally Regulated DRVCC1 Voltage
6V < VIN < 38V
5.0
5.3
5.6
V
DRVCC1 Load Regulation
IDRVCC1 = 0mA to –100mA
–1.5
–3
%
EXTVCC Switchover Voltage
EXTVCC Rising
4.6
4.8
Internal VCC Regulator
VDRVCC1
VEXTVCC
4.4
EXTVCC Switchover Hysteresis
V
200
mV
200
mV
EXTVCC to DRVCC2 Voltage Drop
VEXTVCC = 5V, IDRVCC2 = –100mA
OV
PGOOD Overvoltage Threshold
VFB1,2 Rising, with Respect to Regulated Voltage
5
7.5
10
%
UV
PGOOD Undervoltage Threshold
VFB1,2 Falling, with Respect to Regulated Voltage
–5
–7.5
–10
%
PGOOD Threshold Hysteresis
VFB1,2 Returning to Reference Voltage
VPGOOD(L)1,2
PGOOD Low Voltage
IPGOOD = 2mA
tD(PGOOD)1,2
Delay from VFB Fault (OV/UV) to PGOOD
Falling
Delay from VFB Good to PGOOD Rising
PGood Output
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The junction temperature (TJ, in °C) is calculated from the ambient
temperature (TA, in °C) and power dissipation (PD, in Watts) according to
the formula:
TJ = TA + (PD • θJA)
where θJA (in °C/W) is the package thermal impedance.
Note 3: The LTC3838 is tested under pulsed loading conditions such that
TJ ≈ TA. The LTC3838E is guaranteed to meet specifications over the 0°C
to 85°C operating junction temperature range. Specifications over the
–40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
The LTC3838I is guaranteed to meet specifications over the –40°C to
125°C operating junction temperature range . Note that the maximum
ambient temperature consistent with these specifications is determined by
specific operating conditions in conjunction with board layout, the rated
package thermal impedance and other environmental factors.
2
0.1
%
0.3
V
50
μs
20
μs
Note 4: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. The maximum
rated junction temperature will be exceeded when this protection is active.
Continuous operation above the specified absolute maximum operating
junction temperature may impair device reliability or permanently damage
the device.
Note 5: The LTC3838 is tested in a feedback loop that adjusts
(VOUTSENSE1+ – VOUTSENSE1–) or VFB2 to achieve specified error amplifier
output voltages (ITH1,2).
Note 6: Delay times are measured with top gate (TG) and bottom gate
(BG) driving minimum load, and using 50% levels.
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LTC3838
TYPICAL PERFORMANCE CHARACTERISTICS
Transient Response
(Forced Continuous Mode)
Load Step
(Forced Continuous Mode)
Load Release
(Forced Continuous Mode)
ILOAD
10A/DIV
ILOAD
10A/DIV
ILOAD
10A/DIV
VOUT
50mV/DIV
AC-COUPLED
VOUT
50mV/DIV
AC-COUPLED
VOUT
50mV/DIV
AC-COUPLED
IL
10A/DIV
IL
10A/DIV
IL
10A/DIV
3838 G01
50μs/DIV
LOAD TRANSIENT = 0A TO 15A TO 0A
VIN = 12V
VOUT = 1.2V
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
3838 G02
5μs/DIV
LOAD STEP = 0A TO 15A
VIN = 12V
VOUT = 1.2V
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
3838 G03
5μs/DIV
LOAD RELEASE = 15A TO 0A
VIN = 12V
VOUT = 1.2V
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
Transient Response
(Discontinuous Mode)
Load Step
(Discontinuous Mode)
Load Release
(Discontinuous Mode)
ILOAD
10A/DIV
ILOAD
10A/DIV
VOUT
50mV/DIV
AC-COUPLED
VOUT
50mV/DIV
AC-COUPLED
IL
10A/DIV
IL
10A/DIV
3838 G04
50μs/DIV
LOAD TRANSIENT = 500mA TO 15A TO 500mA
VIN = 12V
VOUT = 1.2V
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
ILOAD
10A/DIV
VOUT
50mV/DIV
AC-COUPLED
IL
10A/DIV
3838 G05
5μs/DIV
LOAD STEP = 500mA TO 15A
VIN = 12V
VOUT = 1.2V
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
Load Release with Detect Transient (DTR)
Feature Enabled
3838 G06
5μs/DIV
LOAD RELEASE = 15A TO 500mA
VIN = 12V
VOUT = 1.2V
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
Load Release with Detect Transient (DTR)
Feature Disabled
SW
3V/DIV
SW
3V/DIV
VOUT
50mV/DIV
AC-COUPLED
ITH
1V/DIV
VOUT
50mV/DIV
AC-COUPLED
IL
10A/DIV
IL
10A/DIV
ITH
1V/DIV
5μs/DIV
LOAD RELEASE = 15A TO 5A
VIN = 5V
VOUT = 0.6V
3838 G07
FIGURE 17 CIRCUIT, CHANNEL 1 MODIFIED:
RFB2 = 0Ω, VRNG2 = SGND, CITH1 = 120pF, CITH2 = 0pF,
FROM DTR1 PIN: RITH1 = 46.4k TO SGND, RITH2 = 42.2k TO INTVCC
5μs/DIV
3838 G08
LOAD RELEASE = 15A TO 5A
VIN = 5V
VOUT = 0.6V
FIGURE 17 CIRCUIT, CHANNEL 1 MODIFIED:
RFB2 = 0Ω, VRNG2 = SGND, CITH1 = 120pF, CITH2 = 0pF,
RITH1/2 = 46.4k TO SGND//42.2k TO INTVCC,
CONNECTION FROM RITH1/2 AND CITH1 TO DTR1 PIN REMOVED.
DTR1 PIN TIED TO INTVCC
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LTC3838
TYPICAL PERFORMANCE CHARACTERISTICS
Soft Start-Up Into
Pre-Biased Output
Regular Soft Start-Up
RUN1
5V/DIV
RUN1
5V/DIV
TRACK/SS1
200mV/DIV
VOUT
500mV/DIV
TRACK/SS1
200mV/DIV
3838 G09
CSS = 10nF
1ms/DIV
VIN = 12V
VOUT = 1.2V
FORCED CONTINUOUS MODE
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
3838 G10
CSS = 10nF
1ms/DIV
VIN = 12V
VOUT = 1.2V
VOUT PRE-BIASED TO 0.75V
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
Overcurrent Protection
Overvoltage Protection
SHORTCIRCUIT
TRIGGER
IL
10A/DIV
VOUT
1V/DIV
FULL CURRENT LIMIT
WHEN VOUT HIGHER
THAN HALF OF REGULATED
CURRENT LIMIT STARTS TO FOLD BACK AS
VOUT DROPS BELOW HALF OF REGULATED
IL
10A/DIV
VOUT
100mV/DIV
AC-COUPLED
3838 G12
VIN = 12V
5ms/DIV
VOUT = 1.2V
FORCED CONTINUOUS MODE
CURRENT LIMIT = 17A
OVERLOAD = 7.5A TO 17.5A
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
OVERVOLTAGE CREATED BY APPLYING
A CHARGED CAPACITOR TO VOUT
VOUT
100mV/DIV
AC-COUPLED
COUT
RECHARGE
3838 G13
VIN = 12V
500μs/DIV
VOUT = 1.2V
ILOAD = 0A
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
Phase Relationship:
PHASMD = Ground
BG1
5V/DIV
3838 G14
VIN = 12V
20μs/DIV
VOUT = 1.2V
BG STAYS ON UNTIL
FORCED CONTINUOUS VOUT IS PULLED
BELOW OVERVOLTAGE
MODE
THRESHOLD
ILOAD = 0A
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
Phase Relationship:
PHASMD = Float
PLLIN
5V/DIV
Phase Relationship:
PHASMD = INTVCC
PLLIN
5V/DIV
SW1
10V/DIV
0°
SW2
10V/DIV
3838 G11
VIN = 12V
10ms/DIV
VOUT = 1.2V
FORCED CONTINUOUS MODE
FIGURE 17 CIRCUIT, CHANNEL 1, VRNG1 = SGND
Short-Circuit Protection
IL
5A/DIV
CLKOUT
5V/DIV
TRACK/SS1
200mV/DIV
VOUT
500mV/DIV
VOUT
500mV/DIV
SW1
10V/DIV
Output Tracking
PLLIN
5V/DIV
SW1
10V/DIV
0°
SW2
10V/DIV
180°
CLKOUT
5V/DIV
60°
3838 G15
500ns/DIV
FIGURE 19 CIRCUIT
VIN = 12V
VOUT1 = 5V, VOUT2 = 3.3V
LOAD = 0A
MODE/PLLIN = 333kHz EXTERNAL CLOCK
0°
SW2
10V/DIV
180°
CLKOUT
5V/DIV
90°
3838 G15
500ns/DIV
FIGURE 19 CIRCUIT
VIN = 12V
VOUT1 = 5V, VOUT2 = 3.3V
LOAD = 0A
MODE/PLLIN = 333kHz EXTERNAL CLOCK
240°
120°
3838 G17
500ns/DIV
FIGURE 19 CIRCUIT
VIN = 12V
VOUT1 = 5V, VOUT2 = 3.3V
LOAD = 0A
MODE/PLLIN = 333kHz EXTERNAL CLOCK
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LTC3838
TYPICAL PERFORMANCE CHARACTERISTICS
Output Regulation
vs Input Voltage
Output Regulation
vs Load Current
0.2
0.2
0
–0.1
5
10
15
20 25
VIN (V)
30
35
0
–0.2
40
2
0
6
4
ILOAD (A)
8
0.2
0
–0.4
CHANNEL 1
CHANNEL 2
–0.6
–50 –25
10
Error Amplifier Transconductance
vs Temperature
1.80
CLKOUT/Switching Frequency
vs Temperature
2
2
1
1
1.60
NORMALIZED Δf (%)
NORMALIZED Δf (%)
1.75
1.65
0
–1
0
–2
25 50 75 100 125 150
TEMPERATURE (°C)
0
5
10
20 25
VIN (V)
15
30
3838 G27
–2
–50 –25
40
100
90
tOFF(MIN)
80
90
tOFF(MIN)
80
70
60
30
50
TIME (ns)
70
60
TIME (ns)
70
tON(MIN)
tON(MIN)
40
30
20
10
0
0
1
2
3
4
0
5
6
3838 G24
tON(MIN)
20
VOUT = 0.6V
RT ADJUSTED FOR fCLKOUT = 2MHz
10
VSENSE– (V)
50
40
30
20
VIN = 38V
RT ADJUSTED FOR fCLKOUT = 2MHz
tOFF(MIN)
80
60
50
25 50 75 100 125 150
TEMPERATURE (°C)
tON(MIN) and tOFF(MIN)
vs Switching Frequency
100
40
0
3838 G23
tON(MIN) and tOFF(MIN)
vs Voltage on VIN Pin
100
90
0
3838 G21
tON(MIN) and tOFF(MIN)
vs VOUT (Voltage on SENSE– Pin)
TIME (ns)
35
VIN = 15V, VOUT = 0.6V
ILOAD = 0A
f = 500kHz
FREQUENCY NORMALIZED AT TA = 25°C
–1
VOUT = 0.6V
ILOAD = 5A
f = 500kHz
FREQUENCY NORMALIZED AT VIN = 15V
1.55
1.50
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3838 G20
CLKOUT/Switching Frequency
vs Input Voltage
1.70
0
3838 G19
3838 G18
TRANSCONDUCTANCE (mS)
VIN = 15V
VOUT = 0.6V
0.4 ILOAD = 0A
VOUT NORMALIZED AT TA = 25°C
–0.2
–0.1
CHANNEL 1
CHANNEL 2
CHANNEL 1
CHANNEL 2
0
0.1
NORMALIZED ΔVOUT (%)
NORMALIZED ΔVOUT (%)
0.1
–0.2
0.6
VIN = 15V
VOUT = 0.6V
VOUT NORMALIZED AT ILOAD = 4A
VOUT = 0.6V
ILOAD = 5A
VOUT NORMALIZED AT VIN = 15V
NORMALIZED ΔVOUT (%)
Output Regulation
vs Temperature
0
5
10
15
20 25
VIN (V)
30
10
35
40
3838 G25
VIN = 38V
VOUT = 0.6V
0
200
500
800 1100 1400 1700 2000
CLKOUT/SWITCHING FREQUENCY (kHz)
3838 G26
3838fa
8
LTC3838
TYPICAL PERFORMANCE CHARACTERISTICS
FORCED CONTINUOUS MODE
CURRENT SENSE VOLTAGE (mV)
100
80
60
40
20
0
–20
VRNG = 2V
VRNG = 1V
VRNG = 0.6V
–40
–60
0
0.8
1.2
1.6
ITH VOLTAGE (V)
0.4
2
120
VRNG = 2V
100
80
60
VRNG = 1V
40
VRNG = 0.6V
20
0
–50 –25
2.4
0
25 50 75 100 125 150
TEMPERATURE (°C)
RUN Pin Thresholds
vs Temperature
0.8
0.6
3
RUN PIN BELOW 1.2V
SWITCHING THRESHOLD
1.05
1.00
0.95
0.85
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
0.80
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
25 50 75 100 125 150
TEMPERATURE (°C)
3838 G32
3838 G31
INTVCC Undervoltage Lockout
Thresholds vs Temperature
Shutdown Current Into VIN Pin
vs Voltage on VIN Pin
Quiescent Current Into VIN Pin
vs Temperature
3.5
40
4.5
UVLO RELEASE
(INTVCC RISING)
5.5
0.90
3838 G30
QUIESCENT CURRENT (mA)
35
30
4.1
CURRENT (μA)
UVLO THRESHOLDS (V)
4.5
1.5
2.5
3.5
SENSE PIN VOLTAGE (V)
1.10
RUN PIN ABOVE 1.2V
SWITCHING THRESHOLD
4
1
3.9
UVLO LOCK
(INTVCC FALLING)
25
20
15
10
3.5
130°C
25°C
–45°C
5
0
0.5
TRACK/SS Pull-Up Currents
vs Temperature
5
0.2
3.3
–50 –25
0
–0.5
1.15
2
3.7
VRNG = 0.6V
20
7
0.4
4.3
VRNG = 1V
40
1.20
SHUTDOWN REGION
0
60
8
6
CURRENT (μA)
RUN PIN THRESHOLDS (V)
1.2
0
–50 –25
80
3838 G22
CURRENT (μA)
SWITCHING REGION
STAND-BY REGION
VRNG = 2V
100
RUN Pull-Up Currents
vs Temperature
1.6
1.0
120
3838 G29
3838 G28
1.4
Maximum Current Sense Voltage
vs Voltage on SENSE– Pin
MAXIMUM CURRENT SENSE– VOLTAGE (mV)
120
Maximum Current Sense Voltage
vs Temperature
MAXIMUM CURRENT SENSE VOLTAGE (mV)
Current Sense Voltage
vs ITH Voltage
25 50 75 100 125 150
TEMPERATURE (°C)
3838 G33
0
0
5
10
15
20
25
30
35
40
VIN (V)
3838 G34
3.0
BOTH CHANNELS ON
2.5
CHANNEL 1 ON ONLY
2.0
CHANNEL 2 ON ONLY
1.5
1.0
–50 –25
0
25 50
75 100 125 150
TEMPERATURE (°C)
3838 G35
3838fa
9
LTC3838
PIN FUNCTIONS
(QFN/TSSOP)
PHASMD (Pin 4/Pin 8): Phase Selector Input. This pin
determines the relative phases of channels and the
CLKOUT signal. With zero phase being defined as the
rising edge of TG1: Pulling this pin to SGND locks TG2 to
180°, and CLKOUT to 60°. Connecting this pin to INTVCC
locks TG2 to 240° and CLKOUT to 120°. Floating this pin
locks TG2 to 180° and CLKOUT to 90°.
MODE/PLLIN (Pin 5/Pin 9): Operation Mode Selection
or External Clock Synchronization Input. When this pin
is tied to INTVCC, forced continuous mode operation is
selected. Tying this pin to SGND allows discontinuous
mode operation. When an external clock is applied at this
pin, both channels operate in forced continuous mode and
synchronize to the external clock.
CLKOUT (Pin 6/Pin 10): Clock Output of Internal Clock
Generator. Its output level swings between INTVCC and
SGND. If clock input is present at the MODE/PLLIN pin, it
will be synchronized to the input clock, with phase set by
the PHASMD pin. If no clock is present at MODE/PLLIN, its
frequency will be set by the RT pin. To synchronize other
controllers, it can be connected to their MODE/PLLIN pins.
SGND (Pin 7/Pin 11): Signal Ground. All small-signal analog
and compensation components should be connected to
this ground. Connect SGND to the exposed pad and PGND
pin using a single PCB trace.
RT (Pin 8/Pin 12): Clock Generator Frequency Programming Pin. Connect an external resistor from RT to SGND
to program the switching frequency between 200kHz and
2MHz. An external clock applied to MODE/PLLIN should
be within ±30% of this programmed frequency to ensure
frequency lock. When the RT pin is floating, the frequency
is internally set to be slightly under 200kHz.
VRNG1, VRNG2 (Pins 9, 3/Pins 13, 7): Current Sense Voltage Range Inputs. When programmed between 0.6V and
2V, the voltage applied to VRNG1,2 is twenty times (20×)
the maximum sense voltage between SENSE1,2+ and
SENSE1,2–, i.e., for either channel, (VSENSE+ – VSENSE–) =
0.05 • VRNG. If a VRNG is tied to SGND, the channel operates with a maximum sense voltage of 30mV, equivalent
to a VRNG of 0.6V; If tied to INTVCC, a maximum sense
voltage of 50mV, equivalent to a VRNG of 1V.
ITH1, ITH2 (Pins 10, 2/Pins 14, 6): Current Control
Threshold. This pin is the output of the error amplifier and
the switching regulator’s compensation point. The current
comparator threshold increases with this control voltage.
The voltage ranges from 0V to 2.4V, with 0.8V corresponding to zero sense voltage (zero inductor valley current).
TRACK/SS1, TRACK/SS2 (Pins 11, 1/Pins 15, 5): External
Tracking and Soft-Start Input. The LTC3838 regulates the
feedback voltages (VOUTSENSE1+ – VOUTSENSE1–) and VFB2
to the smaller of 0.6V or the voltage on the TRACK/SS1,2
pins respectively. An internal 1μA temperature-independent
pull-up current source is connected to each TRACK/SS
pin. A capacitor to ground at this pin sets the ramp time
to the final regulated output voltage. Alternatively, another
voltage supply connected to this pin allows the output to
track the other supply during start-up.
VOUTSENSE1+ (Pin 12/Pin 16): Differential Output Sense
Amplifier (+) Input of Channel 1. Connect this pin to a
feedback resistor divider between the positive and negative
output capacitor terminals of VOUT1. In nominal operation
the LTC3838 will attempt to regulate the differential output
voltage VOUT1 to 0.6V divided by the feedback resistor
divider ratio.
3838fa
10
LTC3838
PIN FUNCTIONS
(QFN/TSSOP)
VOUTSENSE1– (Pin 13/Pin 17): Differential Output Sense
Amplifier (–) Input of Channel 1. Connect this pin to the
negative terminal of the output load capacitor of VOUT1.
SENSE1+, SENSE2+ (Pins 14, 37/Pins 18, 3): Differential
Current Sense Comparator (+) Inputs. The ITH pin voltage
and controlled offsets between the SENSE+ and SENSE–
pins set the current trip threshold. The comparator can
be used for RSENSE sensing or inductor DCR sensing. For
RSENSE sensing, Kelvin (4-wire) connect the SENSE+ pin
to the (+) terminal of RSENSE. For DCR sensing, tie the
SENSE+ pins to the connection between the DCR sense
capacitor and sense resistor tied across the inductor.
SENSE1–, SENSE2– (Pins 15, 36/Pins 19, 2): Differential
Current Sense Comparator (–) Input. The comparator can
be used for RSENSE sensing or inductor DCR sensing.
For RSENSE sensing, Kelvin (4-wire) connect the SENSE–
pin to the (–) terminal of RSENSE. For DCR sensing, tie
the SENSE– pin to the DCR sense capacitor tied to the
inductor VOUT node connection. These pins also function as output voltage sense pins for the top MOSFET
on-time adjustment. The impedance looking into these
pins is different from the SENSE+ pins because there
is an additional 500k internal resistor from each of the
SENSE– pins to SGND.
DTR1, DTR2 (Pins 16, 35/Pins 20, 1): Detect LoadRelease Transient for Overshoot Reduction. When load
current suddenly drops, if voltage on this DTR pin drops
below half of INTVCC, the bottom gate (BG) could turn
off, allowing the inductor current to drop to zero faster,
thus reducing the VOUT overshoot. (Refer to Load-Release
Transient Detection in the Applications Information section
for more details.) An internal 2.5μA current source pulls
this pin toward INTVCC. To disable the DTR feature, simply
tie the DTR pin to INTVCC.
RUN1, RUN2 (Pins 17, 34/Pins 21, 38): Run Control
Inputs. An internal proportional-to-absolute-temperature
(PTAT) pull-up current source (~1.2μA at 25°C) is constantly
connected to this pin. Taking both RUN1 and RUN2 pins
below a threshold voltage (~0.8V at 25°C) shuts down all
bias of INTVCC and DRVCC and places the LTC3838 into
micropower shutdown mode. Allowing either RUN pin to
rise above this threshold would turn on the internal bias
supply and the circuitry for the particular channel. When
a RUN pin rises above 1.2V, its corresponding channel’s
TG and BG drivers are turned on and an additional 5μA
temperature-independent pull-up current is connected
internally to the RUN pin. Either RUN pin can sink up to
50μA, or be forced no higher than 6V.
PGOOD1, PGOOD2 (Pins 18, 33/Pins 22, 37): Power Good
Indicator Outputs. This open-drain logic output is pulled
to ground when the output voltage goes out of a ±7.5%
window around the regulation point, after a 50μs powerbad-masking delay. Returning to the regulation point, there
is a much shorter delay to power good, and a hysteresis
of around 2% on both sides of the voltage window.
BOOST1, BOOST2 (Pins 19, 32/Pins 23, 36): Boosted
Floating Supplies for Top MOSFET Drivers. The (+) terminal
of the bootstrap capacitor, CB, connects to this pin. The
BOOST pins swing by a VIN between a diode drop below
DRVCC, or (DRVCC – VD) and (VIN + DRVCC – VD).
TG1, TG2 (Pins 20, 31/Pins 24, 35): Top Gate Driver
Outputs. The TG pins drive the gates of the top N-channel
power MOSFET with a voltage swing of VDRVCC between
SW and BOOST.
3838fa
11
LTC3838
PIN FUNCTIONS
(QFN/TSSOP)
SW1, SW2 (Pins 21, 30/Pins 25, 34): Switch Node Connection to Inductors. Voltage swings are from a diode
voltage below ground to VIN. The (–) terminal of the
bootstrap capacitor, CB, connects to this node.
INTVCC (Pin 26/Pin 30): Supply Input for Internal Circuitry
(Not Including Gate Drivers). Normally powered from the
DRVCC pins through a decoupling RC filter to SGND (typically 2Ω and 1μF).
BG1, BG2 (Pins 22, 29/Pins 26, 33): Bottom Gate
Driver Outputs. The BG pins drive the gates of the bottom
N-channel power MOSFET between PGND and DRVCC.
EXTVCC (Pin 27/Pin 31): External Power Input. When
EXTVCC exceeds the switchover voltage (typically 4.6V),
an internal switch connects this pin to DRVCC2 and shuts
down the internal regulator so that INTVCC and gate drivers draw power from EXTVCC. The VIN pin still needs to
be powered up but draws minimum current.
DRVCC1, DRVCC2 (Pins 23, 28/Pins 27, 32): Supplies of
Bottom Gate Drivers. DRVCC1 is also the output of an internal
5.3V regulator. DRVCC2 is also the output of the EXTVCC
switch. Normally the two DRVCC pins are shorted together
on the PCB, and decoupled to PGND with a minimum of
4.7μF ceramic capacitor, CDRVCC.
VIN (Pin 24/Pin 28): Input Voltage Supply. The supply
voltage can range from 4.5V to 38V. For increased noise
immunity decouple this pin to SGND with an RC filter.
Voltage at this pin is also used to adjust top gate on-time,
therefore it is recommended to tie this pin to the main
power input supply through an RC filter.
PGND (Pin 25, Exposed Pad Pin 39/Pin 29, Exposed
Pad Pin 39): Power Ground. Connect this pin as close
as practical to the source of the bottom N-channel power
MOSFET, the (–) terminal of CDRVCC and the (–) terminal
of CIN. Connect the exposed pad and PGND pin to SGND
pin using a single PCB trace under the IC. The exposed
pad must be soldered to the circuit board for electrical
and rated thermal performance.
VFB2 (Pin 38/Pin 4): Error Amplifier Feedback Input for
Channel 2. This pin connects the error amplifier to an
external feedback resistor divider from VOUT2. In nominal
operation the LTC3838 will attempt to regulate the VOUT2
to 0.6V divided by the feedback resistor divider ratio.
Shorting this pin to INTVCC will disable Channel 2’s error amplifier, and internally connect ITH2 to ITH1. (As a
result, TRACK/SS2 is no longer functional and PGOOD2
is always pulling low.) By doing so, this part can function
as a dual phase, single VOUT step-down controller, and
the two channels use a single Channel 1’s error amplifier
for the ITH output and compensation.
3838fa
12
LTC3838
FUNCTIONAL DIAGRAM
VIN
VIN
IN
EN LDO
OUT SD
1-2μA
PTAT
5μA
+
UVLO
4.2V
+
1.2V
BOOST
TG
DRV
–
RUN
MT
L
SW
EN_DRV
–
CB
DB
TG
RSENSE
VOUT
DRVCC
~0.8V
EXTVCC
+
+
–
–
LOGIC
CONTROL
SENSE–
VIN
250k
~4.6V
CINTVCC
COUT
DRVCC2
DRVCC1
START
ONE-SHOT
TIMER
INTVCC
CDRVCC
STOP
RFB2
BG DRV
250k
BG
FORCED
CONTINUOUS
MODE
ON-TIME
ADJUST
MODE/PLLIN
IREV
ICMP
+
+
–
–
CLK1
CLOCK PLL/
GENERATOR
RT
RFB1
PGND
PHASE
DETECTOR
MODE/CLK
DETECT
RT
MB
CLK2
SENSE+
SENSE–
TO CHANNEL 2
CLKOUT
gm
INTVCC
1μA
gm
EA
+
RPGD
+
+
–
0.555V
UV
PGOOD
TRACK/SS
0.6V
CSS
–
+
DIFFAMP
(A = 1)
DELAY
–
+
VOUTSENSE1+
VOUTSENSE1–
3838 FD
OV
VFB2
–
0.645V
CHANNEL 2 ONLY
SGND
INTVCC
1/2 INTVCC
+
LOAD
RELEASE
DETECTION
TO LOGIC
CONTROL
–
DUPLICATE DASHED
LINE BOX FOR
CHANNEL 2
VRNG
ITH
INTVCC
DTR
INTVCC
CITH1
CITH2
RITH2
RITH1
3838fa
13
LTC3838
OPERATION (Refer to Functional Diagram)
Main Control Loop
The LTC3838 is a controlled on-time, valley current mode
step-down DC/DC dual controller with two channels
operating out of phase. Each channel drives both main
and synchronous N-channel MOSFETs. The two channels
can be either configured to two independently regulated
outputs, or combined into a single output.
The top MOSFET is turned on for a time interval determined
by a one-shot timer. The duration of the one-shot timer is
controlled to maintain a fixed switching frequency. As the
top MOSFET is turned off, the bottom MOSFET is turned
on after a small delay. The delay, or dead time, is to avoid
both top and bottom MOSFETs being on at the same time,
causing shoot-through current from VIN directly to power
ground. The next switching cycle is initiated when the current comparator, ICMP, senses that inductor current falls
below the trip level set by voltages at the ITH and VRNG
pins. The bottom MOSFET is turned off immediately and
the top MOSFET on again, restarting the one-shot timer
and repeating the cycle. In order to avoid shoot-through
current, there is also a small dead-time delay before the
top MOSFET turns on. At this moment, the inductor current hits its “valley” and starts to rise again.
Inductor current is determined by sensing the voltage
between SENSE+ and SENSE–, either by using an explicit
resistor connected in series with the inductor or by implicitly sensing the inductor’s DC resistive (DCR) voltage drop
through an RC filter connected across the inductor. The
trip level of the current comparator, ICMP , is proportional
to the voltage at the ITH pin, with a zero-current threshold
corresponding to an ITH voltage of around 0.8V.
The error amplifier (EA) adjusts this ITH voltage by comparing the feedback signal to the internal 0.6V reference
voltage. On Channel 1, the difference amplifier (DIFFAMP)
converts the differential feedback signal (VOUTSENSE1+ –
VOUTSENSE1–) to a single-ended input for the EA; Channel 2
uses VFB2 directly with respect to SGND. Output voltage is
regulated so that the feedback voltage is equal to the internal
0.6V reference. If the load current increases/decreases, it
causes a momentary drop/rise in the differential feedback
voltage relative to the reference. The EA then moves ITH
voltage, or inductor valley current setpoint, higher/lower
until the average inductor current again matches the load
current, so that the output voltage comes back to the
regulated voltage.
The LTC3838 features a detect transient (DTR) pin to detect
“load-release”, or a transient where the load current suddenly drops, by monitoring the first derivative of the ITH
voltage. When detected, the bottom gate (BG) is turned
off and inductor current flows through the body diode
in the bottom MOSFET, allowing the SW node voltage to
drop below PGND by the body diode’s forward-conduction
voltage. This creates a more negative differential voltage
(VSW – VOUT) across the inductor, allowing the inductor
current to drop faster to zero, thus creating less overshoot
on VOUT. See Load-Release Transient Detection in Applications Information for details.
Differential Output Sensing
This dual controller’s first channel features differential
output voltage sensing. The output voltage is resistively
divided externally to create a feedback voltage for the controller. The internal difference amplifier (DIFFAMP) senses
this feedback voltage with respect to the output’s remote
ground reference to create a differential feedback voltage.
This scheme eliminates any ground offsets between local
ground and remote output ground, resulting in a more
accurate output voltage. Channel 1 allows remote output
ground to deviate as much as ±500mV with respect to
local ground (SGND).
DRVCC/EXTVCC/INTVCC Power
DRVCC1,2 are the power for the bottom MOSFET drivers.
Normally the two DRVCC pins are shorted together on
the PCB, and decoupled to PGND with a minimum 4.7μF
ceramic capacitor, CDRVCC. The top MOSFET drivers are
biased from the floating bootstrap capacitors (CB1,2)
which are recharged during each cycle through an external
Schottky diode when the top MOSFET turns off and the
SW pin swings down.
The DRVCC can be powered on two ways: an internal lowdropout (LDO) linear voltage regulator that is powered
3838fa
14
LTC3838
OPERATION (Refer to Functional Diagram)
from VIN and can output 5.3V to DRVCC1. Alternatively,
an internal EXTVCC switch (with on-resistance of around
2Ω) can short the EXTVCC pin to DRVCC2.
If the EXTVCC pin is below the EXTVCC switchover voltage
(typically 4.6V with 200mV hysteresis, see the Electrical Characteristics Table), then the internal 5.3V LDO is
enabled. If the EXTVCC pin is tied to an external voltage
source greater than this EXTVCC switchover voltage, then
the LDO is shut down and the internal EXTVCC switch
shorts the EXTVCC pin to the DRVCC2 pin, thereby powering DRVCC and INTVCC with the external voltage source
and helping to increase overall efficiency and decrease
internal self heating from power dissipated in the LDO.
This external power source could be the output of the
step-down converter itself (if the output is programmed
to higher than the switchover voltage’s higher limit, 4.8V).
The VIN pin still needs to be powered up but now draws
minimum current.
Power for most internal control circuitry other than gate
drivers is derived from the INTVCC pin. INTVCC can be powered from the combined DRVCC pins through an external
RC filter to SGND to filter out noises due to switching.
Shutdown and Start-Up
Each of the RUN1 and RUN2 pins has an internal proportional-to-absolute-temperature (PTAT) current source (around
1.2μA at 25°C) to pull up the pins. Taking both RUN1 and
RUN2 pins below a certain threshold voltage (around 0.8V
at 25°C) shuts down all bias of INTVCC and DRVCC and
places the LTC3838 into micropower shutdown mode
with a minimum IQ at the VIN pin. The LTC3838’s DRVCC
(through the internal 5.3V LDO regulator or EXTVCC) and
the corresponding channel’s internal circuitry off INTVCC
will be biased up when either or both RUN pins are pulled
up above the 0.8V threshold, either by the internal pull-up
current or driven directly by external voltage source such
as logic gate output.
A channel of the LTC3838 will not start switching until the
RUN pin of the respective channel is pulled up to 1.2V.
When a RUN pin rises above 1.2V, the corresponding
channel’s TG and BG drivers are enabled, and TRACK/
SS released. An additional 5μA temperature-independent
pull-up current is connected internally to the channel’s
respective RUN pin. To turn off TG, BG and the additional 5μA pull-up current, RUN needs to be pulled down
below 1.2V by about 100mV. These built-in current and
voltage hystereses prevent false jittery turn-on and turn-off
due to noise. Such features on the RUN pins allow input
undervoltage lockout (UVLO) to be set up using external
voltage dividers from VIN.
The start-up of a channel’s output voltage (VOUT) is
controlled by the voltage on its TRACK/SS pin. When the
voltage on the TRACK/SS pin is less than the 0.6V internal
reference, the (differential) feedback voltage is regulated to
the TRACK/SS voltage instead of the 0.6V reference. The
TRACK/SS pin can be used to program the output voltage
soft-start ramp-up time by connecting an external capacitor from a TRACK/SS pin to signal ground. An internal
temperature-independent 1μA pull-up current charges
this capacitor, creating a voltage ramp on the TRACK/SS
pin. As the TRACK/SS voltage rises linearly from ground
to 0.6V, the switching starts, VOUT ramps up smoothly to
its final value and the feedback voltage to 0.6V. TRACK/
SS will keep rising beyond 0.6V, until being clamped to
around 3.7V.
Alternatively, the TRACK/SS pin can be used to track an
external supply like in a master slave configuration. Typically, this requires connecting a resistor divider from the
master supply to the TRACK/SS pin (see the Applications
Information section).
TRACK/SS is pulled low internally when the corresponding channel’s RUN pin is pulled below the 1.2V threshold
(hysteresis applies), or when INTVCC or either of the
DRVCC1,2 pins drop below their respective undervoltage
lockout (UVLO) thresholds.
Light Load Current Operation
If the MODE/PLLIN pin is tied to INTVCC or an external clock
is applied to MODE/PLLIN, the LTC3838 will be forced to
operate in continuous mode. With load current less than
one-half of the full load peak-to-peak ripple, the inductor
current valley can drop to zero or become negative. This
allows constant-frequency operation but at the cost of low
efficiency at light loads.
3838fa
15
LTC3838
OPERATION (Refer to Functional Diagram)
If the MODE/PLLIN pin is left open or connected to signal
ground, the channel will transition into discontinuous mode
operation, where a current reversal comparator (IREV) shuts
off the bottom MOSFET (MB) as the inductor current approaches zero, thus preventing negative inductor current
and improving light-load efficiency. In this mode, both
switches can remain off for extended periods of time. As
the output capacitor discharges by load current and the
output voltage droops lower, EA will eventually move the
ITH voltage above the zero current level (0.8V) to initiate
another switching cycle.
Power Good and Fault Protection
Each PGOOD pin is connected to an internal open-drain
N-channel MOSFET. An external resistor or current source
can be used to pull this pin up to 6V (e.g., VOUT1,2 or DRVCC).
Overvoltage or undervoltage comparators (OV, UV) turn on
the MOSFET and pull the PGOOD pin low when the feedback
voltage is outside the ±7.5% window of the 0.6V reference voltage. The PGOOD pin is also pulled low when the
channel’s RUN pin is below the 1.2V threshold (hysteresis
applies), or in undervoltage lockout (UVLO). Note that feedback voltage of Channel 1 is sensed differentially through
VOUTSENSE1+ with respect to VOUTSENSE1–, while Channel 2
is sensed through VFB2 with respect to SGND.
When the feedback voltage is within the ±7.5% window,
the open-drain NMOS is turned off and the pin is pulled
up by the external source. The PGOOD pin will indicate
power good immediately after the feedback is within the
window. But when a feedback voltage of a channel goes
out of the window, there is an internal 50μs delay before
its PGOOD is pulled low. In an overvoltage (OV) condition,
MT is turned off and MB is turned on immediately without
delay and held on until the overvoltage condition clears.
Foldback current limiting is provided if the output is below
one-half of the regulated voltage, such as being shorted to
ground. As the feedback approaches 0V, the internal clamp
voltage for the ITH pin drops from 2.4V to around 1.27V,
which reduces the inductor valley current level to about
30% of its maximum value. Foldback current limiting is
disabled at start-up.
Frequency Selection and External Clock
Synchronization
An internal oscillator (clock generator) provides phaseinterleaved internal clock signals for individual channels
to lock up to. The switching frequency and phase of each
switching channel is independently controlled by adjusting the top MOSFET turn-on time (on-time) through the
one-shot timer. This is achieved by sensing the phase
relationship between a top MOSFET turn-on signal and
its internal reference clock through a phase detector, and
the time interval of the one-shot timer is adjusted on a
cycle-by-cycle basis, so that the rising edge of the top
MOSFET turn-on is always trying to synchronize to the
internal reference clock signal for the respective channel.
The frequency of the internal oscillator can be programmed
from 200kHz to 2MHz by connecting a resistor, RT, from the
RT pin to signal ground (SGND). The RT pin is regulated
to 1.2V internally.
For applications with stringent frequency or interference
requirements, an external clock source connected to the
MODE/PLLIN pin can be used to synchronize the internal
clock signals through a clock phase-locked loop (Clock
PLL). The LTC3838 operates in forced continuous mode
of operation when it is synchronized to the external clock.
The external clock frequency has to be within ±30% of the
internal oscillator frequency for successful synchronization. The clock input levels should be no less than 2V for
“high” and no greater than 0.5V for “low”. The MODE/
PLLIN pin has an internal 600k pull-down resistor.
Multichip Operations
The PHASMD pin determines the relative phases between
the internal reference clock signals for the two channels
as well as the CLKOUT signal, as shown in Table 1. The
phases tabulated are relative to zero degree (0°) being
defined as the rising edge of the internal reference clock
signal of Channel 1. The CLKOUT signal can be used to
synchronize additional power stages in a multiphase power
supply solution feeding either a single high current output,
or separate outputs.
3838fa
16
LTC3838
OPERATION (Refer to Functional Diagram)
The system can be configured for up to 12-phase operation with a multichip solution. Typical configurations are
shown in Table 2 to interleave the phases of the channels.
Table 1
PHASMD
SGND
FLOAT
INTVCC
Channel 1
0°
0°
0°
Channel 2
180°
180°
240°
CLKOUT
60°
90°
120°
Table 2
NUMBER OF
PHASES
externally for single-output applications. Set PHASMD
to SGND or FLOAT so that the two channels are 180°
out-of-phase. Efficiency losses may be substantially
reduced because the peak current drawn from the input
capacitor is effectively divided by the number of phases
used and power loss is proportional to the RMS current
squared. A 2-phase implementation can reduce the input
path power loss by up to 75%.
To make a single-output converter of three or more phases,
additional LTC3838 or LTC3833 chips can be used. The first
chip should be tied the same way as the 2-phase above. If
only one more channel of an additional LTC3838 is needed,
use Channel 1 for the additional phase:
NUMBER OF
LTC3838
PIN CONNECTIONS
[PIN NAME (CHIP NUMBER)]
2
1
PHASMD(1) = FLOAT or SGND
3
2, or 1 +
LTC3833
PHASMD(1) = INTVCC
MODE/PLLIN(2) = CLKOUT(1)
• Tie the ITH1 pin to the ITH1 pin of the first chip
4
2
PHASMD(1) = FLOAT
PHASMD(2) = FLOAT or SGND
MODE/PLLIN(2) = CLKOUT(1)
• Tie the RUN1 pin to the RUN pins of the first chip
6
12
3
6
PHASMD(1) = SGND
PHASMD(2) = SGND
MODE/PLLIN(2) = CLKOUT(1)
PHASMD(3) = FLOAT or SGND
MODE/PLLIN(3) = CLKOUT(2)
PHASMD(1) = SGND
PHASMD(2) = SGND
MODE/PLLIN(2) = CLKOUT(1)
PHASMD(3) = FLOAT
MODE/PLLIN(3) = CLKOUT(2)
PHASMD(4) = SGND
MODE/PLLIN(4) = CLKOUT(3)
PHASMD(5) = SGND
MODE/PLLIN(5) = CLKOUT(4)
PHASMD(6) = FLOAT or SGND
MODE/PLLIN(6) = CLKOUT(5)
• Tie the VOUTSENSE1+ pin to the VOUTSENSE1+ pin of the
first chip
• Tie the VOUTSENSE1– pin to the VOUTSENSE1– pin of the
first chip
• Tie the TRACK/SS1 pin to the TRACK/SS1 pin of the
first chip
If both channels are needed, the additional LTC3838 chip
should be tied the same way as the first LTC3838 chip to
disable the second channel’s EA:
• Tie the VFB2 pin to the chip’s own INTVCC
• Tie the ITH1 pin to the ITH1 pin of the first chip
• Tie the RUN pins to the RUN pins of the first chip
Single-Output PolyPhase Configurations
To use LTC3838 for a 2-phase single output step-down
controller: Tie the VFB2 pin to INTVCC, which will disable
Channel 2’s error amplifier and internally connect ITH2 to
ITH1. Tie the compensation R-C components to the ITH1
pin. The ITH2 pin can be either left open or shorted to ITH1
externally. The TRACK/SS2 and PGOOD2 pins become
defunct and can be left open. Note that the RUN1, RUN2,
DTR1, DTR2, VRNG1 and VRNG2 pins still function for
the two channels individually, therefore should be shorted
• Tie the VOUTSENSE1+ pin to the VOUTSENSE1+ pin of the
first chip
• Tie the VOUTSENSE1– pin to the VOUTSENSE1– pin of the
first chip
• Tie the TRACK/SS1 pin to the TRACK/SS1 pin of the
first chip
3838fa
17
LTC3838
APPLICATIONS INFORMATION
Once the required output voltage and operating frequency
have been determined, external component selection is
driven by load requirement, and begins with the selection of inductors and current sense method (either sense
resistors RSENSE or inductor DCR sensing). Next, power
MOSFETs are selected. Finally, input and output capacitors are selected.
Output Voltage Programming
As shown in Figure 1, external resistor dividers are used
from the regulated outputs to their respective ground references to program the output voltages. On Channel 1, the
resistive divider is tapped by the VOUTSENSE1+ pin, and the
ground reference is remotely sensed by the VOUTSENSE1–
pin, this voltage is sensed differentially. On Channel 2, the
resistive divider is tapped by the VFB2 pin, with respect to
signal ground at the SGND pin. By regulating the tapped
(differential) feedback voltages to the internal reference
0.6V, the resulting output voltages are:
VOUT1 – VOUTSENSE1– = 0.6V • (1 + RFB2/RFB1)
and
VOUT2 = 0.6V • (1 + RFB2/RFB1)
For example, if VOUT1 is programmed to 5V and the output ground reference is sitting at –0.5V with respect to
SGND, then the absolute value of the output will be 4.5V
with respect to SGND. The minimum (differential) output
voltages are limited to the internal reference 0.6V, and the
maximum are 5.5V.
The VOUTSENSE1+ pin is a high impedance pin with no
input bias current other than leakage in the nA range. The
VOUTSENSE1– pin has about 30μA of current flowing out
of the pin. The VFB2 pin is quasi-high impedance pin with
minimum bias current out of the pin.
Differential output sensing allows for more accurate output
regulation in high power distributed systems having large
line losses. Figure 2 illustrates the potential variations in
the power and ground lines due to parasitic elements.
The variations may be exacerbated in multi-application
systems with shared ground planes. Without differential
output sensing, these variations directly reflect as an error
in the regulated output voltage. The LTC3838 Channel 1’s
differential output sensing can correct for up to ±500mV
of variation in the output’s power and ground lines.
The LTC3838 Channel 1’s differential output sensing
scheme is distinct from conventional schemes where the
regulated output and its ground reference are directly
sensed with a difference amplifier whose output is then
divided down with an external resistor divider and fed
into the error amplifier input. This conventional scheme
is limited by the common mode input range of the difference amplifier and typically limits differential sensing to
the lower range of output voltages.
The LTC3838’s Channel 1 allows for seamless differential
output sensing by sensing the resistively divided feedback
voltage differentially. This allows for differential sensing
in the full output range from 0.6V to 5.5V. The difference
amplifier (DIFFAMP) has a bandwidth of 8MHz, high
VOUT
LTC3838
RFB2
COUT
VOUTSENSE1+/VFB2
RFB1
VOUTSENSE1–/SGND
3838 F01
Figure 1. Setting Output Voltage
3838fa
18
LTC3838
APPLICATIONS INFORMATION
CIN
MT
+
–
VIN
POWER TRACE
PARASITICS
L
LTC3838
VOUTSENSE1+ VOUTSENSE1–
RFB2
±VDROP(PWR)
MB
RFB1
COUT1
ILOAD
COUT2 I
LOAD
GROUND TRACE
PARASITICS
±VDROP(GND)
OTHER CURRENTS
FLOWING IN
SHARED GROUND
PLANE
3838 F02
Figure 2. Differential Output Sensing Used to Correct Line Loss Variations
in a High Power Distributed System with a Shared Ground Plane
enough so that it will not affect main loop compensation
and transient behavior.
To avoid noise coupling into the feedback voltages
(VOUTSENSE1+ or VFB2), the resistor dividers should be
placed close to the VOUTSENSE1+ and VOUTSENSE1–, or
VFB2 and SGND pins. Remote output and ground traces
should be routed together as a differential pair to the
remote output. For best accuracy, these traces to the
remote output and ground should be connected as close
as possible to the desired regulation point.
Switching Frequency Programming
The choice of operating frequency is a trade-off between
efficiency and component size. Lowering the operating frequency improves efficiency by reducing MOSFET switching
losses but requires larger inductance and/or capacitance
to maintain low output ripple voltage. Conversely, raising
the operating frequency degrades efficiency but reduces
component size.
The switching frequency of the LTC3838 can be programmed from 200kHz to 2MHz by connecting a resistor
from the RT pin to signal ground. The value of this resistor
can be chosen according to the following formula:
R T [kΩ ] =
41550
– 2.2
f [kHz ]
The overall controller system, including the clock PLL
and switching channels, has a synchronization range of
no less than ±30% around this programmed frequency.
Therefore, during external clock synchronization be sure
that the external clock frequency is within this ±30% range
of the RT programmed frequency. It is advisable that the
RT programmed frequency be equal the external clock for
maximum synchronization margin. Refer to the “Phase
and Frequency Synchronization” section for more details.
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of
smaller inductor and capacitor values. A higher frequency
generally results in lower efficiency because of MOSFET
gate charge losses. In addition to this basic trade-off, the
effect of inductor value on ripple current and low current
operation must also be considered.
3838fa
19
LTC3838
APPLICATIONS INFORMATION
The inductor value has a direct effect on ripple current.
The inductor ripple current ∆IL decreases with higher
inductance or frequency and increases with higher VIN:
⎛V ⎞⎛ V ⎞
ΔIL = ⎜ OUT ⎟ ⎜ 1– OUT ⎟
⎝ f •L ⎠ ⎝
VIN ⎠
Accepting larger values of ∆IL allows the use of low inductances, but results in higher output voltage ripple, higher
ESR losses in the output capacitor, and greater core losses.
A reasonable starting point for setting ripple current is ∆IL
= 0.4 • IMAX. The maximum ∆IL occurs at the maximum
input voltage. To guarantee that ripple current does not
exceed a specified maximum, the inductance should be
chosen according to:
⎛ V
⎞⎛
VOUT ⎞
OUT
1–
L=⎜
⎟⎜
⎟
⎝ f • ΔIL(MAX) ⎠ ⎝ VIN(MAX) ⎠
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. The two basic types are iron powder and ferrite. The iron powder types have a soft saturation curve
which means they do not saturate hard like ferrites do.
However, iron powder type inductors have higher core
losses. Ferrite designs have very low core loss and are
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Sumida, Panasonic, Coiltronics, Coilcraft, Toko, Vishay,
Pulse and Würth.
Current Sense Pins
Inductor current is sensed through voltage between
SENSE+ and SENSE– pins, the inputs of the internal current
comparators. The input voltage range of the SENSE pins is
–0.5V to 5.5V. Care must be taken not to float these pins
during normal operation. The SENSE+ pins are quasi-high
impedance inputs. There is no bias current into a SENSE+
pin when its corresponding channel’s SENSE– pin ramps
up from below 1.1V and stays below 1.4V. But there is a
small (~1μA) current flowing into a SENSE+ pin when its
corresponding SENSE– pin ramps down from 1.4V and
stays above 1.1V. Such currents also exist on SENSE– pins.
But in addition, each SENSE– pin has an internal 500k
resistor to SGND. The resulted current (VOUT/500k) will
dominate the total current flowing into the SENSE– pins.
SENSE+ and SENSE– pin currents have to be taken into
account when designing either RSENSE or DCR inductor
current sensing.
Current Limit Programming
The current sense comparators’ maximum trip voltage
between SENSE+ and SENSE– (or “sense voltage”), when
ITH is clamped at its maximum 2.4V, is set by the voltage
applied to the VRNG pin and is given by:
VSENSE(MAX) = 0.05VRNG
Core loss is independent of core size for a fixed inductor
value, but it is very dependent on inductance selected. As
inductance increases, core losses go down. Unfortunately,
increased inductance requires more turns of wire and
therefore copper losses will increase.
The valley current mode control loop does not allow the
inductor current valley to exceed 0.05VRNG. In practice,
one should allow sufficient margin, to account for tolerance
of the parts and external component values. Note that ITH
is close to 2.4V when in current limit.
Ferrite core material saturates hard, which means that inductance collapses abruptly when the peak design current
is exceeded. This results an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
An external resistive divider from INTVCC can be used to set
the voltage on a VRNG pin between 0.6V and 2V, resulting
in a maximum sense voltage between 30mV and 100mV.
Such wide voltage range allows for variety of applications.
The VRNG pin can also be tied to either SGND or INTVCC
to force internal defaults. When VRNG is tied to SGND, the
device has an equivalent VRNG of 0.6V. When the VRNG pin
is tied to INTVCC, the device has an equivalent VRNG of 2V.
A variety of inductors designed for high current, low voltage applications are available from manufacturers such as
3838fa
20
LTC3838
APPLICATIONS INFORMATION
RSENSE Inductor Current Sensing
The LTC3838 can be configured to sense the inductor
currents through either low value series current sensing
resistors (RSENSE) or inductor DC resistance (DCR). The
choice between the two current sensing schemes is largely
a design trade-off between cost, power consumption and
accuracy. DCR sensing is becoming popular because it
saves expensive current sensing resistors and is more
power efficient, especially in high current applications.
However, current sensing resistors provide the most accurate current limits for the controller.
A typical RSENSE inductor current sensing scheme is shown
in Figure 3a. The filter components (RF , CF) need to be
placed close to the IC. The positive and negative sense
traces need to be routed as a differential pair close together and Kelvin (4-wire) connected underneath the sense
resistor, as shown in Figure 3b. Sensing current elsewhere
can effectively add parasitic inductance to the current sense
element, degrading the information at the sense terminals
and making the programmed current limit unpredictable.
RSENSE is chosen based on the required maximum output
current. Given the maximum current, IOUT(MAX), maximum
sense voltage, VSENSE(MAX), set by VRNG, and maximum
inductor ripple current ∆IL(MAX), the value of RSENSE can
be chosen as:
RSENSE =
VSENSE(MAX)
ΔIL(MAX)
IOUT(MAX) –
2
Conversely, given RSENSE and IOUT(MAX), VSENSE(MAX) and
thus VRNG voltage can be determined from the above equation. To ensure the maximum output current, sufficient
margin should be built in the calculations to account for
variations of LTC3838 under different operating conditions
and tolerances of external components.
Because of possible PCB noise in the current sensing
loop, the current sensing voltage ripple ∆VSENSE = ∆IL •
RSENSE also needs to be checked in the design to get a
RSENSE RESISTOR
AND
PARASITIC INDUCTANCE
R
ESL
VOUT
LTC3838
SENSE+
CF
SENSE–
CF • 2RF ≤ ESL/RS
POLE-ZERO
CANCELLATION
RF
RF
3838 F03a
FILTER COMPONENTS
PLACED NEAR SENSE PINS
Figure 3a. RSENSE Current Sensing
TO SENSE FILTER,
NEXT TO THE CONTROLLER
COUT
RSENSE
3838 F03b
Figure 3b. Sense Lines Placement with Sense Resistor
good signal-to-noise ratio. In general, for a reasonably
good PCB layout, 10mV of ∆VSENSE is recommended as
a conservative number to start with, either for RSENSE or
Inductor DCR sensing applications.
For today’s highest current density solutions the value
of the sense resistor can be less than 1mΩ and the
peak sense voltage can be as low as 20mV. In addition,
inductor ripple currents greater than 50% with operation
up to 2MHz are becoming more common. Under these
conditions, the voltage drop across the sense resistor’s
parasitic inductance becomes more relevant. A small RC
filter placed near the IC has been traditionally used to reduce the effects of capacitive and inductive noise coupled
in the sense traces on the PCB. A typical filter consists of
two series 10Ω resistors connected to a parallel 1000pF
capacitor, resulting in a time constant of 20ns.
This same RC filter, with minor modifications, can be
used to extract the resistive component of the current
sense signal in the presence of parasitic inductance.
3838fa
21
LTC3838
APPLICATIONS INFORMATION
For example, Figure 4a illustrates the voltage waveform
across a 2mΩ sense resistor with a 2010 footprint for a
1.2V/15A converter operating at 100% load. The waveform
is the superposition of a purely resistive component and a
purely inductive component. It was measured using two
scope probes and waveform math to obtain a differential
measurement. Based on additional measurements of the
inductor ripple current and the on-time and off-time of
the top switch, the value of the parasitic inductance was
determined to be 0.5nH using the equation:
ESL =
VESL(STEP) tON • tOFF
•
ΔIL
tON + tOFF
where VESL(STEP) is the voltage step caused by the ESL
and shown in Figure 4a, and tON and tOFF are top MOSFET
on-time and off-time respectively. If the RC time constant
VSENSE
20mV/DIV
VESL(STEP)
500ns/DIV
3838 F04a
Figure 4a. Voltage Waveform Measured
Directly Across the Sense Resistor
VSENSE
20mV/DIV
is chosen to be close to the parasitic inductance divided by
the sense resistor (L/R), the resulting waveform looks resistive again, as shown in Figure 4b. For applications using
low VSENSE(MAX), check the sense resistor manufacturer’s
data sheet for information about parasitic inductance. In
the absence of data, measure the voltage drop directly
across the sense resistor to extract the magnitude of the
ESL step and use the equation above to determine the ESL.
However, do not over filter. Keep the RC time constant less
than or equal to the inductor time constant to maintain a
high enough ripple voltage on VRSENSE.
Note that the SENSE1– and SENSE2– pins are also used
for sensing the output voltage for the adjustment of top
gate on time, tON. For this purpose, there is an additional
internal 500k resistor from each SENSE– pin to SGND,
therefore there is an impedance mismatch with their corresponding SENSE+ pins. The voltage drop across the
RF causes an offset in sense voltage. For example, with
RF = 100Ω, at VOUT = VSENSE– = 5V, the sense-voltage
offset VSENSE(OFFSET) = VSENSE– • RF/500k = 1mV. Such
small offset may seem harmless for current limit, but
could be significant for current reversal detection (IREV),
causing excess negative inductor current at discontinuous
mode. Also, at VSENSE(MAX) = 30mV, a mere 1mV offset
will cause a significant shift of zero-current ITH voltage
by (2.4V – 0.8V) • 1mV/30mV = 53mV. Too much shift
may not allow the output voltage to return to its regulated
value after the output is shorted due to ITH foldback.
Therefore, when a larger filter resistor RF value is used,
it is recommended to use an external 500k resistor from
each SENSE+ pin to SGND, to balance the internal 500k
resistor at its corresponding SENSE– pin.
The previous discussion generally applies to high density/
high current applications where IOUT(MAX) > 10A and low
inductor values are used. For applications where IOUT(MAX)
500ns/DIV
3838 F04b
Figure 4b. Voltage Waveform Measured After the
Sense Resistor Filter. CF = 1000pF, RF = 100Ω
3838fa
22
LTC3838
APPLICATIONS INFORMATION
< 10A, set RF to 10Ω and CF to 1000pF. This will provide
a good starting point.
The filter components need to be placed close to the IC.
The positive and negative sense traces need to be routed
as a differential pair and Kelvin (4-wire) connected to the
sense resistor.
DCR Inductor Current Sensing
For applications requiring higher efficiency at high load
currents, the LTC3838 is capable of sensing the voltage
drop across the inductor DCR, as shown in Figure 5. The
DCR of the inductor represents the small amount of DC
winding resistance, which can be less than 1mΩ for today’s
low value, high current inductors.
In a high current application requiring such an inductor,
conduction loss through a sense resistor would cost several
points of efficiency compared to DCR sensing.
The inductor DCR is sensed by connecting an RC filter
across the inductor. This filter typically consists of one or
two resistors (R1 and R2) and one capacitor (C1) as shown
in Figure 5. If the external (R1||R2) • C1 time constant is
chosen to be exactly equal to the L/DCR time constant, the
voltage drop across the external capacitor is equal to the
voltage drop across the inductor DCR multiplied by R2/
(R1 + R2). Therefore, R2 may be used to scale the voltage
across the sense terminals when the DCR is greater than
INDUCTOR
L
DCR
VOUT
COUT
L/DCR = (R1||R2) C1
LTC3838
the target sense resistance. With the ability to program
current limit through the VRNG pin, R2 may be optional. C1
is usually selected in the range of 0.01μF to 0.47μF. This
forces R1||R2 to around 2k to 4k, reducing error that might
have been caused by the SENSE pins’ input bias currents.
Resistor R1 should be placed close to the switching node,
to prevent noise from coupling into sensitive small-signal
nodes. Capacitor C1 should be placed close to the IC pins.
The first step in designing DCR current sensing is to
determine the DCR of the inductor. Where provided, use
the manufacturer’s maximum value, usually given at 25°C.
Increase this value to account for the temperature coefficient of resistance, which is approximately 0.4%/°C. A
conservative value for inductor temperature TL is 100°C.
The DCR of the inductor can also be measured using a good
RLC meter, but the DCR tolerance is not always the same
and varies with temperature; consult the manufacturers’
data sheets for detailed information.
From the DCR value, VSENSE(MAX) is easily calculated as:
VSENSE(MAX) = DCRMAX(25°C)
(
)
• ⎡1+ 0.4% TL(MAX) – 25°C ⎤
⎣
⎦
ΔI ⎞
⎛
• ⎜ IOUT(MAX) – L ⎟
⎝
2 ⎠
If VSENSE(MAX) is within the maximum sense voltage (30mV
to 100mV) of the LTC3838 as programmed by the VRNG
pin, then the RC filter only needs R1. If VSENSE(MAX) is
higher, then R2 may be used to scale down the maximum
sense voltage so that it falls within range.
R1
SENSE+
C1
SENSE–
R2
(OPT)
3838 F05
C1 NEAR SENSE PINS
Figure 5. DCR Current Sensing
3838fa
23
LTC3838
APPLICATIONS INFORMATION
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at the maximum input
voltage:
PLOSS
(V
(R1) =
IN(MAX) – VOUT
)• V
OUT
CMILLER ≅
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
RSENSE sensing. Light load power loss can be modestly
higher with a DCR network than with a sense resistor due
to the extra switching losses incurred through R1. However,
DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
To maintain a good signal-to-noise ratio for the current
sense signal, start with a ∆VSENSE of 10mV. For a DCR
sensing application, the actual ripple voltage will be determined by:
ΔVSENSE =
horizontal axis while the curve is approximately flat (or
the parameter QGD if specified on a manufacturer’s data
sheet), divided by the specified VDS test voltage:
VIN – VOUT VOUT
•
R1• C1 VIN • f
Power MOSFET Selection
Two external N-channel power MOSFETs must be selected
for each channel of the LTC3838 controller: one for the
top (main) switch and one for the bottom (synchronous)
switch. The gate drive levels are set by the DRVCC voltage.
This voltage is typically 5.3V. Pay close attention to the
BVDSS specification for the MOSFETs as well; most of the
logic-level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the onresistance, RDS(ON), Miller capacitance, CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data sheet.
CMILLER is equal to the increase in gate charge along the
QGD
VDS(TEST)
When the IC is operating in continuous mode, the duty
cycles for the top and bottom MOSFETs are given by:
VOUT
VIN
V
DBOT = 1– OUT
VIN
DTOP =
The MOSFET power dissipations at maximum output
current are given by:
PTOP = DTOP •IOUT(MAX)2 •RDS(ON)(MAX) (1+ δ ) + VIN 2
R TG(UP)
R TG(DOWN) ⎤
⎡
⎛ IOUT(MAX) ⎞
•CMILLER ⎢
•⎜
+
⎥•f
⎟
2
VMILLER ⎦
⎝
⎠
⎣ VDRVCC – VMILLER
PBOT = DBOT • IOUT(MAX)2 • RDS(ON)(MAX) • (1 + δ )
where δ is the temperature dependency of RDS(ON), RTG(UP)
is the TG pull-up resistance, and RTG(DOWN) is the TG pulldown resistance. VMILLER is the Miller effect VGS voltage
and is taken graphically from the MOSFET ’s data sheet.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V,
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V, the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
short-circuit when the synchronous switch is on close to
100% of the period.
3838fa
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LTC3838
APPLICATIONS INFORMATION
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs temperature curve in the
power MOSFET data sheet. For low voltage MOSFETs,
0.5% per degree (°C) can be used to estimate δ as an
approximation of percentage change of RDS(ON):
δ = 0.005/°C • (TJ – TA)
where TJ is estimated junction temperature of the MOSFET
and TA is ambient temperature.
CIN Selection
In continuous mode, the source current of the top Nchannel MOSFET is a square wave of duty cycle VOUT/
VIN. To prevent large voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The worst-case RMS current occurs by assuming
a single-phase application. The maximum RMS capacitor
current is given by:
IRMS ≅IOUT(MAX) •
VOUT
•
VIN
VIN
–1
VOUT
This formula has a maximum at VIN = 2VOUT , where
IRMS = IOUT(MAX)/2. This simple worst-case condition
is commonly used for design because even significant
deviations do not offer much relief. Note that capacitor
manufacturers’ ripple current ratings are often based on
only 2000 hours of life. This makes it advisable to further
derate the capacitor or to choose a capacitor rated at a
higher temperature than required. Several capacitors may
also be paralleled to meet size or height requirements in
the design. Due to the high operating frequency of the
LTC3838, additional ceramic capacitors should also be
used in parallel for CIN close to the IC and power switches
to bypass the high frequency switching noises. Typically
multiple X5R or X7R ceramic capacitors are put in parallel
with either conductive-polymer or aluminum-electrolytic
types of bulk capacitors. Because of its low ESR, the
ceramic capacitors will take most of the RMS ripple current. Vendors do not consistently specify the ripple current
rating for ceramics, but ceramics could also fail due to
excessive ripple current. Always consult the manufacturer
if there is any question.
Figure 6 represents a simplified circuit model for calculating the ripple currents in each of these capacitors. The
input inductance (LIN) between the input source and the
input of the converter will affect the ripple current through
the capacitors. A lower input inductance will result in less
ripple current through the input capacitors since more
ripple current will now be flowing out of the input source.
For simulations with this model, look at the ripple current
during steady-state for the case where one phase is fully
loaded and the other was not loaded. This will in general
be the worst case for ripple current since the ripple current from one phase will not be cancelled by ripple current
from the other phase.
Note that the bulk capacitor also has to be chosen for
RMS rating with ample margin beyond its RMS current
per simulation with the circuit model provided. For a lower
VIN range, a conductive-polymer type (such as Sanyo
OS-CON) can be used for its higher ripple current rating
and lower ESR. For a wide VIN range that also require
higher voltage rating, aluminum-electrolytic capacitors are
LIN
1μH
+
–
VIN
ESR(BULK)
ESR(CERAMIC)
ESL(BULK)
ESL(CERAMIC)
IPULSE(PHASE1)
IPULSE(PHASE2)
+
CIN(BULK)
CIN(CERAMIC)
3838 F06
Figure 6. Circuit Model for Input Capacitor
Ripple Current Simulation
3838fa
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LTC3838
APPLICATIONS INFORMATION
more attractive since it can provide a larger capacitance
for more damping. An aluminum-electrolytic capacitor
with a ripple current rating that is high enough to handle
all of the ripple current by itself will be very large. But
when in parallel with ceramics, an aluminum-electrolytic
capacitor will take a much smaller portion of the RMS
ripple current due to its high ESR. However, it is crucial
that the ripple current through the aluminum-electrolytic
capacitor should not exceed its rating since this will
produce significant heat, which will cause the electrolyte
inside the capacitor to dry over time and its capacitance
to go down and ESR to go up.
The benefit of PolyPhase operation is reduced RMS currents and therefore less power loss on the input capacitors. Also, the input protection fuse resistance, battery
resistance, and PC board trace resistance losses are also
reduced due to the reduced peak currents in a PolyPhase
system. The details of a close form equation can be found
in Application Note 77 “High Efficiency, High Density,
PolyPhase Converters for High Current Applications”.
Figure 7 shows the input capacitor RMS ripple currents
normalized against the DC output currents with respect
to the duty cycle. This graph can be used to estimate the
maximum RMS capacitor current for a multiple-phase
0.5
DC LOAD CURRENT
RMS INPUT RIPPLE CURRNET
0.6
0.3
0.2
0.1
0
0.1
0.2
0.3 0.4 0.5 0.6 0.7
DUTY FACTOR (VO/VIN)
Figure 7 shows that the use of more phases will reduce
the ripple current through the input capacitors due to
ripple current cancellation. However, since LTC3838 is
only truly phase-interleaved at steady state, transient RMS
currents could be higher than the curves for the designated
number of phase. Therefore, it is advisable to choose
capacitors by taking account the specific load situations
of the applications. It is always the safest to choose input
capacitors’ RMS current rating closer to the worst case of
a single-phase application discussed above, calculated by
assuming the loss that would have resulted if controller
channels switched on at the same time.
However, it is generally not needed to size the input capacitor for such worst-case conditions where on-times of the
phases coincide all the time. During a load step event, the
overlap of on-time will only occur for a small percentage
of time, especially when duty cycles are low. A transient
event where the switch nodes align for several cycles at
a time should not damage the capacitor. In most applications, sizing the input capacitors for 100% steady-state
load should be adequate. For example, a microprocessor
load may cause frequent overlap of the on-times, which
makes the ripple current higher, but the load current may
rarely be at 100% of IOUT(MAX). Using the worst-case load
current should already have margin built in for transient
conditions.
The VIN sources of the top MOSFETs should be placed
close to each other and share common CIN(s). Separating
the sources and CIN may produce undesirable voltage and
current resonances at VIN.
1-PHASE
2-PHASE
3-PHASE
4-PHASE
6-PHASE
0.4
application, assuming the channels are identical and their
phases are fully interleaved.
0.8
0.9
3838 F07
A small (0.1μF to 1μF) bypass capacitor between the IC’s
VIN pin and ground, placed close to the IC, is suggested.
A 2.2Ω to 10Ω resistor placed between CIN and the VIN
pin is also recommended as it provides further isolation
from switching noise of the two channels.
Figure 7. Normalized RMS Input Ripple Current
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APPLICATIONS INFORMATION
COUT Selection
The selection of output capacitance COUT is primarily
determined by the effective series resistance, ESR, to
minimize voltage ripple. The output voltage ripple ∆VOUT ,
in continuous mode is determined by:
⎛
⎞
1
ΔVOUT ≤ ΔIL ⎜ RESR +
8 • f • COUT ⎟⎠
⎝
where f is operating frequency, and ∆IL is ripple current
in the inductor. The output ripple is highest at maximum
input voltage since ∆IL increases with input voltage. Typically, once the ESR requirement for COUT has been met,
the RMS current rating generally far exceeds that required
from ripple current.
In multiphase single-output applications, it is advisable to
consider ripple requirements at specific load conditions.
At steady state, the LTC3838’s individual phases are interleaved, and their ripples cancel each other at the output,
so ripple on COUT is reduced. During transient, when the
phases are not fully interleaved, the ripple cancellation
may not be as effective. While the worst-case ∆IL is the
sum of the ∆ILs of individual phases aligned during a
fast transient, such ripple tends to counteract the effect
of load transient itself and lasts for only a short time. For
example, during sudden load current increase, the phases
align to ramp up the total inductor current to quickly pull
the VOUT up from the droop.
The choice of using smaller output capacitance increases
the ripple voltage due to the discharging term but can be
compensated for by using capacitors of very low ESR to
maintain the ripple voltage.
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but
have lower capacitance density than other types. Tantalum
capacitors have the highest capacitance density but it is
important to only use types that have been surge tested
for use in switching power supplies. Aluminum electrolytic
capacitors have significantly higher ESR, but can be used
in cost-sensitive applications provided that consideration
is given to ripple current ratings and long-term reliability.
Ceramic capacitors have excellent low ESR characteristics
but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace
inductance can also lead to significant ringing. When used
as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an
overvoltage hazard to the power switches and controller.
For high switching frequencies, reducing output ripple and
better EMI filtering may require small value capacitors that
have low ESL (and correspondingly higher self-resonant
frequencies) to be placed in parallel with larger value
capacitors that have higher ESL. This will ensure good
noise and EMI filtering in the entire frequency spectrum
of interest. Even though ceramic capacitors generally
have good high frequency performance, small ceramic
capacitors may still have to be parallel connected with
large ones to optimize performance.
High performance through-hole capacitors may also be
used, but an additional ceramic capacitor in parallel is
recommended to reduce the effect of their lead inductance.
Remember also to place high frequency decoupling capacitors as close as possible to the power pins of the load.
Top MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor, CB, connected to the
BOOST pin supplies the gate drive voltage for the topside
MOSFET. This capacitor is charged through diode DB from
DRVCC when the switch node is low. When the top MOSFET
turns on, the switch node rises to VIN and the BOOST pin
rises to approximately VIN + INTVCC. The boost capacitor
needs to store approximately 100 times the gate charge
required by the top MOSFET. In most applications a 0.1μF
to 0.47μF, X5R or X7R dielectric capacitor is adequate. It
is recommended that the BOOST capacitor be no larger
than 10% of the DRVCC capacitor, CDRVCC, to ensure that
the CDRVCC can supply the upper MOSFET gate charge
3838fa
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LTC3838
APPLICATIONS INFORMATION
and BOOST capacitor under all operating conditions. Variable frequency in response to load steps offers superior
transient performance but requires higher instantaneous
gate drive. Gate charge demands are greatest in high
frequency low duty factor applications under high load
steps and at start-up.
DRVCC Regulator and EXTVCC Power
The LTC3838 features a PMOS low dropout (LDO) linear
regulator that supplies power to DRVCC from the VIN supply.
The LDO regulates its output at the DRVCC1 pin to 5.3V.
The LDO can supply a maximum current of 100mA and
must be bypassed to ground with a minimum of 4.7μF
ceramic capacitor. Good bypassing is needed to supply
the high transient currents required by the MOSFET gate
drivers and to minimize interaction between the channels.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3838 to be
exceeded, especially if the LDO is active and provides
DRVCC. Power dissipation for the IC in this case is highest and is approximately equal to VIN • IDRVCC. The gate
charge current is dependent on operating frequency as
discussed in the Efficiency Considerations section. The
junction temperature can be estimated by using the equation given in Note 2 of the Electrical Characteristics. For
example, when using the LDO, LTC3838’s DRVCC current
is limited to less than 52mA from a 38V supply at TA =
70°C in the FE package:
TJ = 70°C + (52mA)(38V)(28°C/W) = 125°C
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked while
operating in continuous conduction mode at maximum VIN.
When the voltage applied to the EXTVCC pin rises above
the switchover voltage (typically 4.6V), the VIN LDO is
turned off and the EXTVCC is connected to DRVCC2 pin with
an internal switch. This switch remains on as long as the
voltage applied to EXTVCC remains above the hysteresis
(around 200mV) below the switchover voltage. Using
EXTVCC allows the MOSFET driver and control power to
be derived from the LTC3838’s switching regulator output
VOUT during normal operation and from the LDO when the
output is out of regulation (e.g., start up, short circuit). If
more current is required through the EXTVCC than is specified, an external Schottky diode can be added between the
EXTVCC and DRVCC pins. Do not apply more than 6V to the
EXTVCC pin and make sure that EXTVCC is less than VIN.
Significant efficiency and thermal gains can be realized
by powering DRVCC from the switching converter output,
since the VIN current resulting from the driver and control
currents will be scaled by a factor of (Duty Cycle)/(Switcher
Efficiency).
Tying the EXTVCC pin to a 5V supply reduces the junction
temperature in the previous example from 125°C to:
TJ = 70°C + (52mA)(5V)(28°C/W) = 77°C
However, for 3.3V and other low voltage outputs, additional circuitry is required to derive DRVCC power from
the converter output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5.3V LDO resulting
in an efficiency penalty of up to 10% at high input
voltages.
2. EXTVCC connected directly to switching converter output
VOUT is higher than the switchover voltage’s higher limit
(4.8V). This provides the highest efficiency.
3. EXTVCC connected to an external supply. If a 4.8V or
greater external supply is available, it may be used to
power EXTVCC providing that the external supply is
sufficient for MOSFET gate drive requirements.
4. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage converters, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage that has been boosted to greater
than 4.8V.
3838fa
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LTC3838
APPLICATIONS INFORMATION
For applications where the main input power never exceeds
5.3V, tie the DRVCC1 and DRVCC2 pins to the VIN input
through a small resistor, (such as 1Ω to 2Ω) as shown
in Figure 8 to minimize the voltage drop caused by the
gate charge current. This will override the LDO and will
prevent DRVCC from dropping too low due to the dropout
voltage. Make sure the DRVCC voltage exceeds the RDS(ON)
test voltage for the external MOSFET which is typically at
4.5V for logic-level devices.
LTC3838
DRVCC2
DRVCC1
RDRVCC
VIN
CDRVCC
CIN
3838 F08
Figure 8. Setup for VIN ≤ 5.3V
Input Undervoltage Lockout (UVLO)
The LTC3838 has two functions that help protect the controller in case of input undervoltage conditions. An internal
UVLO comparator constantly monitors the INTVCC and
DRVCC voltages to ensure that adequate voltages are present. The comparator enables internal UVLO signal, which
locks out the switching action of both channels, until the
INTVCC and DRVCC1,2 pins are all above their respective
UVLO thresholds. The rising threshold (to release UVLO)
of the INTVCC is typically 4.2V, with 0.5V falling hysteresis
(to re-enable UVLO). The UVLO thresholds for DRVCC1,2 are
lower than that of INTVCC but higher than typical threshold
voltages of power MOSFETs, to prevent them from turning
on without sufficient gate drive voltages.
Generally for VIN > 6V, a UVLO can be set through monitoring
the VIN supply by using external voltage dividers at the RUN
pins from VIN to SGND. To design the voltage divider, note
that both RUN pins have two levels of threshold voltages.
The precision gate-drive-enable threshold voltage of 1.2V
can be used to set a VIN to turn on a channel’s switching.
If resistor dividers are used on both RUN pins, when VIN
is low enough and both RUN pins are pulled below the
~0.8V threshold, the part will shut down all bias of INTVCC
and DRVCC and be put in micropower shutdown mode.
The RUN pins’ bias currents depend on the RUN voltages.
The bias current changes should be taken into account
when designing the external voltage divider UVLO circuit.
An internal proportional-to-absolute-temperature (PTAT)
pull-up current source (~1.2μA at 25°C) is constantly connected to this pin. When a RUN pin rises above 1.2V, the
corresponding channel’s TG and BG drives are turned on
and an additional 5μA temperature-independent pull-up
current is connected internally to the RUN pin. Pulling the
RUN pin to fall below 1.2V by more than an 80mV hysteresis turns off TG and BG of the corresponding channel,
and the additional 5μA pull-up current is disconnected.
As voltage on a RUN pin increases, typically beyond 3V,
its bias current will start to reverse direction and flow into
the RUN pin. Keep in mind that neither of the RUN pins
can sink more than 50μA; Even if a RUN pin may slightly
exceed 6V when sinking 50μA, a RUN pin should never
be forced to higher than 6V by a low impedance voltage
source to prevent faulty conditions.
Soft-Start and Tracking
The LTC3838 has the ability to either soft-start by itself
with a capacitor or track the output of another channel or
an external supply. Note that the soft-start and tracking
features are achieved not by limiting the maximum output
current of the controller, but by controlling the output ramp
voltage according to the ramp rate on the TRACK/SS pin.
3838fa
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LTC3838
APPLICATIONS INFORMATION
When a channel is configured to soft-start by itself, a capacitor should be connected to its TRACK/SS pin. TRACK/
SS is pulled low until the RUN pin voltage exceeds 1.2V
and UVLO is released, at which point an internal current
of 1μA charges the soft-start capacitor, CSS, connected
to the TRACK/SS pin. Current-limit foldback is disabled
during this phase to ensure smooth soft-start or tracking. The soft-start or tracking range is defined to be the
voltage range from 0V to 0.6V on the TRACK/SS pin. The
total soft-start time can be calculated as:
t SS (SEC) = 0.6(V)•
CSS (µF)
1(µA)
When one particular channel is configured to track an
external supply, a voltage divider can be used from the
external supply to the TRACK/SS pin to scale the ramp
rate appropriately. Two common implementations are coincidental tracking and ratiometric tracking. For coincident
tracking, make the divider ratio from the external supply
the same as the divider ratio for the differential feedback
voltage. Ratiometric tracking could be achieved by using
a different ratio than the differential feedback.
Note that the 1μA soft-start capacitor charging current is
still flowing, producing a small offset error. To minimize
this error, select the tracking resistive divider values to be
small enough to make this offset error negligible.
The LTC3838 allows the user to program how its two
channels’ outputs track each other ramping up or down.
In the following discussions, VOUT1 refers to the LTC3838’s
output 1 as a master channel and VOUT2 refers to the
LTC3838’s output 2 as a slave channel. In practice though,
either channel can be used as the master.
By selecting different resistors, the LTC3838 can achieve
different modes of tracking including the two in Figure 9.
To implement the coincident tracking, connect an additional resistive divider to VOUT1 and connect its midpoint
to the TRACK/SS pin of the slave channel. The ratio of this
divider should be the same as that of the slave channel’s
feedback divider shown in Figure 9b. In this tracking mode,
VOUT2
VOUT1
OUTPUT VOLTAGE
OUTPUT VOLTAGE
VOUT1
VOUT2
TIME
TIME
Coincident Tracking
3838 F09a
Ratiometric Tracking
Figure 9a. Two Different Modes of Output Tracking
VOUT1
TO
TRACK/SS2
PIN
RFB2(2)
RFB1(2)
RFB2(1)
TO
VOUTSENSE1+
PIN
RFB1(1)
TO
VOUTSENSE1–
PIN
TO
VFB2
PIN
Coincident Tracking Setup
VOUT2
RFB2(2)
RFB1(2)
VOUT1
TO
TRACK/SS2
PIN
RFB2(1)
RFB1(1)
TO
VOUTSENSE1+
PIN
TO
VFB2
PIN
TO
VOUTSENSE1–
PIN
VOUT2
RFB2(2)
RFB1(2)
3838 F09b
Ratiometric Tracking Setup
Figure 9b. Setup for Coincident and Ratiometric Tracking
3838fa
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LTC3838
APPLICATIONS INFORMATION
VOUT1 must be set higher than VOUT2. To implement the
ratiometric tracking, the master channel’s feedback divider
can be also used to provide TRACK/SS voltage for the slave
channel, since the additional divider, if used, should be of
the same ratio as the master channel’s feedback divider.
So which mode should be programmed? While either
mode satisfies most practical applications, some tradeoffs exist. The ratiometric mode saves a pair of resistors,
but the coincident mode offers better output regulation.
When the master channel’s output experiences dynamic
excursion (under load transient, for example), the slave
channel output will be affected as well. For better output
regulation, use the coincident tracking mode instead of
ratiometric.
Phase and Frequency Synchronization
For applications that require better control of EMI and
switching noise or have special synchronization needs, the
LTC3838 can synchronize the turn-on of the top MOSFET
to an external clock signal applied to the MODE/PLLIN pin.
The applied clock signal needs to be within ±30% of the
RT programmed frequency to ensure proper frequency
and phase lock. The clock signal levels should generally
comply to VPLLIN(H) > 2V and VPLLIN(L) < 0.5V. The MODE/
PLLIN pin has an internal 600k pull-down resistor to ensure
discontinuous current mode operation if the pin is left open.
The LTC3838 uses the voltages on VIN and VOUT as well
as RT to adjust the top gate on-time in order to maintain
phase and frequency lock for wide ranges of VIN, VOUT
and RT-programmed switching frequency f:
tON ≈
VOUT
VIN • f
As the on-time is a function of the switching regulator’s
output voltage, this output is measured by the SENSE– pin
to set the required on-time. The SENSE– pin is tied to the
regulator’s local output point to the IC for most applications, as the remotely regulated output point could be
significantly different from the local output point due to
line losses, and local output versus local ground is typically
the VOUT required for the calculation of tON.
However, there could be circumstances where this VOUT
programmed on-time differs significantly different from
the on-time required in order to maintain frequency
and phase lock. For example, lower efficiencies in the
switching regulator can cause the required on-time to be
substantially higher than the internally set on-time (see
Efficiency Considerations). If a regulated VOUT is relatively
low, proportionally there could be significant error caused
by the difference between the local ground and remote
ground, due to other currents flowing through the shared
ground plane.
If necessary, the RT resistor value, voltage on the VIN pin,
or even the common mode voltage of the SENSE pins may
be programmed externally to correct for such systematic
errors. The goal is to set the on-time programmed by VIN,
VOUT and RT close to the stready-state on-time so that the
system will have sufficient range to correct for component
and operating condition variations, or to synchronize to the
external clock. Note that there is an internal 500k resistor
on each SENSE– pin to SGND, but not on the SENSE+ pin.
During dynamic transient conditions either in the line
voltage or load current (e.g., load step or release), the top
switch will turn on more or less frequently in response
to achieve faster transient response. This is the benefit
of the LTC3838’s controlled on-time, valley current mode
architecture. However, this process may understandably
lose phase and even frequency lock momentarily. For
relatively slow changes, phase and frequency lock can
still be maintained. For large load current steps with fast
slew rates, phase lock will be lost until the system returns
back to a steady-state condition (see Figure 10). It may
take up to several hundred microseconds to fully resume
the phase lock, but the frequency lock generally recovers
quickly, long before phase lock does.
For light load conditions, the phase and frequency synchronization depends on the MODE/PLLIN pin setting. If
the external clock is applied, synchronization will be active
and switching in continuous mode. If MODE/PLLIN is tied
to INTVCC, it will operate in forced continuous mode at
the RT-programmed frequency. If the MODE/PLLIN pin is
tied to SGND, the LTC3838 will operate in discontinuous
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APPLICATIONS INFORMATION
ILOAD
CLOCK
INPUT
PHASE AND
FREQUENCY
LOCKED
PHASE AND
FREQUENCY
LOCK LOST
DUE TO FAST
LOAD STEP
FREQUENCY
RESTORED
QUICKLY
PHASE LOCK
RESUMED
PHASE AND
FREQUENCY
LOCK LOST
DUE TO FAST
LOAD STEP
FREQUENCY
RESTORED
QUICKLY
SW
VOUT
3838 F10
Figure 10. Phase and Frequency Locking Behavior During Transient Conditions
mode at light load and switch into continuous conduction
at the RT programmed frequency as load increases. The TG
on-time during discontinuous conduction is intentionally
slightly extended (approximately 1.2 times the continuous
conduction on-time as calculated from VIN, VOUT and f) to
create hysteresis at the load-current boundary of continuous/discontinuous conduction.
If an application requires very low (approaching minimum)
on-time, the system may not be able to maintain its full
frequency synchronization range. Getting closer to minimum on-time, it may even lose phase/frequency lock at no
load or light load conditions, under which the SW on-time
is effectively longer than TG on-time due to TG/BG dead
times. This is discussed further under Minimum On-Time,
Minimum Off-Time and Dropout Operation.
Minimum On-Time, Minimum Off-Time
and Dropout Operation
The minimum on-time is the smallest duration that
LTC3838’s TG (top gate) pin can be in high or “on” state.
It has dependency on the operating conditions of the
switching regulator, and is a function of voltages on the
VIN and VOUT pins, as well as the value of external resistor
RT. A minimum on-time of 30ns can be achieved when the
VOUT pin is tied to its minimum value of 0.6V while the VIN
is tied to its maximum value of 38V. For larger values of
VOUT and/or smaller values of VIN, the minimum achievable
on-time will be longer. The valley mode control architecture
allows low on-time, making the LTC3838 suitable for high
step-down ratio applications.
The effective on-time, as determined by the SW node
pulse width, can be different from this TG on-time, as it
also depends on external components, as well as loading
conditions of the switching regulator. One of the factors that
contributes to this discrepancy is the characteristics of the
power MOSFETs. For example, if the top power MOSFET’s
turn-on delay is much smaller than the turn-off delay,
the effective on-time will be longer than the TG on-time,
limiting the effective minimum on-time to a larger value.
Light-load operation, in forced continuous mode, will
further elongate the effective on-time due to the dead
times between the “on” states of TG and BG, as shown in
Figure 11. During the dead time from BG turn-off to TG
turn-on, the inductor current flows in the reverse direction,
charging the SW node high before the TG actually turns
on. The reverse current is typically small, causing a slow
rising edge. On the falling edge, after the top FET turns off
and before the bottom FET turns on, the SW node lingers
high for a longer duration due to a smaller peak inductor
current available in light load to pull the SW node low. As
a result of the sluggish SW node rising and falling edges,
the effective on-time is extended and not fully controlled
by the TG on-time. Closer to minimum on-time, this may
cause some phase jitter to appear at light load. As load
current increase, the edges become sharper, and the phase
locking behavior improves.
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LTC3838
APPLICATIONS INFORMATION
and/or programmed f = 2MHz (i.e., RT = 18k). In applications with different VIN, VOUT and/or f, the tON(MIN) that
can be achieved will generally be larger. Also, to guarantee
frequency and phase locking at light load, sufficient margin
needs to be added to account for the dead times (tD(TG/BG)
+ tD(TG/BG) in the Electrical Characteristics).
TG-SW
(VGS OF TOP
MOSFET)
DEAD-TIME
DELAYS
BG
(VGS OF
BOTTOM
MOSFET)
IL 0
NEGATIVE
INDUCTOR
CURRENT
IN FCM
VIN
SW
3838 F11
DURING BG-TG DEAD TIME,
NEGATIVE INDUCTOR CURRENT
WILL FLOW THROUGH TOP MOSFET’S
BODY DIODE TO PRECHARGE SW NODE
IL
+
–
DURING TG-BG DEAD TIME,
THE RATE OF SW NODE DISCHARGE
WILL DEPEND ON THE CAPACITANCE
ON THE SW NODE AND INDUCTOR
CURRENT MAGNITUDE
VIN
L
L
SW
IL
TOTAL CAPACITANCE
ON THE SW NODE
Figure 11. Light Loading On-Time Extension for Forced
Continuous Mode Operation
In continuous mode operation, the minimum on-time limit
imposes a minimum duty cycle of:
DMIN = f • tON(MIN)
where tON(MIN) is the effective minimum on-time for the
switching regulator. As the equation shows, reducing the
operating frequency will alleviate the minimum duty cycle
constraint. If the minimum on-time that LTC3838 can
provide is longer than the on-time required by the duty
cycle to maintain the switching frequency, the switching
frequency will have to decrease to maintain the duty cycle,
but the output voltage will still remain in regulation. This is
generally more preferable to skipping cycles and causing
larger ripple at the output, which is typically seen in fixed
frequency switching regulators.
For applications that require relatively low on-time, proper
caution has to be taken when choosing the power MOSFET.
If the gate of the MOSFET is not able to fully turn on due
to insufficient on-time, there could be significant heat dissipation and efficiency loss as a result of larger RDS(ON).
This may even cause early failure of the power MOSFET.
The minimum off-time is the smallest duration of time
that the TG pin can be turned low and then immediately
turned back high. This minimum off-time includes the
time to turn on the BG (bottom gate) and turn it back off,
plus the dead-time delays from TG off to BG on and from
BG off to TG on. The minimum off-time that the LTC3838
can achieve is 90ns.
The effective minimum off-time of the switching regulator,
or the shortest period of time that the SW node can stay
low, can be different from this minimum off-time. The main
factor impacting the effective minimum off-time is the top
and bottom power MOSFETs’ electrical characteristics,
such as Qg and turn-on/off delays. These characteristics
can either extend or shorten the SW nodes’ effective
minimum off-time. Large size (high Qg) power MOSFETs
generally tend to increase the effective minimum off-time
due to longer gate charging and discharging times. On
the other hand, imbalances in turn-on and turn-off delays
could reduce the effective minimum off-time.
The minimum off-time limit imposes a maximum duty
cycle of:
DMAX = 1 – f • tOFF(MIN)
where tOFF(MIN) is the effective minimum off-time of the
switching regulator. Reducing the operating frequency can
alleviate the maximum duty cycle constraint.
The tON(MIN) curves in the Typical Performance Characteristics are measured with minimum load on TG and
BG, at extreme cases of VIN = 38V, and/or VOUT = 0.6V,
3838fa
33
LTC3838
APPLICATIONS INFORMATION
If the maximum duty cycle is reached, due to a drooping
input voltage for example, the output will drop out of
regulation. The minimum input voltage to avoid dropout is:
VIN(MIN) =
VOUT
DMAX
At the onset of drop-out, there is a region of VIN of about
500mV that generates two discrete off-times, one being
the minimum off time and the other being an off-time that
is about 40ns to 60ns longer than the minimum off-time.
This secondary off-time is due to the extra delay in tripping the internal current comparator. The two off-times
average out to the required duty cycle to keep the output
in regulation. There may be higher SW node jitter, apparent
especially when synchronized to an external clock, but the
output voltage ripple remains relatively small.
Fault Conditions: Current Limiting and Overvoltage
The maximum inductor current is inherently limited in a
current mode controller by the maximum sense voltage.
In the LTC3838, the maximum sense voltage is controlled
by the voltage on the VRNG pin. With valley current mode
control, the maximum sense voltage and the sense resistance determine the maximum allowed inductor valley
current. The corresponding output current limit is:
ILIMIT =
VSENSE(MAX)
RSENSE
1
+ • ΔIL
2
The current limit value should be checked to ensure that
ILIMIT(MIN) > IOUT(MAX). The current limit value should
be greater than the inductor current required to produce
maximum output power at the worst-case efficiency.
Worst-case efficiency typically occurs at the highest VIN
and highest ambient temperature. It is important to check
for consistency between the assumed MOSFET junction
temperatures and the resulting value of ILIMIT which heats
the MOSFET switches.
To further limit current in the event of a short circuit to
ground, the LTC3838 includes foldback current limiting.
If the output falls by more than 50%, the maximum sense
voltage is progressively lowered, to about one-fourth of
its full value as the feedback voltage reaches 0V.
A feedback voltage exceeding 7.5% of the regulated target
of 0.6V is considered as overvoltage (OV). In such an OV
condition, the top MOSFET is immediately turned off and
the bottom MOSFET is turned on indefinitely until the OV
condition is removed, i.e., the feedback voltage falling
back below the 7.5% threshold by more than a hysteresis
of typical 2%. Current limiting is not active during an OV.
If the OV persists, and the BG turns on for a longer time,
the current through the inductor and the bottom MOSFET
may exceed their maximum ratings, sacrificing themselves
to protect the load.
OPTI-LOOP Compensation
OPTI-LOOP compensation, through the availability of the
ITH pin, allows the transient response to be optimized for
a wide range of loads and output capacitors. The ITH pin
not only allows optimization of the control-loop behavior
but also provides a DC-coupled and AC-filtered closed-loop
response test point. The DC step, rise time and settling
at this test point truly reflects the closed-loop response.
Assuming a predominantly 2nd order system, phase
margin and/or damping factor can be estimated using the
percentage of overshoot seen at this pin.
The external series RITH-CITH1 filter at the ITH pin sets the
dominant pole-zero loop compensation. The values can
be adjusted to optimize transient response once the final
PCB layout is done and the particular output capacitor type
and value have been determined. The output capacitors
need to be selected first because their various types and
values determine the loop feedback factor gain and phase.
An additional small capacitor, CITH2, can be placed from
the ITH pin to SGND to attenuate high frequency noise.
Note this CITH2 contributes an additional pole in the loop
gain therefore can affect system stability if too large. It
should be chosen so that the added pole is higher than
the loop bandwidth by a significant margin.
3838fa
34
LTC3838
APPLICATIONS INFORMATION
The regulator loop response can also be checked by
looking at the load transient response. An output current
pulse of 20% to 100% of full-load current having a rise
time of 1μs to 10μs will produce VOUT and ITH voltage
transient-response waveforms that can give a sense of the
overall loop stability without breaking the feedback loop.
For a detailed explanation of OPTI-LOOP compensation,
refer to Application Note 76.
Switching regulators take several cycles to respond to
a step in load current. When a load step occurs, VOUT
immediately shifts by an amount equal to ∆ILOAD • ESR,
where ESR is the effective series resistance of COUT. ∆ILOAD
also begins to charge or discharge COUT , generating a
feedback error signal used by the regulator to return VOUT
to its steady-state value. During this recovery time, VOUT
can be monitored for overshoot or ringing that would
indicate a stability problem.
Connecting a resistive load in series with a power MOSFET,
then placing the two directly across the output capacitor
and driving the gate with an appropriate signal generator
is a practical way to produce a realistic load step condition. The initial output voltage step resulting from the step
change in load current may not be within the bandwidth
of the feedback loop, so it cannot be used to determine
phase margin. The output voltage settling behavior is more
related to the stability of the closed-loop system. However,
it is better to look at the filtered and compensated feedback
loop response at the ITH pin.
The gain of the loop increases with the RITH and the bandwidth of the loop increases with decreasing CITH1. If RITH
is increased by the same factor that CITH1 is decreased, the
zero frequency will be kept the same, thereby keeping the
phase the same in the most critical frequency range of the
feedback loop. In addition, a feedforward capacitor, CFF ,
can be added to improve the high frequency response, as
shown in Figure 1. Capacitor CFF provides phase lead by
creating a high frequency zero with RFB2 which improves
the phase margin.
A more severe transient can be caused by switching in
loads with large supply bypass capacitors. The discharged
bypass capacitors of the load are effectively put in parallel
with the converter’s COUT , causing a rapid drop in VOUT .
No regulator can deliver current quick enough to prevent
this sudden step change in output voltage, if the switch
connecting the COUT to the load has low resistance and is
driven quickly. The solution is to limit the turn-on speed of
the load switch driver. Hot Swap™ controllers are designed
specifically for this purpose and usually incorporate current
limiting, short-circuit protection and soft starting.
Load-Release Transient Detection
As the output voltage requirement of step-down switching
regulators becomes lower, VIN to VOUT step-down ratio
increases, and load transients become faster, a major
challenge is to limit the overshoot in VOUT during a fast
load current drop, or “load-release” transient.
Inductor current slew rate diL/dt = VL/L is proportional
to voltage across the inductor VL = VSW – VOUT. When
the top MOSFET is turned on, VL = VIN – VOUT, inductor
current ramps up. When bottom MOSFET turns on, VL =
VSW – VOUT = –VOUT, inductor current ramps down. At
very low VOUT, the low differential voltage, VL, across the
inductor during the ramp down makes the slew rate of the
inductor current much slower than needed to follow the
load current change. The excess inductor current charges
up the output capacitor, which causes overshoot at VOUT.
If the bottom MOSFET could be turned off during the loadrelease transient, the inductor current would flow through
the body diode of the bottom MOSFET, and the equation
can be modified to include the bottom MOSFET body
diode drop to become VL = –(VOUT + VBD). Obviously the
benefit increases as the output voltage gets lower, since
VBD would increase the sum significantly, compared to a
single VOUT only.
The load-release overshoot at VOUT causes the error amplifier output, ITH, to drop quickly. ITH voltage is proportional
3838fa
35
LTC3838
APPLICATIONS INFORMATION
to the inductor current setpoint. A load transient will
result in a quick change of this load current setpoint, i.e.,
a negative spike of the first derivative of the ITH voltage.
The LTC3838 uses a detect transient (DTR) pin to monitor
the first derivative of the ITH voltage, and detect the loadrelease transient. Referring to the Functional Diagram, the
DTR pin is the input of a DTR comparator, and the internal
reference voltage for the DTR comparator is half of INTVCC.
To use this pin for transient detection, ITH compensation
needs an additional RITH resistor tied to INTVCC, and connects the junction point of ITH compensation components
CITH1, RITH1 and RITH2 to the DTR pin as shown in the
Functional Diagram. The DTR pin is now proportional to
the first derivative of the inductor current setpoint, through
the highpass filter of CITH1 and (RITH1//RITH2).
The two RITH resistors establish a voltage divider from
INTVCC to SGND, and bias the DC voltage on DTR pin (at
steady-state load or ITH voltage) slightly above half of
INTVCC. Compensation performance will be identical by
using the same CITH1 and make RITH1//RITH2 equal the
RITH as used in conventional single resistor OPTI-LOOP
compensation. This will also provide the R-C time constant
needed for the DTR duration. The DTR sensitivity can be
adjusted by the DC bias voltage difference between DTR
and half INTVCC. This difference could be set as low as
200mV, as long as the ITH ripple voltage with DC load
current does not trigger the DTR.
Note the internal 2.5μA pull-up current from the DTR pin
will generate an additional offset on top of the resistor
divider itself, making the total difference between the DC
bias voltage on the DTR pin and half INTVCC:
⎡
⎤
RITH1
VDTR – 0.5VINTVCC = ⎢
– 0.5 ⎥ • 5.3V
⎣ (RITH1+RITH2)
⎦
+ 2.5µA • (RITH1 / /RITH2)
As illustrated in Figure 12, when load current suddenly
drops, VOUT overshoots, and ITH drops quickly. The voltage
on the DTR pin will also drop quickly, since it is coupled
to the ITH pin through a capacitor. If the load transient
is fast enough that the DTR voltage drops below half of
INTVCC, a load release event is detected. The bottom gate
(BG) will be turned off, so that the inductor current flows
through the body diode in the bottom MOSFET. This allows the SW node to drop below PGND by a voltage of
a forward-conducted silicon diode. This creates a more
negative differential voltage (VSW – VOUT) across the
inductor, allowing the inductor current to drop at a faster
rate to zero, therefore creating less overshoot on VOUT.
The DTR comparator output is overridden by reverse
inductor current detection (IREV) and overvoltage (OV)
condition. This means BG will be turned off when SENSE+
is higher than SENSE– (i.e., inductor current is positive),
SW
5V/DIV
SW
5V/DIV
BG
5V/DIV
BG
5V/DIV
DTR
1V/DIV
BG TURNS BACK ON, INDUCTOR
CURRENT (IL) GOES NEGATIVE
IL
10A/DIV
DTR DETECTS LOAD
RELEASE, TURNS OFF BG
FOR FASTER INDUCTOR
CURRENT (IL) DECAY
ITH
1V/DIV
BG REMAINS ON
DURING THE LOAD
RELEASE EVENT
IL
10A/DIV
5μs/DIV
5μs/DIV
LOAD RELEASE = 15A TO 0A
VIN = 5V
VOUT = 0.6V
(12a) DTR Enabled
3838 F12
LOAD RELEASE = 15A TO 0A
VIN = 5V
VOUT = 0.6V
(12b) DTR Disabled
Figure 12. Comparison of VOUT Overshoot with Detect Transient (DTR) Feature Enabled and Disabled
3838fa
36
LTC3838
APPLICATIONS INFORMATION
as long as the OV condition is not present. When inductor
current drops to zero and starts to reverse, BG will turn
back on in forced continuous mode (e.g., the MODE/
PLLIN pin tied to INTVCC, or an input clock is present),
even if DTR is still below half INTVCC. This is to allow the
inductor current to go negative to quickly pull down the
VOUT overshoot. Of course, if the MODE/PLLIN pin is set
to discontinuous mode (i.e., tied to SGND), BG will stay
off as inductor current reverse, as it would with the DTR
feature disabled.
Also, if VOUT gets higher than the OV window (7.5%
typical), the DTR function is defeated and BG will turn
on regardless. Therefore, in order for the DTR feature
to reduce VOUT overshoot effectively,sufficient output
capacitance needs to be used in the application so that
OV is not triggered with the amount of load step desired
to have its overshoot suppressed.
Experimenting with a 0.6V output application (modified
from the design example circuit by setting VOUT to 0.6V
and ITH compensation adjusted accordingly) shows this
detect transient feature significantly reduces the overshoot
peak voltage, as well as time to resume regulation during
load release steps (see application examples in Typical
Performance Characteristics).
Note that it is expected that this DTR feature will cause
additional loss on the bottom MOSFET, due to its body
diode conduction. The bottom MOSFET temperature may
be higher with a load of frequent and large load steps. This
is an important design consideration. Experiments on the
demo board shows a 20°C increase when a continuous
100% to 50% load step pulse train with 50% duty cycle
and 100kHz frequency is applied to the output.
If not needed, this DTR feature can be disabled by tying
the DTR pin to INTVCC, or simply leave the DTR pin open
so that an internal 2.5μA current source will pull itself up
to INTVCC.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percentage efficiency
can be expressed as:
%Efficiency = 100% – (L1% + L2% + L3% + ...)
where L1%, L2%, etc. are the individual losses as a percentage of input power. Although all dissipative elements
in the circuit produce power losses, several main sources
usually account for most of the losses in LTC3838 circuits:
1. I2R loss. These arise from the DC resistances of the
MOSFETs, inductor, current sense resistor and is the majority of power loss at high output currents. In continuous mode the average output current flows though the
inductor L, but is chopped between the top and bottom
MOSFETs. If the two MOSFETs have approximately the
same RDS(ON), then the resistance of one MOSFET can
simply be summed with the inductor’s DC resistances
(DCR) and the board traces to obtain the I2R loss. For
example, if each RDS(ON) = 8mΩ, RL = 5mΩ, and RSENSE
= 2mΩ the loss will range from 15mW to 1.5W as the
output current varies from 1A to 10A. This results in loss
from 0.3% to 3% a 5V output, or 1% to 10% for a 1.5V
output. Efficiency varies as the inverse square of VOUT
for the same external components and output power
level. The combined effects of lower output voltages
and higher currents load demands greater importance
of this loss term in the switching regulator system.
2. Transition loss. This loss mostly arises from the brief
amount of time the top MOSFET spends in the saturation (Miller) region during switch node transitions. It
depends upon the input voltage, load current, driver
strength and MOSFET capacitance, among other factors, and can be significant at higher input voltages or
higher switching frequencies.
3838fa
37
LTC3838
APPLICATIONS INFORMATION
3. DRVCC current. This is the sum of the MOSFET driver
and INTVCC control currents. The MOSFET driver currents result from switching the gate capacitance of the
power MOSFETs. Each time a MOSFET gate is switched
from low to high to low again, a packet of charge dQ
moves from DRVCC to ground. The resulting dQ/dt is a
current out of DRVCC that is typically much larger than
the controller IQ current. In continuous mode,
IGATECHG = f • (Qg(TOP) + Qg(BOT)),
where Qg(TOP) and Qg(BOT) are the gate charges of the
top and bottom MOSFETs, respectively.
Supplying DRVCC power through EXTVCC could save
several percents of efficiency, especially for high VIN
applications. Connecting EXTVCC to an output-derived
source will scale the VIN current required for the driver
and controller circuits by a factor of (Duty Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA
of DRVCC current results in approximately 2.5mA of VIN
current. This reduces the mid-current loss from 10%
or more (if the driver was powered directly from VIN)
to only a few percent.
4. CIN loss. The input capacitor filters large square-wave
input current drawn by the regulator into an averaged
DC current from the supply. The capacitor itself has
a zero average DC current, but square-wave-like AC
current flows through it. Therefore the input capacitor
must have a very low ESR to minimize the RMS current
loss on ESR. It must also have sufficient capacitance
to filter out the AC component of the input current to
prevent additional RMS losses in upstream cabling,
fuses or batteries. The LTC3838 2-phase architecture
improves the ESR loss.
“Hidden” copper trace, fuse and battery resistance, even
at DC current, can cause a significant amount of efficiency
degradation, so it is important to consider them during
the design phase. Other losses, which include the COUT
ESR loss, bottom MOSFET ’s body diode reverse-recovery
loss, and inductor core loss generally account for less
than 2% additional loss.
Power losses in the switching regulator will reflect as
a higher than ideal duty cycle, or a longer on-time for a
constant frequency. This efficiency accounted on-time
can be calculated as:
tON ≈ tON(IDEAL)/Efficiency
When making adjustments to improve efficiency, the input
current is the best indicator of changes in efficiency. If you
make a change and the input current decreases, then the
efficiency has increased.
Design Example
Consider a channel of step-down converter from VIN =
4.5V to 26V to VOUT = 1.2V, with IOUT(MAX) = 15A, and
f = 350kHz (see Figure 13, Channel 1.
The regulated output voltage is determined by:
⎛ R ⎞
VOUT = 0.6V • ⎜ 1+ FB2 ⎟
⎝ RFB1 ⎠
Using a 10k resistor for RFB1, RFB2 is also 10k.
The frequency is programmed by:
R T [kΩ ] =
41550
41550
– 2.2 =
– 2.2 ≈ 116.5
f [kHz ]
350
Use the nearest 1% resistor standard value of 115k.
The minimum on-time occurs for maximum VIN. Using the
tON(MIN) curves in the Typical Performance Characteristics
as references, make sure that the tON(MIN) at maximum VIN
is greater than that the LTC3838 can achieve, and allow
sufficient margin to account for the extension of effective
on-time at light load due to the dead times (tD(TG/BG) +
tD(TG/BG) in the Electrical Characteristics). The minimum
on-time for this application is:
tON(MIN) =
VOUT
VIN(MAX) • f
=
1.2V
= 143ns
24V • 350kHz
3838fa
38
LTC3838
APPLICATIONS INFORMATION
VIN
4.5V TO 26V
+
CIN1
220μF
CIN2
10μF
w3
2.2Ω
LTC3838
1μF
VIN
SENSE1–
SENSE2–
SENSE1+
SENSE2+
BOOST1
BOOST2
0.1μF
15k
0.1μF
0.1μF
0.1μF
3.57k
L1
0.56μH
VOUT1
1.2V
15A
TG1
MT1
+
MT2
DB2
SW1
L2
0.56μH
VOUT2
1.5V
15A
SW2
DRVCC1
INTVCC
COUT2
330μF
w2
3.57k
TG2
DB1
2.2Ω
COUT1
100μF
w2
15k
DRVCC2
EXTVCC
COUT4 +
330μF
w2
4.7μF
1μF
BG1
MB1
BG2
MB2
COUT3
100μF
w2
PGND
10k
VOUTSENSE1+
15k
VFB2
10k
100k
PGOOD1
0.01μF
PGOOD1
PGOOD2
100k
PGOOD2
0.01μF
TRACK/SS1 TRACK/SS2
22pF
220pF
90.9k
10k
VOUTSENSE1–
ITH1
ITH2
220pF
90.9k
82.5k
82.5k
115k
DTR1
VRNG1
RT
SGND
RUN1
DTR2
VRNG2
3838 F13a
PHASMD
MODE/PLLIN
CLKOUT
RUN2
3.0
100
3.0
100
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
90
EFFICIENCY
1.5
70
POWER
LOSS
1.0
60
0.5
50
VIN = 12V
VOUT = 1.2V
0.1
1
LOAD CURRENT (A)
10
0
2.5
2.0
80
EFFICIENCY
1.5
70
POWER
LOSS
1.0
60
POWER LOSS (W)
2.0
80
EFFICIENCY (%)
2.5
POWER LOSS (W)
EFFICIENCY (%)
90
40
CIN1: PANASONIC EEEFK1V221P
CIN2: TAIYO YUDEN GMK325BJ106MN-T
COUT2, COUT4: SANYO 2R5TPE330M9
COUT1, COUT3: MURATA GRM31CR60J107ME39L
DB1, DB2: CENTRAL SEMI CMDSH-4ETR
L1, L2: VISHAY IHLP4040DZERR56M01
MT1, MT2: RENESAS RJK0305DPB
MB1, MB2: RENESAS RJK0330DPB
22pF
0.5
50
VIN = 12V
VOUT = 1.5V
40
0.1
1
LOAD CURRENT (A)
3838 F13b
0
10
3838 F13c
Figure 13. Design Example: 4.5V to 26V Input, 1.2V/15A and 1.5V/15A Dual Outputs,
350kHz, DCR Sense, DTR Enabled, Step-Down Converter
3838fa
39
LTC3838
APPLICATIONS INFORMATION
Set the inductor value to give 40% ripple current at maximum VIN using the adjusted operating frequency:
1.2V
⎛
⎞ ⎛ 1.2V ⎞
L=⎜
1–
= 0.54µH
⎝ 350kHz • 40% • 15A ⎟⎠ ⎜⎝ 24V ⎟⎠
Select 0.56μH which is the nearest standard value.
The resulting maximum ripple current is:
⎛
⎞ ⎛ 1.2V ⎞
1.2V
1–
= 5.8A
ΔIL = ⎜
⎝ 350kHz • 0.56µH ⎟⎠ ⎜⎝ 24V ⎟⎠
Often in a high power application, DCR current sensing is
preferred over RSENSE in order to maximize efficiency. In
order to determine the DCR filter values, first the inductor
manufacturer has to be chosen. For this design, the Vishay
IHLP-4040DZ-01 model is chosen with a value of 0.56μH
and a DCRMAX =1.8mΩ. This implies that:
VSENSE(MAX) = 1.8mΩ • [1 + (100°C – 25°C) • 0.4%/°C]
• (15A – 5.8A/2) = 28mV
The maximum sense voltage, VSENSE(MAX), is within the
range that LTC3838 can handle without any additional
scaling. Therefore, the DCR filter can use a simple RC
filter across the inductor. If the C is chosen to be 0.1μF,
then the R can be calculated as:
RDCR =
L
0.56µH
=
= 3.1kΩ
DCR • CDCR 1.8mΩ • 0.1µF
The resulting VRNG pin voltage is:
VRNG =
VSENSE(MAX)
0.05
= 28mV • 20 = 0.56V
This voltage can be generated with a resistive divider from
the INTVCC pin to signal ground (SGND). To make sure that
the maximum load current of 15A can be supplied under
all conditions, such as lower INTVCC due to a lower VIN,
and account for the range of LTC3838’s own VSENSE(MAX)
variation within specification, a higher VRNG should be
used to provide margin.
while using an additional resistor in the DCR filter, as
discussed in DCR Inductor Current Sensing, to scale the
VSENSE(MAX) down by a comfortable margin below the
lower limit of the LTC3838’s own VSENSE(MAX) specification,
so that the maximum output current can be guaranteed.
In this design example, a 3.57k and 15k resistor divider is
used. The previously calculated VSENSE(MAX) is scaled down
from 28mV to 22.6mV, which is close to the lower limit of
LTC3838’s VSENSE(MAX) specification at VRNG = 0.6V. Note
the equivalent RDCR = 3.57k//15k = 2.9k, slightly lower
than the 3.1k calculated above for a matched RDCR-CDCR
and L-DCR network. The resulted mismatch allows for a
slightly higher ripple in VSENSE.
Remember to check the peak inductor current, considering
the upper spec limit of VSENSE(MAX) and the DCR(MIN) at
lowest operating temperature, is not going to saturate the
inductor or exceed the rating of power MOSFETs.
For the external N-channel MOSFETs, Renesas
RJK0305DBP (RDS(ON) = 13mΩ max, CMILLER = 150pF, VGS
= 4.5V, θJA = 40°C/W, TJ(MAX) = 150°C) is chosen for the top
MOSFET (main switch). RJK0330DBP (RDS(ON) = 3.9mΩ
max, VGS = 4.5V, θJA = 40°C/W, TJ(MAX) = 150°C) is chosen for
the bottom MOSFET (synchronous switch). The power dissipation for each MOSFET can be calculated for VIN = 24V and
typical TJ = 125°C:
⎛ 1.2V ⎞
PTOP = ⎜
(15A )2 (13mΩ) ⎡⎣1+ 0.4%(125°C – 25°C)⎤⎦
⎝ 24V ⎟⎠
1.2Ω ⎤
2.5Ω
2 ⎛ 15A ⎞
+
+ ( 24V ) ⎜
(150pF ) ⎡⎢ 5.3V
(350kHz )
⎝ 2 ⎟⎠
3V ⎥⎦
–
3V
⎣
= 0.54W
⎛ 24V – 1.2V ⎞
PBOT = ⎜
(15A )2 (3.9mΩ) ⎡⎣1+ 0.4%(125°C – 25°C)⎤⎦
⎝
24V ⎟⎠
= 1.2W
The resulted junction temperatures at an ambient temperature TA = 75°C are:
TJ(TOP) = 75°C + (0.54W)(40°C/W) = 97°C
TJ(BOT) = 75°C + (1.2W)(40°C/W) = 123°C
A better and the recommended way to set VRNG is to simply
tie the VRNG pin to SGND for an equivalent of VRNG = 0.6V,
3838fa
40
LTC3838
APPLICATIONS INFORMATION
These numbers show that careful attention should be paid
to proper heat sinking when operating at higher ambient
temperatures.
Select the CIN capacitors to give ample capacitance and
RMS ripple current rating. Consider worst-case duty cycles
per Figure 6: If operated at steady-state with SW nodes
fully interleaved, the two channels would generate not
more than 7.5A RMS at full load. In this design example,
3 × 10μF 35V X5R ceramic capacitors are put in parallel
to take the RMS ripple current, with a 220μF aluminumelectrolytic bulk capacitor for stability. For 10μF 1210 X5R
ceramic capacitors, try to keep the ripple current less
than 3A RMS through each device. The bulk capacitor
is chosen for RMS rating per simulation with the circuit
model provided.
The output capacitor COUT is chosen for a low ESR of
4.5mΩ to minimize output voltage changes due to inductor
ripple current and load steps. The output voltage ripple
is given as:
∆VOUT(RIPPLE) = ∆IL(MAX) • ESR = 5.85A • 4.5mΩ = 26mV
However, a 10A load step will cause an output change of
up to:
∆VOUT(STEP) = ∆ILOAD • ESR = 10A • 4.5mΩ = 45mV
Optional 2 × 100μF ceramic output capacitors are included
to minimize the effect of ESR and ESL in the output ripple
and to improve load step response.
The ITH compensation resistor RITH of 40k and a CITH of
220pF are chosen empirically for fast transient response,
and an additional CITH2 = 22pF is added directly from
ITH pin to SGND, to roll off the system gain at switching
frequency and attenuate high frequency noise.
To set up the detect transient (DTR) feature, pick resistors
for an equivalent RITH = RITH1//RITH2 close to the 40k.
Here, 1% resistors RITH1 = 90.9k (low side) and RITH2
= 82.5k (high side) are used, which yields an equivalent
RITH of 43.2k, and a DC-bias threshold of 128mV above
one-half of INTVCC. Note that even though the accuracy
of the equivalent compensation resistance RITH is not as
important, always use 1% or better resistors for the resistor divider from INTVCC to SGND to guarantee the relative
accuracy of this DC-bias threshold. To disable the DTR
feature, simply use a single RITH resistor to SGND, and
tie the DTR pin to INTVCC.
PCB Layout Checklist
The printed circuit board layout is illustrated graphically
in Figure 14. Figure 15 illustrates the current waveforms
present in the various branches of 2-phase synchronous
regulators operating in continuous mode. Use the following checklist to ensure proper operation of the LTC3838:
• A multilayer printed circuit board with dedicated ground
planes is generally preferred to reduce noise coupling
and improve heat sinking. The ground plane layer
should be immediately next to the routing layer for the
power components, e.g., MOSFETs, inductors, sense
resistors, input and output capacitors etc.
• Keep SGND and PGND separate. Upon finishing the
layout, connect SGND and PGND together with a single
PCB trace underneath the IC from the SGND pin through
the exposed PGND pad to the PGND pin.
• All power train components should be referenced to
PGND; all components connected to noise-sensitive
pins, e.g., ITH, RT , TRACK/SS and VRNG, should return
to the SGND pin. Keep PGND ample, but SGND area
compact. Use a modified “star ground” technique: a low
impedance, large copper area central PCB point on the
same side of the as the input and output capacitors.
• Place power components, such as CIN, COUT , MOSFETs,
DB and inductors, in one compact area. Use wide but
shortest possible traces for high current paths (e.g.,
VIN, VOUT , PGND etc.) to this area to minimize copper
loss.
3838fa
41
LTC3838
APPLICATIONS INFORMATION
DTR2
SENSE2–
SENSE2+
VFB2
RFB2(2)
CSS2
RFB1(2)
TRACK/SS2
RITH2(2)
CITH1(2)
RITH1(2)
ITH2
CITH2(2)
L2
VOUT2
DB2
DRVCC2
EXTVCC
MT2
CITH2(1)
MB2
RINTVCC
CERAMIC
CINTVCC
COUT2
PGND
CVIN
VIN
VRNG1
CDRVCC
VIN
RVIN
PGND
+
+
RT
RSENSE2
+
VRNG2
PHASMD
MODE/PLLIN
CLKOUT
SGND
RT
CIN
CERAMIC
COUT1
DRVCC1
RITH1(1)
DB1
ITH1
CITH1(1)
TRACK/SS1
RFB2(1)
CB2
TG2
SW2
BG2
INTVCC
LOCALIZED
SGND TRACE
RITH2(1)
RUN2
PGOOD2
BOOST2
CSS1
MT1
MB1
BG1
SW1
RSENSE1
VOUT1
TG1
VOUTSENSE1+
RFB1(1)
L1
CB1
BOOST1
VOUTSENSE1–
SENSE1+
SENSE1–
PGOOD1
RUN1
DTR1
BOLD LINES INDICATE HIGH SWITCHING CURRENT. KEEP LINES TO A MINIMUM LENGTH.
Figure 14. Recommended PCB Layout Diagram
• Keep the switch nodes (SW1,2), top gates (TG1,2) and
boost nodes (BOOST1,2) away from noise-sensitive
small-signal nodes, especially from the opposite
channel’s voltage and current sensing feedback pins.
These nodes have very large and fast moving signals
and therefore should be kept on the “output side” of
the LTC3838 (power-related pins are toward the right
hand side of the IC), and occupy minimum PC trace
area. Use compact switch node (SW) planes to improve
cooling of the MOSFETs and to keep EMI down. If DCR
sensing is used, place the top filter resistor (R1 only in
Figure 5) close to the switch node.
3838fa
42
LTC3838
APPLICATIONS INFORMATION
SW2
L2
RSENSE2
VOUT2
COUT2
RL2
VIN
RIN
CIN
SW1
L1
RSENSE1 VOUT1
COUT1
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
RL1
3838 F15
Figure 15. Branch Current Waveforms
• The top N-channel MOSFETs of the two channels have
to be located within a short distance from (preferably
<1cm) each other with a common drain connection at
CIN. Do not attempt to split the input decoupling for the
two channels as it can result in a large resonant loop.
• Connect the input capacitor(s), CIN, close to the power
MOSFETs. This capacitor provides the MOSFET transient
spike current. Connect the drain of the top MOSFET as
close as possible to the (+) plate of the ceramic portion
of input capacitors CIN. Connect the source of the bottom MOSFET as close as possible to the (–) terminal
of the same ceramic CIN capacitor(s). These ceramic
capacitor(s) bypass the high di/dt current locally, and
both top and bottom MOSFET should have short PCB
trace lengths to minimize high frequency EMI and
prevent MOSFET voltage stress from inductive ringing.
3838fa
43
LTC3838
APPLICATIONS INFORMATION
• The path formed by the top and bottom N-channel
MOSFETs, and the CIN capacitors should have short
leads and PCB trace. The (–) terminal of output capacitors should be connected close to the (–) terminal of
CIN, but away from the loop described above. This is
to achieve an effect of Kevin (4-wire) connection to the
input ground so that the “chopped” switching current
will not flow through the path between the input ground
and the output ground, and cause common mode output
voltage ripple.
• Use sufficient isolation when routing a clock signal into
the MODE/PLLIN pin or out of the CLKOUT pin, so that
the clock does not couple into sensitive pins.
• Several smaller sized ceramic output capacitors, COUT ,
can be placed close to the sense resistors and before
the rest bulk output capacitors.
• Place the ceramic decoupling capacitor CINTVCC between
the INTVCC pin and SGND and as close as possible to
the IC.
• The filter capacitor between the SENSE+ and SENSE– pins
should always be as close as possible to these pins.
Ensure accurate current sensing with Kevin (4-wire)
connections to the soldering pads from underneath
the sense resistors or inductor. A pair of sense traces
should be routed together with minimum spacing.
RSENSE, if used, should be connected to the inductor
on the noiseless output side, and its filter resistors
close to the SENSE+/SENSE– pins. For DCR sensing,
however, filter resistor should be placed close to the
inductor, and away from the SENSE+/SENSE– pins, as
its terminal is the SW node.
• Place the ceramic decoupling capacitor CDRVCC close
to the IC, between the combined DRVCC1,2 pins and
PGND.
• Keep small-signal components connected noise-sensitive pins (give priority to SENSE+/SENSE–, VOUTSENSE1+/
VOUTSENSE1–, VFB2, RT , ITH, VRNG pins) on the left hand
side of the IC as close to their respective pins as possible. This minimizes the possibility of noise coupling
into these pins. If the LTC3838 can be placed on the
bottom side of a multilayer board, use ground planes
to isolate from the major power components on the top
side of the board, and prevent noise coupling to noise
sensitive components on the bottom side.
• Place the resistor feedback divider RFB1, RFB2 close to
VOUTSENSE1+ and VOUTSENSE1– pins for Channel 1, or
VFB2 pin for Channel 2, so that the feedback voltage
tapped from the resistor divider will not be disturbed by
noise sources. Route remote sense PCB traces (use a
pair of wires closely together for differential sensing in
Channel 1) directly to the terminals of output capacitors
for best output regulation.
• Place decoupling capacitors CITH2 next to the ITH and
SGND pins with short, direct trace connections.
• Filter the VIN input to the LTC3838 with an RC filter.
Place the filter capacitor close to the VIN pin.
• If vias have to be used, use immediate vias to connect
components to the SGND and PGND planes of LTC3838.
Use multiple large vias for power components.
• Flood all unused areas on all layers with copper. Flooding
with copper will reduce the temperature rise of power
components. Connect the copper areas to DC rails only,
e.g., PGND.
PCB Layout Debugging
Only after each controller is checked for its individual
performance should both controllers be turned on at the
same time. It is helpful to use a current probe to monitor
the current in the inductor while testing the circuit. Monitor
the output switching node (SW pin) to synchronize the
oscilloscope to the internal oscillator output CLKOUT, or
external clock if used. Probe the actual output voltage as
well. Check for proper performance over the operating
voltage and current range expected in the application.
The frequency of operation should be maintained over
the input voltage range. The phase should be maintained
3838fa
44
LTC3838
APPLICATIONS INFORMATION
from cycle to cycle in a well designed, low noise PCB
implementation. Variation in the phase of SW node pulse
can suggest noise pickup at the current or voltage sensing
inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PCB layout if
regulator bandwidth optimization is not required.
Pay special attention to the region of operation when one
controller channel is turning on (right after its current
comparator trip point) while the other channel is turning
off its top MOSFET at the end of its on-time. This may
cause minor phase-lock jitter at either channel due to
noise coupling.
Reduce VIN from its nominal level to verify operation of
the regulator in dropout. Check the operation of the undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins.
The capacitor placed across the current sensing pins needs
to be placed immediately adjacent to the pins of the IC.
This capacitor helps to minimize the effects of differential
noise injection due to high frequency capacitive coupling.
If problems are encountered with high current output
loading at lower input voltages, look for inductive coupling
between CIN, top and bottom MOSFET components to the
sensitive current and voltage sensing traces.
In addition, investigate common ground path voltage pickup
between these components and the SGND pin of the IC.
High Switching Frequency Operation
At high switching frequencies there may be an increased
sensitivity to noise. Special care may need to be taken to
prevent cycle-by-cycle instability and/or phase-lock jitter.
First, carefully follow the recommended layout techniques
to reduce coupling from the high switching voltage/current
traces. Additionally, use low ESR and low impedance X5R
or X7R ceramic input capacitors: up to 5μF per Ampere of
load current may be needed. If necessary, increase ripple
sense voltage by increasing sense resistance value and
VRNG setting, to improve noise immunity.
3838fa
45
LTC3838
TYPICAL APPLICATIONS
VIN
4.5V TO 38V
+
CIN1
100μF
CIN2
10μF
w3
2.2Ω
LTC3838
1μF
VIN
SENSE1–
SENSE2–
SENSE1+
SENSE2+
BOOST1
BOOST2
0.1μF
15k
0.1μF
0.1μF
0.1μF
3.57k
L1
0.56μH
VOUT1
1.2V
15A
TG1
MT1
+
MT2
DB2
SW1
L2
0.56μH
VOUT2
1.5V
15A
SW2
DRVCC1
INTVCC
COUT2
330μF
w2
3.57k
TG2
DB1
2.2Ω
COUT1
100μF
w2
15k
DRVCC2
EXTVCC
COUT4 +
330μF
w2
4.7μF
1μF
BG1
MB1
BG2
MB2
COUT3
100μF
w2
PGND
10k
VOUTSENSE1+
15k
VFB2
10k
100k
PGOOD1
0.01μF
PGOOD1
220pF
115k
ITH1
DTR1
VRNG1
RT
SGND
RUN1
100
0.01μF
ITH2
220pF
DTR2
VRNG2
3838 F16a
PHASMD
MODE/PLLIN
CLKOUT
RUN2
90
1.0
60
0.5
50
VIN = 12V
VOUT = 1.2V
40
0.1
1
LOAD CURRENT (A)
10
3838 F16b
0
EFFICIENCY (%)
POWER
LOSS
3.0
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
2.5
2.0
80
EFFICIENCY
POWER
LOSS
70
1.5
1.0
60
POWER LOSS (W)
70
40.2k
100
POWER LOSS (W)
1.5
EFFICIENCY
CIN1: NICHICON UCJ1H101MCL165
CIN2: MURATA GRM32ER71H106K
COUT2, COUT4: SANYO 2R5TPE330M9
COUT1, COUT3: MURATA GRM31CR60J107ME39L
DB1, DB2: DIODES INC. SDM10K45
L1, L2: TOKO FDA1055-R56M
MT1, MT2: INFINEON BSC093N04LSG
MB1, MB2: INFINEON BSC035N04LSG
22pF
2.0
80
100k
PGOOD2
2.5
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
90
EFFICIENCY (%)
PGOOD2
TRACK/SS1 TRACK/SS2
22pF
40.2k
10k
VOUTSENSE1–
0.5
50
VIN = 12V
VOUT = 1.5V
40
0.1
1
LOAD CURRENT (A)
0
10
3838 F16c
Figure 16. 4.5V to 38V Input, 1.2V/15A and 1.5V/15A Dual Output, 350kHz, DCR Sense, Step-Down Converter
3838fa
46
LTC3838
TYPICAL APPLICATIONS
VIN
6V TO 26V
+
CIN1
220μF
CIN2
10μF
w3
2.2Ω
LTC3838
1μF
VIN
100Ω
SENSE1–
SENSE2–
SENSE1+
SENSE2+
100Ω
1nF
100Ω
1nF
0.1μF
BOOST1
RS1
0.002Ω
VOUT1
1.2V
12A
L1
0.47μH
MT1
+
COUT2
330μF
w2
BOOST2
TG1
MT2
TG2
DB1
DB2
SW1
2.2Ω
COUT1
100μF
w2
100Ω
0.1μF
DRVCC2
EXTVCC
4.7μF
MB1
BG1
MB2
BG2
PGND
10k
VOUTSENSE1+
0.01μF
39.2k
60.4k
220pF
115k
10k
+
COUT4
100μF
w2
10k
PGOOD1
PGOOD2
TRACK/SS1 TRACK/SS2
22pF
COUT3
330μF
w2
15k
VOUTSENSE1–
PGOOD1
VOUT2
1.5V
12A
VFB2
10k
100k
RS2
0.002Ω
SW2
DRVCC1
INTVCC
1μF
L2
0.47μH
ITH1
DTR1
VRNG1
RT
SGND
RUN1
ITH2
100k
PGOOD2
0.01μF
CIN1: PANASONIC EEEFK1V221P
CIN2: TAIYO YUDEN GMK325BJ106MN-T
COUT1, COUT4: MURATA GRM31CR60J107ME39L
COUT2, COUT3: SANYO 2R5TPE330M9
DB1, DB2: CENTRAL SEMI CMDSH-4ETR
L1, L2: WÜRTH 7443330047
MT1, MT2: RENESAS RJK0305DPB
MB1, MB2: RENESAS RJK0330DPB
22pF
220pF
39.2k
DTR2
VRNG2
PHASMD
MODE/PLLIN
CLKOUT
RUN2
10k
60.4k
3838 F17
Figure 17. 6V to 26V Input, 1.2V/15A and 1.5V/15A Dual Output, 350kHz, RSENSE, Step-Down Converter
3838fa
47
LTC3838
TYPICAL APPLICATIONS
VIN
4.5V TO 14V
+
CIN2
22μF
w4
CIN1
180μF
2.2Ω
LTC3838
1μF
VIN
SENSE1–
SENSE2–
SENSE1+
SENSE2+
0.1μF
0.1μF
0.1μF
0.1μF
BOOST1
2.55k
L1
0.36μH
VOUT
1.2V
50A
MT1
+
COUT2
330μF
w2
2.55k
MT2
TG2
DB1
L2
0.36μH
DB2
SW1
2.2Ω
COUT1
100μF
w2
BOOST2
TG1
SW2
DRVCC1
INTVCC
DRVCC2
EXTVCC
COUT3 +
330μF
w2
4.7μF
1μF
MB1
BG1
MB2
BG2
COUT4
100μF
w2
PGND
10k
VOUTSENSE1+
VFB2
10k
VOUTSENSE1–
100k
PGOOD
0.01μF
PGOOD2
TRACK/SS1 TRACK/SS2
47pF
41.2k
PGOOD1
470pF
ITH1
ITH2
35.7k
CIN1: SANYO 16SVP180MX
CIN2: TDK C3225X5R1C226MT
COUT1, COUT4: MURATA GRM31CR60J107ME39L
COUT2, COUT3: SANYO 2R5TPE330M9
DB1, DB2: CENTRAL SEMI CMDSH-4ETR
L1, L2: VISHAY IHLP4040DZERR36M01
MT1, MT2: INFINEON BSC050NE2LS
MB1, MB2: INFINEON BSC010NE2LS
RT
SGND
RUN1
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
8
90
70
4
60
0.1
1
10
VIN = 12V LOAD CURRENT (A)
VOUT = 1.2V
3838 F18b
10
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
8
6
80
EFFICIENCY
POWER
LOSS
70
4
2
60
2
0
50
POWER LOSS (W)
POWER
LOSS
50
3838 F18
PHASMD
MODE/PLLIN
CLKOUT
RUN2
100
6
80
DTR2
VRNG2
10
POWER LOSS (W)
EFFICIENCY (%)
90
EFFICIENCY
DTR1
VRNG1
EFFICIENCY (%)
100
137k
0
0.1
1
10
LOAD CURRENT (A)
VIN = 5V
VOUT = 1.2V
3838 F18c
Figure 18. 4.5V to 14V Input, 1.2V/50A 2-Phase Single Output, 300kHz, DCR Sense, DTR Enabled, Step-Down Converter
3838fa
48
LTC3838
TYPICAL APPLICATIONS
VIN
6.5V TO 34V
+
CIN2
10μF
w3
CIN1
220μF
2.2Ω
LTC3838
1μF
VIN
20Ω
SENSE1–
SENSE2–
SENSE1+
SENSE2+
BOOST1
BOOST2
20Ω
1nF
20Ω
1nF
0.1μF
0.1μF
RS1
0.002Ω
VOUT1
5V
12A
L1
2.2μH
MT1
TG1
+
COUT2
150μF
w2
MT2
TG2
DB1
DB2
SW1
2.2Ω
COUT1
100μF
20Ω
L2
1.3μH
DRVCC1
INTVCC
DRVCC2
EXTVCC
VOUT1
COUT3
330μF
4.7μF
1μF
MB1
RS2
0.002Ω
VOUT2
3.3V
12A
SW2
BG1
+
COUT4
100μF
MB2
BG2
PGND
73.2k
VOUTSENSE1+
45.3k
VFB2
10k
10k
VOUTSENSE1–
100k
PGOOD1
0.01μF
PGOOD1
ITH1
RT
SGND
RUN1
45.3k
DTR2
VRNG2
PHASMD
MODE/PLLIN
CLKOUT
RUN2
3838 F19
3.0
100
3.0
90
2.5
90
2.5
EFFICIENCY
70
1.5
POWER
LOSS
1.0
60
50
VIN = 12V
VOUT = 5V
40
0.1
1
LOAD CURRENT (A)
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
10
3838 F19b
2.0
80
EFFICIENCY
70
1.5
POWER
LOSS
1.0
60
0.5
50
0
40
VIN = 12V
VOUT = 3.3V
0.1
1
LOAD CURRENT (A)
POWER LOSS (W)
2.0
80
EFFICIENCY (%)
100
POWER LOSS (W)
EFFICIENCY (%)
22pF
220pF
DTR1
VRNG1
CIN1: PANASONIC EEEFK1V221P
CIN2: TAIYO YUDEN GMK325BJ106MN-T
COUT1, COUT4: MURATA GRM31CR60J107ME39L
COUT2: SANYO 6TPE150MIC2
COUT3: SANYO 4TPD330M
DB1, DB2: DIODES INC. SDM10K45
L1: WÜRTH 7443320220
L2: WÜRTH 7443551130
MT1, MT2: INFINEON BSC093N04LSG
MB1, MB2: INFINEON BSC035N04LSG
0.01μF
ITH2
220pF
137k
100k
PGOOD2
TRACK/SS1 TRACK/SS2
22pF
53.6k
PGOOD2
0.5
0
10
3838 F19c
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
Figure 19. 6.5V to 34V Input, 5V/12A and 3.3V/12A Dual Output, 300kHz, RSENSE, 5V Output Tied to EXTVCC, Step-Down Converter
3838fa
49
LTC3838
TYPICAL APPLICATIONS
VIN
7V TO 14V
+
CIN1
39μF
CIN2
10μF
w3
2.2Ω
LTC3838
1μF
VIN
10Ω
SENSE1–
SENSE2–
SENSE1+
SENSE2+
10Ω
1nF
10Ω
1nF
0.1μF
BOOST1
VOUT1
5V
5A
RS1
0.008Ω
L1
0.8μH
MT1
BOOST2
TG1
MT2
TG2
DB1
L2
0.8μH
DB2
SW1
2.2Ω
COUT1
47μF
w2
10Ω
0.1μF
SW2
DRVCC1
INTVCC
DRVCC2
EXTVCC
MB1
BG1
MB2
BG2
VOUT2
3.3V
5A
COUT2
47μF
w2
VOUT1
4.7μF
1μF
RS2
0.008Ω
PGND
73.2k
330pF
18.7k
ITH1
ITH2
DTR1
VRNG1
RT
SGND
RUN1
100
2.0
POWER
LOSS
0.01μF
22pF
330pF
CIN1: SANYO 16SVP39M
CIN2: MURATA GRM32DR61E106K
COUT1, COUT2: TAIYO YUDEN LMK325BJ476MM-T
DB1, DB2: CENTRAL CMDSH-3
L1, L2: COILCRAFT XAL5030-801MEB
MT1, MB1, MT2, MB2: VISHAY SI7114ADN
35.7k
DTR2
VRNG2
3838 F20a
PHASMD
MODE/PLLIN
CLKOUT
RUN2
100
90
2.0
90
2.0
80
1.5
80
1.5
EFFICIENCY
70
1.0
60
0.5
60
50
0.01
0.1
1
VIN = 12V LOAD CURRENT (A)
VOUT = 5V
EXTVCC TIED TO VOUT
0
0
50
0.01
0.1
1
VIN = 12V LOAD CURRENT (A)
VOUT = 3.3V
3838 F20c
EXTVCC TIED TO 5V BIAS SUPPLY
EFFICIENCY MEASUREMENTS INCLUDE POWER
FROM THE 5V BIAS SUPPLY
3838 F20b
2.5
2.5
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
1.0
70
POWER
LOSS
0.5
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
EFFICIENCY
1.0
70
60
POWER
LOSS
50
0.01
0.1
1
VIN = 12V LOAD CURRENT (A)
VOUT = 3.3V
EXTVCC FLOATING
POWER LOSS (W)
EFFICIENCY
100k
PGOOD2
POWER LOSS (W)
1.5
80
EFFICIENCY (%)
2.5
POWER LOSS (W)
EFFICIENCY (%)
90
PGOOD2
EFFICIENCY (%)
24.9k
PGOOD1
–
TRACK/SS1 TRACK/SS2
22pF
68pF
10k
VOUTSENSE1
PGOOD1
0.01μF
FORCED CONTINUOUS MODE
DISCONTINUOUS MODE
VFB2
10k
100k
100
45.3k
VOUTSENSE1+
47pF
0.5
0
3838 F20d
Figure 20. 7V to 14V Input, 5V/5A and 3.3V/5A Dual Output, 2MHz, RSENSE, Step-Down Converter with EXTVCC Tied to 5V Output
3838fa
50
LTC3838
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UHF Package
38-Lead Plastic QFN (5mm × 7mm)
(Reference LTC DWG # 05-08-1701 Rev C)
0.70 p 0.05
5.50 p 0.05
5.15 ± 0.05
4.10 p 0.05
3.00 REF
3.15 ± 0.05
PACKAGE
OUTLINE
0.25 p 0.05
0.50 BSC
5.5 REF
6.10 p 0.05
7.50 p 0.05
RECOMMENDED SOLDER PAD LAYOUT
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
5.00 p 0.10
0.75 p 0.05
PIN 1 NOTCH
R = 0.30 TYP OR
0.35 s 45o CHAMFER
3.00 REF
37
0.00 – 0.05
38
0.40 p0.10
PIN 1
TOP MARK
(SEE NOTE 6)
1
2
5.15 ± 0.10
7.00 p 0.10
5.50 REF
3.15 ± 0.10
(UH) QFN REF C 1107
0.200 REF 0.25 p 0.05
0.50 BSC
R = 0.125
TYP
R = 0.10
TYP
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE
OUTLINE M0-220 VARIATION WHKD
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3838fa
51
LTC3838
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
FE Package
38-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1772 Rev C)
Exposed Pad Variation AA
4.75 REF
38
9.60 – 9.80*
(.378 – .386)
4.75 REF
(.187)
20
6.60 ±0.10
2.74 REF
4.50 REF
SEE NOTE 4
6.40
2.74
REF (.252)
(.108)
BSC
0.315 ±0.05
1.05 ±0.10
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
2. DIMENSIONS ARE IN MILLIMETERS
(INCHES)
3. DRAWING NOT TO SCALE
1
0.25
REF
19
1.20
(.047)
MAX
0s – 8s
0.50
(.0196)
BSC
0.17 – 0.27
(.0067 – .0106)
TYP
0.05 – 0.15
(.002 – .006)
FE38 (AA) TSSOP REV C 0910
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3838fa
52
LTC3838
REVISION HISTORY
REV
DATE
DESCRIPTION
PAGE NUMBER
A
6/12
Electrical specs clarification, 4.6V EXTVCC switch over
3, 4, 5, 13
3838fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
53
LTC3838
TYPICAL APPLICATION
4.5V to 26V Input, 2.5V/4A and 1.8V/4A Dual Output, 1MHz, RSENSE, Dual Channel Power FETs, Step-Down Converter
VIN
4.5V TO 26V
+
CIN1
47μF
CIN2
10μF
w3
2.2Ω
LTC3838
1μF
VIN
20Ω
20Ω
SENSE1–
SENSE2–
SENSE1+
SENSE2+
1nF
20Ω
1nF
0.1μF
BOOST1
M1
VOUT1
2.5V
4A
RS1
0.006Ω
BOOST2
TG1
L1
1.2μH
M2
TG2
DB1
L2
1.2μH
DB2
SW1
2.2Ω
SW2
DRVCC1
INTVCC
COUT1
100μF
w2
20Ω
0.1μF
DRVCC2
EXTVCC
BG1
VOUT2
1.8V
4A
COUT2
100μF
w2
4.7μF
1μF
RS2
0.006Ω
BG2
PGND
31.6k
20k
VOUTSENSE1+
VFB2
10k
10k
VOUTSENSE1–
100k
PGOOD1
0.01μF
75k
100pF
PGOOD1
PGOOD2
TRACK/SS1 TRACK/SS2
ITH1
39.2k
SGND
RUN1
0.01μF
100pF
61.9k
ITH2
DTR1
VRNG1
RT
CIN1: SUN CON 35HVP47M
CIN2: TAIYO YUDEN GMK325BJ106MN-T
COUT1,COUT2: MURATA GRM31CR60J107ME39L
COUT2, COUT4: SANYO 4TPC150M
DB1, DB2: CENTRAL SEMI CMDSH-4ETR
L1, L2: COILCRAFT XAL5030-122MEB
M1, M2: SILICONIX SI4816DY
100k
PGOOD2
DTR2
VRNG2
3838 TA02
PHASMD
MODE/PLLIN
CLKOUT
RUN2
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC3833
Fast Controlled On-Time, High Frequency Synchronous
Step-Down Controller with Diff Amp
200kHz to 2MHz Operating Frequency, 4.5V ≤ VIN ≤ 38V,
0.6V ≤ VOUT ≤ 5.5V, , 3mm × 4mm QFN-20, TSSOP-20
LTC3880/LTC3880-1 Dual Output PolyPhase Step-Down DC/DC Controller with
Digital Power System Management
I2C/PMBus Interface with EEPROM and 16-Bit ADC, VIN Up to 24V,
0.5V ≤ VOUT ≤ 5.5V, Analog Control Loop
LTC3869/LTC3869-2 Dual Output, 2-Phase Synchronous Step-Down DC/DC
Controller, with Accurate Current Sharing
PLL Fixed 250kHz to 750kHz Frequency, 4V ≤ VIN ≤ 38V,
VOUT3 Up to 12.5V
LTC3855
Dual Output, 2-phase, Synchronous Step-Down DC/DC
PLL Fixed Frequency 250kHz to 770kHz, 4.5V ≤ VIN ≤ 38V,
Controller with Diff Amp and DCR Temperature Compensation 0.8V ≤ VOUT ≤ 12V
LTC3856
Single Output 2-Channel Synchronous Step-Down DC/DC
Controller with Diff Amp and Up to 12-Phase Operation
PLL Fixed 250kHz to 770kHz Frequency, 4.5V ≤ VIN ≤ 38V,
0.8V ≤ VOUT ≤ 5V
LTC3860
Dual, Multiphase, Synchronous Step-Down DC/DC Controller
with Diff Amp and Three-State Output Drive
Operates with Power Blocks, DRMOS Devices or External Drivers/
MOSFETs, 3V ≤ VIN ≤ 24V, tON(MIN) = 20ns
LTC3850/LTC3850-1 Dual Output, 2-Phase Synchronous Step-Down DC/DC
LTC3850-2
Controller, RSENSE or DCR Current Sensing
PLL Fixed 250kHz to 780kHz Frequency, 4V ≤ VIN ≤ 30V,
0.8V ≤ VOUT ≤ 5.25V
LTC3829
Single Output 3-Channel Synchronous Step-Down DC/DC with Controller Phase-Lockable Fixed 250kHz to 770kHz Frequency,
Diff Amp and Up to 6-phase Operation
4.5V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 5V
LTC3853
Triple Output, Multiphase Synchronous Step-Down DC/DC
Controller, RSENSE or DCR Current Sensing and Tracking
PLL Fixed 250kHz to 750kHz Frequency, 4V ≤ VIN ≤ 24V,
VOUT3 Up to 13.5V
3838fa
54 Linear Technology Corporation
LT 0612 REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2011
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