Continuous Rate 12.3 Mb/s to 675 Mb/s Clock and Data Recovery IC with Integrated Limiting Amp ADN2814 Preliminary Technical Data FEATURES PRODUCT DESCRIPTION Serial data input: 12.3 Mb/s to 675 Mb/s Exceeds SONET requirements for jitter transfer/ generation/tolerance Quantizer sensitivity: 6 mV typical Adjustable slice level: ±100 mV Patented clock recovery architecture Loss of signal (LOS) detect range: 3 mV to 15 mV Independent slice level adjust and LOS detector No reference clock required Loss of lock indicator I2C™ interface to access optional features Single-supply operation: 3.3 V Low power: 350 mW typical 5 mm × 5 mm 32-lead LFCSP, Pb Free The ADN2814 provides the receiver functions of quantization, signal level detect, and clock and data recovery for continuous data rates from 12.3 Mb/s to 675 Mb/s. The ADN2814 automatically locks to all data rates without the need for an external reference clock or programming. All SONET jitter requirements are met, including jitter transfer, jitter generation, and jitter tolerance. All specifications are quoted for −40°C to +85°C ambient temperature, unless otherwise noted. This device, together with a PIN diode and a TIA preamplifier, can implement a highly integrated, low cost, low power fiber optic receiver. The receiver front end loss of signal (LOS) detector circuit indicates when the input signal level has fallen below a useradjustable threshold. The LOS detect circuit has hysteresis to prevent chatter at the output. APPLICATIONS SONET OC-1/3/12 and all associated FEC rates ESCON, Fast Ethernet, Serial Digital Interface (DTV) WDM transponders Regenerators/repeaters Test equipment Broadband cross-connects and routers The ADN2814 is available in a compact 5 mm × 5 mm 32-lead chip scale package. FUNCTIONAL BLOCK DIAGRAM REFCLKP/N (OPTIONAL) SLICEP/N LOL 2 CF1 CF2 FREQUENCY DETECT LOOP FILTER PHASE DETECT LOOP FILTER VCC VEE PIN NIN QUANTIZER PHASE SHIFTER VCO VREF LOS DETECT DATA RE-TIMING 2 THRADJ LOS DATAOUTP/N CLKOUTP/N 04949-0-001 2 Figure 1. Rev. PrA Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.326.8703 © 2004 Analog Devices, Inc. All rights reserved. ADN2814 Preliminary Technical Data TABLE OF CONTENTS Specifications..................................................................................... 3 Functional Description.................................................................. 16 Jitter Specifications....................................................................... 4 Frequency Acquisition............................................................... 16 Output and Timing Specifications ............................................. 5 Limiting Amplifier ..................................................................... 16 Absolute Maximum Ratings............................................................ 6 Slice Adjust.................................................................................. 16 Thermal Characteristics .............................................................. 6 Loss of Signal (LOS) Detector .................................................. 16 ESD Caution.................................................................................. 6 Lock Detector Operation .......................................................... 16 Timing Characteristics..................................................................... 7 Harmonic Detector .................................................................... 17 Pin Configuration and Function Descriptions............................. 8 Squelch Mode ............................................................................. 17 Typical Performance Characteristics ............................................. 9 I2C Interface ................................................................................ 18 I2C Interface Timing and Internal Register Description........... 10 Reference Clock (Optional) ...................................................... 18 Terminology .................................................................................... 12 Applications Information .............................................................. 21 Jitter Specifications ......................................................................... 13 PCB Design Guidelines ............................................................. 21 Jitter Generation ......................................................................... 13 DC-Coupled Application .......................................................... 23 Jitter Transfer............................................................................... 13 Coarse Data Rate Readback Look-Up Table............................... 24 Jitter Tolerance ............................................................................ 13 Outline Dimensions ....................................................................... 26 Theory of Operation ...................................................................... 14 Ordering Guide .......................................................................... 26 REVISION HISTORY 6/04—Revision PrA: Initial Version Rev. PrA | Page 2 of 28 Preliminary Technical Data ADN2814 SPECIFICATIONS TA = TMIN to TMAX, VCC = VMIN to VMAX, VEE = 0 V, CF = 0.47 µF, SLICEP = SLICEN = VEE, Input Data Pattern: PRBS 223 − 1, unless otherwise noted. Table 1. Parameter QUANTIZER—DC CHARACTERISTICS Input Voltage Range Peak-to-Peak Differential Input Input Common Mode Level Differential Input Sensitivity Input Overdrive Input Offset Input RMS Noise QUANTIZER—AC CHARACTERISTICS Data Rate S11 Input Resistance Input Capacitance QUANTIZER—SLICE ADJUSTMENT Gain Differential Control Voltage Input Control Voltage Range Slice Threshold Offset LOSS OF SIGNAL DETECT (LOS) Loss of Signal Detect Range (see Figure 5) Hysteresis (Electrical) LOS Assert Time LOS De-Assert Time LOSS OF LOCK DETECT (LOL) VCO Frequency Error for LOL Assert VCO Frequency Error for LOL De-Assert LOL Response Time ACQUISITION TIME Lock to Data Mode Optional Lock to REFCLK Mode DATA RATE READBACK ACCURACY Coarse Readback 1 2 Conditions Min @ PIN or NIN, dc-coupled PIN – NIN DC-coupled (see Figure 26, Figure 27, and Figure 28) 223 − 1 PRBS, ac-coupled,1 BER = 1 x 10–10 (see Figure 11) 1.8 Typ 2.3 2.5 10 5 6 3 500 290 BER = 1 x 10–10 12.3 @ 2.5 GHz Differential SLICEP – SLICEN = ±0.5 V SLICEP – SLICEN DC level @ SLICEP or SLICEN Max Unit 2.8 2.0 V V 2.8 V mV p-p mV p-p µV µV rms 675 Mb/s dB Ω pF 0.12 +0.95 0.95 V/V V V mV −15 100 0.65 0.08 –0.95 VEE 0.1 1 RThresh = 0 Ω RThresh = 100 kΩ GbE RThresh = 0 Ω RThresh = 100 kΩ OC-1 RThresh = 0 Ω RThresh = 10 kΩ DC-coupled2 DC-coupled2 12 2.0 15 3.0 17 4.0 mV mV 5.0 3.0 6 6 7.0 8.0 dB dB 5.6 2.0 6 4 500 400 7.2 6.7 dB dB ns ns With respect to nominal With respect to nominal 12.3 Mb/s OC-12 1000 250 4 1.0 ppm ppm ms µs OC-12 OC-3 OC-1 12.3 Mb/s 2.0 3.4 9.8 40.0 10.0 ms ms ms ms ms (See Table 13) 10 % PIN and NIN should be differentially driven and ac-coupled for optimum sensitivity. When ac-coupled, the LOS assert and de-assert time is dominated by the RC time constant of the ac coupling capacitor and the 50 Ω input termination of the ADN2814 input stage. Rev. PrA | Page 3 of 28 ADN2814 Parameter Fine Readback Preliminary Technical Data Conditions In addition to REFCLK accuracy Data rate < 20 Mb/s Data rate > 20 Mb/s POWER SUPPLY VOLTAGE POWER SUPPLY CURRENT OPERATING TEMPERATURE RANGE Min 3.0 Typ 3.3 106 –40 Max Unit 200 100 3.6 ppm ppm V mA °C +85 JITTER SPECIFICATIONS TA = TMIN to TMAX, VCC = VMIN to VMAX, VEE = 0 V, CF = 0.47 uF, SLICEP = SLICEN = VEE, Input Data Pattern: PRBS 223 − 1, unless otherwise noted. Table 2. Parameter PHASE-LOCKED LOOP CHARACTERISTICS Jitter Transfer BW Jitter Peaking Jitter Generation Conditions Min OC-12 OC-3 OC-12 OC-3 OC-12, 12 kHz to 5 MHz OC-3, 12 kHz to 1.3 MHz Jitter Tolerance 1 OC-12, 223 − 1 PRBS 30 Hz1 300 Hz1 25 kHz 250 kHz1 OC-3, 223 − 1 PRBS 30 Hz1 300 Hz1 6500 Hz 65 kHz Max Unit 71 23 0 0 0.001 0.01 0.001 0.01 108 35 0.03 0.03 0.002 0.019 0.002 0.011 kHz kHz dB dB UI rms UI p-p UI rms UI p-p 100 44 2.5 1.0 UI p-p UI p-p UI p-p UI p-p 50 24 3.5 1.0 UI p-p UI p-p UI p-p UI p-p Jitter tolerance of the ADN2814 at these jitter frequencies is better than what the test equipment is able to measure. Rev. PrA | Page 4 of 28 Typ Preliminary Technical Data ADN2814 OUTPUT AND TIMING SPECIFICATIONS Table 3. Parameter LVDS OUPUT CHARACTERISTICS (CLKOUTP/N, DATAOUTP/N) Single-Ended Output Swing Differential Output Swing Output Offset Voltage Output Impedance LVDS Ouputs Timing Rise Time Fall Time Setup Time Hold Time 2 I C INTERFACE DC CHARACTERISTICS Input High Voltage Input Low Voltage Input Current Output Low Voltage I2C INTERFACE TIMING SCK Clock Frequency SCK Pulse Width High SCK Pulse Width Low Start Condition Hold Time Start Condition Setup Time Data Setup Time Data Hold Time SCK/SDA Rise/Fall Time Stop Condition Setup Time Bus Free Time between a Stop and a Start REFCLK CHARACTERISTICS Input Voltage Range Minimum Differential Input Drive Reference Frequency Required Accuracy LVTTL DC INPUT CHARACTERISTICS Input High Voltage Input Low Voltage Input High Current Input Low Current LVTTL DC OUTPUT CHARACTERISTICS Output High Voltage Output Low Voltage 1 Conditions Min VSE (see Figure 3) VDIFF (see Figure 3) 250 500 1125 Differential 20% to 80% 80% to 20% TS (see Figure 2), OC12 TH (see Figure 2), OC12 LVCMOS VIH VIL VIN = 0.1 VCC or VIN = 0.9 VCC VOL, IOL = 3.0 mA (See Figure 10) Typ 1200 100 Max Unit 400 800 1275 mV mV mV Ω TBD TBD 800 800 ps ps ps ps 0.7 VCC 0.3 VCC +10.0 0.4 −10.0 400 tHIGH tLOW tHD;STA tSU;STA tSU;DAT tHD;DAT TR/TF tSU;STO tBUF Optional lock to REFCLK mode @ REFCLKP or REFCLKN VIL VIH 600 1300 600 600 100 300 20 + 0.1 Cb1 600 1300 300 0 VCC 100 12.3 200 100 VIH VIL IIH, VIN = 2.4 V IIL, VIN = 0.4 V 2.0 VOH, IOH = −2.0 mA VOL, IOL = 2.0 mA 2.4 0.8 5 −5 Cb = total capacitance of one bus line in pF. If mixed with Hs-mode devices, faster fall-times are allowed. Rev. PrA | Page 5 of 28 0.4 V V µA V kHz ns ns ns ns ns ns ns ns ns V V mV p-p MHz ppm V V µA µA V V ADN2814 Preliminary Technical Data ABSOLUTE MAXIMUM RATINGS TA = TMIN to TMAX, VCC = VMIN to VMAX, VEE = 0 V, CF = 0.47 µF, SLICEP = SLICEN = VEE, unless otherwise noted. Table 4. Parameter Supply Voltage (VCC) Minimum Input Voltage (All Inputs) Maximum Input Voltage (All Inputs) Maximum Junction Temperature Storage Temperature Lead Temperature (Soldering 10 s) Rating 4.2 V VEE − 0.4 V VCC + 0.4 V 125°C −65°C to +150°C 300°C Stress above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. THERMAL CHARACTERISTICS Thermal Resistance 32-LFCSP, 4-layer board with exposed paddle soldered to VEE θJA = 28°C/W. ESD CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. Rev. PrA | Page 6 of 28 Preliminary Technical Data ADN2814 TIMING CHARACTERISTICS CLKOUTP TH 04949-0-002 TS DATAOUTP/N Figure 2. Output Timing OUTP VCML VSE OUTN OUTP–OUTN VDIFF 04949-0-003 VSE 0V Figure 3. Single-Ended vs. Differential Output Specifications Rev. PrA | Page 7 of 28 ADN2814 Preliminary Technical Data 32 TEST2 31 VCC 30 VEE 29 DATAOUTP 28 DATAOUTN 27 SQUELCH 26 CLKOUTP 25 CLKOUTN PIN CONFIGURATION AND FUNCTION DESCRIPTIONS ADN2814* TOP VIEW (Not to Scale) 24 VCC 23 VEE 22 LOS 21 SDA 20 SCK 19 SADDR5 18 VCC 17 VEE 04949-0-004 PIN 1 INDICATOR THRADJ 9 REFCLKP 10 REFCLKN 11 VCC 12 VEE 13 CF2 14 CF1 15 LOL 16 TEST1 1 VCC 2 VREF 3 NIN 4 PIN 5 SLICEP 6 SLICEN 7 VEE 8 * THERE IS AN EXPOSED PAD ON THE BOTTOM OF THE PACKAGE THAT MUST BE CONNECTED TO GND. Figure 4. Pin Configuration Table 5. Pin Function Descriptions Pin No. 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 Exposed Pad Mnemonic TEST1 VCC VREF NIN PIN SLICEP SLICEN VEE THRADJ REFCLKP REFCLKN VCC VEE CF2 CF1 LOL VEE VCC SADDR5 SCK SDA LOS VEE VCC CLKOUTN CLKOUTP SQUELCH DATAOUTN DATAOUTP VEE VCC TEST2 Pad Type1 P AO AI AI AI AI P AI DI DI P P AO AO DO P P DI DI DI DO P P DO DO DI DO DO P P P Description Connect to VCC. Power for Limamp, LOS. Internal VREF Voltage. Decouple to GND with a 0.1 µF capacitor. Differential Data Input. CML. Differential Data Input. CML. Differential Slice Level Adjust Input. Differential Slice Level Adjust Input. GND for Limamp, LOS. LOS Threshold Setting Resistor. Differential REFCLK Input. 12.3 MHz to 200 MHz. Differential REFCLK Input. 12.3 MHz to 200 MHz. VCO Power. VCO GND. Frequency Loop Capacitor. Frequency Loop Capacitor. Loss of Lock Indicator. LVTTL active high. FLL Detector GND. FLL Detector Power. Slave Address Bit 5. I2C Clock Input. I2C Data Input. Loss of Signal Detect Output. Active high. LVTTL. Output Buffer, I2C GND. Output Buffer, I2C Power. Differential Recovered Clock Output. LVDS. Differential Recovered Clock Output. LVDS. Disable Clock and Data Outputs. Active high. LVTLL. Differential Recovered Data Output. LVDS. Differential Recovered Data Output. LVDS. Phase Detector, Phase Shifter GND. Phase Detector, Phase Shifter Power. Connect to VCC. Connect to GND 1 Type: P = power, AI = analog input, AO = analog output, DI = digital input, DO = digital output. Rev. PrA | Page 8 of 28 Preliminary Technical Data ADN2814 TYPICAL PERFORMANCE CHARACTERISTICS 16 12 10 8 6 4 2 04949-0-005 TRIP POINT (mV p-p) 14 1 10 100 1k 10k 100k RTH (Ω) Figure 5. LOS Comparator Trip Point Programming Rev. PrA | Page 9 of 28 ADN2814 Preliminary Technical Data I2C INTERFACE TIMING AND INTERNAL REGISTER DESCRIPTION R/W CTRL. SLAVE ADDRESS [6...0] A5 SET BY PIN 19 0 0 0 0 0 X 04949-0-007 1 MSB = 1 0 = WR 1 = RD S SLAVE ADDR, LSB = 0 (WR) A(S) SUB ADDR A(S) DATA A(S) DATA A(S) P 04949-0-008 Figure 6. Slave Address Configuration S SLAVE ADDR, LSB = 0 (WR) A(S) SUB ADDR A(S) S SLAVE ADDR, LSB = 1 (RD) A(S) DATA A(M) S = START BIT A(S) = ACKNOWLEDGE BY SLAVE DATA A(M) P P = STOP BIT A(M) = LACK OF ACKNOWLEDGE BY MASTER A(M) = ACKNOWLEDGE BY MASTER 04949-0-009 Figure 7. I2C Write Data Transfer Figure 8. I2C Read Data Transfer SDA SLAVE ADDRESS A6 SUB ADDRESS A5 STOP BIT DATA A7 A0 D7 D0 SCK S WR ACK ACK SLADDR[4...0] ACK SUB ADDR[6...1] DATA[6...1] Figure 9. I2C Data Transfer Timing tF tSU;DAT tHD;STA tBUF SDA tR tR tSU;STO tF tLOW tHIGH tHD;STA S tSU;STA tHD;DAT S Figure 10. I2C Port Timing Diagram Rev. PrA | Page 10 of 28 P S 04949-0-011 SCK P 04949-0-010 START BIT Preliminary Technical Data ADN2814 Table 6. Internal Register Map1 1 Reg Name FREQ0 FREQ1 FREQ2 RATE R/W R R R R Address 0x0 0x1 0x2 0x3 D7 D6 MSB MSB 0 MSB COARSE_RD[8] MSB MISC R 0x4 CTRLA W 0x8 CTRLB W 0x9 Config LOL Reset MISC[4] System reset 0 Reset MISC[2] CTRLC W 0x11 0 0 0 0 0 x D5 D4 D3 D2 D1 D0 LSB LSB LSB COARSE_RD[1] Datarate measure complete x COARSE_ RD[0] LSB Measure datarate Lock to reference 0 0 0 Config LOS Squelch mode 0 Coarse datarate readback LOS status x FREF range Static LOL LOL status Datarate/DIV_FREF ratio All writeable registers default to 0x00. Table 7. Miscellaneous Register, MISC D7 x D6 x LOS Status D5 0 = No loss of signal 1 = Loss of signal Static LOL D4 0 = Waiting for next LOL 1 = Static LOL until reset LOL Status D3 0 = Locked 1 = Acquiring Datarate Measurement Complete D2 0 = Measuring datarate 1 = Measurement complete D1 x Coarse Rate Readback LSB D0 COARSE_RD[0] Table 8. Control Register, CTRLA1 FREF Range D7 D6 0 0 0 1 1 0 1 1 1 12.3 MHz to 25 MHz 25 MHz to 50 MHz 50 MHz to 100 MHz 100 MHz to 200 MHz Datarate/Div_FREF Ratio D5 D4 D3 D2 0 0 0 0 0 0 0 1 0 0 1 0 n 1 0 0 0 1 2 4 2n 256 Measure Datarate D1 Set to 1 to measure datarate Lock to Reference D0 0 = Lock to input data 1 = Lock to reference clock Where DIV_FREF is the divided down reference referred to the 12.3 MHz to 25 MHz band (see the Reference Clock (Optional) section). Table 9. Control Register, CTRLB Config LOL D7 0 = LOL pin normal operation 1 = LOL pin is static LOL Reset MISC[4] D6 Write a 1 followed by 0 to reset MISC[4] System Reset D5 Write a 1 followed by 0 to reset ADN2814 D4 Set to 0 Reset MISC[2] D3 Write a 1 followed by 0 to reset MISC[2] D2 Set to 0 D1 Set to 0 D0 Set to 0 Table 10. Control Register, CTRLC D7 Set to 0 D6 Set to 0 D5 Set to 0 D4 Set to 0 D3 Set to 0 Config LOS D2 0 = Active high LOS 1 = Active low LOS Rev. PrA | Page 11 of 28 Squelch Mode D1 0 = Squelch CLK and DATA 1 = Squelch CLK or DATA D0 Set to 0 ADN2814 Preliminary Technical Data TERMINOLOGY 10mV p-p Input Sensitivity and Input Overdrive OUTPUT NOISE 1 VREF SCOPE PROBE ADN2814 PIN + QUANTIZER – 50Ω 2.5V 3kΩ Figure 12. Single-Ended Sensitivity Measurement Driving the ADN2814 differentially (see Figure 13), sensitivity seems to improve from observing the quantizer input with an oscilloscope probe. This is an illusion caused by the use of a single-ended probe. A 5 mV p-p signal appears to drive the ADN2814 quantizer. However, the single-ended probe measures only half the signal. The true quantizer input signal is twice this value, because the other quantizer input is a complementary signal to the signal being observed. 0 5mV p-p SCOPE PROBE VREF OFFSET 50Ω VREF 04949-0-013 Sensitivity and overdrive specifications for the quantizer involve offset voltage, gain, and noise. The relationship between the logic output of the quantizer and the analog voltage input is shown in Figure 11. For sufficiently large positive input voltage, the output is always Logic 1 and, similarly for negative inputs, the output is always Logic 0. However, the transitions between output Logic Levels 1 and 0 are not at precisely defined input voltage levels, but occur over a range of input voltages. Within this range of input voltages, the output might be either 1 or 0, or it might even fail to attain a valid logic state. The width of this zone is determined by the input voltage noise of the quantizer. The center of the zone is the quantizer input offset voltage. Input overdrive is the magnitude of signal required to guarantee the correct logic level with 1 × 10−10 confidence level. INPUT (V p-p) OVERDRIVE PIN 04949-0-012 + QUANTIZER NIN Figure 11. Input Sensitivity and Input Overdrive – Single-Ended vs. Differential 50Ω VREF AC coupling is typically used to drive the inputs to the quantizer. The inputs are internally dc biased to a commonmode potential of ~2.5 V. Driving the ADN2814 single-ended and observing the quantizer input with an oscilloscope probe at the point indicated in Figure 12 shows a binary signal with an average value equal to the common-mode potential and instantaneous values both above and below the average value. It is convenient to measure the peak-to-peak amplitude of this signal and call the minimum required value the quantizer sensitivity. Referring to Figure 12, because both positive and negative offsets need to be accommodated, the sensitivity is twice the overdrive. The ADN2814 quantizer typically has 6 mV p-p sensitivity. 50Ω VREF 5mV p-p 3kΩ 2.5V 04949-0-014 SENSITIVITY (2× OVERDRIVE) Figure 13. Differential Sensitivity Measurement LOS Response Time LOS response time is the delay between removal of the input signal and indication of loss of signal (LOS) at the LOS output, Pin 22. When the inputs are dc-coupled, the LOS assert time of the AD2814 is 500 ns typically and the de-assert time is 400 ns typically,. In practice, the time constant produced by the ac coupling at the quantizer input and the 50 Ω on-chip input termination determines the LOS response time. Rev. PrA | Page 12 of 28 Preliminary Technical Data ADN2814 JITTER SPECIFICATIONS JITTER GENERATION The jitter generation specification limits the amount of jitter that can be generated by the device with no jitter and wander applied at the input. For SONET devices, the jitter generated must be less than 0.01 UI rms, and must be less than 0.1 UIp-p. JITTER TRANSFER The jitter transfer function is the ratio of the jitter on the output signal to the jitter applied on the input signal versus the frequency. This parameter measures the limited amount of the jitter on an input signal that can be transferred to the output signal (see Figure 14). SLOPE = –20dB/DECADE 04949-0-015 ACCEPTABLE RANGE fC JITTER FREQUENCY (kHz) Figure 14. Jitter Transfer Curve JITTER TOLERANCE The jitter tolerance is defined as the peak-to-peak amplitude of the sinusoidal jitter applied on the input signal, which causes a 1 dB power penalty. This is a stress test intended to ensure that no additional penalty is incurred under the operating conditions (see Figure 15). 15.00 SLOPE = –20dB/DECADE 1.50 0.15 f0 f1 f2 f3 JITTER FREQUENCY (kHz) Figure 15. SONET Jitter Tolerance Mask Rev. PrA | Page 13 of 28 f4 04949-0-016 The following sections briefly summarize the specifications of jitter generation, transfer, and tolerance in accordance with the Telcordia document (GR-253-CORE, Issue 3, September 2000) for the optical interface at the equipment level and the ADN2814 performance with respect to those specifications. JITTER GAIN (dB) Jitter is the dynamic displacement of digital signal edges from their long-term average positions, measured in unit intervals (UI), where 1 UI = 1 bit period. Jitter on the input data can cause dynamic phase errors on the recovered clock sampling edge. Jitter on the recovered clock causes jitter on the retimed data. 0.1 INPUT JITTER AMPLITUDE (UI p-p) The ADN2814 CDR is designed to achieve the best bit-errorrate (BER) performance and exceeds the jitter transfer, generation, and tolerance specifications proposed for SONET/SDH equipment defined in the Telcordia Technologies specification. ADN2814 Preliminary Technical Data THEORY OF OPERATION Another view of the circuit is that the phase shifter implements the zero required for frequency compensation of a second-order phase-locked loop, and this zero is placed in the feedback path and, thus, does not appear in the closed-loop transfer function. Jitter peaking in a conventional second-order phase-locked loop is caused by the presence of this zero in the closed-loop transfer function. Because this circuit has no zero in the closed-loop transfer, jitter peaking is minimized. The delay- and phase-loops together simultaneously provide wide-band jitter accommodation and narrow-band jitter filtering. The linearized block diagram in Figure 16 shows that the jitter transfer function, Z(s)/X(s), is a second-order low-pass providing excellent filtering. Note that the jitter transfer has no zero, unlike an ordinary second-order phase-locked loop. This means that the main PLL loop has virtually zero jitter peaking (see Figure 17). This makes this circuit ideal for signal regenerator applications, where jitter peaking in a cascade of regenerators can contribute to hazardous jitter accumulation. The error transfer, e(s)/X(s), has the same high-pass form as an ordinary phase-locked loop. This transfer function is free to be optimized to give excellent wide-band jitter accommodation, because the jitter transfer function, Z(s)/X(s), provides the narrow-band jitter filtering. INPUT DATA e(s) X(s) d/sc o/s 1/n Z(s) RECOVERED CLOCK d = PHASE DETECTOR GAIN o = VCO GAIN c = LOOP INTEGRATOR psh = PHASE SHIFTER GAIN n = DIVIDE RATIO JITTER TRANSFER FUNCTION Z(s) 1 = cn n psh X(s) s2 +s +1 do o TRACKING ERROR TRANSFER FUNCTION 04949-0-017 e(s) s2 = d psh do X(s) + s2 + s c cn Figure 16. ADN2814 PLL/DLL Architecture JITTER PEAKING IN ORDINARY PLL ADN2814 Z(s) X(s) o n psh d psh c FREQUENCY (kHz) 04949-0-018 The delay- and phase-loops together track the phase of the input data signal. For example, when the clock lags input data, the phase detector drives the VCO to higher frequency, and also increases the delay through the phase shifter; both these actions serve to reduce the phase error between the clock and data. The faster clock picks up phase, while the delayed data loses phase. Because the loop filter is an integrator, the static phase error is driven to zero. psh JITTER GAIN (dB) The ADN2814 is a delay- and phase-locked loop circuit for clock recovery and data retiming from an NRZ encoded data stream. The phase of the input data signal is tracked by two separate feedback loops, which share a common control voltage. A high speed delay-locked loop path uses a voltage controlled phase shifter to track the high frequency components of input jitter. A separate phase control loop, comprised of the VCO, tracks the low frequency components of input jitter. The initial frequency of the VCO is set by yet a third loop, which compares the VCO frequency with the input data frequency and sets the coarse tuning voltage. The jitter tracking phase-locked loop controls the VCO by the fine-tuning control. Figure 17. ADN2814 Jitter Response vs. Conventional PLL The delay- and phase-loops contribute to overall jitter accommodation. At low frequencies of input jitter on the data signal, the integrator in the loop filter provides high gain to track large jitter amplitudes with small phase error. In this case, the VCO is frequency modulated and jitter is tracked as in an ordinary phase-locked loop. The amount of low frequency jitter that can be tracked is a function of the VCO tuning range. A wider tuning range gives larger accommodation of low frequency jitter. The internal loop control voltage remains small for small phase errors, so the phase shifter remains close to the center of its range and thus contributes little to the low frequency jitter accommodation. Rev. PrA | Page 14 of 28 Preliminary Technical Data ADN2814 At medium jitter frequencies, the gain and tuning range of the VCO are not large enough to track input jitter. In this case, the VCO control voltage becomes large and saturates, and the VCO frequency dwells at one extreme of its tuning range or the other. The size of the VCO tuning range, therefore, has only a small effect on the jitter accommodation. The delay-locked loop control voltage is now larger, and so the phase shifter takes on the burden of tracking the input jitter. The phase shifter range, in UI, can be seen as a broad plateau on the jitter tolerance curve. The phase shifter has a minimum range of 2 UI at all data rates. cannot be tolerated. In this region, the gain of the integrator determines the jitter accommodation. Because the gain of the loop integrator declines linearly with frequency, jitter accommodation is lower with higher jitter frequency. At the highest frequencies, the loop gain is very small, and little tuning of the phase shifter can be expected. In this case, jitter accommodation is determined by the eye opening of the input data, the static phase error, and the residual loop jitter generation. The jitter accommodation is roughly 0.5 UI in this region. The corner frequency between the declining slope and the flat region is the closed loop bandwidth of the delay-locked loop, which is roughly 1.0 MHz at 622 Mb/s. The gain of the loop integrator is small for high jitter frequencies, so that larger phase differences are needed to make the loop control voltage big enough to tune the range of the phase shifter. Large phase errors at high jitter frequencies Rev. PrA | Page 15 of 28 ADN2814 Preliminary Technical Data FUNCTIONAL DESCRIPTION FREQUENCY ACQUISITION The ADN2814 acquires frequency from the data over a range of data frequencies from 12.3 Mb/s to 675 Mb/s. The lock detector circuit compares the frequency of the VCO and the frequency of the incoming data. When these frequencies differ by more than 1000 ppm, LOL is asserted. This initiates a frequency acquisition cycle. The VCO frequency is reset to the bottom of its range, which is 12.3 MHz. The frequency detector then compares this VCO frequency and the incoming data frequency and increments the VCO frequency, if necessary. Initially, the VCO frequency is incremented in large steps to aid fast acquisition. As the VCO frequency approaches the data frequency, the step size is reduced until the VCO frequency is within 250 ppm of the data frequency, at which point LOL is de-asserted. ADN2814 drops below the programmed LOS threshold, the output of the LOS detector, LOS Pin 22, is asserted to a Logic 1. The LOS detector’s response time is ~500 ns by design, but is dominated by the RC time constant in ac-coupled applications. The LOS pin defaults to active high. However, by setting Bit CTRLC[2] to 1, the LOS pin is configured as active low. There is typically 6 dB of electrical hysteresis designed into the LOS detector to prevent chatter on the LOS pin. This means that, if the input level drops below the programmed LOS threshold causing the LOS pin to assert, the LOS pin is not deasserted until the input level has increased to 6 dB (2×) above the LOS threshold (see Figure 18). INPUT LEVEL HYSTERESIS LOS THRESHOLD t 04949-0-019 The frequency loop requires a single external capacitor between CF1 and CF2, Pins 14 and 15. A 0.47 µF ± 20%, X7R ceramic chip capacitor with < 10 nA leakage current is recommended. Leakage current of the capacitor can be calculated by dividing the maximum voltage across the 0.47 µF capacitor, ~3 V, by the insulation resistance of the capacitor. The insulation resistance of the 0.47 uF capacitor should be greater than 300 MΩ. INPUT VOLTAGE (VDIFF) LOS OUTPUT Once LOL is de-asserted, the frequency-locked loop is turned off. The PLL/DLL pulls in the VCO frequency the rest of the way until the VCO frequency equals the data frequency. Figure 18. LOS Detector Hysteresis LIMITING AMPLIFIER The limiting amplifier has differential inputs (PIN/NIN), which are internally terminated with 50 Ω to an on-chip voltage reference (VREF = 2.5 V typically). The inputs are typically ac-coupled externally, although dc coupling is possible as long as the input common mode voltage remains above 2.5 V (see Figure 26, Figure 27, and Figure 28 in the Applications Information section). Input offset is factory trimmed to achieve better than 6 mV typical sensitivity with minimal drift. The limiting amplifier can be driven differentially or single-ended. The LOS detector and the SLICE level adjust can be used simultaneously on the ADN2814. This means that any offset added to the input signal by the SLICE adjust pins does not affect the LOS detector’s measurement of the absolute input level. LOCK DETECTOR OPERATION The lock detector on the ADN2814 has three modes of operation: normal mode, REFCLK mode, and static LOL mode. Normal Mode SLICE ADJUST The quantizer slicing level can be offset by ±100 mV to mitigate the effect of amplified spontaneous emission (ASE) noise or duty cycle distortion by applying a differential voltage input of up to ±0.95 V to SLICEP/N inputs. If no adjustment of the slice level is needed, SLICEP/N should be tied to VEE. The gain of the slice adjustment is ~0.1 V/V. LOSS OF SIGNAL (LOS) DETECTOR The receiver front end LOS detector circuit detects when the input signal level has fallen below a user-adjustable threshold. The threshold is set with a single external resistor from Pin 9, THRADJ, to VEE. The LOS comparator trip point-versusresistor value is illustrated in Figure 5. If the input level to the In normal mode, the ADN2814 is a continuous rate CDR that locks onto any data rate from 12.3 Mb/s to 675 Mb/s without the use of a reference clock as an acquisition aid. In this mode, the lock detector monitors the frequency difference between the VCO and the input data frequency, and de-asserts the loss of lock signal, which appears on LOL Pin 16, when the VCO is within 250 ppm of the data frequency. This enables the D/PLL, which pulls the VCO frequency in the remaining amount and also acquires phase lock. Once locked, if the input frequency error exceeds 1000 ppm (0.1%), the loss of lock signal is reasserted and control returns to the frequency loop, which begins a new frequency acquisition starting at the lowest point in the VCO operating range, 12.3 MHz. The LOL pin remains asserted until the VCO locks onto a valid input data stream to Rev. PrA | Page 16 of 28 Preliminary Technical Data ADN2814 within 250 ppm frequency error. This hysteresis is shown in Figure 19. LOL –250 –1000 0 250 1000 fVCO ERROR (ppm) 04949-0-020 1 Figure 19. Transfer Function of LOL LOL Detector Operation Using a Reference Clock In this mode, a reference clock is used as an acquisition aid to lock the ADN2814 VCO. Lock to reference mode is enabled by setting CTRLA[0] to 1. The user also needs to write to the CTRLA[7:6] and CTRLA[5:2] bits in order to set the reference frequency range and the divide ratio of the data rate with respect to the reference frequency. For more details, see the Reference Clock (Optional) section. In this mode, the lock detector monitors the difference in frequency between the divided down VCO and the divided down reference clock. The loss of lock signal, which appears on the LOL Pin 16, is deasserted when the VCO is within 250 ppm of the desired frequency. This enables the D/PLL, which pulls the VCO frequency in the remaining amount with respect to the input data and also acquires phase lock. Once locked, if the input frequency error exceeds 1000 ppm (0.1%), the loss of lock signal is re-asserted and control returns to the frequency loop, which re-acquires with respect to the reference clock. The LOL pin remains asserted until the VCO frequency is within 250 ppm of the desired frequency. This hysteresis is shown in Figure 19. Static LOL Mode The ADN2814 implements a static LOL feature, which indicates if a loss of lock condition has ever occurred and remains asserted, even if the ADN2814 regains lock, until the static LOL bit is manually reset. The I2C register bit, MISC[4], is the static LOL bit. If there is ever an occurrence of a loss of lock condition, this bit is internally asserted to logic high. The MISC[4] bit remains high even after the ADN2814 has reacquired lock to a new data rate. This bit can be reset by writing a 1 followed by 0 to I2C Register Bit CTRLB[6]. Once reset, the MISC[4] bit remains de-asserted until another loss of lock condition occurs. Writing a 1 to I2C Register Bit CTRLB[7] causes the LOL pin, Pin 16, to become a static LOL indicator. In this mode, the LOL pin mirrors the contents of the MISC[4] bit and has the functionality described in the previous paragraph. The CTRLB[7] bit defaults to 0. In this mode, the LOL pin operates in the normal operating mode, that is, it is asserted only when the ADN2814 is in acquisition mode and de-asserts when the ADN2814 has re-acquired lock. HARMONIC DETECTOR The ADN2814 provides a harmonic detector, which detects whether or not the input data has changed to a lower harmonic of the data rate that the VCO is currently locked onto. For example, if the input data instantaneously changes from OC-12, 622.08Mb/s, to an OC-3, 155.52 Mb/s bit stream, this could be perceived as a valid OC-12 bit stream, because the OC-3 data pattern is exactly 4× slower than the OC-12 pattern. So, if the change in data rate is instantaneous, a 101 pattern at OC-3 would be perceived by the ADN2814 as a 111100001111 pattern at OC-12. If the change to a lower harmonic is instantaneous, a typical CDR could remain locked at the higher data rate. The ADN2814 implements a harmonic detector that automatically identifies whether or not the input data has switched to a lower harmonic of the data rate that the VCO is currently locked onto. When a harmonic is identified, the LOL pin is asserted and a new frequency acquisition is initiated. The ADN2814 automatically locks onto the new data rate, and the LOL pin is de-asserted. However, the harmonic detector does not detect higher harmonics of the data rate. If the input data rate switches to a higher harmonic of the data rate the VCO is currently locked onto, the VCO loses lock, the LOL pin is asserted, and a new frequency acquisition is initiated. The ADN2814 automatically locks onto the new data rate. The time to detect lock to harmonic is 16,384 × (Td/ρ) where: 1/Td is the new data rate. For example, if the data rate is switched from OC-12 to OC-3, then Td = 1/155.52 MHz. ρ is the data transition density. Most coding schemes seek to ensure that ρ = 0.5, for example, PRBS, 8B/10B. When the ADN2814 is placed in lock to reference mode, the harmonic detector is disabled. SQUELCH MODE Two squelch modes are available with the ADN2814. Squelch DATAOUT AND CLKOUT mode is selected when CTRLC[1] = 0 (default mode). In this mode, when the squelch input, Pin 27, is driven to a TTL high state, both the clock and data outputs are set to the zero state to suppress downstream processing. If the squelch function is not required, Pin 27 should be tied to VEE. Squelch DATAOUT OR CLKOUT mode is selected when CTRLC[1] is 1. In this mode, when the squelch input is driven to a high state, the DATAOUT pins are squelched. When the squelch input is driven to a low state, the CLKOUT pins are squelched. This is especially useful in repeater applications, where the recovered clock may not be needed. Rev. PrA | Page 17 of 28 ADN2814 Preliminary Technical Data A reference clock is not required to perform clock and data recovery with the ADN2814. However, support for an optional reference clock is provided. The reference clock can be driven differentially or single-ended. If the reference clock is not being used, then REFCLKP should be tied to VCC, and REFCLKN can be left floating or tied to VEE (the inputs are internally terminated to VCC/2). See Figure 20 through Figure 22 for sample configurations. The REFCLK input buffer accepts any differential signal with a peak-to-peak differential amplitude of greater than 100 mV (for example, LVPECL or LVDS) or a standard single-ended low voltage TTL input, providing maximum system flexibility. Phase noise and duty cycle of the reference clock are not critical and 100 ppm accuracy is sufficient. The ADN2814 acts as a standard slave device on the bus. The data on the SDA pin is 8 bits long supporting the 7-bit addresses plus the R/W bit. The ADN2814 has 8 subaddresses to enable the user-accessible internal registers (see Table 1 through Table 7). It, therefore, interprets the first byte as the device address and the second byte as the starting subaddress. Autoincrement mode is supported, allowing data to be read from or written to the starting subaddress and each subsequent address without manually addressing the subsequent subaddress. A data transfer is always terminated by a stop condition. The user can also access any unique subaddress register on a one-by-one basis without updating all registers. Stop and start conditions can be detected at any stage of the data transfer. If these conditions are asserted out of sequence with normal read and write operations, then they cause an immediate jump to the idle condition. During a given SCK high period, the user should issue one start condition, one stop condition, or a single stop condition followed by a single start condition. If an invalid subaddress is issued by the user, the ADN2814 does not issue an acknowledge and returns to the idle Rev. PrA | Page 18 of 28 ADN2814 REFCLKP 10 BUFFER 11 REFCLKN 100kΩ 04949-0-021 To control the device on the bus, the following protocol must be followed. First, the master initiates a data transfer by establishing a start condition, defined by a high to low transition on SDA while SCK remains high. This indicates that an address/data stream follows. All peripherals respond to the start condition and shift the next eight bits (the 7-bit address and the R/W bit). The bits are transferred from MSB to LSB. The peripheral that recognizes the transmitted address responds by pulling the data line low during the ninth clock pulse. This is known as an acknowledge bit. All other devices withdraw from the bus at this point and maintain an idle condition. The idle condition is where the device monitors the SDA and SCK lines waiting for the start condition and correct transmitted address. The R/W bit determines the direction of the data. Logic 0 on the LSB of the first byte means that the master writes information to the peripheral. Logic 1 on the LSB of the first byte means that the master reads information from the peripheral. REFERENCE CLOCK (OPTIONAL) 100kΩ VCC/2 Figure 20. Differential REFCLK Configuration VCC CLK OSC REFCLKP ADN2814 OUT BUFFER REFCLKN 100kΩ 100kΩ VCC/2 Figure 21. Single-Ended REFCLK Configuration VCC ADN2814 10 REFCLKP BUFFER NC 11 REFCLKN 100kΩ 100kΩ VCC/2 Figure 22. No REFCLK Configuration 04949-0-022 The ADN2814 supports a 2-wire, I2C compatible, serial bus driving multiple peripherals. Two inputs, serial data (SDA) and serial clock (SCK), carry information between any devices connected to the bus. Each slave device is recognized by a unique address. The ADN2814 has two possible 7-bit slave addresses for both read and write operations. The MSB of the 7-bit slave address is factory programmed to 1. B5 of the slave address is set by Pin 19, SADDR5. Slave address bits [4:0] are defaulted to all 0s. The slave address consists of the 7 MSBs of an 8-bit word. The LSB of the word sets either a read or write operation (see Figure 6). Logic 1 corresponds to a read operation, while Logic 0 corresponds to a write operation. condition. If the user exceeds the highest subaddress while reading back in autoincrement mode, then the highest subaddress register contents continue to be output until the master device issues a no-acknowledge. This indicates the end of a read. In a no-acknowledge condition, the SDATA line is not pulled low on the ninth pulse. See Figure 7 and Figure 8 for sample read and write data transfers and Figure 9 for a more detailed timing diagram. 04949-0-023 I2C INTERFACE Preliminary Technical Data ADN2814 The two uses of the reference clock are mutually exclusive. The reference clock can be used either as an acquisition aid for the ADN2814 to lock onto data, or to measure the frequency of the incoming data to within 0.01%. (There is the capability to measure the data rate to approximately ±10% without the use of a reference clock.) The modes are mutually exclusive, because, in the first use, the user knows exactly what the data rate is and wants to force the part to lock onto only that data rate; in the second use, the user does not know what the data rate is and wants to measure it. Lock to reference mode is enabled by writing a 1 to I2C Register Bit CTRLA[0]. Fine data rate readback mode is enabled by writing a 1 to I2C Register Bit CTRLA[1]. Writing a 1 to both of these bits at the same time causes an indeterminate state and is not supported. Using the Reference Clock to Lock onto Data In this mode, the ADN2814 locks onto a frequency derived from the reference clock according to the following equation: Data Rate/2 CTRLA[5:2] = REFCLK/2 CTRLA[7:6] The user must know exactly what the data rate is, and provide a reference clock that is a function of this rate. The ADN2814 can still be used as a continuous rate device in this configuration, provided that the user has the ability to provide a reference clock that has a variable frequency (see Application Note AN-632). The reference clock can be anywhere between 12.3 MHz and 200 MHz. By default, the ADN2814 expects a reference clock of between 12.3 MHz and 25 MHz. If it is between 25 MHz and 50 MHz, 50 MHz and 100 MHz, or 100 MHz and 200 MHz, the user needs to configure the ADN2814 to use the correct reference frequency range by setting two bits of the CTRLA register, CTRLA[7:6]. Table 11. CTRLA Settings CTRLA[7:6] 00 01 10 11 Range (MHz) 12.3 to 25 25 to 50 50 to 100 100 to 200 CTRLA[5:2] 0000 0001 n 1000 Ratio 1 2 2n 256 The user can specify a fixed integer multiple of the reference clock to lock onto using CTRLA[5:2], where CTRLA should be set to the data rate/DIV_FREF, where DIV_FREF represents the divided-down reference referred to the 12.3 MHz to 25 MHz band. For example, if the reference clock frequency was 38.88 MHz and the input data rate was 622.08 Mb/s, then CTRLA[7:6] would be set to [01] to give a divided-down reference clock of 19.44 MHz. CTRLA[5:2] would be set to [0101], that is, 5, because In this mode, if the ADN2814 loses lock for any reason, it relocks onto the reference clock and continues to output a stable clock. While the ADN2814 is operating in lock to reference mode, if the user ever changes the reference frequency, the FREF range (CTRLA[7:6]), or the FREF ratio (CTRLA[5:2]), this must be followed by writing a 0 to 1 transition into the CTRLA[0] bit to initiate a new lock to reference command. Using the Reference Clock to Measure Data Frequency The user can also provide a reference clock to measure the recovered data frequency. In this case, the user provides a reference clock, and the ADN2814 compares the frequency of the incoming data to the incoming reference clock and returns a ratio of the two frequencies to 0.01% (100 ppm). The accuracy error of the reference clock is added to the accuracy of the ADN2814 data rate measurement. For example, if a 100-ppm accuracy reference clock is used, the total accuracy of the measurement is within 200 ppm. The reference clock can range from 12.3 MHz and 200 MHz. The ADN2814 expects a reference clock between 12.3 MHz and 25 MHz by default. If it is between 25 MHz and 50 MHz, 50 MHz and 100 MHz, or 100 MHz and 200 MHz, the user needs to configure the ADN2814 to use the correct reference frequency range by setting two bits of the CTRLA register, CTRLA[7:6]. Using the reference clock to determine the frequency of the incoming data does not affect the manner in which the part locks onto data. In this mode, the reference clock is used only to determine the frequency of the data. For this reason, the user does not need to know the data rate to use the reference clock in this manner. Prior to reading back the data rate using the reference clock, the CTRLA[7:6] bits must be set to the appropriate frequency range with respect to the reference clock being used. A fine data rate readback is then executed as follows: Step 1: Write a 1 to CTRLA[1]. This enables the fine data rate measurement capability of the ADN2814. This bit is level sensitive and does not need to be reset to perform subsequent frequency measurements. Step 2: Reset MISC[2] by writing a 1 followed by a 0 to CTRLB[3]. This initiates a new data rate measurement. Step 3: Read back MISC[2]. If it is 0, then the measurement is not complete. If it is 1, then the measurement is complete and the data rate can be read back on FREQ[22:0]. The time for a data rate measurement is typically 80 ms. Step 4: Read back the data rate from registers FREQ2[6:0], FREQ1[7:0], and FREQ0[7:0]. 622.08 Mb/s/19.44 MHz = 25 Rev. PrA | Page 19 of 28 ADN2814 Preliminary Technical Data Additional Features Available via the I2C Interface Coarse Data Rate Readback Use the following equation to determine the data rate: f DATARATE = (FREQ [22..0]× f REFCLK )/ 2 (14 + SEL _ RATE ) The data rate can be read back over the I2C interface to approximately +10% without the need of an external reference clock. A 9-bit register, COARSE_RD[8:0], can be read back when LOL is de-asserted. The 8 MSBs of this register are the contents of the RATE[7:0] register. The LSB of the COARSE_RD register is Bit MISC[0]. where: FREQ[22:0] is the reading from FREQ2[6:0] (MSByte), FREQ1[7:0], and FREQ0[7:0] (LSByte). Table 12. D22 D21...D17 FREQ2[6:0] D16 D15 D14...D9 D8 FREQ1[7:0] D7 D6...D1 D0 FREQ0[7:0] fDATARATE is the data rate (Mb/s). fREFCLK is the REFCLK frequency (MHz). SEL_RATE is the setting from CTRLA[7:6]. For example, if the reference clock frequency is 32 MHz, SEL_RATE = 1, since the CTRLA[7:6] setting would be [01], because the reference frequency would fall into the 25 MHz to 50 MHz range. Assume for this example that the input data rate is 622.08 Mb/s (OC12). After following Steps 1 through 4, the value that is read back on FREQ[22:0] = 0x9B851, which is equal to 637 x 103. Plugging this value into the equation yields (637e3 × 32e6)/ (2(14+1) ) = 622.08Mb/s Table 13 provides coarse data rate readback to within ±10%. LOS Configuration The LOS detector output, LOS Pin 22, can be configured to be either active high or active low. If CTRLC[2] is set to Logic 0 (default), the LOS pin is active high when a loss of signal condition is detected. Writing a 1 to CTRLC[2] configures the LOS pin to be active low when a loss of signal condition is detected. System Reset A frequency acquisition can be initiated by writing a 1 followed by a 0 to the I2C Register Bit CTRLB[5]. This initiates a new frequency acquisition while keeping the ADN2814 in the operating mode that it was previously programmed to in registers CTRL[A], CTRL[B], and CTRL[C]. If subsequent frequency measurements are required, CTRLA[1] should remain set to 1. It does not need to be reset. The measurement process is reset by writing a 1 followed by a 0 to CTRLB[3]. This initiates a new data rate measurement. Follow Steps 2 through 4 to read back the new data rate. Note: A data rate readback is valid only if LOL is low. If LOL is high, the data rate readback is invalid. Rev. PrA | Page 20 of 28 Preliminary Technical Data ADN2814 APPLICATIONS INFORMATION PCB DESIGN GUIDELINES If connections to the supply and ground are made through vias, the use of multiple vias in parallel helps to reduce series inductance, especially on Pin 24, which supplies power to the high speed CLKOUTP/CLKOUTN and DATAOUTP/ DATAOUTN output buffers. Refer to the schematic in Figure 23 for recommended connections. Proper RF PCB design techniques must be used for optimal performance. Power Supply Connections and Ground Planes Use of one low impedance ground plane is recommended. The VEE pins should be soldered directly to the ground plane to reduce series inductance. If the ground plane is an internal plane and connections to the ground plane are made through vias, multiple vias can be used in parallel to reduce the series inductance, especially on Pin 23, which is the ground return for the output buffers. The exposed pad should be connected to the GND plane using plugged vias so that solder does not leak through the vias during reflow. By using adjacent power supply and GND planes, excellent high frequency decoupling can be realized by using close spacing between the planes. This capacitance is given by Cplane = 0.88ε r A/d (pF ) where: Use of a 22 µF electrolytic capacitor between VCC and VEE is recommended at the location where the 3.3 V supply enters the PCB. When using 0.1 µF and 1 nF ceramic chip capacitors, they should be placed between the IC power supply VCC and VEE, as close as possible to the ADN2814 VCC pins. εr is the dielectric constant of the PCB material. A is the area of the overlap of power and GND planes (cm2). d is the separation between planes (mm). For FR-4, εr = 4.4 mm and 0.25 mm spacing, C ~15 pF/cm2. 50Ω TRANSMISSION LINES VCC DATAOUTP + 22µF 0.1µF DATAOUTN 1nF CLKOUTP TEST2 VCC VEE DATAOUTP DATAOUTN SQUELCH CLKOUTP CLKOUTN CLKOUTN 0.1µF TIA 50Ω CIN 50Ω 32 31 30 29 28 27 26 25 1nF VCC VEE LOS SDA SCK SADDR5 VCC VEE 1nF 0.1µF I2C CONTROLLER I2C CONTROLLER µC VCC 0.1µF µC RTH VCC 0.1µF 1nF 0.47µF ±20% >300MΩ INSULATION RESISTANCE Figure 23. Typical ADN2814 Applications Circuit Rev. PrA | Page 21 of 28 04949-0-031 VCC 24 EXPOSED PAD 23 TIED OFF TO 22 VEE PLANE 21 20 WITH VIAS 19 18 17 9 10 11 12 13 14 15 16 1nF 1 2 3 4 5 6 7 8 THRADJ REFCLKP REFCLKN NC VCC VEE CF2 CF1 LOL 0.1µF TEST1 VCC VREF NIN PIN SLICEP SLICEN VEE VCC ADN2814 Preliminary Technical Data Transmission Lines Use of 50 Ω transmission lines is required for all high frequency input and output signals to minimize reflections: PIN, NIN, CLKOUTP, CLKOUTN, DATAOUTP, DATAOUTN (also REFCLKP, REFCLKN, if a high frequency reference clock is used, such as 155 MHz). It is also necessary for the PIN/NIN input traces to be matched in length, and the CLKOUTP/N and DATAOUTP/N output traces to be matched in length to avoid skew between the differential traces. The high speed inputs, PIN and NIN, are internally terminated with 50 Ω to an internal reference voltage (see Figure 24). A 0.1 µF is recommended between VREF, Pin 3, and GND to provide an ac ground for the inputs. As with any high speed mixed-signal design, take care to keep all high speed digital traces away from sensitive analog nodes. rates used in the application. When choosing the capacitors, the time constant formed with the two 50 Ω resistors in the signal path must be considered. When a large number of consecutive identical digits (CIDs) are applied, the capacitor voltage can droop due to baseline wander (see Figure 25), causing patterndependent jitter (PDJ). The user must determine how much droop is tolerable and choose an ac coupling capacitor based on that amount of droop. The amount of PDJ can then be approximated based on the capacitor selection. The actual capacitor value selection may require some trade-offs between droop and PDJ. Example: Assuming that 2% droop can be tolerated, then the maximum differential droop is 4%. Normalizing to Vpp: Droop = ∆ V = 0.04 V = 0.5 Vpp (1 − e–t/τ) ; therefore, τ = 12t where: τ is the RC time constant (C is the ac coupling capacitor, R = 100 Ω seen by C). VCC ADN2814 50Ω CIN PIN 50Ω CIN NIN n is the number of CIDs. 50Ω 0.1µF VREF T is the bit period. 50Ω 3kΩ 2.5V The capacitor value can then be calculated by combining the equations for τ and t: 04949-0-026 TIA t is the total discharge time, which is equal to nΤ. C = 12nT/R Figure 24. ADN2814 AC-Coupled Input Configuration Soldering Guidelines for Chip Scale Package The lands on the 32 LFCSP are rectangular. The printed circuit board pad for these should be 0.1 mm longer than the package land length and 0.05 mm wider than the package land width. The land should be centered on the pad. This ensures that the solder joint size is maximized. The bottom of the chip scale package has a central exposed pad. The pad on the printed circuit board should be at least as large as this exposed pad. The user must connect the exposed pad to VEE using plugged vias so that solder does not leak through the vias during reflow. This ensures a solid connection from the exposed pad to VEE. Choosing AC Coupling Capacitors AC coupling capacitors at the input (PIN, NIN) and output (DATAOUTP, DATAOUTN) of the ADN2814 must be chosen such that the device works properly over the full range of data Once the capacitor value is selected, the PDJ can be approximated as ( ) PDJ pspp = 0.5t r 1 − e (−nT/RC ) / 0.6 where: PDJpspp is the amount of pattern-dependent jitter allowed; < 0.01 UI p-p typical. tr is the rise time, which is equal to 0.22/BW, where BW ~ 0.7 (bit rate). Note that this expression for tr is accurate only for the inputs. The output rise time for the ADN2814 is ~100 ps regardless of data rate. Rev. PrA | Page 22 of 28 Preliminary Technical Data ADN2814 VCC V1 CIN V2 ADN2814 PIN TIA V1b CIN V2b COUT + 50Ω 50Ω VREF DATAOUTP CDR LIMAMP DATAOUTN COUT – NIN V1 1 2 3 4 V1b V2 VREF V2b VTH VDIFF VDIFF = V2–V2b VTH = ADN2814 QUANTIZER THRESHOLD 04949-0-027 NOTES: 1. DURING DATA PATTERNS WITH HIGH TRANSITION DENSITY, DIFFERENTIAL DC VOLTAGE AT V1 AND V2 IS ZERO. 2. WHEN THE OUTPUT OF THE TIA GOES TO CID, V1 AND V1b ARE DRIVEN TO DIFFERENT DC LEVELS. V2 AND V2b DISCHARGE TO THE VREF LEVEL, WHICH EFFECTIVELY INTRODUCES A DIFFERENTIAL DC OFFSET ACROSS THE AC COUPLING CAPACITORS. 3. WHEN THE BURST OF DATA STARTS AGAIN, THE DIFFERENTIAL DC OFFSET ACROSS THE AC COUPLING CAPACITORS IS APPLIED TO THE INPUT LEVELS CAUSING A DC SHIFT IN THE DIFFERENTIAL INPUT. THIS SHIFT IS LARGE ENOUGH SUCH THAT ONE OF THE STATES, EITHER HIGH OR LOW DEPENDING ON THE LEVELS OF V1 AND V1b WHEN THE TIA WENT TO CID, IS CANCELED OUT. THE QUANTIZER DOES NOT RECOGNIZE THIS AS A VALID STATE. 4. THE DC OFFSET SLOWLY DISCHARGES UNTIL THE DIFFERENTIAL INPUT VOLTAGE EXCEEDS THE SENSITIVITY OF THE ADN2814. THE QUANTIZER CAN RECOGNIZE BOTH HIGH AND LOW STATES AT THIS POINT. Figure 25. Example of Baseline Wander PIN VPP = PIN – NIN = 2 × VSE = 10mV AT SENSITIVITY VSE = 5mV MIN NIN VCM = 2.3V MIN (DC-COUPLED) 04949-0-029 The inputs to the ADN2814 can also be dc-coupled. This might be necessary in burst mode applications, where there are long periods of CIDs, and baseline wander cannot be tolerated. If the inputs to the ADN2814 are dc-coupled, care must be taken not to violate the input range and common-mode level requirements of the ADN2814 (see Figure 26 through Figure 28). If dc coupling is required, and the output levels of the TIA do not adhere to the levels shown in Figure 27, then level shifting and/or an attenuator must be between the TIA outputs and the ADN2814 inputs. INPUT (V) DC-COUPLED APPLICATION Figure 27. Minimum Allowed DC-Coupled Input Levels VCC 50Ω PIN 50Ω NIN ADN2814 PIN VPP = PIN – NIN = 2 × VSE = 2.0V MAX VSE = 1.0V MAX VREF 3kΩ 2.5V VCM = 2.3V (DC-COUPLED) NIN 04949-0-030 0.1µF 50Ω 04949-0-028 50Ω INPUT (V) TIA Figure 26. DC-Coupled Application Figure 28. Maximum Allowed DC-Coupled Input Levels Rev. PrA | Page 23 of 28 ADN2814 Preliminary Technical Data COARSE DATA RATE READBACK LOOK-UP TABLE Code is the 9-bit value read back from COARSE_RD[8:0]. Table 13. Look-Up Table Code 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40 41 42 43 44 45 46 47 FMID 5.1934e+06 5.1930e+06 5.2930e+06 5.3989e+06 5.5124e+06 5.6325e+06 5.7612e+06 5.8995e+06 6.0473e+06 6.2097e+06 6.3819e+06 6.5675e+06 6.7688e+06 6.9874e+06 7.2262e+06 7.4863e+06 7.4139e+06 7.4135e+06 7.5606e+06 7.7173e+06 7.8852e+06 8.0633e+06 8.2548e+06 8.4586e+06 8.6784e+06 8.9180e+06 9.1736e+06 9.4481e+06 9.7464e+06 1.0068e+07 1.0417e+07 1.0791e+07 1.0387e+07 1.0386e+07 1.0586e+07 1.0798e+07 1.1025e+07 1.1265e+07 1.1522e+07 1.1799e+07 1.2095e+07 1.2419e+07 1.2764e+07 1.3135e+07 1.3538e+07 1.3975e+07 1.4452e+07 1.4973e+07 Code 48 49 50 51 52 53 54 55 56 57 58 59 60 61 62 63 64 65 66 67 68 69 70 71 72 73 74 75 76 77 78 79 80 81 82 83 84 85 86 87 88 89 90 91 92 93 94 95 FMID 1.4828e+07 1.4827e+07 1.5121e+07 1.5435e+07 1.5770e+07 1.6127e+07 1.6510e+07 1.6917e+07 1.7357e+07 1.7836e+07 1.8347e+07 1.8896e+07 1.9493e+07 2.0136e+07 2.0833e+07 2.1582e+07 2.0774e+07 2.0772e+07 2.1172e+07 2.1596e+07 2.2049e+07 2.2530e+07 2.3045e+07 2.3598e+07 2.4189e+07 2.4839e+07 2.5527e+07 2.6270e+07 2.7075e+07 2.7950e+07 2.8905e+07 2.9945e+07 2.9655e+07 2.9654e+07 3.0242e+07 3.0869e+07 3.1541e+07 3.2253e+07 3.3019e+07 3.3834e+07 3.4714e+07 3.5672e+07 3.6694e+07 3.7792e+07 3.8985e+07 4.0273e+07 4.1666e+07 4.3164e+07 Code 96 97 98 99 100 101 102 103 104 105 106 107 108 109 110 111 112 113 114 115 116 117 118 119 120 121 122 123 124 125 126 127 128 129 130 131 132 133 134 135 136 137 138 139 140 141 142 143 Rev. PrA | Page 24 of 28 FMID 4.1547e+07 4.1544e+07 4.2344e+07 4.3191e+07 4.4099e+07 4.5060e+07 4.6090e+07 4.7196e+07 4.8378e+07 4.9678e+07 5.1055e+07 5.2540e+07 5.4150e+07 5.5899e+07 5.7810e+07 5.9890e+07 5.9311e+07 5.9308e+07 6.0485e+07 6.1739e+07 6.3081e+07 6.4506e+07 6.6038e+07 6.7669e+07 6.9427e+07 7.1344e+07 7.3388e+07 7.5585e+07 7.7971e+07 8.0546e+07 8.3333e+07 8.6328e+07 8.3095e+07 8.3087e+07 8.4689e+07 8.6383e+07 8.8198e+07 9.0120e+07 9.2179e+07 9.4392e+07 9.6757e+07 9.9356e+07 1.0211e+08 1.0508e+08 1.0830e+08 1.1180e+08 1.1562e+08 1.1978e+08 Code 144 145 146 147 148 149 150 151 152 153 154 155 156 157 158 159 160 161 162 163 164 165 166 167 168 169 170 171 172 173 174 175 176 177 178 179 180 181 182 183 184 185 186 187 188 189 190 191 FMID 1.1862e+08 1.1862e+08 1.2097e+08 1.2348e+08 1.2616e+08 1.2901e+08 1.3208e+08 1.3534e+08 1.3885e+08 1.4269e+08 1.4678e+08 1.5117e+08 1.5594e+08 1.6109e+08 1.6667e+08 1.7266e+08 1.6619e+08 1.6617e+08 1.6938e+08 1.7277e+08 1.7640e+08 1.8024e+08 1.8436e+08 1.8878e+08 1.9351e+08 1.9871e+08 2.0422e+08 2.1016e+08 2.1660e+08 2.2360e+08 2.3124e+08 2.3956e+08 2.3724e+08 2.3723e+08 2.4194e+08 2.4695e+08 2.5233e+08 2.5802e+08 2.6415e+08 2.7067e+08 2.7771e+08 2.8538e+08 2.9355e+08 3.0234e+08 3.1188e+08 3.2218e+08 3.3333e+08 3.4531e+08 Preliminary Technical Data Code 192 193 194 195 196 197 198 199 200 FMID 3.3238e+08 3.3235e+08 3.3876e+08 3.4553e+08 3.5279e+08 3.6048e+08 3.6872e+08 3.7757e+08 3.8703e+08 Code 201 202 203 204 205 206 207 208 209 ADN2814 FMID 3.9742e+08 4.0844e+08 4.2032e+08 4.3320e+08 4.4719e+08 4.6248e+08 4.7912e+08 4.7449e+08 4.7447e+08 Code 210 211 212 213 214 215 216 217 218 Rev. PrA | Page 25 of 28 FMID 4.8388e+08 4.9391e+08 5.0465e+08 5.1605e+08 5.2831e+08 5.4135e+08 5.5542e+08 5.7075e+08 5.8711e+08 Code 219 220 221 222 223 224 225 226 227 FMID 6.0468e+08 6.2377e+08 6.4437e+08 6.6666e+08 6.9062e+08 6.6476e+08 6.6470e+08 6.7751e+08 6.9106e+08 ADN2814 Preliminary Technical Data OUTLINE DIMENSIONS 5.00 BSC SQ 0.60 MAX 0.50 BSC 4.75 BSC SQ TOP VIEW 0.50 0.40 0.30 1.00 0.85 0.80 32 25 24 PIN 1 INDICATOR 12° MAX PIN 1 INDICATOR 0.60 MAX 1 3.25 3.10 SQ 2.95 BOTTOM VIEW 17 16 9 8 0.25 MIN 3.50 REF 0.80 MAX 0.65 TYP 0.05 MAX 0.02 NOM SEATING PLANE 0.30 0.23 0.18 0.20 REF COPLANARITY 0.08 COMPLIANT TO JEDEC STANDARDS MO-220-VHHD-2 Figure 29. 32-Lead Frame Chip Scale Package [LFCSP] (CP-32) Dimensions shown in millimeters ORDERING GUIDE Model ADN2814XCPZ Temperature Range −40°C to 85°C Package Description Pb Free 32-LFCSP Rev. PrA | Page 26 of 28 Package Option CP-32 Preliminary Technical Data ADN2814 NOTES Rev. PrA | Page 27 of 28 ADN2814 Preliminary Technical Data NOTES Purchase of licensed I2C components of Analog Devices or one of its sublicensed Associated Companies conveys a license for the purchaser under the Philips I2C Patent Rights to use these components in an I2C system, provided that the system conforms to the I2C Standard Specification as defined by Philips. © 2004 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. PR04949-0-8/04(PrA) Rev. PrA | Page 28 of 28