LINER MMSZ5266BT1G 100vin micropower isolated flyback converter with 150v/260ma switch Datasheet

LT8300
100VIN Micropower Isolated
Flyback Converter with
150V/260mA Switch
Description
Features
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6V to 100V Input Voltage Range
260mA, 150V Internal DMOS Power Switch
Low Quiescent Current:
70µA in Sleep Mode
330µA in Active Mode
Boundary Mode Operation at Heavy Load
Low-Ripple Burst Mode® Operation at Light Load
Minimum Load <0.5% (Typ) of Full Output
VOUT Set with a Single External Resistor
No Transformer Third Winding or Opto-Isolator
Required for Regulation
Accurate EN/UVLO Threshold and Hysteresis
Internal Compensation and Soft-Start
5-Lead TSOT-23 Package
Applications
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Isolated Telecom, Automotive, Industrial, Medical
Power Supplies
Isolated Auxiliary/Housekeeping Power Supplies
The LT®8300 is a micropower high voltage isolated flyback
converter. By sampling the isolated output voltage directly
from the primary-side flyback waveform, the part requires
no third winding or opto-isolator for regulation. The output
voltage is programmed with a single external resistor. Internal compensation and soft-start further reduce external
component count. Boundary mode operation provides a
small magnetic solution with excellent load regulation.
Low ripple Burst Mode operation maintains high efficiency
at light load while minimizing the output voltage ripple.
A 260mA, 150V DMOS power switch is integrated along
with all high voltage circuitry and control logic into a 5-lead
ThinSOT™ package.
The LT8300 operates from an input voltages range of 6V
to 100V and can deliver up to 2W of isolated output power.
The high level of integration and the use of boundary
and low ripple burst modes result in a simple to use, low
component count, and high efficiency application solution
for isolated power delivery.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497,
and 7471522.
Typical Application
5V Micropower Isolated Flyback Converter
Efficiency vs Load Current
100
VOUT+
5V
1mA TO 300mA
4:1
2.2µF
1M
300µH
VIN
19µH
•
LT8300
EN/UVLO
•
VOUT–
SW
40.2k
210k
RFB
GND
8300 TA01a
47µF
90
VIN = 36V
80
EFFICIENCY (%)
VIN
36V TO 72V
70
VIN = 72V
60
VIN = 48V
50
40
30
20
10
0
0
50
100
150
200
LOAD CURRENT (mA)
250
300
8300 TA01b
8300f
1
LT8300
Absolute Maximum Ratings
Pin Configuration
(Note 1)
TOP VIEW
SW (Note 2)............................................................ 150V
VIN.......................................................................... 100V
EN/UVLO.................................................................... VIN
RFB....................................................... VIN – 0.5V to VIN
Current into RFB.................................................... 200µA
Operating Junction Temperature Range (Notes 3, 4)
LT8300E, LT8300I.............................. –40°C to 125°C
LT8300H............................................. –40°C to 150°C
LT8300MP.......................................... –55°C to 150°C
Storage Temperature Range................... –65°C to 150°C
EN/UVLO 1
5 VIN
GND 2
RFB 3
4 SW
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
TJMAX = 150°C, θJA = 150°C/W
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT8300ES5#PBF
LT8300ES5#TRPBF
LTGFF
5-Lead Plastic TSOT-23
–40°C to 125°C
LT8300IS5#PBF
LT8300IS5#TRPBF
LTGFF
5-Lead Plastic TSOT-23
–40°C to 125°C
LT8300HS5#PBF
LT8300HS5#TRPBF
LTGFF
5-Lead Plastic TSOT-23
–40°C to 150°C
LT8300MPS5#PBF
LT8300MPS5#TRPBF
LTGFF
5-Lead Plastic TSOT-23
–55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
8300f
2
LT8300
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted.
SYMBOL
PARAMETER
VIN
Input Voltage Range
CONDITIONS
MIN
TYP
6
MAX
UNIT
100
V
VIN UVLO Threshold
Rising
Falling
5.8
3.2
6
V
V
VIN Quiescent Current
VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
Sleep Mode (Switch Off)
Active Mode (Switch On)
1.2
200
70
330
2
µA
µA
µA
µA
EN/UVLO Shutdown Threshold
For Lowest Off IQ
l
0.3
0.75
EN/UVLO Enable Threshold
Falling
Hysteresis
l
1.199
1.223
0.016
1.270
V
V
IHYS
EN/UVLO Hysteresis Current
VEN/UVLO = 0.3V
VEN/UVLO = 1.1V
VEN/UVLO = 1.3V
–0.1
2.2
–0.1
0
2.5
0
0.1
2.8
0.1
µA
µA
µA
fMAX
Maximum Switching Frequency
720
750
780
kHz
fMIN
Minimum Switching Frequency
6
7.5
9
kHz
IQ
V
tON(MIN)
Minimum Switch-On Time
160
ns
tOFF(MIN)
Minimum Switch-Off Time
350
ns
tOFF(MAX)
Maximum Switch-Off Time
200
µs
ISW(MAX)
ISW(MIN)
SW Over Current Limit
To Initiate Soft-Start
520
mA
RDS(ON)
Switch On-Resistance
ISW = 100mA
10
Ω
ILKG
Switch Leakage Current
VIN = 100V, VSW = 150V
0.1
IRFB
RFB Regulation Current
Maximum SW Current Limit
l
228
260
292
mA
Minimum SW Current Limit
l
34
52
70
mA
RFB Regulation Current Line Regulation
tSS
Backup Timer
l
6V ≤ VIN ≤ 100V
Soft-Start Timer
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The SW pin is rated to 150V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 150V as shown
in Figure 5.
Note 3: The LT8300E is guaranteed to meet performance specifications
from 0°C to 125°C operating junction temperature. Specifications over
the –40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
98
0.5
µA
100
102
µA
0.001
0.01
%/V
2.7
ms
The LT8300I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT8300H is guaranteed over the full –40°C to
150°C operating junction temperature range. The LT8300MP is guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperature greater than 125°C.
Note 4: The LT8300 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 150°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
8300f
3
LT8300
Typical Performance Characteristics
Output Load and Line Regulation
5.20
5.5
FRONT PAGE APPLICATION
5.2
5.00
4.95
4.90
5.1
5.0
4.9
4.8
0
50
100
150
200
LOAD CURRENT (mA)
250
4.5
–50 –25
300
VSW
50V/DIV
VOUT
50mV/DIV
2µs/DIV
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 300mA
ILPRI
100mA/DIV
VSW
50V/DIV
VSW
50V/DIV
VOUT
50mV/DIV
VOUT
50mV/DIV
2µs/DIV
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 60mA
8300 G06
360
IQ (µA)
TJ = 150°C
80
IQ (µA)
300
380
90
8
TJ = 25°C
250
VIN Quiescent Current,
Active Mode
100
4
100
150
200
LOAD CURRENT (mA)
20µs/DIV
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 1mA
8300 G05
VIN Quiescent Current,
Sleep Mode
6
50
Burst Mode Waveforms
ILPRI
100mA/DIV
8300 G04
VIN Shutdown Current
0
8300 G03
Discontinuous Mode Waveforms
ILPRI
100mA/DIV
IQ (µA)
0
0
25 50 75 100 125 150
AMBIENT TEMPERATURE (°C)
8300 G02
Boundary Mode Waveforms
2
200
4.6
8300 G01
TJ = 150°C
300
100
4.7
VIN = 36V
VIN = 48V
VIN = 72V
4.85
FREQUENCY (kHz)
5.05
FRONT PAGE APPLICATION
VIN = 48V
400
5.3
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
500
FRONT PAGE APPLICATION
VIN = 48V, IOUT = 200mA
5.4
5.10
10
Switching Frequency
vs Load Current
Output Temperature Variation
5.15
4.80
TA = 25°C, unless otherwise noted.
TJ = 25°C
70
60
TJ = 150°C
340
TJ = 25°C
320
TJ = –55°C
TJ = –55°C
300
50
TJ = –55°C
0
0
20
40
60
VIN (V)
80
100
8300 G07
40
0
20
40
60
VIN (V)
80
100
8300 G08
280
0
20
40
60
VIN (V)
80
100
8300 G09
8300f
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LT8300
Typical Performance Characteristics
EN/UVLO Enable Threshold
TA = 25°C, unless otherwise noted.
EN/UVLO Hysteresis Current
1.240
105
5
104
1.235
103
4
1.230
102
1.220
1.215
3
IRFB (µA)
1.225
IHYS (µA)
VEN/UVLO (V)
RFB Regulation Current
2
101
100
99
98
1.210
97
1
1.205
96
1.200
–50 –25
0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
95
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
8300 G10
0
25 50 75 100 125 150
TEMPERATURE (°C)
8300 G12
8300 G11
RDS(ON)
Switch Current Limit
Maximum Switching Frequency
300
25
1000
MAXIMUM CURRENT LIMIT
250
800
10
FREQUENCY (kHz)
200
15
ISW (mA)
RESISTANCE (Ω)
20
150
100
5
MINIMUM CURRENT LIMIT
50
0
–50 –25
0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
Minimum Switching Frequency
0
–50 –25
400
300
300
TIME (ns)
FREQUENCY (kHz)
TIME (ns)
25 50 75 100 125 150
TEMPERATURE (°C)
200
8300 G16
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
Minimum Switch-Off Time
400
100
0
0
8300 G15
Minimum Switch-On Time
16
4
0
–50 –25
8300 G14
20
8
400
200
25 50 75 100 125 150
TEMPERATURE (°C)
8300 G13
12
600
200
100
0
25 50 75 100 125 150
TEMPERATURE (°C)
8300 G17
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
8300 G18
8300f
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LT8300
Pin Functions
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8300. Pull the pin
below 0.3V to shut down the LT8300. This pin has an accurate 1.223V threshold and can be used to program a VIN
undervoltage lockout (UVLO) threshold using a resistor
divider from VIN to ground. A 2.5µA current hysteresis
allows the programming of VIN UVLO hysteresis. If neither
function is used, tie this pin directly to VIN.
mary SW pin. The ratio of the RFB resistor to the internal
trimmed 12.23k resistor, times the internal bandgap
reference, determines the output voltage (plus the effect
of any non-unity transformer turns ratio). Minimize trace
area at this pin.
GND (Pin 2): Ground. Tie this pin directly to local ground
plane.
VIN (Pin 5): Input Supply. The VIN pin supplies current
to internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the RFB pin. Locally
bypass this pin to ground with a capacitor.
RFB (Pin 3): Input Pin for External Feedback Resistor.
Connect a resistor from this pin to the transformer pri-
SW (Pin 4): Drain of the 150V Internal DMOS Power
Switch. Minimize trace area at this pin to reduce EMI and
voltage spikes.
8300f
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LT8300
Block Diagram
T1
NPS:1
VIN
CIN
LPRI
•
•
DOUT
VOUT+
LSEC
COUT
RFB
5
3
VIN
VOUT–
4
RFB
SW
BOUNDARY
DETECTOR
1:4
M3
M2
OSCILLATOR
–
25µA
RREF
12.23kΩ
1.223V
+
–
gm
+
S
A3
R
Q
DRIVER
M1
R1
1
–
EN/UVLO
2.5µA
R2
1.223V
M4
+
+
A2
A1
VIN
REFERENCE
REGULATORS
–
RSENSE
0.3Ω
GND
2
8300 BD
8300f
7
LT8300
Operation
The LT8300 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key
problem in isolated topologies is how to communicate the
output voltage information from the isolated secondary
side of the transformer to the primary side for regulation.
Historically, opto-isolators or extra transformer windings
communicate this information across the isolation boundary. Opto-isolator circuits waste output power, and the
extra components increase the cost and physical size of
the power supply. Opto-isolators can also cause system
issues due to limited dynamic response, nonlinearity, unitto-unit variation and aging over lifetime. Circuits employing
extra transformer windings also exhibit deficiencies, as
using an extra winding adds to the transformer’s physical
size and cost, and dynamic response is often mediocre.
The LT8300 samples the isolated output voltage through
the primary-side flyback pulse waveform. In this manner,
neither opto-isolator nor extra transformer winding is required for regulation. Since the LT8300 operates in either
boundary conduction mode or discontinuous conduction
mode, the output voltage is always sampled on the SW
pin when the secondary current is zero. This method improves load regulation without the need of external load
compensation components.
The LT8300 is a simple to use micropower isolated flyback
converter housed in a 5-lead TSOT-23 package. The output
voltage is programmed with a single external resistor. By
integrating the loop compensation and soft-start inside, the
part further reduces the number of external components.
As shown in the Block Diagram, many of the blocks are
similar to those found in traditional switching regulators
including reference, regulators, oscillator, logic, current
amplifier, current comparator, driver, and power switch.
The novel sections include a flyback pulse sense circuit,
a sample-and-hold error amplifier, and a boundary mode
detector, as well as the additional logic for boundary
conduction mode, discontinuous conduction mode, and
low ripple Burst Mode operation.
Boundary Conduction Mode Operation
The LT8300 features boundary conduction mode operation
at heavy load, where the chip turns on the primary power
switch when the secondary current is zero. Boundary
conduction mode is a variable frequency, variable peakcurrent switching scheme. The power switch turns on
and the transformer primary current increases until an
internally controlled peak current limit. After the power
switch turns off, the voltage on the SW pin rises to the
output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
the SW pin voltage collapses and rings around VIN. A
boundary mode detector senses this event and turns the
power switch back on.
Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduction mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit sub-harmonic oscillation.
Discontinuous Conduction Mode Operation
As the load gets lighter, boundary conduction mode increases the switching frequency and decreases the switch
peak current at the same ratio. Running at a higher switching
frequency up to several MHz increases switching and gate
charge losses. To avoid this scenario, the LT8300 has an
additional internal oscillator, which clamps the maximum
switching frequency to be less than 750kHz. Once the
switching frequency hits the internal frequency clamp,
the part starts to delay the switch turn-on and operates
in discontinuous conduction mode.
Low Ripple Burst Mode Operation
Unlike traditional flyback converters, the LT8300 has to
turn on and off at least for a minimum amount of time
and with a minimum frequency to allow accurate sampling
of the output voltage. The inherent minimum switch current limit and minimum switch-off time are necessary to
guarantee the correct operation of specific applications.
As the load gets very light, the LT8300 starts to fold back
the switching frequency while keeping the minimum switch
current limit. So the load current is able to decrease while
still allowing minimum switch-off time for the sampleand-hold error amplifier. Meanwhile, the part switches
between sleep mode and active mode, thereby reducing the
8300f
8
LT8300
Operation
effective quiescent current to improve light load efficiency.
In this condition, the LT8300 operates in low ripple Burst
Mode. The typical 7.5kHz minimum switching frequency
determines how often the output voltage is sampled and
also the minimum load requirement.
Applications Information
Output Voltage
The RFB resistor as depicted in the Block Diagram is the
only external resistor used to program the output voltage.
The LT8300 operates similar to traditional current mode
switchers, except in the use of a unique flyback pulse
sense circuit and a sample-and-hold error amplifier, which
sample and therefore regulate the isolated output voltage
from the flyback pulse.
Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the VIN supply. The
amplitude of the flyback pulse, i.e., the difference between
the SW pin voltage and VIN supply, is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = Output diode forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current IRFB by
the flyback pulse sense circuit (M2 and M3). This current IRFB also flows through the internal trimmed 12.23k
RREF resistor to generate a ground-referred voltage. The
resulting voltage feeds to the inverting input of the sampleand-hold error amplifier. Since the sample-and-hold error
amplifier samples the voltage when the secondary current
is zero, the (ISEC • ESR) term in the VFLBK equation can be
assumed to be zero.
The bandgap reference voltage VBG, 1.223V, feeds to the
non-inverting input of the sample-and-hold error amplifier. The relatively high gain in the overall loop causes
the voltage across RREF resistor to be nearly equal to the
bandgap reference voltage VBG. The resulting relationship
between VFLBK and VBG can be expressed as:
 VFLBK 
 R
 • RREF = VBG
FB 
or
 V 
VFLBK =  BG  • RFB = I RFB • RFB
 RREF 
VBG = Bandgap reference voltage
IRFB = RFB regulation current = 100µA
Combination with the previous VFLBK equation yields an
equation for VOUT, in terms of the RFB resistor, transformer
turns ratio, and diode forward voltage:
R 
VOUT = 100µA •  FB  − VF
 NPS 
Output Temperature Coefficient
The first term in the VOUT equation does not have temperature dependence, but the output diode forward voltage VF
has a significant negative temperature coefficient (–1mV/°C
to –2mV/°C). Such a negative temperature coefficient produces approximately 200mV to 300mV voltage variation
on the output voltage across temperature.
For higher voltage outputs, such as 12V and 24V, the output
diode temperature coefficient has a negligible effect on the
output voltage regulation. For lower voltage outputs, such
as 3.3V and 5V, however, the output diode temperature
coefficient does count for an extra 2% to 5% output voltage
regulation. For customers requiring tight output voltage
regulation across temperature, please refer to other LTC
parts with integrated temperature compensation features.
8300f
9
LT8300
Applications Information
Selecting Actual RFB Resistor Value
Output Power
The LT8300 uses a unique sampling scheme to regulate
the isolated output voltage. Due to the sampling nature,
the scheme contains repeatable delays and error sources,
which will affect the output voltage and force a re-evaluation
of the RFB resistor value. Therefore, a simple two-step
process is required to choose feedback resistor RFB.
A flyback converter has a complicated relationship between
the input and output currents compared to a buck or a
boost converter. A boost converter has a relatively constant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum output
current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flyback converter has both discontinuous input and output
currents which make it similar to a non-isolated buck-boost
converter. The duty cycle will affect the input and output
currents, making it hard to predict output power. In addition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.
Rearrangement of the expression for VOUT in the Output
Voltage section yields the starting value for RFB:
RFB =
(
NPS • VOUT + VF
100µA
)
VOUT = Output voltage
VF = Output diode forward voltage = ~0.3V
NPS = Transformer effective primary-to-secondary
turns ratio
Power up the application with the starting RFB value and
other components connected, and measure the regulated
output voltage, VOUT(MEAS). The final RFB value can be
adjusted to:
VOUT
RFB(FINAL) =
• RFB
VOUT(MEAS)
Once the final RFB value is selected, the regulation accuracy
from board to board for a given application will be very
consistent, typically under ±5% when including device
variation of all the components in the system (assuming
resistor tolerances and transformer windings matching
within ±1%). However, if the transformer or the output
diode is changed, or the layout is dramatically altered,
there may be some change in VOUT.
The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,
12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 120V during the switch-off time. 30V of margin is left for leakage
inductance voltage spike. To achieve this power level at
a given input, a winding ratio value must be calculated
to stress the switch to 120V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
One design example would be a 5V output converter with
a minimum input voltage of 36V and a maximum input
voltage of 72V. A six-to-one winding ratio fits this design
example perfectly and outputs equal to 2.44W at 72V but
lowers to 1.87W at 36V.
The following equations calculate output power:
POUT = η • VIN • D •ISW(MAX) • 0.5
η = Efficiency = 85%
( VOUT + VF ) • NPS
D = DutyCycle =
( VOUT + VF ) • NPS + VIN

ISW(MAX) = Maximum switch current limit = 260mA
8300f
10
LT8300
Applications Information
3.5
3.5
MAXIMUM
OUTPUT
POWER
2.5
N = 8:1
2.0
N = 6:1
1.5
N = 4:1
1.0
0.5
0
MAXIMUM
OUTPUT
N = 8:1
POWER
3.0
N = 12:1
OUTPUT POWER (W)
OUTPUT POWER (W)
3.0
2.5
N = 6:1
N = 4:1
2.0
1.5
N = 2:1
1.0
0.5
0
20
40
60
INPUT VOLTAGE (V)
0
100
80
0
20
40
60
INPUT VOLTAGE (V)
8300 F01
8300 F02
Figure 1. Output Power for 3.3V Output
3.5
3.5
N = 2:1
3.0
3.0
N = 3:1
MAXIMUM
OUTPUT
POWER
OUTPUT POWER (W)
OUTPUT POWER (W)
Figure 2. Output Power for 5V Output
N = 4:1
2.5
N = 2:1
2.0
1.5
N = 1:1
1.0
0.5
0
N = 3:2
MAXIMUM
OUTPUT
POWER
2.5
2.0
N = 1:1
1.5
N = 1:2
1.0
0.5
0
20
40
60
INPUT VOLTAGE (V)
0
100
80
0
20
40
60
INPUT VOLTAGE (V)
8300 F03
Primary Inductance Requirement
The LT8300 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction
of secondary current reflects the output voltage on the
primary SW pin. The sample-and-hold error amplifier needs
a minimum 350ns to settle and sample the reflected output
voltage. In order to ensure proper sampling, the secondary winding needs to conduct current for a minimum of
350ns. The following equation gives the minimum value
for primary-side magnetizing inductance:
(
tOFF(MIN) • NPS • VOUT + VF
ISW(MIN)
)
tOFF(MIN) = Minimum switch-off time = 350ns
ISW(MIN) = Minimum switch current limit = 52mA
80
100
8300 F04
Figure 3. Output Power for 12V Output
LPRI ≥
100
80
Figure 4. Output Power for 24V Output
In addition to the primary inductance requirement for
the minimum switch-off time, the LT8300 has minimum
switch-on time that prevents the chip from turning on
the power switch shorter than approximately 160ns. This
minimum switch-on time is mainly for leading-edge blanking the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loop will lose its ability to regulate. Therefore, the following
equation relating to maximum input voltage must also be
followed in selecting primary-side magnetizing inductance:
LPRI ≥
tON(MIN) • VIN(MAX)
ISW(MIN)
tON(MIN) = Minimum Switch-On Time = 160ns
8300f
11
LT8300
Applications Information
In general, choose a transformer with its primary magnetizing inductance about 20% to 40% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed
flyback transformers for use with the LT8300. Table 1
shows the details of these transformers.
Selecting a Transformer
Note that when choosing the RFB resistor to set output
voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast,
the use of simple ratios of small integers, e.g., 4:1, 2:1,
1:1, provides more freedom in settling total turns and
mutual inductance.
Transformer specification and design is perhaps the most
critical part of successfully applying the LT8300. In addition
to the usual list of guidelines dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
Turns Ratio
Table 1. Predesigned Transformers — Typical Specifications
TRANSFORMER
PART NUMBER
LPRI
(µH)
LLEAKAGE
(µH)
NP:NS:NB
VENDOR
750312367
400
4.5
8:1
Würth Elektronik
48V to 3.3V/0.51A, 24V to 3.3V/0.37A, 12V to 3.3V/0.24A
750312557
300
2.5
6:1
Würth Elektronik
48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
750312365
300
1.8
4:1
Würth Elektronik
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
750312558
300
1.75
2:1:1
Würth Elektronik
48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA
48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA
750312559
300
2
1:1
Würth Elektronik
48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA
750311019
400
5
6:1:2
Würth Elektronik
48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
750311558
300
1.5
4:1:1
Würth Elektronik
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
750311660
350
3
2:1:0.33
Würth Elektronik
48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A
48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A
750311838
350
3
2:1:1
Würth Elektronik
48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA
48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA
48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA
TARGET APPLICATIONS
750311659
300
2
1:1:0.2
Würth Elektronik
10396-T026
300
2.5
6:1:2
Sumida
48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
10396-T024
300
2
4:1:1
Sumida
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
10396-T022
300
2
2:1:0.33
Sumida
48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A
48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A
10396-T028
300
2.5
2:1:1
Sumida
48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA
48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA
L10-0116
500
7.3
6:1
BH Electronics
48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A
48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A
L10-0112
230
3.38
4:1
BH Electronics
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
L11-0067
230
2.16
4:1
BH Electronics
48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A
* All the transformers are rated for 1.5kV Isolation.
8300f
12
LT8300
Applications Information
Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V or
5V), a larger N:1 turns ratio can be used with multiple
primary windings relative to the secondary to maximize the
transformer’s current gain (and output power). However,
remember that the SW pin sees a voltage that is equal
to the maximum input supply voltage plus the output
voltage multiplied by the turns ratio. In addition, leakage
inductance will cause a voltage spike (VLEAKAGE) on top of
this reflected voltage. This total quantity needs to remain
below the 150V absolute maximum rating of the SW pin
to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns
ratio, NPS, for a given application. Choose a turns ratio
low enough to ensure:
NPS <
150V − VIN(MAX) − VLEAKAGE
VOUT + VF
For lower output power levels, choose a smaller N:1 turns
ratio to alleviate the SW pin voltage stress. Although a
1:N turns ratio makes it possible to have very high output
voltages without exceeding the breakdown voltage of the
internal power switch, the multiplied parasitic capacitance
through turns ratio coupled with the relatively resistive
150V internal power switch may cause the switch turn-on
current spike ringing beyond 160ns leading-edge blanking,
thereby producing light load instability in certain applications. So any 1:N turns ratio should be fully evaluated
before its use with the LT8300.
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer specifies turns ratio accuracy within ±1%.
Saturation Current
The current in the transformer windings should not exceed
its rated saturation current. Energy injected once the core is
saturated will not be transferred to the secondary and will
instead be dissipated in the core. When designing custom
transformers to be used with the LT8300, the saturation
current should always be specified by the transformer
manufacturers.
Winding Resistance
Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output voltage
regulation will be maintained independent of winding resistance due to the boundary/discontinuous conduction
mode operation of the LT8300.
Leakage Inductance and Snubbers
Transformer leakage inductance on either the primary or
secondary causes a voltage spike to appear on the primary
after the power switch turns off. This spike is increasingly
prominent at higher load currents where more stored energy must be dissipated. It is very important to minimize
transformer leakage inductance.
When designing an application, adequate margin should
be kept for the worst-case leakage voltage spikes even
under overload conditions. In most cases shown in Figure
5, the reflected output voltage on the primary plus VIN
should be kept below 120V. This leaves at least 30V margin
for the leakage spike across line and load conditions. A
larger voltage margin will be required for poorly wound
transformers or for excessive leakage inductance.
In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trigger boundary mode detector, the LT8300 internally blanks
the boundary mode detector for approximately 250ns. Any
remaining voltage ringing after 250ns may turn the power
switch back on again before the secondary current falls
to zero. So the leakage inductance spike ringing should
be limited to less than 250ns.
8300f
13
LT8300
Applications Information
VSW
VSW
<150V
VSW
<150V
<150V
VLEAKAGE
VLEAKAGE
<120V
VLEAKAGE
<120V
<120V
tOFF > 350ns
tOFF > 350ns
tOFF > 350ns
tSP < 250ns
tSP < 250ns
tSP < 250ns
TIME
TIME
No Snubber
TIME
with DZ Snubber
with RC Snubber
8300 F05
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
Lℓ
Lℓ
•
Z
D
•
C
•
R
8300 F06a
•
8300 F06b
DZ Snubber
RC Snubber
Figure 6. Snubber Circuits
A snubber circuit is recommended for most applications.
Two types of snubber circuits shown in Figure 6 that can
protect the internal power switch include the DZ (diodeZener) snubber and the RC (resistor-capacitor) snubber. The
DZ snubber ensures well defined and consistent clamping
voltage and has slightly higher power efficiency, while the
RC snubber quickly damps the voltage spike ringing and
provides better load regulation and EMI performance.
Figure 5 shows the flyback waveforms with the DZ and
RC snubbers.
For the DZ snubber, proper care must be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leakage inductance spike. Choose a diode that has a reversevoltage rating higher than the maximum SW pin voltage.
The Zener diode breakdown voltage should be chosen to
balance power loss and switch voltage protection. The best
compromise is to choose the largest voltage breakdown.
Use the following equation to make the proper choice:
VZENER(MAX) ≤ 150V – VIN(MAX)
For an application with a maximum input voltage of 72V,
choose a 68V Zener diode, the VZENER(MAX) of which is
around 72V and below the 78V maximum.
The power loss in the clamp will determine the power rating of the Zener diode. Power loss in the clamp is highest
at maximum load and minimum input voltage. The switch
current is highest at this point along with the energy stored
in the leakage inductance. A 0.5W Zener will satisfy most
applications when the highest VZENER is chosen.
8300f
14
LT8300
Applications Information
Tables 2 and 3 show some recommended diodes and
Zener diodes.
Table 2. Recommended Zener Diodes
VZENER
(V)
POWER
(W)
CASE
VENDOR
MMSZ5266BT1G
68
0.5
SOD-123
On Semi
MMSZ5270BT1G
91
0.5
SOD-123
CMHZ5266B
68
0.5
SOD-123
CMHZ5267B
75
0.5
SOD-123
BZX84J-68
68
0.5
SOD323F NXP
BZX100A
100
0.5
SOD323F
PART
Central
Semiconductor
Table 3. Recommended Diodes
PART
I (A)
VREVERSE
(V)
BAV21W
0.625
200
SOD-123 Diodes Inc.
BAV20W
0.625
150
SOD-123
CASE
VENDOR
The recommended approach for designing an RC snubber
is to measure the period of the ringing on the SW pin when
the power switch turns off without the snubber and then
add capacitance (starting with 100pF) until the period of
the ringing is 1.5 to 2 times longer. The change in period
will determine the value of the parasitic capacitance, from
which the parasitic inductance can be determined from
the initial period, as well. Once the value of the SW node
capacitance and inductance is known, a series resistor can
be added to the snubber capacitance to dissipate power
and critically dampen the ringing. The equation for deriving
the optimal series resistance using the observed periods
( tPERIOD and tPERIOD(SNUBBED)) and snubber capacitance
(CSNUBBER) is:
CPAR =
CSNUBBER
2
Note that energy absorbed by the RC snubber will be
converted to heat and will not be delivered to the load.
In high voltage or high current applications, the snubber
may need to be sized for thermal dissipation.
Undervoltage Lockout (UVLO)
A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). The EN/UVLO pin
falling threshold is set at 1.223V with 16mV hysteresis.
In addition, the EN/UVLO pin sinks 2.5µA when the voltage at the pin is below 1.223V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:
1.239V • (R1+ R2)
+ 2.5µA • R1
R2
1.223V • (R1+ R2)
VIN(UVLO−) =
R2
VIN(UVLO+) =
Figure 7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8300 in shutdown with quiescent current less than 2µA.
VIN
R1
EN/UVLO
LT8300
R2
RUN/STOP
CONTROL
(OPTIONAL)
GND
8300 F07
Figure 7. Undervoltage Lockout (UVLO)
 tPERIOD(SNUBBED) 

 − 1
t
PERIOD
L PAR =
tPERIOD 2
CPAR • 4π 2
RSNUBBER =
LPAR
CPAR
8300f
15
LT8300
Applications Information
Minimum Load Requirement
Design Example
The LT8300 samples the isolated output voltage from the
primary-side flyback pulse waveform. The flyback pulse
occurs once the primary switch turns off and the secondary
winding conducts current. In order to sample the output
voltage, the LT8300 has to turn on and off at least for a
minimum amount of time and with a minimum frequency.
The LT8300 delivers a minimum amount of energy even
during light load conditions to ensure accurate output voltage information. The minimum energy delivery creates a
minimum load requirement, which can be approximately
estimated as:
Use the following design example as a guide to design
applications for the LT8300. The design example involves
designing a 12V output with a 120mA load current and an
input range from 36V to 72V.
I LOAD(MIN) =
2
L PRI • I SW(MIN)
• f MIN
2 • VOUT
LPRI = Transformer primary inductance
ISW(MIN) = Minimum switch current limit = 52mA
fMIN = Minimum switching frequency = 7.5kHz
The LT8300 typically needs less than 0.5% of its full output
power as minimum load. Alternatively, a Zener diode with its
breakdown of 20% higher than the output voltage can serve
as a minimum load if pre-loading is not acceptable. For a 5V
output, use a 6V Zener with cathode connected to the output.
Output Short Protection
When the output is heavily overloaded or shorted, the
reflected SW pin waveform rings longer than the internal
blanking time. After the 350ns minimum switch-off time,
the excessive ring falsely trigger the boundary mode
detector and turn the power switch back on again before
the secondary current falls to zero. Under this condition,
the LT8300 runs into continuous conduction mode at
750kHz maximum switching frequency. Depending on the
VIN supply voltage, the switch current may run away and
exceed 260mA maximum current limit. Once the switch
current hits 520mA over current limit, a soft-start cycle
initiates and throttles back both switch current limit and
switch frequency. This output short protection prevents the
switch current from running away and limits the average
output diode current.
VIN(MIN) = 36V, VIN(NOM) = 48V, VIN(MAX) = 72V,
VOUT = 12V, IOUT = 120mA
Step 1: Select the Transformer Turns Ratio.
NPS <
150V − VIN(MAX) − VLEAKAGE
VOUT + VF
VLEAKAGE = Margin for transformer leakage spike = 30V
VF = Output diode forward voltage = ~0.3V
Example:
NPS <
150V − 72V − 30V
= 3.9
12V + 0.3V
The choice of transformer turns ratio is critical in determining output current capability of the converter. Table 4
shows the switch voltage stress and output current capability at different transformer turns ratio.
Table 4. Switch Voltage Stress and Output Current Capability
vs Turns Ratio
NPS
VSW(MAX) at
VIN(MAX) (V)
IOUT(MAX) at
VIN(MIN) (mA)
DUTY CYCLE (%)
1:1
84.3
84
15-25
2:1
96.6
135
25-41
3:1
108.9
168
34-51
Since both NPS = 2 and NPS = 3 can meet the 120mA output
current requirement, NPS = 2 is chosen in this example
to allow more margin for transformer leakage inductance
voltage spike.
8300f
16
LT8300
Applications Information
Step 2: Determine the Primary Inductance.
Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:
LPRI ≥
LPRI ≥
(
tOFF(MIN) • NPS • VOUT + VF
ISW(MIN)
)
tON(MIN) • VIN(MAX)
ISW(MIN)
tOFF(MIN) = 350ns
tON(MIN) = 160ns
ISW(MIN) = 52mA
Example:
350ns • 2 • (12V + 0.3V)
= 166µH
52mA
160ns • 72V
LPRI ≥
= 222µH
52mA
LPRI ≥
Most transformers specify primary inductance with a tolerance of ±20%. With other component tolerance considered,
choose a transformer with its primary inductance 20% to
40% larger than the minimum values calculated above.
LPRI = 300µH is then chosen in this example.
Once the primary inductance has been determined, the
maximum load switching frequency can be calculated as:
fSW =
ISW =
1
1
=
LPRI •ISW
tON + tOFF LPRI •ISW +
VIN
NPS • (VOUT + VF )
VOUT •IOUT • 2
η • VIN • D
Example:
(12V + 0.3V) • 2
= 0.34
(12V + 0.3V) • 2 + 48V
12V • 0.12A • 2
= 0.21A
ISW =
0.85 • 48V • 0.34
fSW = 260kHz
D=
The transformer also needs to be rated for the correct
saturation current level across line and load conditions. A
saturation current rating larger than 400mA is necessary
to work with the LT8300. The 10396-T022 from Sumida
is chosen as the flyback transformer.
Step 3: Choose the Output Diode.
Two main criteria for choosing the output diode include
forward current rating and reverse voltage rating. The
maximum load requirement is a good first-order guess
as the average current requirement for the output diode.
A conservative metric is the maximum switch current limit
multiplied by the turns ratio,
IDIODE(MAX) = ISW(MAX) • NPS
Example:
IDIODE(MAX) = 0.52A
Next calculate reverse voltage requirement using maximum VIN:
VREVERSE = VOUT +
VIN(MAX)
NPS
Example:
VREVERSE = 12V +
72V
= 48V
2
The SBR0560S1 (0.5A, 60V diode) from Diodes Inc. is
chosen.
8300f
17
LT8300
Applications Information
Step 4: Choose the Output Capacitor.
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
and cost of a larger capacitor. Use the equation below to
calculate the output capacitance:
COUT =
L PRI • I SW
2
2 • VOUT • ∆VOUT
Example:
Design for output voltage ripple less than 1% of VOUT,
i.e., 120mV.
COUT
300µH • (0.21A)2
=
= 4.6µF
2 • 12V • 0.12V
Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop to 40% of quoted
capacitance at the maximum voltage rating. So a 10uF, 16V
rating ceramic capacitor is chosen.
Step 5: Design Snubber Circuit.
The snubber circuit protects the power switch from leakage
inductance voltage spike. A DZ snubber is recommended
for this application because of lower leakage inductance
and larger voltage margin. The Zener and the diode need
to be selected.
The maximum Zener breakdown voltage is set according
to the maximum VIN:
VZENER(MAX) ≤ 150V – VIN(MAX)
A 68V Zener with a maximum of 72V will provide optimal
protection and minimize power loss. So a 68V, 0.5W Zener
from On Semiconductor (MMSZ5266BT1G) is chosen.
Choose a diode that is fast and has sufficient reverse
voltage breakdown:
VREVERSE > VSW(MAX)
VSW(MAX) = VIN(MAX) + VZENER(MAX)
Example:
VREVERSE > 144V
A 150V, 0.6A diode from Diodes Inc. (BAV20W) is chosen.
Step 6: Select the RFB Resistor.
Use the following equation to calculate the starting value
for RFB:
RFB =
NPS • (VOUT + VF )
100µA
Example:
RFB =
2 • (12V + 0.3V)
= 246k
100µA
Depending on the tolerance of standard resistor values,
the precise resistor value may not exist. For 1% standard
values, a 243k resistor in series with a 3.01k resistor
should be close enough. As discussed in the Application
Information section, the final RFB value should be adjusted
on the measured output voltage.
Example:
VZENER(MAX) ≤ 150V – 72V = 78V
8300f
18
LT8300
Applications Information
Step 7: Select the EN/UVLO Resistors.
Step 8: Ensure minimum load.
Determine the amount of hysteresis required and calculate
R1 resistor value:
The theoretical minimum load can be approximately
estimated as:
VIN(HYS) = 2.5µA • R1
Example:
Choose 2.5V of hysteresis,
R1 = 1M
Determine the UVLO thresholds and calculate R2 resistor
value:
VIN(UVLO+) =
1.239V • (R1+ R2)
+ 2.5µA • R1
R2
ILOAD(MIN) =
300µH • (52mA)2 • 7.5kHz
= 0.25mA
2 • 12V
Remember to check the minimum load requirement in real
application. The minimum load occurs at the point where
the output voltage begins to climb up as the converter
delivers more energy than what is consumed at the output. The real minimum load for this application is about
0.6mA, 0.5% of 120mA maximum load. In this example,
a 20k resistor is selected as the minimum load.
Example:
Set VIN UVLO rising threshold to 34.5V,
R2 = 40.2k
VIN(UVLO+) = 34.1V
VIN(UVLO–) = 31.6V
8300f
19
LT8300
Typical Applications
5V Micropower Isolated Flyback Converter
D1
T1
6:1
VIN
36V TO 72V
2.2µF
1M
8µH
47µF
•
LT8300
EN/UVLO
•
300µH
VIN
VOUT+
5V
1mA TO 330mA
VOUT–
SW
40.2k
316k
RFB
GND
T1: WÜRTH 750312557
D1: DIODES INC. SBR2A30P1
8300 TA02
12V Micropower Isolated Flyback Converter
T1
2:1
VIN
36V TO 72V
2.2µF
1M
75µH
•
LT8300
EN/UVLO
•
300µH
VIN
D1
VOUT+
12V
0.6mA TO 120mA
10µF
VOUT–
SW
243k
40.2k
RFB
GND
8300 TA03
T1: SUMIDA 10396-TO22
D1: DIODES INC. SBR0560S1
8300f
20
LT8300
Typical Applications
24V Micropower Isolated Flyback Converter
T1
1:1
VIN
36V TO 72V
2.2µF
1M
300µH
•
LT8300
EN/UVLO
•
300µH
VIN
D1
VOUT+
24V
0.3mA TO 60mA
4.7µF
VOUT–
SW
243k
40.2k
RFB
GND
T1: WÜRTH 750311559
D1: DIODES DFLS 1200-7
8300 TA04
3.3V Micropower Isolated Flyback Converter
T1
8:1
VIN
36V TO 72V
2.2µF
1M
6µH
•
LT8300
EN/UVLO
•
400µH
VIN
D1
VOUT+
3.3V
2mA TO 440mA
100µF
VOUT–
SW
40.2k
287k
RFB
GND
8300 TA05
T1: WÜRTH 750312367
D1: NXP PMEG2020EH
8300f
21
LT8300
Typical Applications
VIN to (VIN + 10V) Micropower Converter
VOUT+
10V
50mA
4.7µF
VOUT–
VIN
15V TO 80V
1µF
L1
330µH
VIN
1M
Z1
LT8300
D1
SW
EN/UVLO
102k
118k
RFB
GND
L1: COILTRONICS DR73-331-R
D1: DIODES INC. SBR1U150SA
Z1: CENTRAL CMDZ12L
8300 TA06
VIN to (VIN – 10V) Micropower Converter
VIN
15V TO 80V
VOUT+
10V
100mA
1µF
4.7µF
Z1
–
VOUT
1M
L1
330µH
VIN
LT8300
EN/UVLO
D1
SW
102k
118k
RFB
GND
L1: COILTRONICS DR73-331-R
D1: DIODES INC. SBR1U150SA
Z1: CENTRAL CMDZ12L
8300 TA07
8300f
22
LT8300
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
S5 Package
5-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1635 Rev B)
0.62
MAX
0.95
REF
2.90 BSC
(NOTE 4)
1.22 REF
1.4 MIN
3.85 MAX 2.62 REF
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45 TYP
5 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1.90 BSC
S5 TSOT-23 0302 REV B
8300f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT8300
Typical Application
3.3V Isolated Converter (Conforming to DEF-STAN61-5)
L1
1:1
VIN
18V TO 32V
1µF
1M
VOUT+
3.3V
0mA TO 20mA
OUT
LT3009-3.3
150µH
Z1
1µF
•
LT8300
EN/UVLO
IN
•
150µH
VIN
D1
SHDN
1µF
GND
VOUT–
SW
42.2k
93.1k
RFB
GND
D1: DIODES INC. SBR0560S1-7
L1: DRQ73-151-R
Z1: CENTRAL CMDZ4L7
8300 TA08a
Input Current with No Load
400
IVIN (µA)
300
200
100
0
18
20
22
24
26
VIN (V)
28
30
32
8300 TA08b
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LT3511/LT3512
100V Isolated Flyback Converters
Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch,
MSOP-16(12)
LT3748
100V Isolated Flyback Controller
5V ≤ VIN ≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing
LT3798
Off-Line Isolated No Opto-Coupler Flyback Controller
with Active PFC
VIN and VOUT Limited Only by External Components
LT3573/LT3574/LT3575
40V Isolated Flyback Converters
Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch
LT3757/LT3759/LT3758
40V/100V Flyback/Boost Controllers
Universal Controllers with Small Package and Powerful Gate Drive
LT3957/LT3958
40V/100V Flyback/Boost Converters
Monolithic with Integrated 5A/3.3A Switch
LTC3803/LTC3803-3/
LTC3803-5
200kHz/300kHz Flyback Controllers in SOT-23
VIN and VOUT Limited by External Components
LTC3805/LTC3805-5
Adjustable Frequency Flyback Controllers
VIN and VOUT Limited by External Components
8300f
24 Linear Technology Corporation
LT 0812 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2012
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