ON NCP81044DR2G Low voltage synchronous buck controller Datasheet

NCP81044
Product Preview
Low Voltage Synchronous
Buck Controller
The NCP81044 is a PWM controller designed to operate from a 5 V
or 12 V supply. These devices are capable of producing an output
voltage as low as 0.8 V. These 8−pin devices provide an optimal level
of integration to reduce size and cost of the power supply. The
NCP81044 provides a 1 A gate driver design and an internally set
275 kHz oscillator. In addition to the 1 A gate drive capability, other
efficiency enhancing features of the gate driver include adaptive
non−overlap circuitry. The devices also incorporate an externally
compensated error amplifier and a capacitor programmable soft−start
function. Protection features include programmable short circuit
protection and under voltage lockout (UVLO). The NCP81044 comes
in an 8−pin SOIC package.
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MARKING DIAGRAM
8
SOIC−8
D SUFFIX
CASE 751
8
1
1
81044 = Specific Device Code
A
= Assembly Location
L
= Wafer Lot
Y
= Year
W
= Work Week
G
= Pb−Free Device
Features
•
•
•
•
•
•
•
•
•
•
•
•
Input Voltage Range from 4.5 to 13.2 V
275 kHz Internal Oscillator
Boost Pin Operates to 30 V
Voltage Mode PWM Control
0.8 V ±1.0 % Internal Reference Voltage
Adjustable Output Voltage
Capacitor Programmable Soft−Start
Internal 1 A Gate Drivers
80% Max Duty Cycle
Input Under Voltage Lockout
Programmable Current Limit
These Devices are Pb−Free, Halogen Free/BFR Free and are RoHS
Compliant
Applications
•
•
•
•
•
Graphics Cards
Desktop Computers
Servers / Networking
DSP & FPGA Power Supply
DC−DC Regulator Modules
81044
ALYW
G
PIN CONNECTIONS
BST 1
8 PHASE
TG 2
7 COMP/DIS
GND 3
6 FB
BG 4
5 VCC
(Top View)
ORDERING INFORMATION
Device
Package
Shipping†
NCP81044DR2G
SOIC−8
(Pb−Free)
2500/Tape & Reel
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specifications
Brochure, BRD8011/D.
This document contains information on a product under development. ON Semiconductor
reserves the right to change or discontinue this product without notice.
© Semiconductor Components Industries, LLC, 2012
June, 2012 − Rev. P0
1
Publication Order Number:
NCP81044/D
NCP81044
12 V
3.3 V
VCC
BST
FB
COMP/DIS
TG
VOUT
PHASE
BG
GND
Figure 1. Typical Application Diagram
POR
UVLO
5
VCC
1
BST
2
TG
8
PHASE
VOCTH
FAULT
FB
6
+
LATCH
GM
+
-
0.8 V
(VREF)
SCP
FAULT
R
S
PWM
OUT
Q
+
-
Clock
2V
+
-
Ramp
COMP/DIS
7
+
-
OSC
VCC
4
OSC
FAULT
Figure 2. Detailed Block Diagram
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2
3
BG
Rset
GND
NCP81044
PIN FUNCTION DESCRIPTION
Pin No.
Symbol
Description
1
BST
Supply rail for the floating top gate driver. To form a boost circuit, use an external diode to bring the
desired input voltage to this pin (cathode connected to BST pin). Connect a capacitor (CBST) between this pin
and the PHASE pin. Typical values for CBST range from 0.1 mF to 1 mF. Ensure that CBST is placed near the IC.
2
TG
Top gate MOSFET driver pin. Connect this pin to the gate of the top N−Channel MOSFET.
3
GND
4
BG
Bottom gate MOSFET driver pin. Connect this pin to the gate of the bottom N−Channel MOSFET.
5
VCC
Supply rail for the internal circuitry. Operating supply range is 4.5 V to 13.2 V. Decouple with a 1 mF
capacitor to GND. Ensure that this decoupling capacitor is placed near the IC.
6
FB
This pin is the inverting input to the error amplifier. Use this pin in conjunction with the COMP pin to
compensate the voltage−control feedback loop. Connect this pin to the output resistor divider (if used) or directly to Vout.
7
COMP/DIS
8
PHASE
IC ground reference. All control circuits are referenced to this pin.
Compensation Pin. This is the output of the error amplifier (EA) and the non−inverting input of the PWM comparator. Use this pin in conjunction with the FB pin to compensate the voltage−control feedback loop. The compensation capacitor also acts as a soft−start capacitor. Pull this pin low for disable.
Switch node pin. This is the reference for the floating top gate driver. Connect this pin to the source of the top
MOSFET.
ABSOLUTE MAXIMUM RATINGS
Pin Name
Symbol
VMAX
VMIN
Main Supply Voltage Input
VCC
15 V
−0.3 V
Bootstrap Supply Voltage Input
BST
35 V wrt/PGND
40 V < 50 ns wrt/PGND
15 V wrt/SW
−0.3 V
−0.3 V
−0.3 V
PHASE
35 V
40 V < 50 ns
−5.0 V
−10 V for < 200 ns
High−Side Driver Output (Top Gate)
TG
30 V wrt/GND
15 V wrt/PHASE
−0.3 V wrt/PHASE
−2 V < 200 ns wrt/PHASE
Low−Side Driver Output (Bottom Gate)
BG
15 V
−0.3 V
−5.0 V for < 200 ns
Feedback
FB
5.5 V
−0.3 V
COMP/DIS
5.5 V
−0.3 V
Switching Node (Bootstrap Supply Return)
COMP/DISABLE
MAXIMUM RATINGS
Symbol
Value
Unit
Thermal Resistance, Junction−to−Ambient
Rating
RqJA
165
°C/W
Thermal Resistance, Junction−to−Case
RqJC
45
°C/W
Operating Junction Temperature Range
TJ
0 to 125
°C
Operating Ambient Temperature Range
TA
0 to 70
°C
Storage Temperature Range
Tstg
−55 to +150
°C
260
°C
Lead Temperature Soldering (10 sec): Reflow (SMD styles only) Pb−Free
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
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NCP81044
ELECTRICAL CHARACTERISTICS (0_C < TA < 70_C; 4.5 V < VCC < 13.2 V, 4.5 V < [BST−PHASE] < 13.2 V, 4.5 V < BST < 30 V,
0 V < PHASE < 21 V, CTG = CBG = 1.0 nF, for min/max values unless otherwise noted.)
Characteristic
Conditions
Min
Typ
Max
Unit
Input Voltage Range
−
4.5
−
13.2
V
Boost Voltage Range
−
4.5
−
26.5
V
Quiescent Supply Current
VFB = 1.0 V, No Switching, VCC = 13.2 V
1.0
−
8.0
mA
Boost Quiescent Current
VFB = 1.0 V, No Switching, VCC = 13.2 V
0.1
−
1.0
mA
UVLO Threshold
VCC Rising Edge
3.8
−
4.2
V
UVLO Hysteresis
−
−
350
−
mV
VFB Feedback Voltage,
Control Loop in Regulation
TA = 0 to 70°C
792
800
808
mV
Oscillator Frequency
TA = 0 to 70°C
250
275
300
kHz
Supply Current
Under Voltage Lockout
Switching Regulator
Ramp−Amplitude Voltage
0.8
1.1
1.4
V
Minimum Duty Cycle
0
−
−
%
Maximum Duty Cycle
80
88
93
%
3.0
−
4.4
mmho
55
70
−
DB
80
80
120
120
−
−
mA
−
0.1
1.0
mA
Error Amplifier (GM)
Transconductance
Open Loop DC Gain
Output Source Current
Output Sink Current
VFB < 0.8 V
VFB > 0.8 V
Input Bias Current
Soft−Start
SS Source Current
VFB < 0.8 V
8.49
11
13.3
mA
Switch Over Threshold
VFB = 0.8 V
−
100
−
% of Vref
−
1.0
−
A
−
1.0
−
A
−
1.0
−
A
−
2.0
−
A
Gate Drivers
Upper Gate Source
Upper Gate Sink
Lower Gate Source
VCC = 12 V, VTG = VBG = 2.0 V
Lower Gate Sink
TG Falling to BG Rising Delay
VCC = 12 V, TG < 2.0 V, BG > 2.0 V
−
40
90
ns
BG Falling to TG Rising Delay
VCC = 12 V, BG < 2.0 V, TG > 2.0 V
−
35
90
ns
0.3
0.4
0.5
V
Enable Threshold
Over−Current Protection
8.9
10
11.1
mA
OC Switch−Over Threshold
−
700
−
mV
Fixed OC Threshold
−
−375
−
mV
OCSET Current Source
Sourced from BG pin, before SS
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4
NCP81044
5.0
203
4.7
202
FSW, FREQUENCY (Khz)
ICC (mA)
TYPICAL CHARACTERISTICS (TA = 25°C unless otherwise noted)
4.4
4.1
3.8
0
10
20
30
40
50
60
VCC = 12 V
199
VCC = 5 V
198
70
0
10
20
30
40
50
60
TJ, JUNCTION TEMPERATURE (°C)
Figure 3. ICC vs. Temperature
Figure 4. Oscillator Frequency (FSW) vs.
Temperature
70
375
14
SCP THRESHOLD (mV)
13
12
11
10
9
8
200
TJ, JUNCTION TEMPERATURE (°C)
0
10
20
30
40
50
60
365
355
345
335
325
70
0
10
20
30
40
50
60
TJ, JUNCTION TEMPERATURE (°C)
TJ, JUNCTION TEMPERATURE (°C)
Figure 5. Soft Start Sourcing Current vs.
Temperature
Figure 6. SCP Threshold vs. Temperature
808
806
Vref, REFERENCE (mV)
SOFT START SOURCING CURRENT (mA)
3.5
201
804
802
800
798
796
794
792
0
10
20
30
40
50
60
TJ, JUNCTION TEMPERATURE (°C)
Figure 7. Reference Voltage (Vref) vs.
Temperature
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5
70
70
NCP81044
DETAILED OPERATING DESCRIPTION
General
External Enable/Disable
The NCP81044 is a PWM controller intended for DC−DC
conversion from 5.0 V & 12 V buses. The devices have a 1
A internal gate driver circuit designed to drive N−channel
MOSFETs in a synchronous−rectifier buck topology. The
output voltage of the converter can be precisely regulated
down to 800 mV ±1.0% when the VFB pin is tied to VOUT.
The switching frequency, is internally set to 275 kHz. A high
gain operational transconductance error amplifier (OTA) is
used.
When the Comp pin voltage falls or is pulled externally
below the 400 mV threshold, it disables the PWM Logic and
the gate drive outputs. In this disabled mode, the operational
transconductance amplifier (EOTA) output source current is
reduced and limited to the Soft−Start mode of 10 mA.
Normal Shutdown Behavior
Normal shutdown occurs when the IC stops switching
because the input supply reaches UVLO threshold. In this
case, switching stops, the internal SS is discharged, and all
GATE pins go low. The switch node enters a high impedance
state and the output capacitors discharge through the load
with no ringing on the output voltage.
Duty Cycle and Maximum Pulse Width Limits
In steady state DC operation, the duty cycle will stabilize
at an operating point defined by the ratio of the input to the
output voltage. The devices can achieve an 80% duty cycle.
There is a built in off−time which ensures that the bootstrap
supply is charged every cycle. This part can allow a 12 V to
0.8 V conversion at 275 kHz.
External Soft−Start
The NCP81044 features an external soft−start function,
which reduces inrush current and overshoot of the output
voltage. Soft−start is achieved by using the internal current
source of 10 mA (typ), which charges the external integrator
capacitor of the transconductance amplifier. Figure 8 is a
typical soft−start sequence. This sequence begins once VCC
surpasses its UVLO threshold and OCP programming is
complete. During soft−start, as the Comp Pin rises through
400 mV, the PWM Logic and gate drives are enabled. When
the feedback voltage crosses 800 mV, the EOTA will be
given control to switch to its higher regulation mode output
current of 120 mA.
Input Voltage Range (VCC and BST)
The input voltage range for both VCC and BST is 4.5 V to
13.2 V with respect to GND and PHASE, respectively.
Although BST is rated at 13.2 V with respect to PHASE, it
can also tolerate 26.4 V with respect to GND.
4.2 V
VCC
0.9 V
Comp
0.8 V
Vfb
550 mV
BG
50 mV
OCP
Program
ming
TG
Vout
POR
UVLO
SS
NORMAL
Figure 8. Soft−Start Implementation
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NCP81044
UVLO
go through a Power On Reset (POR) cycle to reset the OCP
fault.
Undervoltage Lockout (UVLO) is provided to ensure that
unexpected behavior does not occur when VCC is too low to
support the internal rails and power the converter. For the
NCP81044, the UVLO is set to permit operation when
converting from a 5.0 input voltage.
Drivers
The NCP81044 includes gate drivers to switch external
N−channel MOSFETs. This allows the devices to address
high−power as well as low−power conversion requirements.
The gate drivers also include adaptive non−overlap
circuitry. The non−overlap circuitry increase efficiency,
which minimizes power dissipation, by minimizing the
body diode conduction time.
A detailed block diagram of the non−overlap and gate
drive circuitry used in the chip is shown in Figure 9.
Overcurrent Threshold Setting
NCP81044 can easily program an Overcurrent Threshold
ranging from 50 mV to 550 mV, simply by adding a resistor
(RSET) between BG and GND. During a short period of
time following VCC rising over UVLO threshold, an internal
10 mA current (IOCSET) is sourced from BG pin,
determining a voltage drop across ROCSET. This voltage
drop will be sampled and internally held by the device as
Overcurrent Threshold. The OC setting procedure overall
time length is about 6 ms. Connecting a ROCSET resistor
between BG and GND, the programmed threshold will be:
I
@ ROCSET
IOCth + OCSET
RDS(on)
FAULT
1
BST
2
TG
8
PHASE
(eq. 1)
RSET values range from 5 kW to 55 kW. In case ROCSET
is not connected, the device switches the OCP threshold to
a fixed 375 mV value: an internal safety clamp on BG is
triggered as soon as BG voltage reaches 700 mV, enabling
the 375 mV fixed threshold and ending OC setting phase.
The current trip threshold tolerance is ±25 mV. The accuracy
of the set point is best at the highest set point (550 mV). The
accuracy will decrease as the set point decreases.
+
2V
+
-
VCC
4
BG
Rset
Current Limit Protection
In case of a short circuit or overload, the low−side (LS)
FET will conduct large currents. The controller will shut
down the regulator in this situation for protection against
overcurrent. The low−side RDS(on) sense is implemented at
the end of each of the LS−FET turn−on duration to sense the
over current trip point. While the LS driver is on, the Phase
voltage is compared to the internally generated OCP trip
voltage. If the phase voltage is lower than OCP trip voltage,
an overcurrent condition occurs and a counter is initiated.
When the counter completes, the PWM logic and both
HS−FET and LS−FET are turned off. The controller has to
3
FAULT
GND
Figure 9. Block Diagram
Careful selection and layout of external components is
required, to realize the full benefit of the onboard drivers.
The capacitors between VCC and GND and between BST
and SWN must be placed as close as possible to the IC. The
current paths for the TG and BG connections must be
optimized. A ground plane should be placed on the closest
layer for return currents to GND in order to reduce loop area
and inductance in the gate drive circuit.
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NCP81044
APPLICATION SECTION
Input Capacitor Selection
ESR. ((neglecting the effect of the effective series
inductance (ESL)):
The input capacitor has to sustain the ripple current
produced during the on time of the upper MOSFET, so it
must have a low ESR to minimize the losses. The RMS value
of this ripple is:
Iin RMS + I OUT ǸD
DV OUT−ESR + DI OUT
where VOUT-ESR is the voltage deviation of VOUT due to the
effects of ESR and the ESRCOUT is the total effective series
resistance of the output capacitors.
A minimum capacitor value is required to sustain the
current during the load transient without discharging it. The
voltage drop due to output capacitor discharge is given by
the following equation:
(1 * D) ,
where D is the duty cycle, IinRMS is the input RMS current,
& IOUT is the load current. The equation reaches its
maximum value with D = 0.5. Loss in the input capacitors
can be calculated with the following equation:
P CIN + ESR CIN
Iin RMS 2 ,
DV OUT−DISCHARGE +
where PCIN is the power loss in the input capacitors &
ESRCIN is the effective series resistance of the input
capacitance. Due to large dI/dt through the input capacitors,
electrolytic or ceramics should be used. If a tantalum must
be used, it must by surge protected. Otherwise, capacitor
failure could occur.
To calculate the input start up current, the following
equation can be used.
C OUT
t SS
V OUT
2
DI OUT 2 L OUT
,
C OUT (V IN D * V OUT)
where VOUT-DISCHARGE is the voltage deviation of VOUT
due to the effects of discharge, LOUT is the output inductor
value & VIN is the input voltage.
It should be noted that ΔVOUT-DISCHARGE and
ΔVOUT-ESR are out of phase with each other, and the larger
of these two voltages will determine the maximum deviation
of the output voltage (neglecting the effect of the ESL).
Calculating Input Start-up Current
I inrush +
ESR COUT
Inductor Selection
Both mechanical and electrical considerations influence
the selection of an output inductor. From a mechanical
perspective, smaller inductor values generally correspond to
smaller physical size. Since the inductor is often one of the
largest components in the regulation system, a minimum
inductor value is particularly important in space-constrained
applications. From an electrical perspective, the maximum
current slew rate through the output inductor for a buck
regulator is given by:
,
where Iinrush is the input current during start-up, COUT is the
total output capacitance, VOUT is the desired output voltage,
and tSS is the soft start interval.
If the inrush current is higher than the steady state input
current during max load, then the input fuse should be rated
accordingly, if one is used.
Calculating Soft Start Time
SlewRate LOUT +
To calculate the soft start time, the following equation can
be used.
V IN * V OUT
L OUT
This equation implies that larger inductor values limit the
regulator’s ability to slew current through the output
inductor in response to output load transients. Consequently,
output capacitors must supply the load current until the
inductor current reaches the output load current level. This
results in larger values of output capacitance to maintain
tight output voltage regulation. In contrast, smaller values of
inductance increase the regulator’s maximum achievable
slew rate and decrease the necessary capacitance, at the
expense of higher ripple current. The peak-to-peak ripple
current for NCP81044 is given by the following equation:
(C p ) C c) * DV
t ss +
I ss
Where Cc is the compensation as well as the soft start
capacitor,
Cp is the additional capacitor that forms the second pole.
Iss is the soft start current
DV is the comp voltage from 0.9 V to until it reaches
regulation: ((d * ramp) + 0.9)
Output Capacitor Selection
The output capacitor is a basic component for the fast
response of the power supply. In fact, during load transient,
for the first few microseconds it supplies the current to the
load. The controller immediately recognizes the load
transient and sets the duty cycle to maximum, but the current
slope is limited by the inductor value.
During a load step transient the output voltage initial
drops due to the current variation inside the capacitor and the
Ipk * pk LOUT +
V OUT(1 * D)
,
L OUT 275 kHz
where Ipk-pkLOUT is the peak to peak current of the output.
From this equation it is clear that the ripple current increases
as LOUT decreases, emphasizing the trade-off between
dynamic response and ripple current.
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NCP81044
Feedback and Compensation
network around the EOTA, the output capacitor, output
inductor and the output divider. Figure 11 shows the open
loop and closed loop gain plots.
The NCP81044 allows the output of the DC-DC converter
to be adjusted from 0.8 V to 5.0 V via an external resistor
divider network. The controller will try to maintain 0.8 V at
the feedback pin. Thus, if a resistor divider circuit was
placed across the feedback pin to VOUT, the controller will
regulate the output voltage proportional to the resistor
divider network in order to maintain 0.8 V at the FB pin.
Compensation Network Frequency:
The inductor and capacitor form a double pole at the
frequency
F LC +
VOUT
1
2p
Co
The ESR of the output capacitor creates a “zero” at the
frequency,
R1
FB
F ESR +
2p
1
ESR
Co
The zero of the compensation network is formed as,
R2
FZ +
ǒ
2p
1
R cC c
The pole of the compensation network is calculated as,
The relationship between the resistor divider network above
and the output voltage is shown in the following equation:
R2 + R1
ǸLo
Fp +
Ǔ
V REF
V OUT * V REF
2p
1
Rc
Cp
Resistor R1 is selected based on a design tradeoff between
efficiency and output voltage accuracy. For high values of
R1 there is less current consumption in the feedback
network, However the trade off is output voltage accuracy
due to the bias current in the error amplifier. The output
voltage error of this bias current can be estimated using the
following equation (neglecting resistor tolerance):
Error% +
0.1 mA R 1
V REF
100%
Once R1 has been determined, R2 can be calculated.
Figure 11. Gain Plot of the Error Amplifier
R1
EA
Thermal Considerations
Gm
Cc
Rc
Cp
Vref
+
The power dissipation of the NCP81044 varies with the
MOSFETs used, VCC, and the boost voltage (VBST). The
average MOSFET gate current typically dominates the
control IC power dissipation. The IC power dissipation is
determined by the formula:
R2
−
P IC + (I CC
Figure 10. Type II Transconductance Error
Amplifier
V CC) ) P TG ) P BG
Where:
PIC = control IC power dissipation,
ICC = IC measured supply current,
VCC = IC supply voltage,
PTG = top gate driver losses,
PBG = bottom gate driver losses.
The upper (switching) MOSFET gate driver losses are:
Figure 10 shows a typical Type II transconductance error
amplifier (EOTA). The compensation network consists of
the internal error amplifier and the impedance networks ZIN
(R1, R2) and external ZFB (Rc, Cc and Cp). The
compensation network has to provide a closed loop transfer
function with the highest 0 dB crossing frequency to have
fast response (but always lower than FSW/8) and the highest
gain in DC conditions to minimize the load regulation. A
stable control loop has a gain crossing with -20 dB/decade
slope and a phase margin greater than 45°. Include
worst-case component variations when determining phase
margin. Loop stability is defined by the compensation
P TG + Q TG
f SW
V BST
Where:
QTG = total upper MOSFET gate charge at VBST,
fSW = the switching frequency,
VBST = the BST pin voltage.
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NCP81044
DESIGN EXAMPLE I: Type II Compensation
(Electrolytic Cap. with large ESR)
The lower (synchronous) MOSFET gate driver losses are:
P BG + Q BG
f SW
V CC
Switching Frequency
FSW = 275 KHz
Output Capacitance
RESR = 45 mW/Each
Output Capacitance
Cout = 2×1800 mF
Output Inductance
Lout = 1 mH
Input Voltage
Vin = 12 V
Output Voltage
Vout = 1.6 V
Choose the loop gain crossover frequency;
Where:
QBG = total lower MOSFET gate charge at VCC.
The junction temperature of the control IC can then be
calculated as:
T J + T A ) P IC
q JA
Where:
TJ = the junction temperature of the IC,
TA = the ambient temperature,
θJA = the junction−to−ambient thermal resistance of the
IC package.
The package thermal resistance can be obtained from the
specifications section of this data sheet and a calculation can
be made to determine the IC junction temperature. However,
it should be noted that the physical layout of the board, the
proximity of other heat sources such as MOSFETs and
inductors, and the amount of metal connected to the IC,
impact the temperature of the device. Use these calculations
as a guide, but measurements should be taken in the actual
application.
F co + 1
5
F sw + 55 KHz
The corner frequency of the output filter is calculated below;
F LC +
1
2
Ǹ1 mH
p
+ 2.65 KHz
3600 mF
Check that the ESR zero frequency is not too high;
F ESR +
F ESR +
2
p
1
R ESR
2
p
45 mW
2
CO
t
F co
10
1
(1800 mF
2)
+ 2 KHz
If ESR zero is larger than Fco/10, Type III compensation
is necessary.
Choose CC for the crossover frequency and the soft start
Layout Considerations
As in any high frequency switching converter, layout is
very important. Switching current from one power device to
another can generate voltage transients across the
impedances of the interconnecting bond wires and circuit
traces. These interconnecting impedances should be
minimized by using wide, short printed circuit traces. The
critical components should be located as close together as
possible using ground plane construction or single point
grounding. The figure below shows the critical power
components of the converter. To minimize the voltage
overshoot the interconnecting wires indicated by heavy lines
should be part of ground or power plane in a printed circuit
board. The components shown in the figure below should be
located as close together as possible. Please note that the
capacitors CIN and COUT each represent numerous physical
capacitors. It is desirable to locate the NCP81044 within 1
inch of the MOSFETs, Q1 and Q2. The circuit traces for the
MOSFETs’ gate and source connections from the
NCP81044 must be sized to handle up to 2 A peak current.
C C + 100 nF
The compensation capacitor (CC) is related to the loop
gain magnitude, zero position and the soft start. By adjusting
the value of this compensation capacitor, the crossover
frequency and the soft start time can be adjusted.
Zero of the compensation network is calculated as follows;
F Z + F LC + 2.65 KHz
RC +
+
2
p
2
p
1
Fz
CC
1
2.65 kHz
100 nF
+ 600.6 W
Pole of the compensation network is calculated as follows;
F p + F sw + 275 KHz
1
Cp +
2 p Fp RC
1
+
2 p 275 kHz
600.6
+ 963.6 pF
The recommended compensation values are; RC = 604,
CC = 100 nF, CP = 1000 pF
Figure 12. Components to be Considered for
Layout Specifications
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10
NCP81044
Blue curve: Gain-Frequency
Red curve: Gain-Frequency
(Phase margin = 61.417 degree, Gain margin = 9.347 dB)
Figure 13. Closed-loop Voltage Loop-gain of the NCP81044
DESIGN EXAMPLE II: Type III Compensation
(Oscon Cap. with small ESR; Do not place RC, CC, CP)
Switching Frequency
Fsw = 275 KHz
Output Capacitance
RESR = 7 mW/Each
Output Capacitance
Cout = 2×560 mF
Output Inductance
Lout = 1 mH
Input Voltage
Vin = 12 V
Output Voltage
Vout = 1.6 V
Choose the loop gain crossover frequency;
F co + 1
5
R C1 +
+
F sw + 55 KHz
2
1
Ǹ1 mH
p
1120 mF
F ESR +
F ESR +
2
F z2 + F LC + 4.7 KHz
2
C20 +
p
CO
p
1
7 mW
560 mF
2
R3 + 10 kW
+ 4.7 KHz
Check the ESR zero frequency;
1
R ESR
2
RC1 should be much larger than 2/gm in order to get the
stable system with transconductance amplifier. Ù choose
RC1 = 12.1 kW
2nd zero;
Choose R3 for the crossover frequency. R3 should be
much larger than 2/gm for the stable system.
The corner frequency of the output filter is calculated below;
F LC +
F LC
+ 470 Hz
10
1
p F z1 C C1
1
+ 11.3 kW
p 470 Hz 30 nF
F Z1 +
2
p
2
p
+
+ 40.6 KHz
1
F z2
R3
1
+ 3.4 nF
4.7 KHz 10 kW
Choose C20 = 3.3 nF
Poles of the compensation network are calculated as follows;
1st pole;
Choose R4 to cancel the output capacitor ESR zero.
Choose CC1 for the soft start
C C1 + 33 nF
The compensation capacitor (CC1) is related to the loop
gain magnitude, one zero position and the soft start. By
adjusting the value of this compensation capacitor, the
crossover frequency and the soft start time can be adjusted.
Zeros of the compensation network are calculated as follows;
1st zero;
F P1 + F ESR + 40.6 KHz
R4 +
+
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11
2
2
p
1
F P1
p
1
40.6 kHz
C20
3.3 n
+ 1.2 kW
NCP81044
After choose R4 value, adjust R4 to get enough phase
margin Ù R4 = 665 W
2nd pole;
Choose CP1 to eliminate the noise;
Choose CP1 = 47 pF
The recommended compensation values are;
R2 = 10 kW, R3 = 10 kW, R4 = 665 W,
RC1 = 12.1 kW, CC1 = 33 nF, CP1= 47 pF,
C20 = 3.3 nF
F P2 + F sw + 275 KHz
C P1 +
+
2
p
2
p
1
F P2
R C1
1
+ 48.23 pF
275 kHz 12 kW
Blue curve: Gain-Frequency
Red curve: Gain-Frequency
(Phase margin = 80.285 degree, Gain margin = 19.362 dB)
Figure 14. Closed-loop Voltage Loop-gain of the NCP81044
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12
TP96
TP95
MH1
GND
VCC
GND
TP2
MH2
MH3
VCC
Vbst
Figure 15. Demo Board PCB Layout
http://onsemi.com
MH4
Rc
604
TP106
R3
1.02K
FB
Cp
100pF
TP105
R2
1.02K
CC
0.1uF
TP98
TP97
COMP
1
TP94
R4
DNP
C20
DNP
FB
Cp1
DNP
COMP
TP93
7
TG
BST
U1
10
R1
2 TG
Note : gating
length s
1 BST
R5
5.11
R639
0
TP108
TP110
TP112
TP111
BG
R6
0
R7
0
TP107
TP109
length sho
DNP
+ C5
Vin
DPAK
Q6
NTD4806
DPAK
Q2
NTD4815
C19
10uF
DUAL PLACEMENT SITE
DPAK
Q5
NTD4806
DPAK
Q1
NTD4815
C22
10uF
IPAK
Q8
NTD4806
IPAK
Q4
NTD4815
DUAL PLACEMENT SITE
IPAK
Q7
NTD4806
IPAK
Q3
NTD4815
G
C23
10uF
DUAL PLACEMENT SITE
1500uF
+ C4
DUAL PLACEMENT SITE
TP113
TP20
C18
10uF
GND
Vin
C11
0.1uF
TP101
TG
TP99
CR1
BAS116LT1
8 SWITCH_NODE
PHASE Note
: gating
NCP81044
COMP
C8
1uF
C9
1uF
6 FBBG 4 BG
Cc1
DNP
Rc1
DNP
R90
1
3
TP1
5
VCC
GND
3
G
SO8−FL SO8−FL
Q12
DNP
G
Q10
DNP
C21
DNP
R8
DNP
L11uH
L2DNP
1800uF
+C12
SWITCH_NODE
SO8−FL
DUAL PLACEMENT SITE
Q11
DNP
G
SO8−FL
Q9
DNP
DUAL PLACEMENT SITE
C25
10uF
D
S
D
D
S
D
S
13
S
Vbst
1800uF
+C13
TP104
TP100
DNP
+ C15
DNP
+ C24
C16
10uF
C17
10uF
VOUT
GND
GND
TP9
OUTPU
TP7
TP103
TP102
VOUT
NCP81044
NCP81044
Bill of Materials
Item Number
Part Reference
Value
Quantity
MFG
1
C4
1500 mF
1
PANASONIC
2
C5
DNP
1
-
3
C8,C9
1 mF
2
TAIYO YUDEN
4
C11
0.1 mF
1
AVX
5
C12,C13
1800 mF
2
PANASONIC
6
C15,C24
DNP
2
-
7
C16,C17,C18,C19,C22,C23,C25
10 mF
7
PANASONIC
8
C20,CC1,CP1
DNP
3
-
9
C21
DNP
1
-
10
CC
0.1 mF
1
TDK
11
CR1
BAS116LT1
1
ON SEMICONDUCTOR
12
CP
100 pF
1
PANASONIC
13
J9
20PIN 2ROW
1
MOLEX
14
J23
5PIN
1
PASTERNACK
ENTERPRISES
15
L1
1 mH
1
PANASONIC
16
L2
DNP
1
-
17
Q1,Q2
NTD4815
2
ON SEMICONDUCTOR
18
Q3,Q4
NTD4815
2
ON SEMICONDUCTOR
19
Q5,Q6
NTD4806
2
ON SEMICONDUCTOR
20
Q7,Q8
NTD4806
2
ON SEMICONDUCTOR
21
Q9,Q10,Q11,Q12
DNP
4
-
22
Q17,Q18,Q19,Q20,Q21,Q22,Q23,Q24,Q25,Q26,
Q27,Q28,Q29,Q30,Q31,Q32,Q33,Q34,Q35,Q36,
Q37,Q38,Q39,Q40
NTHS5404T1
24
ON SEMICONDUCTOR
23
R1
10
1
PANASONIC
24
R2,R3
1.02 K
2
DALE
25
R4,RC1
DNP
2
-
26
R5
5.11
1
DALE
27
R6,R7,R639
0
3
PANASONIC
28
R8
DNP
1
-
29
R9
0
1
DALE
30
R551,R552,R553,R569,R570,R571,R584,R585,
R586,R599,R600,R601,R608,R609,R610,R617,
R618,R619,R626,R627,R628,R635,R636,R637
100 K
24
DALE
31
R602,R603,R604,R605,R606,R607,R611,R612,
R613,R614,R615,R616,R620,R621,R622,R623,
R624,R625,R629,R630,R631,R632,R633,R634
0.56
24
PANASONIC
32
R638
49.9
1
DALE
33
RC
604
1
DALE
34
TP97,TP98,TP99,TP100,TP101,TP102,TP103,
TP104,TP105,TP106,TP107,TP108,TP109,
TP110,TP111,TP112
TP
16
KEYSTONE
35
U1
NCP81044
1
ON SEMICONDUCTOR
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14
NCP81044
Figure 16. Gate Waveforms 20 A Load Sustaining
Figure 17. Over Current Protection (12.4 A DC Trip)
Figure 18. Start-up Sequence
Figure 19. Transient Response 0-10 A Load Step
Efficiency (%)
Efficiency
90
88
86
84
82
80
78
76
74
72
70
68
66
64
62
60
0
2
4
6
8
10
12
Load Current (A)
Figure 20. Efficiency vs. Load Current
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15
14
16
NCP81044
PACKAGE DIMENSIONS
SOIC−8
D SUFFIX
CASE 751−07
ISSUE AJ
−X−
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEW
STANDARD IS 751−07.
A
8
5
S
B
0.25 (0.010)
M
Y
M
1
4
−Y−
K
G
C
N
DIM
A
B
C
D
G
H
J
K
M
N
S
X 45 _
SEATING
PLANE
−Z−
0.10 (0.004)
H
D
0.25 (0.010)
M
Z Y
S
X
M
J
S
MILLIMETERS
MIN
MAX
4.80
5.00
3.80
4.00
1.35
1.75
0.33
0.51
1.27 BSC
0.10
0.25
0.19
0.25
0.40
1.27
0_
8_
0.25
0.50
5.80
6.20
INCHES
MIN
MAX
0.189
0.197
0.150
0.157
0.053
0.069
0.013
0.020
0.050 BSC
0.004
0.010
0.007
0.010
0.016
0.050
0 _
8 _
0.010
0.020
0.228
0.244
SOLDERING FOOTPRINT*
1.52
0.060
7.0
0.275
4.0
0.155
0.6
0.024
1.270
0.050
SCALE 6:1
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC owns the rights to a number of patents, trademarks,
copyrights, trade secrets, and other intellectual property. A listing of SCILLC’s product/patent coverage may be accessed at www.onsemi.com/site/pdf/Patent−Marking.pdf. SCILLC
reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any
particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without
limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications
and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC
does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for
surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where
personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and
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any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture
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NCP81044/D