DATASHEET ISL6267 FN7801 Rev 1.00 Jan 8, 2013 Multiphase PWM Regulator for AMD Fusion™ Mobile CPUs The ISL6267 is designed to be completely compliant with AMD Fusion™ specifications. The ISL6267 controls two Voltage Regulators (VRs), with three integrated gate drivers. The first VR can be configured as 3-, 2-, or 1-phase VR, while the second output can be configured as 2- or 1-phase VR, providing maximum flexibility. The two VRs share the serial control bus to communicate with the CPU and achieve lower cost and smaller board area compared with two-chip solutions. The PWM modulator of the ISL6267 is based on Intersil’s R3 (Robust Ripple Regulator) Technology™. Compared with the traditional multi-phase buck regulator, the R3 modulator commands variable switching frequency during load transients, achieving faster transient response. With the same modulator, it naturally goes into pulse frequency modulation in light load conditions, which achieves higher light load efficiency and extends battery life. The ISL6267 has several other key features. Both outputs support DCR current sensing with a single NTC thermostat for DCR temperature compensation or accurate resistor current sensing. Both of the outputs utilize remote voltage sense, adjustable switching frequency, current monitor, OC protection, independent power-good indicators, temperature monitors, and a common thermal alert. Applications Features • Supports AMD SVI 1.0 serial data bus interface • Dual output controller with integrated drivers - Core VR configurable 3-, 2-, 1-phase with two integrated drivers - Northbridge VR configurable 2- or 1-phase with one integrated driver • Precision voltage regulation - 0.5% system accuracy over-temperature - 0V to 1.55V in 12.5mV steps - Enhanced load line accuracy • Supports multiple current sensing methods - Lossless inductor DCR current sensing - Precision resistor current sensing • Programmable 1-, 2- or 3-phase for the core output and 1- or 2-phase for the northbridge output • Adaptive body diode conduction time reduction • Superior noise immunity and transient response • Output current monitor and thermal monitor • Differential remote voltage sensing • High efficiency across entire load range • AMD fusion CPU/GPU core power • Programmable +VID offset for both core and NB • Notebook computer • Programmable switching frequency for both outputs • Excellent dynamic current balance between phases • OCP/WOC, OVP, PGOOD, and thermal monitor • Small footprint 48 Ld 6x6 QFN package • Pb-free (RoHS compliant) Core Performance on ISL6267EVAL1Z 100 1.12 90 1.10 VIN = 8V 70 1.08 VIN = 12V 60 VOUT (A) EFFICIENCY (%) 80 VIN = 19V 50 40 30 VIN = 8V 1.04 VIN = 12V 1.02 1.00 20 10 0 0 1.06 5 10 15 20 25 30 35 IOUT (A) 40 45 FIGURE 1. EFFICIENCY vs LOAD FN7801 Rev 1.00 Jan 8, 2013 50 VIN = 19V 0.98 VOUT CORE = 1.1V 55 0.96 VOUT CORE = 1.1V 0 5 10 15 20 25 30 35 40 45 50 55 IOUT (A) FIGURE 2. VOUT vs LOAD Page 1 of 33 ISL6267 VCCP ENABLE VDD NB_PH1 NB_PH2 Simplified Application Circuit For High Power CPU Core BOOT_NB UG1_NB VNB ISEN1_NB PH1_NB ISEN2_NB VNB1 VIN LG1_NB ISUMN_NB NB_PH1 VNB1 VNB2 VW_NB PWM2_NB ISL6208 NB_PH2 NB_PH1 ISUMP_NB NB_PH2 COMP_NB VNB_SENSE FB_NB NTC_NB FB2_NB PROG1 VSEN_NB PROG2 VR_HOT RTN_NB ISL6267 THERMAL INDICATOR NTC VIN PWROK µP VNB2 SVD PWM3 VW ISL6208 SVC PH3 VO3 COMP UG2 VSEN RTN PH1 PH2 VCORE PH2 VCORE_SENSE PH3 VIN BOOT2 FB LG2 ISEN3 ISEN2 UG1 PGOOD_NB PGOOD PH1 LG1 PH1 VO1 PH3 PH2 PH1 ISUMP VIN VO3 GND PAD ISUMN VO2 VO2 VIN BOOT1 ISEN1 VO1 PH2 FIGURE 3. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING FN7801 Rev 1.00 Jan 8, 2013 Page 2 of 33 ISL6267 VCCP ENABLE VDD NB_PH2 NB_PH1 Simplified Application Circuit For AMD Torpedo Platform NTC_NB ISEN1_NB UG1_NB ISUMN_NB VNB1 VIN BOOT_NB ISEN2_NB VNB2 VNB PH1_NB LG1_NB NB_PH1 VIN VW_NB COMP_NB PWM2_NB FB_NB NB_PH2 FB2_NB VSEN_NB VNB_SENSE PWROK VNB2 NTC RTN_NB µP VNB1 ISL6208 NB_PH2 NB_PH1 ISUMP_NB VR_HOT ISL6267 THERMAL INDICATOR PROG1 PROG2 SVD SVC PWM3 VW +5V VIN BOOT2 COMP UG2 FB PH2 VCORE ISEN3/FB2 LG2 PH2 VO2 VSEN RTN VIN BOOT1 PH1 PH2 VCORE_SENSE UG1 PH1 ISEN2 LG1 VO1 PGOOD VO1 PH2 PH1 ISUMP VIN VO2 GND PAD ISUMN PH1 PGOOD_NB ISEN1 FIGURE 4. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING FN7801 Rev 1.00 Jan 8, 2013 Page 3 of 33 ISL6267 Simplified Application Circuit For Low Power CPU Core And NB ISEN1_NB VCCP OPEN VDD ISEN2_NB ENABLE +5V UG1_NB ISUMN_NB VNB1 VIN BOOT_NB VNB PH1_NB NB_PH1 ISUMP_NB LG1_NB NB_PH1 VNB1 NTC_NB VW_NB optional PWM2_NB OPEN COMP_NB FB_NB PROG1 VSEN_NB VNB_SENSE PROG2 RTN_NB NTC VR_HOT THERMAL INDICATOR PWROK µP SVD ISL6267 SVC OPEN +5V PWM3 ISEN1 +5V BOOT2 OPEN UG2 OPEN PH2 OPEN LG2 OPEN PGND2 OPEN ISEN2 OPEN ISEN3 VW optional COMP VIN BOOT1 FB UG1 VCORE VSEN VCORE_SENSE PH1 RTN VO1 LG1 PH1 VO1 PGOOD_NB PGOOD VIN PH1 ISUMP GND PAD ISUMN FIGURE 5. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING FN7801 Rev 1.00 Jan 8, 2013 Page 4 of 33 ISL6267 OPEN VCCP ISEN2_NB VDD +5V ENABLE Simplified Application Circuit Showing Resistor Sensing ISEN1_NB VIN BOOT_NB UG1_NB ISUMN_NB NB_N VNB PH1_NB NB_P ISUMP_NB LG1_NB NB_P NB_N NTC_NB VW_NB optional PWM2_NB OPEN COMP_NB FB_NB PROG1 VSEN_NB VNB_SENSE PROG2 RTN_NB NTC VR_HOT THERMAL INDICATOR PWROK µP SVD ISL6267 SVC OPEN +5V PWM3 ISEN1 ISEN2 OPEN ISEN3 VW optional COMP BOOT2 OPEN UG2 OPEN PH2 OPEN LG2 OPEN PGND2 OPEN BOOT1 FB PGOOD VIN VON VOP GND PAD VOP PGOOD_NB LG1 ISUMN ISUMP VCORE PH1 RTN VON VIN UG1 VSEN VCORE_SENSE +5V FIGURE 6. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING FN7801 Rev 1.00 Jan 8, 2013 Page 5 of 33 ISL6267 PGOOD_NB ISEN1_NB ISEN2_NB ISUMN_NB ISUMP_NB COMP_NB FB_NB FB2_NB VW_NB VSEN_NB RTN_NB VIN ENABLE Block Diagram SVD SERIAL VID INTERFACE SVC PWROK PROG1 PROG2 VCORE VNB A/D BOOT_NB DAC1 NORTHBRIDGE CONTROLLER AND DRIVER UG_NB (SIMILAR ARCHITECTURE TO PH_NB CORE SECTION) SLEEP LG_NB D/A MODE DAC2 (PSI_L) PWM2_NB VDD PROG1 IBAL OFFSET VOLTAGE PROG2 PHASE CURRENT BALANCE ISEN1 ISEN2 ISEN3/FB2 PWM3 NTC_NB T_MONITOR TEMP NTC BOOT2 MONITOR DRIVER UG2 VR_HOT PH2 VW DAC1 + RTN + DRIVER + MODULATOR BOOT1 E/A FB DRIVER - LG2 UG1 PH1 COMP VCCP ISUMP + ISUMN - DRIVER CURRENT OC AND WOC SENSE PROTECTION LG1 PGOOD OV VSEN FN7801 Rev 1.00 Jan 8, 2013 PROTECTION Page 6 of 33 ISL6267 Pin Configuration LG1_NB PH1_NB BOOT1_NB UG1_NB PROG2 ISUMN_NB NTC_NB ISUMP_NB RTN_NB ISEN2_NB VSEN_NB ISEN1_NB ISL6267 (48 LD QFN) TOP VIEW 48 47 46 45 44 43 42 41 40 39 38 37 36 PWM2_NB FB2_NB 1 35 BOOT2 FB_NB 2 34 UG2 COMP_NB 3 VW_NB 4 33 PH2 PGOOD_NB 5 32 LG2 SVD 6 GND PAD 31 VCCP PWROK 7 (BOTTOM) 30 PWM3 SVC 8 29 LG1 ENABLE 9 28 PH1 VDD 23 VIN 24 ISUMN 21 ISUMP 22 VSEN 19 RTN 20 37 ISEN2 17 ISEN1 18 25 PROG1 FB 15 ISEN3/FB2 16 NTC 12 VW 13 27 UG1 26 BOOT1 COMP 14 PGOOD 10 VR_HOT 11 Pin Descriptions PIN NUMBER SYMBOL DESCRIPTION 1 FB2_NB The components connecting to FB2_NB are used to adjust the compensation in 1-phase mode to achieve optimum performance. 2 FB_NB Output voltage feedback to the inverting input of the Northbridge controller error amplifier. 3 COMP_NB 4 VW_NB 5 PGOOD_NB 6 SVD 7 PWROK 8 SVC Serial VID clock input from the CPU processor master device. 9 ENABLE Enable input. A high level logic on this pin enables both VRs. 10 PGOOD Open-drain output to indicate the Core portion of the IC is ready to supply regulated voltage. Pull-up externally to VCCP or 3.3V. 11 VR_HOT Thermal overload open drain output indicator active LOW. 12 NTC Thermistor input to VR_HOT circuit to monitor Core VR temperature. 13 VW Window voltage set pin used to set the switching frequency for the Core controller. A resistor from this pin to COMP programs the switching frequency (8k gives approximately 300kHz). 14 COMP 15 FB FN7801 Rev 1.00 Jan 8, 2013 Northbridge VR error amplifier output. Window voltage set pin used to set the switching frequency for the Northbridge controller. A resistor from this pin to COMP_NB programs the switching frequency (8k gives approximately 300kHz). Open-drain output to indicate the Northbridge portion of the IC is ready to supply regulated voltage. Pull-up externally to VCCP or 3.3V. Serial VID data bi-directional signal from the CPU processor master device to the VR. System power good input. When this pin is high, the SVI interface is active and the I2C protocol is running. While this pin is low, the SVC and SVD input states determine the pre-PWROK metal VID. This pin must be low prior to the ISL6267 PGOOD output going high per the AMD SVI Controller Guidelines. Error amplifier output. Output voltage feedback to the inverting input of the Core controller error amplifier. Page 7 of 33 ISL6267 Pin Descriptions (Continued) PIN NUMBER SYMBOL DESCRIPTION 16 ISEN3/FB2 When the Core VR of ISL6267 is configured in 3-phase mode, this pin is ISEN3. ISEN3 is the individual current sensing for Channel 3. When the Core VR of ISL6267 is configured in 2-phase mode, this pin is FB2. There is a switch between the FB2 pin and the FB pin. The switch is on in 2-phase mode and is off in 1-phase mode. The components connecting to FB2 are used to adjust the compensation in 1-phase mode to achieve optimum performance. 17 ISEN2 Individual current sensing for Channel 2 of the Core VR. When ISEN2 is pulled to 5V VDD, the controller disables Channel 2, and the Core VR runs in single-phase mode. 18 ISEN1 Individual current sensing for Channel 1 of the Core output. 19 VSEN Output voltage sense pin for the Core controller. Connect to the +sense pin of the microprocessor die. 20 RTN Output voltage sense return pin for the Core controller. Connect to the -sense pin of the microprocessor die. 21 ISUMN Inverting input of the transconductance amplifier for current monitor and load line of Core output. 22 ISUMP Non-inverting input of the transconductance amplifier for current monitor and load line of Core output. 23 VDD 5V bias power. 24 VIN Battery supply voltage, used for feed-forward. 25 PROG1 Program pin for setting output voltage offset for Core VR. 26 BOOT1 Connect an MLCC capacitor across the BOOT1 and the phase (PH1) pin. The boot capacitor is charged through an internal boot diode connected from the VCCP pin to the BOOT1 pin, each time the PH1 pin drops below VCCP minus the voltage dropped across the internal boot diode. 27 UG1 Output of the Phase 1 high-side MOSFET gate driver of the Core VR. Connect the UG1 pin to the gate of the Phase 1 high-side MOSFET. 28 PH1 Current return path for the Phase 1 high-side MOSFET gate driver of VR1. Connect the PH1 pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of Phase 1. 29 LG1 Output of the Phase 1 low-side MOSFET gate driver of the Core VR. Connect the LG1 pin to the gate of the Phase 1 low-side MOSFET. 30 PWM3 PWM output for Channel 3 of the Core VR. When PWM3 is pulled to 5V VDD, the controller disables Phase 3 and runs in 2-phase mode. 31 VCCP Input voltage bias for the internal gate drivers. Connect +5V to the VCCP pin. Decouple with at least 1µF of capacitance to GND. A high quality, X7R dielectric MLCC capacitor is recommended. 32 LG2 Output of the Phase 2 low-side MOSFET gate driver of VR1. Connect the LG2 pin to the gate of the Phase 2 low-side MOSFET. 33 PH2 Current return path for the Phase 2 high-side MOSFET gate driver of the Core VR. Connect the PH2 pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of Phase 2. 34 UG2 Output of the Phase 2 high-side MOSFET gate driver of the Core VR. Connect the UG2 pin to the gate of the Phase 2 high-side MOSFET. 35 BOOT2 Connect an MLCC capacitor across the BOOT2 and PH2 pins. The boot capacitor is charged through an internal boot diode connected from the VCCP pin to the BOOT2 pin, each time the PH2 pin drops below VCCP minus the voltage dropped across the internal boot diode. 36 PWM2_NB 37 LG1_NB Output of the low-side MOSFET gate driver of the Northbridge VR. Connect the LG1_NB pin to the gate of the low-side MOSFET of VR2. 38 PH1_NB Current return path for the high-side MOSFET gate driver of the Northbridge VR. Connect the PH1_NB pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of the Northbridge VR. 39 UG1_NB Output of the high-side MOSFET gate driver of the Northbridge VR. Connect the UG1_NB pin to the gate of the high-side MOSFET. FN7801 Rev 1.00 Jan 8, 2013 PWM output for Channel 2 of the Northbridge VR. Page 8 of 33 ISL6267 Pin Descriptions (Continued) PIN NUMBER SYMBOL DESCRIPTION 40 BOOT1_NB Connect an MLCC capacitor across the BOOT1_NB and the PH1_NB pins. The boot capacitor is charged through an internal boot diode connected from the VCCP pin to the BOOT1_NB pin, each time the PH1_NB pin drops below VCCP minus the voltage dropped across the internal boot diode. 41 PROG2 Program pin for setting output voltage offset for Northbridge VR. 42 NTC_NB Thermistor input to VR_HOT circuit to monitor Northbridge VR temperature. 43 ISUMN_NB Inverting input of the transconductance amplifier for current monitor and load line of the Northbridge VR. 44 ISUMP_NB Non-inverting input of the transconductance amplifier for current monitor and load line of the Northbridge VR. 45 RTN_NB Output voltage sense return pin for the Northbridge controller. Connect to the -sense pin of the microprocessor die. 46 VSEN_NB Output voltage sense pin for the Northbridge controller. Connect to the +sense pin of the microprocessor die. 47 ISEN2_NB Individual current sensing for Channel 2 of the Northbridge VR. When ISEN2 is pulled to 5V VDD, the controller will disable Channel 2 and the Northbridge VR will run single-phase. 48 ISEN1_NB Individual current sensing for Channel 1 of the Northbridge VR. GND (Bottom Pad) Signal common of the IC. Unless otherwise stated, signals are referenced to the GND pin. Ordering Information PART NUMBER (Notes 1, 2, 3) PART MARKING TEMP. RANGE (°C) PACKAGE (Pb-free) PKG. DWG. # ISL6267HRZ ISL6267 HRZ -10 to +100 48 Ld 6x6 QFN L48.6x6B ISL6267IRZ ISL6267 IRZ -40 to +100 48 Ld 6x6 QFN L48.6x6B ISL6267EVAL1Z Evaluation Board NOTES: 1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications. 2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6267. For more information on MSL please see tech brief TB363. FN7801 Rev 1.00 Jan 8, 2013 Page 9 of 33 ISL6267 Table of Contents Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 Thermal Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 Recommended Operating Conditions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 Electrical Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 Gate Driver Timing Diagram. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 Theory of Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 Multiphase R3™ Modulator. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 Diode Emulation and Period Stretching . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 Start-up Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 Power-On Reset . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 Serial VID Interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 Pre-PWROK Metal VID . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16 VFIX Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 SVI Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .17 VID-on-the-Fly Transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 SVI WIRE Protocol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 SVI Bus Protocol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .19 VR Offset Programming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .19 Voltage Regulation and Load Line Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 Differential Sensing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .20 Phase Current Balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .20 CCM Switching Frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Modes of Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Dynamic Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 Protections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 FB2 Function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .23 Adaptive Body Diode Conduction Time Reduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 Key Component Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 Inductor DCR Current-Sensing Network. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 Resistor Current-Sensing Network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 Overcurrent Protection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 Load Line Slope . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27 Compensator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27 Current Balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27 NTC Thermal Monitors and VR_HOT Function. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 Layout Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 Revision History . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32 Products . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32 Package Outline Drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 FN7801 Rev 1.00 Jan 8, 2013 Page 10 of 33 ISL6267 Absolute Maximum Ratings Thermal Information Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V Battery Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +28V Boot Voltage (BOOT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V Boot to Phase Voltage (BOOT-PHASE) . . . . . . . . . . . . . . . . -0.3V to +7V(DC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +9V(<10ns) Phase Voltage (PHASE) . . . . . . . . . . . . . . . . -7V (<20ns Pulse Width, 10µJ) UGATE Voltage (UGATE) . . . . . . . . . .PHASE - 0.3V (DC) to BOOTPHASE - 5V . . . . . . . . . . . . . . . . . (<20ns Pulse Width, 10µJ) to BOOT LGATE Voltage . . . . . . . . . . . . . . . . . . . . . -2.5V (<20ns Pulse Width, 5µJ) to VDD + 0.3V All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (VDD + 0.3V) Open Drain Outputs, PGOOD, VR_HOT. . . . . . . . . . . . . . . . . . . -0.3V to +7V Thermal Resistance (Typical) JA (°C/W) JC (°C/W) 48 Ld QFN Package (Notes 4, 5) . . . . . . . . 28 1 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . .+150°C Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . .+150°C Storage Temperature Range. . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp Recommended Operating Conditions Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5% Battery Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5V to 25V Ambient Temperature HRZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-10°C to +100°C IRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +100°C Junction Temperature HRZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-10°C to +125°C IRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty. NOTES: 4. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379. 5. For JC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications Operating Conditions: VDD = 5V, TA = -10°C to +100°C (HRZ), TA = -40°C to +100°C (IRZ), fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. PARAMETER SYMBOL TEST CONDITIONS MIN (Note 6) TYP MAX (Note 6) UNITS 9 10.5 mA INPUT POWER SUPPLY +5V Supply Current IVDD ENABLE = 1V ENABLE = 0V 1 µA Battery Supply Current IVIN ENABLE = 0V 1 µA VIN Input Resistance RVIN ENABLE = 1V 550 k POWER-ON-RESET THRESHOLDS VDD POR Threshold VIN POR Threshold VDD_PORr VDD rising VDD_PORf VDD falling VIN_PORr VIN rising VIN_PORf 4.35 4.00 4.15 VIN falling 2.8 3.30 No load; closed loop, active mode range VID = 0.75V to 1.55V 4.00 4.5 V V 4.35 V V SYSTEM AND REFERENCES System Accuracy HRZ% Error (VCC_CORE) IRZ% Error (VCC_CORE) -0.5 +0.5 % VID = 0.50V to 0.7375V -8 +8 mV VID = 0.25V to 0.4875V -15 +15 mV No load; closed loop, active mode range VID = 0.75V to 1.55V -0.8 +0.8 % VID = 0.50V to 0.7375V -10 +10 mV VID = 0.25V to 0.4875V -18 +18 mV Maximum Output Voltage VCC_CORE(max) VID = [0000000] 1.55 V Minimum Output Voltage VCC_CORE(min) VID = [1111111] 0.0 V FN7801 Rev 1.00 Jan 8, 2013 Page 11 of 33 ISL6267 Electrical Specifications Operating Conditions: VDD = 5V, TA = -10°C to +100°C (HRZ), TA = -40°C to +100°C (IRZ), fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN (Note 6) TYP MAX (Note 6) UNITS 280 300 320 kHz 275 300 CHANNEL FREQUENCY Nominal Channel Frequency HRZ, fSW(nom) IRZ, fSW(nom) Adjustment Range 325 kHz 200 500 kHz -0.15 +0.15 mV AMPLIFIERS IFB = 0A Current-Sense Amplifier Input Offset Error Amp DC Gain Av0 Error Amp Gain-Bandwidth Product GBW CL = 20pF 90 dB 18 MHz ISEN Imbalance Voltage Maximum of ISENs - Minimum of ISENs 1 Input Bias Current 20 mV nA POWER-GOOD AND PROTECTION MONITORS PGOOD Low Voltage VOL IPGOOD = 4mA PGOOD Leakage Current IOH PGOOD = 3.3V PGOOD Delay 0.26 -1 0.4 V 1 µA 460 tPGD µs GATE DRIVER UGATE Pull-Up Resistance RUGPU 200mA Source Current 1.0 UGATE Source Current IUGSRC UGATE - PHASE = 2.5V 2.0 UGATE Sink Resistance RUGPD 250mA Sink Current 1.0 UGATE Sink Current IUGSNK UGATE - PHASE = 2.5V 2.0 LGATE Pull-Up Resistance RLGPU 250mA Source Current 1.0 LGATE Source Current ILGSRC LGATE - VSSP = 2.5V 2.0 1.5 A 1.5 A 1.5 A 0.9 LGATE Sink Resistance RLGPD 250mA Sink Current 0.5 LGATE Sink Current ILGSNK LGATE - VSSP = 2.5V 4.0 A UGATE to LGATE Deadtime tUGFLGR UGATE falling to LGATE rising, no load 23 ns LGATE to UGATE Deadtime tLGFUGR LGATE falling to UGATE rising, no load 28 ns BOOTSTRAP DIODE Forward Voltage VF PVCC = 5V, IF = 2mA 0.58 V Reverse Leakage IR VR = 25V 0.2 µA PROTECTION Overvoltage Threshold OVH VSEN rising above setpoint for > 1ms Severe Overvoltage Threshold OVHS VO rising above threshold > 0.5 µs Undervoltage Threshold OVH Current Imbalance Threshold Core OCP Current Threshold VSEN falls below setpoint for > 1ms HRZ IRZ FN7801 Rev 1.00 Jan 8, 2013 330 260 330 mV V 400 9 mV mV 3-Phase CCM, 2-Phase CCM, 1-Phase 50 60 70 A 3-Phase DE 16 20 24 A 24 30 36 A 49.5 60 70 A 3-Phase DE 16 20 24 A 2-Phase DE 23.5 30 36 A 2-Phase CCM, 1-Phase 50 60 70 A 2-Phase DE 24 30 36 A 2-Phase CCM, 1-Phase 49.5 60 70 A 2-Phase DE 23.5 30 36 A 2-Phase DE HRZ 270 1.800 One ISEN above another ISEN for > 1.2ms IRZ Northbridge OCP Current Threshold 200 3-Phase CCM, 2-Phase CCM, 1-Phase Page 12 of 33 ISL6267 Electrical Specifications Operating Conditions: VDD = 5V, TA = -10°C to +100°C (HRZ), TA = -40°C to +100°C (IRZ), fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN (Note 6) TYP MAX (Note 6) UNITS 0.3 V LOGIC THRESHOLDS ENABLE Input Low VIL ENABLE Input High VIH 0.7 V PWM PWM Output Low V0L Sinking 5mA PWM Output High V0H Sourcing 5mA PWM Tri-State Leakage 1.0 3.5 PWM = 2.5V V V 2 µA THERMAL MONITOR NTC Source Current NTC = 1.3V 57 67 µA NTC_NB Source Current NTC_NB = 1.3V 57 67 µA Thermal Monitor Trip Voltage Falling Threshold 0.87 0.88 0.89 V Thermal Monitor Reset Voltage Rising Threshold 0.91 0.92 0.93 V -1 0 INPUTS ENABLE Leakage Current IENABLE EN = 0V EN = 1V µA 18 36 µA Slew Rate (for VID Change) SR 5 7.5 10 mV/µs Soft-Start Slew Rate SSR 1.25 1.875 2.5 mV/µs PWROK, SVC, SVD Input Logic High VIH 0.798 PWROK, SVC, SVD Input Logic Low VIL 0.57 V EN = 0V, SVC and SVD = 0V 1 µA EN = 5V, SVC and SVD = 1.8V 1 µA SVI INTERFACE SVC, SVD Leakage V NOTE: 6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested. Gate Driver Timing Diagram PWM tLGFUGR 1V UGATE 1V LGATE tFL FN7801 Rev 1.00 Jan 8, 2013 tFU tRU tRL tUGFLGR Page 13 of 33 ISL6267 Theory of Operation out-of-phase. If VR1 is in 1-phase mode, the master clock signal will be distributed to Phase 1 only and be the Clock1 signal. Multiphase R3™ Modulator The ISL6267 is a multiphase regulator implementing two voltage regulators, VDD and VDDNB, on one chip controlled by AMD’s™ SVI1™ protocol. VDD can be programmed for 1-, 2- or 3-phase operation. VDDNB can be configured for 1-phase or 2-phase operation. Both regulators use the Intersil patented R3™ (Robust Ripple Regulator) modulator. The R3™ modulator combines the best features of fixed frequency PWM and hysteretic PWM while eliminating many of their shortcomings. Figure 7 conceptually shows the multiphase R3™ modulator circuit, and Figure 8 shows the operation principles. MASTER CLOCK CIRCUIT MASTER CLOCK COMP PHASE VCRM SEQUENCER VW MASTER CLOCK GMVO COMP MASTER CLOCK CLOCK1 PWM1 PWM2 CLOCK3 SLAVE CIRCUIT 1 VCRS1 HYSTERETIC WINDOW VCRM CLOCK2 CLOCK1 CLOCK2 CLOCK3 CRM VW VW CLOCK1 S R Q PWM1 PHASE1 L1 PWM3 VO VW IL1 CO GM CRS1 VCRS2 VCRS3 VCRS1 SLAVE CIRCUIT 2 VW VCRS2 CLOCK2 S R Q PWM2 PHASE2 L2 IL2 GM CRS2 SLAVE CIRCUIT 3 VW VCRS3 CLOCK3 S R Q PWM3 PHASE3 L3 IL3 GM CRS3 FIGURE 7. R3™ MODULATOR CIRCUIT A current source flows from the VW pin to the COMP pin, creating a voltage window set by the resistor between the two pins. This voltage window is called “VW window” in the following discussion. Inside the IC, the modulator uses the master clock circuit to generate the clocks for the slave circuits. The modulator discharges the ripple capacitor Crm with a current source equal to gmVo, where gm is a gain factor. Crm voltage VCRM is a sawtooth waveform traversing between the VW and COMP voltages. It resets to VW when it hits COMP, and generates a one-shot master clock signal. A phase sequencer distributes the master clock signal to the slave circuits. If VDD is in 3-phase mode, the master clock signal is distributed to the three phases, and the Clock 1~3 signals will be 120° out-of-phase. If VR1 is in 2-phase mode, the master clock signal is distributed to Phases 1 and 2, and the Clock1 and Clock2 signals will be 180° FN7801 Rev 1.00 Jan 8, 2013 FIGURE 8. R3™ MODULATOR OPERATION PRINCIPLES IN STEADY STATE Each slave circuit has its own ripple capacitor Crs, whose voltage mimics the inductor ripple current. A gm amplifier converts the inductor voltage into a current source to charge and discharge Crs. The slave circuit turns on its PWM pulse upon receiving the clock signal, and the current source charges Crs. When Crs voltage VCrs hits VW, the slave circuit turns off the PWM pulse, and the current source discharges Crs. Since the controller works with Vcrs, which are large amplitude and noise-free synthesized signals, it achieves lower phase jitter than conventional hysteretic mode and fixed PWM mode controllers. Unlike conventional hysteretic mode converters, the error amplifier allows the ISL6267 to maintain a 0.5% output voltage accuracy. Figure 9 shows the operation principles during load insertion response. The COMP voltage rises during load insertion, generating the master clock signal more quickly, so the PWM pulses turn on earlier, increasing the effective switching frequency. This allows for higher control loop bandwidth than conventional fixed frequency PWM controllers. The VW voltage rises as the COMP voltage rises, making the PWM pulses wider. During load release response, the COMP voltage falls. It takes the master clock circuit longer to generate the next master clock signal so the PWM pulse is held off until needed. The VW voltage falls as the COMP voltage falls, reducing the current PWM pulse width. This kind of behavior gives the ISL6267 excellent response speed. The fact that all the phases share the same VW window voltage also ensures excellent dynamic current balance among phases. Page 14 of 33 ISL6267 VW PHASE COMP VCRM UG ATE MASTER CLOCK LG ATE CLOCK1 PWM1 IL CLOCK2 FIGURE 10. DIODE EMULATION PWM2 CLOCK3 PWM3 VW VW CCM/DCM BOUNDARY V CRS IL VCRS1 VCRS3 VCRS2 VW LIGHT DCM V CRS FIGURE 9. R3™ MODULATOR OPERATION PRINCIPLES IN LOAD INSERTION RESPONSE Diode Emulation and Period Stretching The ISL6267 can operate in diode emulation (DE) mode to improve light-load efficiency. In DE mode, the low-side MOSFET conducts when the current is flowing from source to drain and does not allow reverse current, thus emulating a diode. As Figure 10 shows, when LGATE is on, the low-side MOSFET carries current, creating negative voltage on the phase node due to the voltage drop across the ON-resistance. The ISL6267 monitors the current by monitoring the phase node voltage. It turns off LGATE when the phase node voltage reaches zero to prevent the inductor current from reversing the direction and creating unnecessary power loss. If the load current is light enough, (see Figure 10). The inductor current reaches and stays at zero before the next phase node pulse, and the regulator is in discontinuous conduction mode (DCM). If the load current is heavy enough, the inductor current will never reaches 0A, and the regulator is in CCM, although the controller is in DE mode. Figure 11 shows the operation principle in diode emulation mode at light load. The load gets incrementally lighter in the three cases from top to bottom. The PWM on-time is determined by the VW window size and therefore is the same, making the inductor current triangle the same in the three cases. The ISL6267 clamps the ripple capacitor voltage VCRS in DE mode to make it mimic the inductor current. It takes the COMP voltage longer to hit VCRS, naturally stretching the switching period. The inductor current triangles move farther apart such that the inductor current average value is equal to the load current. The reduced switching frequency helps increase light-load efficiency. FN7801 Rev 1.00 Jan 8, 2013 IL VW DEEP DCM V CRS IL FIGURE 11. PERIOD STRETCHING Start-up Timing With the controller's VDD and VIN voltages above their POR threshold, the start-up sequence begins when ENABLE exceeds the logic high threshold. Figure 12 shows the typical start-up timing of VR1 and VR2. The ISL6267 uses digital soft-start to rampup DAC to the voltage programmed by the Metal VID. PGOOD is asserted high and low at the end of the rampup. Similar results occur if ENABLE is tied to VDD, with the soft-start sequence starting 800µs after VDD crosses the POR threshold. Page 15 of 33 ISL6267 voltage drops substantially (see “Electrical Specifications” on page 11). VDD SLEW RATE 1.875mV/µs MetalVID VID COMMAND ENABLE Serial VID Interface The on-board Serial VID Interface (SVI) circuitry allows the processor to directly control the Core and Northbridge voltage reference levels within the ISL6267. The SVC and SVD states are decoded according to the PWROK inputs as described in the following sections. The ISL6267 uses a digital-to-analog converter (DAC) to generate a reference voltage based on the decoded SVI value. See Figure 13 for a simple SVI interface timing diagram. VOLTAGE 800µs DAC PGOOD PWROK VIN FIGURE 12. TYPICAL SOFT-START WAVEFORMS Pre-PWROK Metal VID Power-On Reset Typical motherboard start-up begins with the controller decoding the SVC and SVD inputs to determine the pre-PWROK Metal VID setting (see Table 1). Once the ENABLE input exceeds the rising threshold, the ISL6267 decodes and locks the decoded value in an on-board hold register. Before the controller has sufficient bias to guarantee proper operation, the ISL6267 requires both a +5V input supply tied to VCC and PVCC, as well as a battery or other input supply tied to VIN, to exceed their respective rising power-on reset (POR) thresholds. Once these thresholds are reached or exceeded, the ISL6267 has enough bias to begin checking SVI inputs. Hysteresis between the rising and the falling thresholds assure the ISL6267 does not inadvertently turn off unless the bias 1 2 3 4 5 6 7 8 9 10 11 12 VCC SVC SVD ENABLE PWROK METAL_VID VCORE/ VNB V_SVI METAL_VID V_SVI PGOOD Interval 1 to 2: ISL6267 waits to POR. Interval 2 to 3: SVC and SVD are externally set to pre-Metal VID code. Interval 3 to 4: ENABLE locks pre-Metal VID code. All outputs soft-start to this level. Interval 4 to 5: PGOOD signal goes HIGH, indicating proper operation. Interval 5 to 6: CPU detects PGOOD high, and drives PWROK high, to allow ISL6267 to prepare for SVI commands. Interval 6 to 7: SVC and SVD data lines communicate change in VID code. Interval 7 to 8: ISL6267 responds to VID-ON-THE-FLY code change. Interval 8 to 9: PWROK is driven low, and ISL6267 returns all outputs to pre-PWROK Metal VID level. Interval 9 to 10: PWROK driven high once again by CPU, and ISL6267 prepares for SVI commands. Interval 10 to 11: SVC and SVD data lines communicate new VID code. Interval 11 to 12: ISL6267 drives outputs to new VID code level. Post 12: Enable falls, all internal drivers are tri-stated, and PGOOD is driven low. FIGURE 13. SVI INTERFACE TIMING DIAGRAM: TYPICAL PRE-PWROK METAL VID START-UP FN7801 Rev 1.00 Jan 8, 2013 Page 16 of 33 ISL6267 TABLE 1. PRE-PWROK METAL VID CODES SVC SVD OUTPUT VOLTAGE (V) 0 0 1.1 0 1 1.0 1 0 0.9 1 1 0.8 The internal DAC circuitry begins to ramp Core and Northbridge VRs to the decoded pre-PWROK Metal VID output level. The digital soft-start circuitry ramps the internal reference to the target gradually at a fixed rate of approximately 2mV/µs. The controlled ramp of all output voltage planes reduces in-rush current during the soft-start interval. At the end of the soft-start interval, the PGOOD output transitions high, indicating all output planes are within regulation limits. If the ENABLE input falls below the enable falling threshold, the ISL6267 tri-states both outputs. PGOOD is pulled low with the loss of ENABLE. The Core and Northbridge planes decay, based on output capacitance and load leakage resistance. If bias to VCC falls below the POR level, the ISL6267 responds in the manner previously described. Once VCC and ENABLE rise above their respective rising thresholds, the internal DAC circuitry reacquires a pre-PWROK metal VID code, and the controller soft-starts. VFIX Mode The ISL6267 does not support VFIX Mode. In the event a CPU is not present on a motherboard and the ISL6267 is powered on, the state of SVC and SVD sets the pre-PWROK metal VID as the “Pre-PWROK Metal VID” on page 16 and begins soft-starting. SVI Mode Once the controller has successfully soft-starts and PGOOD and PGOOD_NB transition high, the processor can assert PWROK to signal the ISL6267 to prepare for SVI commands. The controller actively monitors the SVI interface for set VID commands to move the plane voltages to start-up VID values. Details of the SVI Bus protocol are provided in the “AMD Design Guide for Voltage Regulator Controllers Accepting Serial VID Codes” specification. A POR event on either VCC or VIN during normal operation shuts down both regulators, and both PGOOD outputs are pulled low. The pre-PWROK metal VID code is not retained. VID-on-the-Fly Transition Once PWROK is high, the ISL6267 detects this flag and begins monitoring the SVC and SVD pins for SVI instructions. The microprocessor follows the protocol outlined in the following sections to send instructions for the VID-on-the-fly transitions. The ISL6267 decodes the instruction and acknowledges the new VID code. For the VID codes higher than the current VID level, the ISL6267 begins stepping the commanded VR outputs to the new VID target with a typical slew rate of 7.5mV/µs, which meets the AMD requirements. When the VID codes are lower than the current VID level, the ISL6267 checks the state of PSI_L. If PSI_L is high, the controller begins stepping the regulator output to the new VID target with a typical slew rate of -7.5mV/µs. If PSI_L is low, the controller allows the output voltage to decay and slowly steps the DAC down with the natural decay of the output. This allows the controller to quickly recover and move to a high VID code if commanded. AMD requirements under these conditions do not require the regulator to meet the minimum slew rate specification of -5mV/µs. In either case, the slew rate is not allowed to exceed 10mV/µs. The ISL6267 does not change the state of PGOOD (VCCPWRGD in AMD specifications), when a VID-on-the-fly transition occurs. SVI WIRE Protocol The SVI WIRE protocol is based on the I2C bus concept. Two wires [serial clock (SVC) and serial data (SVD)], carry information between the AMD processor (master) and the VR controller (slave) on the bus. The master initiates and terminates SVI transactions and drives the clock, SVC, during a transaction. The AMD processor is always the master, and the voltage regulators are the slaves. The slave receives the SVI transactions and acts accordingly. Mobile SVI WIRE protocol timing is based on high-speed mode I2C. See AMD publication #40182 for additional details. Once a set VID command is received, the ISL6267 decodes the information to determine which VR is affected and which VID target is required (see Table 2). The internal DAC circuitry steps the output voltage of the VR commanded to the new VID level. During this time, one or more of the VR outputs could be targeted. In the event either VR is commanded to power-off by serial VID commands, the PGOOD signal remains asserted. If the PWROK input is de-asserted, then the controller steps both the Core and the Northbridge VRs back to the stored pre-PWROK metal VID level in the holding register from initial soft-start. No attempt is made to read the SVC and SVD inputs during this time. If PWROK is re-asserted, then the on-board SVI interface waits for a set VID command. If ENABLE goes low during normal operation, all internal drivers are tri-stated and PGOOD is pulled low. This event clears the pre-PWROK metal VID code and forces the controller to check SVC and SVD upon restart. FN7801 Rev 1.00 Jan 8, 2013 Page 17 of 33 ISL6267 . TABLE 2. SERIAL VID CODES SVID [6:0] VOLTAGE (V) SVID [6:0] VOLTAGE (V) SVID [6:0] VOLTAGE (V) SVID [6:0] VOLTAGE (V) 000_0000b 1.5500 010_0000b 1.1500 100_0000b 0.7500 110_0000b 0.3500* 000_0001b 1.5375 010_0001b 1.1375 100_0001b 0.7375 110_0001b 0.3375* 000_0010b 1.5250 010_0010b 1.1250 100_0010b 0.7250 110_0010b 0.3250* 000_0011b 1.5125 010_0011b 1.1125 100_0011b 0.7125 110_0011b 0.3125* 000_0100b 1.5000 010_0100b 1.1000 100_0100b 0.7000 110_0100b 0.3000* 000_0101b 1.4875 010_0101b 1.0875 100_0101b 0.6875 110_0101b 0.2875* 000_0110b 1.4750 010_0110b 1.0750 100_0110b 0.6750 110_0110b 0.2750* 000_0111b 1.4625 010_0111b 1.0625 100_0111b 0.6625 110_0111b 0.2625* 000_1000b 1.4500 010_1000b 1.0500 100_1000b 0.6500 110_1000b 0.2500* 000_1001b 1.4375 010_1001b 1.0375 100_1001b 0.6375 110_1001b 0.2375* 000_1010b 1.4250 010_1010b 1.0250 100_1010b 0.6250 110_1010b 0.2250* 000_1011b 1.4125 010_1011b 1.0125 100_1011b 0.6125 110_1011b 0.2125* 000_1100b 1.4000 010_1100b 1.0000 100_1100b 0.6000 110_1100b 0.2000* 000_1101b 1.3875 010_1101b 0.9875 100_1101b 0.5875 110_1101b 0.1875* 000_1110b 1.3750 010_1110b 0.9750 100_1110b 0.5750 110_1110b 0.1750* 000_1111b 1.3625 010_1111b 0.9625 100_1111b 0.5625 110_1111b 0.1625* 001_0000b 1.3500 011_0000b 0.9500 101_0000b 0.5500 111_0000b 0.1500* 001_0001b 1.3375 011_0001b 0.9375 101_0001b 0.5375 111_0001b 0.1375* 001_0010b 1.3250 011_0010b 0.9250 101_0010b 0.5250 111_0010b 0.1250* 001_0011b 1.3125 011_0011b 0.9125 101_0011b 0.5125 111_0011b 0.1125* 001_0100b 1.3000 011_0100b 0.9000 101_0100b 0.5000 111_0100b 0.1000* 001_0101b 1.2875 011_0101b 0.8875 101_0101b 0.4875* 111_0101b 0.0875* 001_0110b 1.2750 011_0110b 0.8750 101_0110b 0.4750* 111_0110b 0.0750* 001_0111b 1.2625 011_0111b 0.8625 101_0111b 0.4625* 111_0111b 0.0625* 001_1000b 1.2500 011_1000b 0.8500 101_1000b 0.4500* 111_1000b 0.0500* 001_1001b 1.2375 011_1001b 0.8375 101_1001b 0.4375* 111_1001b 0.0375* 001_1010b 1.2250 011_1010b 0.8250 101_1010b 0.4250* 111_1010b 0.0250* 001_1011b 1.2125 011_1011b 0.8125 101_1011b 0.4125* 111_1011b 0.0125* 001_1100b 1.2000 011_1100b 0.8000 101_1100b 0.4000* 111_1100b OFF 001_1101b 1.1875 011_1101b 0.7875 101_1101b 0.3875* 111_1101b OFF 001_1110b 1.1750 011_1110b 0.7750 101_1110b 0.3750* 111_1110b OFF 001_1111b 1.1625 011_1111b 0.7625 101_1111b 0.3625* 111_1111b OFF NOTE: *Indicates a VID not required for AMD Family 10h processors. FN7801 Rev 1.00 Jan 8, 2013 Page 18 of 33 ISL6267 PSI_L See Table 3 6 5 4 3 2 1 7 0 SVID 6 4 5 3 2 1 0 SVC STOP ACK DATA PHASE ACK SLAVE ADDRESS PHASE WRITE START SVD FIGURE 14. SEND BYTE EXAMPLE SVI Bus Protocol VR Offset Programming The AMD processor bus protocol is compliant with SMBus send byte protocol for VID transactions (see Figure 14). During a send byte transaction, the processor sends the start sequence followed by the slave address of the VR for which the VID command applies. The address byte must be configured according to Table 3. The processor then sends the write bit. After the write bit, if the ISL6267 receives a valid address byte, it sends the acknowledge bit. The processor then sends the PSI-L bit and VID bits during the data phase. The Serial VID 8-bit data field encoding is outlined in Table 4. If the ISL6267 receives a valid 8-bit code during the data phase, it sends the acknowledge bit. Finally, the processor sends the stop sequence. After the ISL6267 has detected the stop, it can then proceed with the VID-on-the-fly transition. A positive or negative offset is programmed for the Core VR using a resistor to ground from the PROG1 pin and the Northbridge in a similar manner from the PROG2 pin. Table 5 provides the resistor value to select the desired output voltage offset TABLE 3. SVI SEND BYTE ADDRESS DESCRIPTION BITS 6:4 DESCRIPTION Always 110b 3 Reserved by AMD for future use 2 VDD1; if set, then the following data byte contains the VID for VDD1 [Note: The ISL6267 does not support VDD1] 1 VDD0; if set, then the following data byte contains the VID for VID0 0 VDDNB; if set then the following data byte contains the VID for VIDNB TABLE 4. SERIAL VID 8-BIT DATA FIELD ENCODING TABLE 5. PROGx PIN RESISTOR VALUE RESISTOR VALUE [] PROG1 VCORE OFFSET [mV] PROG1 VNBOFFSET [mV] 0 50 50 590 43.75 43.75 1100 37.50 37.50 1690 31.25 31.25 2260 25.00 25.00 3160 18.75 18.75 4320 12.50 12.50 5620 6.25 6.25 6650 0.00 0.00 7870 -6.25 -6.25 9530 -12.50 -12.50 11500 -18.75 -18.75 14000 -25.00 -25.00 16500 -31.25 -31.25 BITS DESCRIPTION 18700 -37.50 -37.50 7 PSI_L: =0 means the processor is at an optimal load for the regulators to enter power-saving mode =1 means the processor is not at an optimal load for the regulators to enter power-saving mode OPEN -43.75 -43.75 6:0 SVID[6:0] as defined in Table 2. Operation After the start-up sequence, the ISL6267 begins regulating the Core and Northbridge output voltages to the pre-PWROK metal VID programmed. The controller monitors SVI commands to determine when to enter power-saving mode, implement dynamic VID changes, and shut down individual outputs. FN7801 Rev 1.00 Jan 8, 2013 Voltage Regulation and Load Line Implementation After the start sequence, the ISL6267 regulates the output voltage to the value set by the VID information, per Table 2. The ISL6267 controls the no-load output voltage to an accuracy of ±0.5% over the range of 0.75V to 1.55V. A differential amplifier allows voltage sensing for precise voltage regulation at the microprocessor die. Page 19 of 33 ISL6267 amplifier regulates the inverting and non-inverting input voltages to be equal as shown in Equation 3: Rdroop + FB VCCSENSE Vdroop VCC SENSE + V VR LOCAL VO “CATCH” RESISTOR Idroop COMP - + VDAC INTERNAL TO IC VIDs DAC RTN X1 + - VSSSENSE VCC SENSE – VSS SENSE = V DAC – R droop I droop As the load current increases from zero, the output voltage droops from the VID table value by an amount proportional to the load current, to achieve the load line. The ISL6267 can sense the inductor current through the intrinsic DC Resistance (DCR) of the inductors, as shown in Figures 15 and 16, or through resistors in series with the inductors as shown in Figure 17. In both methods, capacitor Cn voltage represents the inductor total currents. A droop amplifier converts Cn voltage into an internal current source with the gain set by resistor Ri. The current source is used for load line implementation, current monitoring and overcurrent protection. Figure 15 shows the load-line implementation. The ISL6267 drives a current source (Idroop) out of the FB pin, as described by Equation 1. (EQ. 1) When using inductor DCR current sensing, a single NTC element is used to compensate the positive temperature coefficient of the copper winding, thus sustaining the load-line accuracy with reduced cost. Idroop flows through resistor Rdroop and creates a voltage drop as shown in Equation 2. (EQ. 2) Vdroop is the droop voltage required to implement load line. Changing Rdroop or scaling Idroop can change the load line slope. Since Idroop sets the overcurrent protection level, it is recommended to first scale Idroop based on OCP requirement, then select an appropriate Rdroop value to obtain the desired load line slope. Differential Sensing Figure 15 shows the differential voltage sensing scheme. VCCSENSE and VSSSENSE are the remote voltage sensing signals from the processor die. A unity gain differential amplifier senses the VSSSENSE voltage and adds it to the DAC output. The error FN7801 Rev 1.00 Jan 8, 2013 (EQ. 4) VSS FIGURE 15. DIFFERENTIAL SENSING AND LOAD LINE IMPLEMENTATION V droop = R droop I droop (EQ. 3) VID<0:7> “CATCH” RESISTOR 2xV Cn I droop = ---------------Ri = V DAC + VSS SENSE Rewriting Equation 3 and substituting Equation 2 gives Equation 4 is the exact equation required for load-line implementation. + E/A droop The VCCSENSE and VSSSENSE signals come from the processor die. The feedback is an open circuit in the absence of the processor. As Figure 15 shows, it is recommended to add a “catch” resistor to feed the VR local output voltage back to the compensator, and to add another “catch” resistor to connect the VR local output ground to the RTN pin. These resistors, typically 10~100, provide voltage feedback if the system is powered up without a processor installed. Phase Current Balancing Rdcr3 L3 ISEN3 INTERNAL INTERNAL TO IC ISEN2 PHASE3 Risen Cisen ISEN1 IL3 PHASE1 Risen Rdcr2 L2 PHASE2 Risen Cisen Rpcb3 Rpcb2 VO IL2 Rdcr1 L1 Rpcb1 IL1 Cisen FIGURE 16. CURRENT BALANCING CIRCUIT The ISL6267 monitors individual phase average current by monitoring the ISEN1, ISEN2, and ISEN3 voltages. Figure 16 shows the current balancing circuit recommended for the ISL6267. Each phase node voltage is averaged by a low-pass filter consisting of Risen and Cisen, and is presented to the corresponding ISEN pin. Risen should be routed to the inductor phase-node pad in order to eliminate the effect of phase node parasitic PCB DCR. Equations 5 through 7 give the ISEN pin voltages: V ISEN1 = R dcr1 + R pcb1 I L1 (EQ. 5) V ISEN2 = R dcr2 + R pcb2 I L2 (EQ. 6) V ISEN3 = R dcr3 + R pcb3 I L3 (EQ. 7) where Rdcr1, Rdcr2 and Rdcr3 are inductor DCR; Rpcb1, Rpcb2 and Rpcb3 are parasitic PCB DCR between the inductor output side pad and the output voltage rail; and IL1, IL2 and IL3 are inductor average currents. Page 20 of 33 ISL6267 The ISL6267 adjusts the phase pulse-width relative to the other phases to make VISEN1 = VISEN2 = VISEN3, thus to achieve IL1 = IL2 = IL3, when Rdcr1 = Rdcr2 = Rdcr3 and Rpcb1 = Rpcb2 = Rpcb3. Combining Equations 13 and 14 gives: Using the same components for L1, L2 and L3 provides a good match of Rdcr1, Rdcr2 and Rdcr3. Board layout determines Rpcb1, Rpcb2 and Rpcb3. It is recommended to have symmetrical layout for the power delivery path between each inductor and the output voltage rail, such that Rpcb1 = Rpcb2 = Rpcb3. R dcr1 I L1 = R dcr2 I L2 = R dcr3 I L3 PHASE3 R isen ISEN3 Cisen INTERNAL TO IC V3p R dcr3 L3 IL3 R isen Rpcb3 V3n R isen V2p PHASE2 R isen ISEN2 C isen R dcr2 L2 IL2 R isen R pcb2 Vo V2n R isen ISEN1 Cisen PHASE1 V1p R isen R isen R dcr1 L1 IL1 Rpcb1 V1n V 1p – V 1n = V 2p – V 2n = V 3p – V 3n (EQ. 15) Therefore: (EQ. 16) Current balancing (IL1 = IL2 = IL3) is achieved when Rdcr1 = Rdcr2 = Rdcr3. Rpcb1, Rpcb2 and Rpcb3 do not have any effect. Since the slave ripple capacitor voltages mimic the inductor currents, the R3™ modulator can naturally achieve excellent current balancing during steady state and dynamic operations. Figure 18 shows the current balancing performance of the evaluation board with load transient of 12A/51A at different rep rates. The inductor currents follow the load current dynamic change with the output capacitors supplying the difference. The inductor currents can track the load current well at a low repetition rate, but cannot keep up when the repetition rate gets into the hundred-kHz range, where it is out of the control loop bandwidth. The controller achieves excellent current balancing in all cases installed. CCM Switching Frequency R isen FIGURE 17. DIFFERENTIAL-SENSING CURRENT BALANCING CIRCUIT Sometimes, it is difficult to implement symmetrical layout. For the circuit shown in Figure 16, asymmetric layout causes different Rpcb1, Rpcb2 and Rpcb3 values, thus creating a current imbalance. Figure 17 shows a differential sensing current balancing circuit recommended for the ISL6267. The current sensing traces should be routed to the inductor pads so they only pick up the inductor DCR voltage. Each ISEN pin sees the average voltage of three sources: its own, phase inductor phase-node pad, and the other two phases inductor output side pads. Equations 8 through 10 give the ISEN pin voltages: V ISEN1 = V 1p + V 2n + V 3n (EQ. 8) V ISEN2 = V 1n + V 2p + V 3n (EQ. 9) V ISEN3 = V 1n + V 2n + V 3p (EQ. 10) The Rfset resistor between the COMP and the VW pins sets the VW windows size and therefore sets the switching frequency. When the ISL6267 is in continuous conduction mode (CCM), the switching frequency is not absolutely constant due to the nature of the R3™ modulator. As explained in the “Multiphase R3™ Modulator” on page 14, the effective switching frequency increases during load insertion and decreases during load release to achieve fast response. Thus, the switching frequency is relatively constant at steady state. Variation is expected when the power stage condition, such as input voltage, output voltage, load, etc. changes. The variation is usually less than 15% and does not have any significant effect on output voltage ripple magnitude. Equation 17 gives an estimate of the frequency-setting resistor (Rfset) value. A value of 8k Rfset gives approximately 300kHz switching frequency. Lower resistance gives higher switching frequency. R fset k = Period s – 0.29 2.65 (EQ. 17) The ISL6267 will make VISEN1 = VISEN2 = VISEN3 as shown in Equations 11 and 12: V 1p + V 2n + V 3n = V 1n + V 2p + V 3n (EQ. 11) V 1n + V 2p + V 3n = V 1n + V 2n + V 3p (EQ. 12) Rewriting Equation 11 gives Equation 13: V 1p – V 1n = V 2p – V 2n (EQ. 13) Rewriting Equation 12 gives Equation 14: V 2p – V 2n = V 3p – V 3n FN7801 Rev 1.00 Jan 8, 2013 (EQ. 14) Page 21 of 33 ISL6267 Modes of Operation REP RATE = 10kHz TABLE 6. CORE VR MODES OF OPERATION PWM3 ISEN2 To External Driver PSL_L MODE To Power 3-phase Stage CPU VR Config. 1 3-phase CCM 60 0 1-phase DE 20 2-phase CPU VR Config. 1 2-phase CCM 60 0 1-phase DE 30 1-phase CPU VR Config. X 1-phase DE 60 Tied to 5V Tied to 5V REP RATE = 25kHz CONFIG. OCP THRESHOLD (µA) The Core VR can be configured for 3, 2- or 1-phase operation. Table 6 shows Core VR configurations and operational modes, programmed by the PWM3 and ISEN2 pin status and the PS command. For 2-phase configuration, tie the PWM3 pin to 5V. In this configuration, phases 1 and 2 are active. For 1-phase configuration, tie the PWM3 pin and the ISEN2 pin to 5V. In this configuration, only phase-1 is an active, the controller operates in DE mode and the PSI_L input been ignored. REP RATE = 50kHz In 3-phase configuration, Core VR operates in 3-phase CCM, with PSI_L high. It enters 1-phase DE mode when PSI_L is low, dropping phases 3 and 2, and reduces the overcurrent and the way-overcurrent protection levels to one-third of the initial values. In 2-phase configuration, Core VR operates in 2-phase CCM with PSI_L high. It enters 1-phase DE mode with PSI_L low, by dropping phase 2 and reduces the overcurrent and the way overcurrent protection levels to one-half of the initial values. REP RATE = 100kHz In 1-phase configuration, the Core VR operates in 1-phase DE and ignores the PSI_L input. If a resistor is placed from COMP pin to GND with a value less than 150k, then the Core VR operates in 1-phase CCM with PSI_L high and enters 1-phase DE mode when PSI_L is low. A resistor value of 100kis recommended. TABLE 7. NORTHBRIDGE VR MODES OF OPERATION ISEN2_NB REP RATE = 200kHz FIGURE 18. CURRENT BALANCING DURING DYNAMIC OPERATION. CH1: IL1 , CH2: ILOAD, CH3: IL2, CH4: IL3 FN7801 Rev 1.00 Jan 8, 2013 CONFIG. PSL_L MODE OCP THRESHOLD (µA) To Power Stage 2-phase NB VR Config. 1 2-phase CCM 60 0 1-phase DE 30 Tied to 5V 1-phase NB VR Config. X 1-phase DE 60 ISL6267 Northbridge (NB) VR can be configured for 2- or 1-phase operation. Table 7 shows the Northbridge VR configurations and operational modes, which are programmed by the ISEN2 pin status and the PSI_L command. For 1-phase configuration, tie the ISEN2_NB pin to 5V. In 1-phase configuration, the Northbridge VR operates in 1-phase DE and ignores the PSI_L input. If a resistor is placed from COMP_NB pin to GND with a value less than 150k, then the Northbridge VR operates in 1phase CCM with PSI_L high and enters 1-phase DE mode when PSI_L is low. A resistor value of 100k is recommended. The Northbridge VR can be disabled completely by tying ISUMN_NB to 5V. Page 22 of 33 ISL6267 Dynamic Operation Core VR and Northbridge VR behave the same during dynamic operation. The controller responds to VID-on-the-fly changes by slewing to the new voltage at the fixed 7.5mV/µs slew rate. During negative VID transitions, the output voltage decays to the lower VID value at the slew rate determined by the load. SVI_L low command prompts the controller to enter DE mode. Overvoltage protection is blanked during VID down transition in DE mode until the output voltage is within 60mV of the VID value. During load insertion response, the Fast Clock function increases the PWM pulse response speed. The controller monitors the VSEN pin voltage and compares it to 100ns-filtered version. When the unfiltered version is 20mV below the filtered version, the controller knows there is a fast voltage dip due to load insertion, and it issues an additional master clock signal to deliver a PWM pulse immediately. The R3™ modulator intrinsically has voltage feed-forward. The output voltage is insensitive to a fast slew rate input voltage change. threshold. When ENABLE and VDD return to their high operating levels, a soft-start occurs. Table 8 summarizes the fault protections. TABLE 8. FAULT PROTECTION SUMMARY FAULT DURATION BEFORE PROTECTION FAULT TYPE Overcurrent 120µs Phase Current Unbalance FAULT RESET ENABLE toggle or VDD toggle PWM tri-state, PGOOD latched low 1ms Way-Overcurrent (1.5xOC) PROTECTION ACTION Immediately Overvoltage +200mV PGOOD latched low. Actively pulls the output voltage to below VID value, then tri-state. Over-Temperature 400µs N/A Protections FB2 Function Core VR and Northbridge VR both provide overcurrent, current-balance and overvoltage fault protections. The controller also provides over-temperature protection. The following discussion is based on Core VR and also applies to Northbridge VR. The FB2 function is only available for Core VR or Northbridge VR in 2-phase configuration. The controller determines overcurrent protection (OCP) by comparing the average value of the droop current (Idroop) with an internal current source threshold as Table 6 shows. It declares OCP when Idroop is above the threshold for 120µs. For overcurrent conditions above 1.5x the OCP level, the PWM outputs immediately shuts off and PGOOD goes low to maximize protection. This protection is also referred to as way-overcurrent protection or fast overcurrent protection for short-circuit protections. The controller monitors the ISEN pin voltages to determine current-balance protection. If the ISEN pin voltage difference is greater than 9mV for 1ms, the controller will declare a fault and latch off. The controller takes the same actions for all of the previously describe fault protections: de-assertion of PGOOD and turn-off of the high-side and low-side power MOSFETs. Any residual inductor current decays through the MOSFET body diodes. The controller declares an overvoltage fault and de-asserts PGOOD if the output voltage exceeds the VID set value by +250mV. The ISL6267 immediately declares an OV fault, de-asserts PGOOD, and turn on the low-side power MOSFETs. The low-side power MOSFETs remain on until the output voltage is pulled down below the VID set value when all power MOSFETs are turned off. If the output voltage rises above the VID set value +250mV again, the protection process is repeated. This behavior provides the maximum amount of protection against shorted high-side power MOSFETs while preventing output ringing below ground. All of the previously described fault conditions can be reset by bringing ENABLE low or by bringing VDD below the POR FN7801 Rev 1.00 Jan 8, 2013 C1 R2 CONTROLLER IN 2-PHASE MODE C2 R3 VSEN C3.1 CONTROLLER IN 1-PHASE MODE FB2 C3.2 C2 R3 R1 FB VREF E/A VSEN C1 R2 C3.1 FB2 C3.2 R1 FB COMP E/A VREF COMP FIGURE 19. FB2 FUNCTION Figure 19 shows the FB2 function. A switch (called FB2 switch) turns on to short the FB and the FB2 pins when the controller is in 2-phase mode. Capacitors C3.1 and C3.2 are in parallel, serving as part of the compensator. When the controller enters 1-phase mode, the FB2 switch turns off, removing C3.2 and leaving only C3.1 in the compensator. The compensator gain increases with the removal of C3.2. By properly sizing C3.1 and C3.2, the compensator can be optimal for both 2-phase mode and 1-phase mode. When the FB2 switch is off, C3.2 is disconnected from the FB pin. However, the controller still actively drives the FB2 pin voltage to follow the FB pin voltage such that C3.2 voltage always follows C3.1 voltage. When the controller turns on the FB2 switch, C3.2 is reconnected to the compensator smoothly. The FB2 function ensures excellent transient response in both 2-phase and 1-phase mode. If the FB2 function is not used, populate C3.1 only. Adaptive Body Diode Conduction Time Reduction In DCM, the controller turns off the low-side MOSFET when the inductor current approaches zero. During on-time of the low-side MOSFET, phase voltage is negative, and the amount is the Page 23 of 33 ISL6267 MOSFET rDS(ON) voltage drop, which is proportional to the inductor current. A phase comparator inside the controller monitors the phase voltage during on-time of the low-side MOSFET and compares it with a threshold to determine the zero crossing point of the inductor current. If the inductor current has not reached zero when the low-side MOSFET turns off, it will flow through the low-side MOSFET body diode, causing the phase node to have a larger voltage drop until it decays to zero. If the inductor current has crossed zero and reversed the direction when the low-side MOSFET turns off, it will flow through the high-side MOSFET body diode, causing the phase node to have a spike until it decays to zero. The controller continues monitoring the phase voltage after turning off the low-side MOSFET. To minimize the body diode-related loss, the controller also adjusts the phase comparator threshold voltage accordingly in iterative steps such that the low-side MOSFET body diode conducts for approximately 40ns. Key Component Selection Inductor DCR Current-Sensing Network PHASE1 PHASE2 PHASE3 RSUM RSUM ISUM+ RSUM L L L RNTCS + RP DCR DCR DCR RNTC RO CNVCN RI ISUM- RO RO IO FIGURE 20. DCR CURRENT-SENSING NETWORK Figure 20 shows the inductor DCR current-sensing network for a 3-phase solution. An inductor current flows through the DCR and creates a voltage drop. Each inductor has two resistors in Rsum and Ro connected to the pads to accurately sense the inductor current by sensing the DCR voltage drop. The Rsum and Ro resistors are connected in a summing network as shown, and feed the total current information to the NTC network (consisting of Rntcs, Rntc and Rp) and capacitor Cn. Rntc is a negative temperature coefficient (NTC) thermistor, used to temperature compensate the inductor DCR change. The inductor output side pads are electrically shorted in the schematic but have some parasitic impedance in actual board layout, which is why one cannot simply short them together for the current-sensing summing network. It is recommended to use 1~10Ro to create quality signals. Since Ro value is much smaller than the rest of the current sensing circuit, the following analysis ignores it. domain relationship between inductor total current Io(s) and Cn voltage VCn(s): R ntcnet DCR V Cn s = ----------------------------------------- ------------ I o s A cs s R sum N R ntcnet + ------------ N (EQ. 18) R ntcs + R ntc R p R ntcnet = --------------------------------------------------R ntcs + R ntc + R p (EQ. 19) s 1 + -----L A cs s = ---------------------s 1 + ----------- sns (EQ. 20) DCR L = -----------L (EQ. 21) 1 sns = -----------------------------------------------------R sum R ntcnet -------------N ----------------------------------------- C n R sum R ntcnet + -------------N (EQ. 22) where N is the number of phases. Transfer function Acs(s) always has unity gain at DC. The inductor DCR value increases as the winding temperature increases, giving higher reading of the inductor DC current. The NTC Rntc value decrease as its temperature decreases. Proper selection of Rsum, Rntcs, Rp and Rntc parameters ensures that VCn represents the inductor total DC current over the temperature range of interest. There are many sets of parameters that can properly temperature compensate the DCR change. Since the NTC network and the Rsum resistors form a voltage divider, Vcn is always a fraction of the inductor DCR voltage. It is recommended to have a higher ratio of Vcn to the inductor DCR voltage so the droop circuit has a higher signal level to work with. A typical set of parameters that provide good temperature compensation are: Rsum = 3.65k, Rp = 11k, Rntcs = 2.61k and Rntc = 10k (ERT-J1VR103J). The NTC network parameters may need to be fine tuned on actual boards. One can apply full load DC current and record the output voltage reading immediately; then record the output voltage reading again when the board has reached the thermal steady state. A good NTC network can limit the output voltage drift to within 2mV. It is recommended to follow the Intersil evaluation board layout and current sensing network parameters to minimize engineering time. VCn(s) also needs to represent real-time Io(s) for the controller to achieve good transient response. Transfer function Acs(s) has a pole wsns and a zero wL. One needs to match wL and wsns so Acs(s) is unity gain at all frequencies. By forcing wL equal to wsns and solving for the solution. Equation 23 gives Cn value. The summed inductor current information is presented to the capacitor Cn. Equations 18 thru 22 describe the frequency FN7801 Rev 1.00 Jan 8, 2013 Page 24 of 33 ISL6267 L C n = -----------------------------------------------------------R sum R ntcnet -------------N ----------------------------------------- DCR R sum R ntcnet + -------------N (EQ. 23) io For example, given N = 3, Rsum = 3.65k, Rp = 11k, Rntcs = 2.61k, Rntc = 10k, DCR = 0.88m and L = 0.36µH, Equation 23 gives Cn = 0.406µF. Assuming the compensator design is correct, Figure 21 shows the expected load transient response waveforms if Cn is correctly selected. When the load current Icore has a square change, the output voltage Vcore also has a square response. If Cn value is too large or too small, VCn(s) does not accurately represent real-time Io(s) and worsens the transient response. Figure 22 shows the load transient response when Cn is too small. Vcore sags excessively upon load insertion and may create a system failure. Figure 23 shows the transient response when Cn is too large. Vcore is sluggish in drooping to its final value. There is excessive overshoot if load insertion occurs during this time, which may negatively affect the CPU reliability. iL Vo RING BACK FIGURE 24. OUTPUT VOLTAGE RING-BACK PROBLEM ISUM+ Rntcs Cn.1 Cn.2 Vcn Rp Rntc Rn OPTIONAL ISUM- Ri io Rip Cip OPTIONAL Vo FIGURE 21. DESIRED LOAD TRANSIENT RESPONSE WAVEFORMS io Vo FIGURE 22. LOAD TRANSIENT RESPONSE WHEN Cn IS TOO SMALL io Vo FIGURE 23. LOAD TRANSIENT RESPONSE WHEN Cn IS TOO LARGE FN7801 Rev 1.00 Jan 8, 2013 FIGURE 25. OPTIONAL CIRCUITS FOR RING-BACK REDUCTION Figure 24 shows the output voltage ring-back problem during load transient response. The load current io has a fast step change, but the inductor current iL cannot accurately follow. Instead, iL responds in first-order system fashion due to the nature of the current loop. The ESR and ESL effect of the output capacitors makes the output voltage Vo dip quickly upon load current change. However, the controller regulates Vo according to the droop current idroop, which is a real-time representation of iL; therefore, it pulls Vo back to the level dictated by iL, causing the ring-back problem. This phenomenon is not observed when the output capacitor has very low ESR and ESL, as is the case with all ceramic capacitors. Figure 25 shows two optional circuits for reduction of the ring-back. Cn is the capacitor used to match the inductor time constant. It usually takes the parallel of two (or more) capacitors to get the desired value. Figure 25 shows that two capacitors (Cn.1 and Cn.2) are in parallel. Resistor Rn is an optional component to reduce the Vo ring-back. At steady state, Cn.1 + Cn.2 provides the desired Cn capacitance. At the beginning of io change, the effective capacitance is less because Rn increases the impedance of the Cn.1 branch. As Figure 22 shows, Vo tends to dip when Cn is too small, and this effect reduces the Vo ring-back. This effect is more pronounced when Cn.1 is much larger than Cn.2. It is also more pronounced when Rn is bigger. However, the presence of Rn increases the ripple of the Vn signal if Cn.2 is too small. It is recommended to keep Cn.2 greater than 2200pF. Rn value usually is a few ohms. Cn.1, Cn.2 and Rn values should be determined through tuning the load transient response Page 25 of 33 ISL6267 waveforms on an actual board. Rip and Cip form an R-C branch in parallel with Ri, providing a lower impedance path than Ri at the beginning of io change. Rip and Cip do not have any effect at steady state. Through proper selection of Rip and Cip values, idroop can resemble io rather than iL, and Vo will not ring back. The recommended value for Rip is 100. Cip should be determined through tuning the load transient response waveforms on an actual board. The recommended range for Cip is 100pF~2000pF. However, it should be noted that the Rip -Cip branch may distort the idroop waveform. Instead of being triangular as the real inductor current, idroop may have sharp spikes, which may adversely affect idroop average value detection and therefore may affect OCP accuracy. User discretion is advised. Resistor Current-Sensing Network PHASE1 PHASE2 PHASE3 L L L DCR DCR DCR ISUM+ + VCN RSEN RO - CN RI ISUM- RO IO FIGURE 26. RESISTOR CURRENT-SENSING NETWORK Figure 26 shows the resistor current-sensing network for a 2-phase solution. Each inductor has a series current sensing resistor, Rsen. Rsum and Ro are connected to the Rsen pads to accurately capture the inductor current information. The Rsum and Ro resistors are connected to capacitor Cn. Rsum and Cn form a filter for noise attenuation. Equations 24 thru 26 give the VCn(s) expression. R sen V Cn s = ------------ I o s A Rsen s N (EQ. 24) (EQ. 25) (EQ. 27) (EQ. 28) 2R ntcnet DCR I o R i = -------------------------------------------------------------------------------R sum N R ntcnet + -------------- I droop N (EQ. 29) R ntcs + R ntc R p 2 --------------------------------------------------- DCR I omax R ntcs + R ntc + R p R i = ------------------------------------------------------------------------------------------------------------------------ R ntcs + R ntc R p R sum N --------------------------------------------------- + -------------- I droopmax N R ntcs + R ntc + R p (EQ. 30) where Iomax is the full load current and Idroopmax is the corresponding droop current. For example, given N = 3, Rsum = 3.65k, Rp = 11k, Rntcs = 2.61k, Rntc = 10k, DCR = 0.88m, Iomax = 51A and Idroopmax = 40.9µA. Equation 30 gives Ri = 606. For resistor sensing, Equation 31 gives the DC relationship of Vcn(s) and Io(s). R sen V Cn = ------------ I o N (EQ. 31) Substitution of Equation 31 into Equation 1 gives Equation 32: 2 R sen I droop = ----- ------------ I o N Ri (EQ. 32) Therefore: (EQ. 26) Transfer function ARsen(s) always has unity gain at DC. Current-sensing resistor Rsen value does not have significant variation over-temperature, so there is no need for the NTC network. The recommended values are Rsum = 1k and Cn = 5600pF. FN7801 Rev 1.00 Jan 8, 2013 R ntcnet DCR V Cn = ----------------------------------------- ------------ I o N R sum R ntcnet + ------------ N Substitution of Equation 19 and application of the OCP condition in Equation 29 gives Equation 30: RO 1 Rsen = --------------------------R sum -------------- C n N For inductor DCR sensing, Equation 27 gives the DC relationship of Vcn(s) and Io(s): Therefore: RSUM 1 A Rsen s = ---------------------s 1 + ----------- sns For example, the OCP threshold is 60µA for 3-phase solution. Idroop is designed to be 40.9µA at full load. Therefore the OCP trip level is 1.5x of the full load current. R ntcnet DCR 2 I droop = ----- ----------------------------------------- ------------ I o R sum N Ri R ntcnet + -------------N RSUM RSEN Refer to Equation 1 on page 20 and Figures 20, 24 and 26; resistor Ri sets the droop current, Idroop. Tables 6 and 7 show the internal OCP threshold. It is recommended to design Idroop without using the Rcomp resistor. Substitution of Equation 27 into Equation 1 gives Equation 28: RSUM RSEN Overcurrent Protection 2R sen I o R i = --------------------------N I droop (EQ. 33) Substitution of Equation 33 and application of the OCP condition in Equation 29 gives Equation 34: 2R sen I omax R i = -------------------------------------N I droopmax (EQ. 34) where Iomax is the full load current and Idroopmax is the corresponding droop current. For example, given N = 3, Rsen = 1m, Iomax = 51A and Idroopmax = 40.9µA, Equation 34 gives Ri = 831. Page 26 of 33 ISL6267 Load Line Slope See Figure 15 for load-line implementation. For inductor DCR sensing, substitution of Equation 28 into Equation 2 gives the load-line slope expression in Equation 35: 2R droop R ntcnet V droop DCR LL = ------------------ = ---------------------- ----------------------------------------- -----------Io Ri R sum N R ntcnet + -------------N of the sensed output voltage, and then feeds it to the compensator. T1 is measured after the summing node, and T2 is measured in the voltage loop before the summing node. The spreadsheet gives both T1(s) and T2(s) plots. However, only T2(s) can actually be measured on an ISL6267 regulator. (EQ. 35) VO L Q1 For resistor sensing, substitution of Equation 32 into Equation 2 gives the load line slope expression in Equation 36: 2R sen R droop V droop LL = ------------------ = ----------------------------------------Io N Ri VIN Q2 GATE DRIVER iO COUT (EQ. 36) LOAD LINE SLOPE Substitution of Equation 29 and rewriting Equation 35, or substitution of Equation 33 and rewriting Equation 36, gives the same result as in Equation 37: Io R droop = ---------------- LL I droop + 20 EA MOD. + COMP VID + ISOLATION TRANSFORMER (EQ. 37) CHANNEL B LOOP GAIN = One can use the full-load condition to calculate Rdroop. For example, given Iomax = 51A, Idroopmax = 40.9µA and LL = 1.9m, Equation 37 gives Rdroop = 2.37k. It is recommended to start with the Rdroop value calculated by Equation 37 and fine-tune it on the actual board to get accurate load-line slope. One should record the output voltage readings at no load and at full load for load-line slope calculation. Reading the output voltage at lighter load instead of full load will increase the measurement error. Compensator Figure 21 shows the desired load transient response waveforms. Figure 27 shows the equivalent circuit of a voltage regulator (VR) with the droop function. A VR is equivalent to a voltage source (= VID) and output impedance Zout(s). If Zout(s) is equal to the load-line slope LL, i.e., a constant output impedance, then in the entire frequency range, Vo will have a square response when Io has a square change. CHANNEL A NETWORK ANALYZER i CHANNEL B EXCITATION OUTPUT FIGURE 28. LOOP GAIN T1(s) MEASUREMENT SET-UP T1(s) is the total loop gain of the voltage loop and the droop loop. It always has a higher crossover frequency than T2(s), therefore has a higher impact on system stability. T2(s) is the voltage loop gain with closed droop loop, thus having a higher impact on output voltage response. Design the compensator to get stable T1(s) and T2(s) with sufficient phase margin and an output impedance equal to or smaller than the load-line slope. L VO Q1 VIN Zout(s) = LL CHANNEL A GATE Q2 DRIVER IO CO o LOAD LINE SLOPE VID VR LOAD V o EA MOD. + COMP FIGURE 27. VOLTAGE REGULATOR EQUIVALENT CIRCUIT LOOP GAIN = FN7801 Rev 1.00 Jan 8, 2013 VID 20 ISOLATION TRANSFORMER CHANNEL B CHANNEL A Intersil provides a Microsoft Excel-based spreadsheet to help design the compensator and the current sensing network so that VR achieves constant output impedance as a stable system. A VR with active droop function is a dual-loop system consisting of a voltage loop and a droop loop, which is a current loop. However, neither loop alone is sufficient to describe the entire system. The spreadsheet shows two loop gain transfer functions, T1(s) and T2(s), that describe the entire system. Figure 28 conceptually shows T1(s) measurement set-up, and Figure 29 conceptually shows T2(s) measurement set-up. The VR senses the inductor current, multiplies it by a gain of the load-line slope, adds it on top + + - CHANNEL A NETWORK ANALYZER CHANNEL B EXCITATION OUTPUT FIGURE 29. LOOP GAIN T2(s) MEASUREMENT SET-UP Current Balancing Refer to Figures 16 through 20 for information on current balancing. The ISL6267 achieves current balancing through matching the ISEN pin voltages. Risen and Cisen form filters to remove the switching ripple of the phase node voltages. It is recommended to use a rather long RisenCisen time constant such Page 27 of 33 ISL6267 that the ISEN voltages have minimal ripple and represent the DC current flowing through the inductors. Recommended values are Rs = 10k and Cs = 0.22µF. NTC Thermal Monitors and VR_HOT Function The ISL6267 features three pins (NTC, NTC_NB, and VR_HOT) which are allow the IC to monitor board temperature and alert the AMD CPU of a thermal issue. Figure 30 shows the thermal monitor feature of the ISL6267. An NTC network is connected between the NTC and NTC_NB pins and GND. The controller drives a 60µA current source out of the NTC pin and the NTC_NB pin alternatively at 1kHz frequency with 50% duty cycle. The pulsed current flows through the respective NTC resistor network on the pins and creates a voltage that is compared to an over-temperature trip threshold. If the voltage on both NTC pins is higher than the over-temperature trip threshold, then VR_HOT is pulled up by an external resistor on the pin. +V + VNTC - 60µA RNTC VR_HOT R SW1 Rs MONITOR SW2 NTC_NB Rp + VNTC - RNTC 0.88V --------------- = 14.7k 60A (EQ. 38) The equivalent resistance at the typical reset threshold voltage of 0.92V required to change the state of VR_HOT back low, is defined in Equation 39 0.92V --------------- = 15.3k 60A (EQ. 39) The NTC thermistor value correlates this resistance change to the required temperature hysteresis. A standard 1% resistor is typically needed to meet the NTC pin threshold voltage. For example, a Panasonic NTC thermistor with B = 4700 has a resistance ratio of 0.03322 of its nominal value at +105°C and 0.03956 of its nominal value at +100°C. The required resistance of the NTC is defined in Equation 40. 15.3k – 14.7k ------------------------------------------------------- = 94.6k 0.03956 – 0.03322 NTC Rp typical over-temperature threshold voltage of 0.88V, to change the state of VR_HOT, is defined in Equation 38. SW1 SW2 INTERNAL TO ISL6267 Rs (EQ. 40) The closest, larger thermistor value for B = 4700 is 100k. The NTC thermistor part number is ERTJ1VV104. At +105°C, a 100k NTC resistance drops to (0.03322 x 100k) = 3.322k. With a 60µA current flowing out of the NTC pin, the voltage drop across the resistor is only (3.322k x 60µA) = 0.199V. This value is much lower than the threshold voltage of 0.88V. A standard resistor, 1% tolerance, added in series with the thermistor is required to raise the voltage on the pin. The resistance required to meet the trip threshold is calculated in Equation 41. 0.88V --------------- – 3.322k = 11.34k 60A (EQ. 41) The closest, standard 1% tolerance resistor is 11.3k. FIGURE 30. CIRCUITRY ASSOCIATED WITH THE THERMAL MONITOR FEATURE OF THE ISL6267 As the board temperature rises, the NTC thermistor resistance decreases and the voltage at the NTC pin drops. When the voltage on the NTC pin drops below the over-temperature trip threshold, then VR_HOT is pulled low. The VR_HOT signal is used to change the CPU operation and decrease power consumption. With the reduction in power consumption by the CPU, the board temperature decreases and the NTC thermistor voltage rises. Once the over-temperature threshold is tripped and VR_HOT is taken low, the over-temperature threshold changes to the reset level. The addition of hysteresis to the over-temperature threshold prevents nuisance trips. Once both pin voltages exceed the over-temperature reset threshold, the pull-down on VR_HOT is released. The signal changes state and the CPU resumes normal operation. The over-temperature threshold returns to the trip level. The NTC thermistor is placed in a hot spot on the board, typically near the upper MOSFET of channel 1 of the respective output. The standard resistor is placed next to the controller. Selection of the NTC components can vary depending on how the resistor network is configured. The equivalent resistance at the FN7801 Rev 1.00 Jan 8, 2013 Page 28 of 33 ISL6267 Layout Guidelines Table 9 shows layout considerations for the ISL6267 controller. Refer to the reference designators shown in Figure 31. TABLE 9. LAYOUT CONSIDERATIONS FOR THE ISL6267 CONTROLLER ISL6267 SYMBOL LAYOUT GUIDELINES BOTTOM PAD GND Create analog ground plane underneath the controller and the analog signal processing components. Do not let the power ground plane overlap with the analog ground plane. Avoid allowing noisy planes/traces (e.g., phase node) to crossover/overlap the analog plane. 1 FB2_NB 2 FB_NB 3 COMP_NB 4 VW_NB 5 PGOOD_NB 6, 7, 8 SVD, PWROK, SVC 9 ENABLE No special consideration. 10 PGOOD No special consideration. 11 VR_HOT No special consideration. 12 NTC Place the NTC thermistor (R46) close to the thermal source that is monitored to determine CPU VCORE thermal throttling. Usually it is placed close to Core VR phase-1 high-side MOSFET. 13 VW Place the capacitor (C4) across VW and COMP close to the controller. 14 COMP 15 FB 16 FB2 ISEN3 17 ISEN2 18 ISEN1 Place the compensator components (R25, R9, R24, C88, C51, C86, and C153) close to the controller. Place the capacitor (C85) across VW, and place COMP close to the controller. No special consideration. Use good signal integrity practices. Place the compensator components (R7, R10, R11, C3, C6, C11 and C5) in general proximity to the controller. Each ISEN pin has a capacitor (Cisen) decoupling it to VSUMN and then through another capacitor (Cvsumn) to GND. Place Cisen capacitors as close as possible to the controller and keep the following loops small: 1. Any ISEN pin to another ISEN pin 2. Any ISEN pin to GND The red traces in the following drawing show the loops to be minimized. Phase1 L3 Ro Risen ISEN3 Cisen Phase2 Vo L2 Ro Risen ISEN2 Cisen Phase3 Risen ISEN1 GND 19 VSEN 20 RTN FN7801 Rev 1.00 Jan 8, 2013 L1 Ro Vsumn Cisen Cvsumn Place the VSEN/RTN filter (C12, C13) close to the controller for good decoupling. Page 29 of 33 ISL6267 TABLE 9. LAYOUT CONSIDERATIONS FOR THE ISL6267 CONTROLLER (Continued) ISL6267 SYMBOL LAYOUT GUIDELINES 21 ISUMN 22 ISUMP Place the current sensing circuit in general proximity of the controller. Place capacitor Cn very close to the controller. Place the NTC thermistor next to VR1 phase-1 inductor (L1) so it senses the inductor temperature correctly. Each phase of the power stage sends a pair of VSUMP and VSUMN signals to the controller. Run these two signals traces in parallel fashion with decent width (>20mil). IMPORTANT: Sense the inductor current by routing the sensing circuit to the inductor pads. Route R63 and R71 to Core VR phase-1 side pad of inductor L1. Route R88 to the output side pad of inductor L1. Route R65 and R72 to Core VR phase-2 side pad of inductor L2. Route R90 to the output side pad of inductor L2. If possible, route the traces on a different layer from the inductor pad layer and use vias to connect the traces to the center of the pads. If no via is allowed on the pad, consider routing the traces into the pads from the inside of the inductor. The following drawings show the two preferred ways of routing current sensing traces. INDUCTOR INDUCTOR VIAS CURRENT-SENSING TRACES CURRENT-SENSING TRACES 23 VDD A capacitor (C16) decouples it to GND. Place it in close proximity to the controller. 24 VIN A capacitor (C17) decouples it to GND. Place it in close proximity to the controller. 25 PROG1 No special consideration. 26 BOOT1 Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 27 UGATE1 28 PHASE1 Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing PHASE1 trace to VR1 phase-1 high-side MOSFET (Q2 and Q8) source pins instead of general copper. 29 LGATE1 Use decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 30 PWM3 No special consideration. 31 VCCP A capacitor (C22) decouples it to GND. Place it in close proximity to the controller. 32 LGATE2 Use decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 33 PHASE2 34 UGATE2 Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing PHASE2 trace to VR1 phase-2 high-side MOSFET (Q4 and Q10) source pins instead of general copper. 35 BOOT2 Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 36 PWM2_NB No special consideration. 37 LGATE1_NB Use decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 38 PHASE1_NB 39 UGATE1_NB Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing PHASE1G trace to VR2 phase-1 high-side MOSFET source pins instead of general copper. 40 BOOT1_NB Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 41 PROG2 No special consideration. 42 NTC_NB Place the NTC thermistor close to the thermal source that is monitored to determine GT VCORE thermal throttling. Usually it is placed close to Northbridge VR phase-1 high-side MOSFET. 43 ISUMN_NB 44 ISUMP_NB FN7801 Rev 1.00 Jan 8, 2013 Place the current sensing circuit in general proximity to the controller. Place capacitor Cn very close to the controller. Place the NTC thermistor next to Northbridge VR phase-1 inductor (L1) so it senses the inductor temperature correctly. See ISUMN and ISUMP pins for layout guidelines of current-sensing trace routing. Page 30 of 33 ISL6267 TABLE 9. LAYOUT CONSIDERATIONS FOR THE ISL6267 CONTROLLER (Continued) ISL6267 SYMBOL 45 RTN_NB 46 VSEN_NB 47 ISEN2_NB 48 ISEN1_NB LAYOUT GUIDELINES Place the VSEN/RTN filter (C89, C90) in close proximity to the controller for good decoupling. See ISEN1, ISEN2 and ISEN3 pins for layout guidelines of current-balancing circuit trace routing. FIGURE 31. PORTION OF ISL6267EVAL1Z EVALUATION BOARD SCHEMATIC FN7801 Rev 1.00 Jan 8, 2013 Page 31 of 33 ISL6267 Revision History The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you have the latest revision. DATE REVISION CHANGE 1/31/11 FN7801.0 Initial Release. November 5, 2012 FN7801.1 Page 9: Added IRZ parts to ordering information table. About Intersil Intersil Corporation is a leader in the design and manufacture of high-performance analog, mixed-signal and power management semiconductors. The company's products address some of the fastest growing markets within the industrial and infrastructure, personal computing and high-end consumer markets. For more information about Intersil or to find out how to become a member of our winning team, visit our website and career page at www.intersil.com. For a complete listing of Applications, Related Documentation and Related Parts, please see the respective product information page. Also, please check the product information page to ensure that you have the most updated datasheet: ISL6267 To report errors or suggestions for this datasheet, please go to: www.intersil.com/askourstaff Reliability reports are available from our website at: http://rel.intersil.com/reports/search.php © Copyright Intersil Americas LLC 2011-2013. All Rights Reserved. All trademarks and registered trademarks are the property of their respective owners. For additional products, see www.intersil.com/en/products.html Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted in the quality certifications found at www.intersil.com/en/support/qualandreliability.html Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com FN7801 Rev 1.00 Jan 8, 2013 Page 32 of 33 ISL6267 Package Outline Drawing L48.6x6B 48 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 0, 9/09 4X 4.4 6.00 44X 0.40 A B 6 PIN 1 INDEX AREA 6 PIN #1 INDEX AREA 48 37 1 6.00 36 4 .40 ± 0.15 25 12 0.15 (4X) 13 24 0.10 M C A B 0.05 M C TOP VIEW 48X 0.45 ± 0.10 4 48X 0.20 BOTTOM VIEW SEE DETAIL "X" 0.10 C BASE PLANE MAX 1.00 ( SEATING PLANE 0.08 C ( 44 X 0 . 40 ) ( 5. 75 TYP ) C SIDE VIEW 4. 40 ) C 0 . 2 REF 5 ( 48X 0 . 20 ) ( 48X 0 . 65 ) 0 . 00 MIN. 0 . 05 MAX. DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 indentifier may be either a mold or mark feature. FN7801 Rev 1.00 Jan 8, 2013 Page 33 of 33