AD AD9740ARRL 10-bit, 210 msps txdac d/a converter Datasheet

10-Bit, 210 MSPS TxDAC® D/A Converter
AD9740
FEATURES
APPLICATIONS
High performance member of pin-compatible
TxDAC product family
Excellent spurious-free dynamic range performance
SNR @ 5 MHz output, 125 MSPS: 65 dB
Twos complement or straight binary data format
Differential current outputs: 2 mA to 20 mA
Power dissipation: 135 mW @ 3.3 V
Power-down mode: 15 mW @ 3.3 V
On-chip 1.2 V Reference
CMOS-compatible digital interface
28-lead SOIC, 28-lead TSSOP, and 32-lead LFCSP
packages
Edge-triggered latches
Wideband communication transmit channel
Direct IF
Base stations
Wireless local loops
Digital radio links
Direct digital synthesis (DDS)
Instrumentation
FUNCTIONAL BLOCK DIAGRAM
3.3V
RSET
3.3V
REFLO
1.2V REF
REFIO
FS ADJ
CURRENT
SOURCE
ARRAY
DVDD
DCOM
CLOCK
AVDD
150pF
SEGMENTED
SWITCHES
CLOCK
SLEEP
LSB
SWITCHES
ACOM
AD9740
IOUTA
IOUTB
LATCHES
DIGITAL DATA INPUTS (DB9–DB0)
MODE
02911-001
0.1μF
Figure 1.
GENERAL DESCRIPTION
The AD9740 1 is a 10-bit resolution, wideband, third generation
member of the TxDAC series of high performance, low power
CMOS digital-to-analog converters (DACs). The TxDAC
family, consisting of pin-compatible 8-, 10-, 12-, and 14-bit
DACs, is specifically optimized for the transmit signal path
of communication systems. All of the devices share the same
interface options, small outline package, and pinout, providing
an upward or downward component selection path based
on performance, resolution, and cost. The AD9740 offers
exceptional ac and dc performance while supporting update
rates up to 210 MSPS.
The AD9740’s low power dissipation makes it well suited for
portable and low power applications. Its power dissipation
can be further reduced to 60 mW with a slight degradation in
performance by lowering the full-scale current output. In
addition, a power-down mode reduces the standby power
dissipation to approximately 15 mW. A segmented current
source architecture is combined with a proprietary switching
technique to reduce spurious components and enhance
dynamic performance.
Edge-triggered input latches and a 1.2 V temperature-compensated
band gap reference have been integrated to provide a complete
monolithic DAC solution. The digital inputs support 3 V CMOS
logic families.
PRODUCT HIGHLIGHTS
1.
2.
3.
4.
5.
6.
1
The AD9740 is the 10-bit member of the pin-compatible
TxDAC family, which offers excellent INL and DNL
performance.
Data input supports twos complement or straight binary
data coding.
High speed, single-ended CMOS clock input supports
210 MSPS conversion rate.
Low power: Complete CMOS DAC function operates on
135 mW from a 2.7 V to 3.6 V single supply. The DAC fullscale current can be reduced for lower power operation,
and a sleep mode is provided for low power idle periods.
On-chip voltage reference: The AD9740 includes a 1.2 V
temperature-compensated band gap voltage reference.
Industry-standard 28-lead SOIC, 28-lead TSSOP, and 32lead LFCSP packages.
Protected by U.S. Patent Numbers 5568145, 5689257, and 5703519.
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
© 2005 Analog Devices, Inc. All rights reserved.
AD9740
TABLE OF CONTENTS
Features .............................................................................................. 1
DAC Transfer Function ............................................................. 14
Applications....................................................................................... 1
Analog Outputs .......................................................................... 14
Functional Block Diagram .............................................................. 1
Digital Inputs .............................................................................. 15
General Description ......................................................................... 1
Clock Input.................................................................................. 15
Product Highlights ........................................................................... 1
DAC Timing................................................................................ 16
Revision History ............................................................................... 3
Power Dissipation....................................................................... 16
Specifications..................................................................................... 4
Applying the AD9740 ................................................................ 17
DC Specifications ......................................................................... 4
Differential Coupling Using a Transformer............................... 17
Dynamic Specifications ............................................................... 5
Differential Coupling Using an Op Amp................................ 18
Digital Specifications ................................................................... 6
Single-Ended, Unbuffered Voltage Output............................. 18
Absolute Maximum Ratings............................................................ 7
Single-Ended, Buffered Voltage Output Configuration........ 18
Thermal Characteristics .............................................................. 7
Power and Grounding Considerations, Power Supply
Rejection...................................................................................... 19
ESD Caution.................................................................................. 7
Pin Configurations and Function Descriptions ........................... 8
Terminology ...................................................................................... 9
Typical Performance Characteristics ........................................... 10
Functional Description .................................................................. 13
Evaluation Board ............................................................................ 20
General Description................................................................... 20
Outline Dimensions ....................................................................... 30
Ordering Guide .......................................................................... 31
Reference Operation .................................................................. 13
Reference Control Amplifier .................................................... 14
Rev. B | Page 2 of 32
AD9740
REVISION HISTORY
12/05—Rev. A to Rev. B
Updated Format.................................................................. Universal
Changes to General Description and Product Highlights...........1
Changes to Table 1 ............................................................................4
Changes to Table 2 ............................................................................5
Changes to Table 5 ............................................................................8
Changes to Figure 6.........................................................................10
Inserted Figure 11; Renumbered Sequentially ............................10
Changes to Figure 12, Figure 13, Figure 14, and Figure 15 .......11
Changes to Functional Description and Reference
Operation Sections..........................................................................13
Inserted Figure 23; Renumbered Sequentially ............................13
Changes to DAC Transfer Function Section and Figure 25 ......14
Changes to Digital Inputs Section.................................................15
Changes to Figure 30 and Figure 31 .............................................17
Updated Outline Dimensions........................................................30
Changes to Ordering Guide...........................................................31
5/03—Rev. 0 to Rev. A
Added 32-Lead LFCSP Package ....................................... Universal
Edits to Features ................................................................................1
Edits to Product Highlights .............................................................1
Edits to DC Specifications ...............................................................2
Edits to Dynamic Specifications .....................................................3
Edits to Digital Specifications..........................................................4
Edits to Absolute Maximum Ratings..............................................5
Edits to Thermal Characteristics ....................................................5
Edits to Ordering Guide...................................................................5
Edits to Pin Configuration...............................................................6
Edits to Pin Function Descriptions ................................................6
Edits to Figure 2 ................................................................................7
Replaced TPCs 1, 4, 7, and 8............................................................8
Edits to Figure 3 ..............................................................................10
Edits to Functional Description Section ......................................10
Edits to Digital Inputs Section.......................................................12
Added Clock Input Section............................................................12
Added Figure 7 ................................................................................12
Edits to DAC Timing Section........................................................12
Edits to Sleep Mode Operation Section .......................................13
Edits to Power Dissipation Section...............................................13
Renumbered Figures 8 to 26..........................................................13
Added Figure 11 ..............................................................................13
Added Figures 27 to 35...................................................................21
Updated Outline Dimensions........................................................26
5/02—Revision 0: Initial Version
Rev. B | Page 3 of 32
AD9740
SPECIFICATIONS
DC SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, DVDD = 3.3 V, CLKVDD = 3.3 V, IOUTFS = 20 mA, unless otherwise noted.
Table 1.
Parameter
RESOLUTION
DC ACCURACY 1
Integral Linearity Error (INL)
Differential Nonlinearity (DNL)
ANALOG OUTPUT
Offset Error
Gain Error (Without Internal Reference)
Gain Error (With Internal Reference)
Full-Scale Output Current 2
Output Compliance Range
Output Resistance
Output Capacitance
REFERENCE OUTPUT
Reference Voltage
Reference Output Current 3
REFERENCE INPUT
Input Compliance Range
Reference Input Resistance (External Reference)
Small Signal Bandwidth
TEMPERATURE COEFFICIENTS
Offset Drift
Gain Drift (Without Internal Reference)
Gain Drift (With Internal Reference)
Reference Voltage Drift
POWER SUPPLY
Supply Voltages
AVDD
DVDD
CLKVDD
Analog Supply Current (IAVDD)
Digital Supply Current (IDVDD) 4
Clock Supply Current (ICLKVDD)
Supply Current Sleep Mode (IAVDD)
Power Dissipation4
Power Dissipation 5
Power Supply Rejection Ratio—AVDD 6
Power Supply Rejection Ratio—DVDD6
OPERATING RANGE
Min
10
Typ
Max
Unit
Bits
−0.7
−0.5
±0.15
±0.12
+0.7
+0.5
LSB
LSB
+0.02
+2
+2
20
+1.25
% of FSR
% of FSR
% of FSR
mA
V
kΩ
pF
1.26
V
nA
1.25
7
0.5
V
kΩ
MHz
0
±50
±100
±50
ppm of FSR/°C
ppm of FSR/°C
ppm of FSR/°C
ppm/°C
−0.02
−2
−2
2
−1
±0.1
±0.1
100
5
1.14
1.20
100
0.1
2.7
2.7
2.7
−1
−0.04
−40
1
3.3
3.3
3.3
33
8
5
5
135
145
3.6
3.6
3.6
36
9
6
6
145
+1
+0.04
+85
Measured at IOUTA, driving a virtual ground.
Nominal full-scale current, IOUTFS, is 32 times the IREF current.
3
An external buffer amplifier with input bias current <100 nA should be used to drive any external load.
4
Measured at fCLOCK = 25 MSPS and fOUT = 1 MHz.
5
Measured as unbuffered voltage output with IOUTFS = 20 mA, 50 Ω RLOAD at IOUTA and IOUTB, fCLOCK = 100 MSPS, and fOUT = 40 MHz.
6
±5% power supply variation.
2
Rev. B | Page 4 of 32
V
V
V
mA
mA
mA
mA
mW
mW
% of FSR/V
% of FSR/V
°C
AD9740
DYNAMIC SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, DVDD = 3.3 V, CLKVDD = 3.3 V, IOUTFS = 20 mA, differential transformer coupled output, 50 Ω doubly
terminated, unless otherwise noted.
Table 2.
Parameter
DYNAMIC PERFORMANCE
Maximum Output Update Rate (fCLOCK)
Output Settling Time (tST) (to 0.1%) 1
Output Propagation Delay (tPD)
Glitch Impulse
Output Rise Time (10% to 90%)1
Output Fall Time (10% to 90%)1
Output Noise (IOUTFS = 20 mA) 2
Output Noise (IOUTFS = 2 mA)2
Noise Spectral Density 3
AC LINEARITY
Spurious-Free Dynamic Range to Nyquist
fCLOCK = 25 MSPS; fOUT = 1.00 MHz
0 dBFS Output
−6 dBFS Output
−12 dBFS Output
−18 dBFS Output
fCLOCK = 65 MSPS; fOUT = 1.00 MHz
fCLOCK = 65 MSPS; fOUT = 2.51 MHz
fCLOCK = 65 MSPS; fOUT = 10 MHz
fCLOCK = 65 MSPS; fOUT = 15 MHz
fCLOCK = 65 MSPS; fOUT = 25 MHz
fCLOCK = 165 MSPS; fOUT = 21 MHz
fCLOCK = 165 MSPS; fOUT = 41 MHz
fCLOCK = 210 MSPS; fOUT = 40 MHz
fCLOCK = 210 MSPS; fOUT = 69 MHz
Spurious-Free Dynamic Range within a Window
fCLOCK = 25 MSPS; fOUT = 1.00 MHz; 2 MHz Span
fCLOCK = 50 MSPS; fOUT = 5.02 MHz; 2 MHz Span
fCLOCK = 65 MSPS; fOUT = 5.03 MHz; 2.5 MHz Span
fCLOCK = 125 MSPS; fOUT = 5.04 MHz; 4 MHz Span
Total Harmonic Distortion
fCLOCK = 25 MSPS; fOUT = 1.00 MHz
fCLOCK = 50 MSPS; fOUT = 2.00 MHz
fCLOCK = 65 MSPS; fOUT = 2.00 MHz
fCLOCK = 125 MSPS; fOUT = 2.00 MHz
Signal-to-Noise Ratio
fCLOCK = 65 MSPS; fOUT = 5 MHz; IOUTFS = 20 mA
fCLOCK = 65 MSPS; fOUT = 5 MHz; IOUTFS = 5 mA
fCLOCK = 125 MSPS; fOUT = 5 MHz; IOUTFS = 20 mA
fCLOCK = 125 MSPS; fOUT = 5 MHz; IOUTFS = 5 mA
fCLOCK = 165 MSPS; fOUT = 5 MHz; IOUTFS = 20 mA
fCLOCK = 165 MSPS; fOUT = 5 MHz; IOUTFS = 5 mA
fCLOCK = 210 MSPS; fOUT = 5 MHz; IOUTFS = 20 mA
fCLOCK = 210 MSPS; fOUT = 5 MHz; IOUTFS = 5 mA
Min
Typ
Max
210
71
11
1
5
2.5
2.5
50
30
−143
MSPS
ns
ns
pV-s
ns
ns
pA/√Hz
pA/√Hz
dBm/Hz
79
75
67
61
84
80
78
76
75
70
60
67
63
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
90
90
90
dBc
dBc
dBc
dBc
80
−79
−77
−77
−77
68
64
64
62
64
62
63
60
Rev. B | Page 5 of 32
Unit
−71
dBc
dBc
dBc
dBc
dB
dB
dB
dB
dB
dB
dB
dB
AD9740
Parameter
Multitone Power Ratio (8 Tones at 400 kHz Spacing)
fCLOCK = 78 MSPS; fOUT = 15.0 MHz to 18.2 MHz
0 dBFS Output
−6 dBFS Output
−12 dBFS Output
−18 dBFS Output
1
2
3
Min
Typ
Max
65
66
60
55
Unit
dBc
dBc
dBc
dBc
Measured single-ended into 50 Ω load.
Output noise is measured with a full-scale output set to 20 mA with no conversion activity. It is a measure of the thermal noise only.
Noise spectral density is the average noise power normalized to a 1 Hz bandwidth, with the DAC converting and producing an output tone.
DIGITAL SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, DVDD = 3.3 V, CLKVDD = 3.3 V, IOUTFS = 20 mA, unless otherwise noted.
Table 3.
Parameter
DIGITAL INPUTS 1
Logic 1 Voltage
Logic 0 Voltage
Logic 1 Current
Logic 0 Current
Input Capacitance
Input Setup Time (tS)
Input Hold Time (tH)
Latch Pulse Width (tLPW)
CLK INPUTS 2
Input Voltage Range
Common-Mode Voltage
Differential Voltage
2
Typ
2.1
3
0
−10
−10
Max
0.9
+10
+10
5
2.0
1.5
1.5
0
0.75
0.5
1.5
1.5
3
2.25
Includes CLOCK pin on SOIC/TSSOP packages and CLK+ pin on LFCSP package in single-ended clock input mode.
Applicable to CLK+ and CLK− inputs when configured for differential or PECL clock input mode.
DB0–DB9
tS
tH
CLOCK
tLPW
tPD
IOUTA
OR
IOUTB
tST
0.1%
Figure 2. Timing Diagram
Rev. B | Page 6 of 32
0.1%
02911-002
1
Min
Unit
V
V
μA
μA
pF
ns
ns
ns
V
V
V
AD9740
ABSOLUTE MAXIMUM RATINGS
Table 4.
Parameter
AVDD
DVDD
CLKVDD
ACOM
ACOM
DCOM
AVDD
AVDD
DVDD
CLOCK, SLEEP
Digital Inputs, MODE
IOUTA, IOUTB
REFIO, REFLO, FS ADJ
CLK+, CLK−, MODE
Junction
Temperature
Storage
Temperature
Range
Lead Temperature
(10 sec)
THERMAL CHARACTERISTICS 1
With
Respect to
ACOM
DCOM
CLKCOM
DCOM
CLKCOM
CLKCOM
DVDD
CLKVDD
CLKVDD
DCOM
DCOM
ACOM
ACOM
CLKCOM
Min
−0.3
−0.3
−0.3
−0.3
−0.3
−0.3
−3.9
−3.9
−3.9
−0.3
−0.3
−1.0
−0.3
−0.3
Max
+3.9
+3.9
+3.9
+0.3
+0.3
+0.3
+3.9
+3.9
+3.9
DVDD + 0.3
DVDD + 0.3
AVDD + 0.3
AVDD + 0.3
CLKVDD + 0.3
150
Unit
V
V
V
V
V
V
V
V
V
V
V
V
V
V
°C
−65
+150
°C
300
°C
Thermal Resistance
28-Lead 300-Mil SOIC
θJA = 55.9°C/W
28-Lead TSSOP
θJA = 67.7°C/W
32-Lead LFCSP
θJA = 32.5°C/W
1
Thermal impedance measurements were taken on a 4-layer board in still air,
in accordance with EIA/JESD51-7.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
sections of this specification is not implied. Exposure to
absolute maximum ratings for extended periods may effect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. B | Page 7 of 32
AD9740
27 DVDD
DB7 3
26 DCOM
DB6 4
25 MODE
DB5
24 AVDD
DB3 7
TOP VIEW
(Not to Scale)
DB2 8
DB3
DB2
DVDD
DB1
DB0
NC
NC
NC
23 RESERVED
22 IOUTA
21 IOUTB
DB1 9
20 ACOM
DB0 10
19 NC
1
2
3
4
5
6
7
8
PIN 1
INDICATOR
AD9740
TOP VIEW
(Not to Scale)
24
23
22
21
20
19
18
17
FS ADJ
REFIO
ACOM
IOUTA
IOUTB
ACOM
AVDD
AVDD
FS ADJ
NC 11
18
NC 12
17 REFIO
NC 13
16
REFLO
NC 14
15
SLEEP
NC = NO CONNECT
9
10
11
12
13
14
15
16
AD9740
NC
DCOM
CLKVDD
CLK+
CLK–
CLKCOM
CMODE
MODE
DB4 6
02911-003
5
32
31
30
29
28
27
26
25
28 CLOCK
DB8 2
NC = NO CONNECT
02911-004
(MSB) DB9 1
DB4
DB5
DB6
DB7
DB8
DB9 (MSB)
DCOM
SLEEP
PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
Figure 4. 32-Lead LFCSP Pin Configuration
Figure 3. 28-Lead SOIC and TSSOP Pin Configuration
Table 5. Pin Function Descriptions
SOIC/TSSOP
Pin No.
1
2 to 9
10
11 to 14, 19
15
LFCSP
Pin No.
27
28 to 32, 1, 2, 4
5
6 to 9
25
Mnemonic
DB9 (MSB)
DB8 to DB1
DB0 (LSB)
NC
SLEEP
16
N/A
REFLO
17
23
REFIO
18
20
21
22
23
24
25
N/A
24
19, 22
20
21
N/A
17, 18
16
15
FS ADJ
ACOM
IOUTB
IOUTA
RESERVED
AVDD
MODE
CMODE
26
27
28
N/A
N/A
N/A
N/A
10, 26
3
N/A
12
13
11
14
DCOM
DVDD
CLOCK
CLK+
CLK−
CLKVDD
CLKCOM
Description
Most Significant Data Bit (MSB).
Data Bits 8 to 1.
Least Significant Data Bit (LSB).
No Internal Connection.
Power-Down Control Input. Active high. Contains active pull-down circuit; it can be
left unterminated if not used.
Reference Ground when Internal 1.2 V Reference Used. Connect to ACOM for both
internal and external reference operation modes.
Reference Input/Output. Serves as reference input when using external reference.
Serves as 1.2 V reference output when using internal reference. Requires 0.1 μF capacitor
to ACOM when using internal reference.
Full-Scale Current Output Adjust.
Analog Common.
Complementary DAC Current Output. Full-scale current when all data bits are 0s.
DAC Current Output. Full-scale current when all data bits are 1s.
Reserved. Do Not Connect to Common or Supply.
Analog Supply Voltage (3.3 V).
Selects Input Data Format. Connect to DCOM for straight binary, DVDD for twos complement.
Clock Mode Selection. Connect to CLKCOM for single-ended clock receiver (drive CLK+
and float CLK–). Connect to CLKVDD for differential receiver. Float for PECL receiver
(terminations on-chip).
Digital Common.
Digital Supply Voltage (3.3 V).
Clock Input. Data latched on positive edge of clock.
Differential Clock Input.
Differential Clock Input.
Clock Supply Voltage (3.3 V).
Clock Common.
Rev. B | Page 8 of 32
AD9740
TERMINOLOGY
Linearity Error (Also Called Integral Nonlinearity or INL)
Linearity error is defined as the maximum deviation of the
actual analog output from the ideal output, determined by a
straight line drawn from zero to full scale.
Power Supply Rejection
The maximum change in the full-scale output as the supplies
are varied from nominal to minimum and maximum specified
voltages.
Differential Nonlinearity (or DNL)
Settling Time
The time required for the output to reach and remain within a
specified error band about its final value, measured from the
start of the output transition.
DNL is the measure of the variation in analog value, normalized
to full scale, associated with a 1 LSB change in digital input code.
Monotonicity
A DAC is monotonic if the output either increases or remains
constant as the digital input increases.
Glitch Impulse
Asymmetrical switching times in a DAC give rise to undesired
output transients that are quantified by a glitch impulse. It is
specified as the net area of the glitch in pV-s.
Offset Error
The deviation of the output current from the ideal of zero is
called the offset error. For IOUTA, 0 mA output is expected
when the inputs are all 0s. For IOUTB, 0 mA output is expected
when all inputs are set to 1s.
Spurious-Free Dynamic Range
The difference, in dB, between the rms amplitude of the output
signal and the peak spurious signal over the specified bandwidth.
Gain Error
The difference between the actual and ideal output span. The
actual span is determined by the output when all inputs are set
to 1s minus the output when all inputs are set to 0s.
Total Harmonic Distortion (THD)
THD is the ratio of the rms sum of the first six harmonic
components to the rms value of the measured input signal. It is
expressed as a percentage or in decibels (dB).
Output Compliance Range
The range of allowable voltage at the output of a current output
DAC. Operation beyond the maximum compliance limits can
cause either output stage saturation or breakdown, resulting in
nonlinear performance.
Multitone Power Ratio
The spurious-free dynamic range containing multiple carrier
tones of equal amplitude. It is measured as the difference
between the rms amplitude of a carrier tone to the peak
spurious signal in the region of a removed tone.
T
Temperature Drift
Temperature drift is specified as the maximum change from the
ambient (25°C) value to the value at either TMIN or TMAX. For
offset and gain drift, the drift is reported in ppm of full-scale
range (FSR) per °C. For reference drift, the drift is reported in
ppm per °C.
3.3V
REFLO
1.2V REF
REFIO
PMOS
CURRENT SOURCE
ARRAY
FS ADJ
RSET
2kΩ
3.3V
DVDD
DCOM
50Ω
RETIMED
CLOCK
OUTPUT*
LECROY 9210
PULSE GENERATOR
MINI-CIRCUITS
T1-1T
IOUTA
LSB
SWITCHES
SEGMENTED SWITCHES
FOR DB9–DB1
CLOCK
DVDD
DCOM
ACOM
AD9740
LATCHES
IOUTB
ROHDE & SCHWARZ
FSEA30
SPECTRUM
ANALYZER
MODE
50Ω
SLEEP
50Ω
CLOCK
OUTPUT
DIGITAL
DATA
TEKTRONIX AWG-2021
WITH OPTION 4
Figure 5. Basic AC Characterization Test Setup (SOIC/TSSOP Packages)
Rev. B | Page 9 of 32
*AWG2021 CLOCK RETIMED
SO THAT THE DIGITAL DATA
TRANSITIONS ON FALLING EDGE
OF 50% DUTY CYCLE CLOCK.
02911-005
0.1μF
AVDD
150pF
AD9740
TYPICAL PERFORMANCE CHARACTERISTICS
95
95
90
210MSPS (LFCSP)
125MSPS
90
0dBFS
85
85
165MSPS (LFCSP)
80
75
SFDR (dBc)
65MSPS
70
125MSPS (LFCSP)
65
60
–6dBFS
70
65
–12dBFS
60
210MSPS
55
55
165MSPS
50
10
100
fOUT (MHz)
45
0
10
95
90
90
85
85
SFDR (dBc)
75
–6dBFS
70
–12dBFS
55
50
50
15
20
25
5mA
65
55
fOUT (MHz)
10mA
70
60
10
45
0
10
15
20
25
80
Figure 10. SFDR vs. fOUT and IOUTFS @ 65 MSPS and 0 dBFS
95
95
90
90
0dBFS
0dBFS (LFCSP)
85
80
80
75
SFDR (dBc)
85
–6dBFS
70
–12dBFS
65
0dBFS
65
55
55
50
50
5
10
15
20
25
fOUT (MHz)
30
35
40
45
–6dBFS (LFCSP)
70
60
0
–12dBFS (LFCSP)
75
60
02911-008
SFDR (dBc)
5
fOUT (MHz)
Figure 7. SFDR vs. fOUT @ 65 MSPS
45
60
20mA
75
60
5
50
80
0dBFS
02911-007
SFDR (dBc)
80
0
40
Figure 9. SFDR vs. fOUT @ 165 MSPS
95
45
30
fOUT (MHz)
Figure 6. SFDR vs. fOUT @ 0 dBFS
65
20
02911-010
0
02911-009
50
02911-006
45
75
02911-054
SFDR (dBc)
80
45
–12dBFS
0
10
20
30
–6dBFS
40
50
60
fOUT (MHz)
Figure 11. SFDR vs. fOUT @ 210 MSPS
Figure 8. SFDR vs. fOUT @ 125 MSPS
Rev. B | Page 10 of 32
70
AD9740
95
95
90
125MSPS
125MSPS
85
65MSPS
165MSPS
80
65MSPS
75
SFDR (dBc)
SFDR (dBc)
85
165MSPS
210MSPS
(LFCSP)
70
210MSPS
65
75
210MSPS (29, 31)
210MSPS (29, 31) LFCSP
65
78MSPS
60
55
55
–20
–15
–10
AOUT (dBFS)
–5
0
45
–25
02911-011
45
–25
Figure 12. Single-Tone SFDR vs. AOUT @ fOUT = fCLOCK/11
–20
–15
–10
AOUT (dBFS)
–5
02911-014
50
0
Figure 15. Dual-Tone IMD vs. AOUT @ fOUT = fCLOCK/7
95
0.25
90
85
210MSPS (LFCSP)
75
ERROR (LSB)
SFDR (dBc)
80
0.15
65MSPS
125MSPS
70
65
0.05
–0.05
165MSPS
60
–0.15
210MSPS
55
–15
–10
AOUT (dBFS)
–5
0
–0.25
0
256
Figure 13. Single-Tone SFDR vs. AOUT @ fOUT = fCLOCK/5
512
CODE
768
1024
02911-015
–20
02911-012
45
–25
768
1024
02911-016
50
Figure 16. Typical INL
90
0.25
85
0.15
80
20mA (LFCSP)
70
65
5mA
10mA (LFCSP)
60
0.05
–0.05
–0.15
10mA
5mA (LFCSP)
55
50
0
30
60
90
120
fCLOCK (MSPS)
150
180
210
02911-013
SNR (dB)
ERROR (LSB)
20mA
75
–0.25
0
256
512
CODE
Figure 17. Typical DNL
Figure 14. SNR vs. fCLOCK and IOUTFS @ fOUT = 5 MHz and 0 dBFS
Rev. B | Page 11 of 32
AD9740
90
0
fCLOCK = 78MSPS
fOUT1 = 15.0MHz
fOUT2 = 15.4MHz
–10
85
–20
SFDR = 77dBc
AMPLITUDE = 0dBFS
80
MAGNITUDE (dBm)
SFDR (dBc)
4MHz
75
70
19MHz
65
–30
34MHz
–40
–50
–60
–70
60
–80
49MHz
55
0
20
40
TEMPERATURE (°C)
60
80
–100
1
6
Figure 18. SFDR vs. Temperature @ 165 MSPS, 0 dBFS
26
31
36
36
0
fCLOCK = 78MSPS
fOUT = 15.0MHz
–10
–20
–30
MAGNITUDE (dBm)
–30
–40
–50
–60
–70
–50
–60
–70
–80
–90
–90
11
16
21
FREQUENCY (MHz)
26
31
36
–100
02911-018
6
SFDR = 72dBc
AMPLITUDE = 0dBFS
–40
–80
1
fCLOCK = 78MSPS
fOUT1 = 15.0MHz
fOUT2 = 15.4MHz
fOUT3 = 15.8MHz
fOUT4 = 16.2MHz
–10
SFDR = 77dBc
AMPLITUDE = 0dBFS
–20
1
6
Figure 19. Single-Tone SFDR
11
16
21
FREQUENCY (MHz)
26
31
Figure 21. Four-Tone SFDR
3.3V
REFLO
AVDD
150pF
1.2V REF
VREFIO
REFIO
IREF
0.1μF
RSET
2kΩ
3.3V
FS ADJ
AD9740
PMOS
CURRENT SOURCE
ARRAY
DVDD
DCOM
CLOCK
ACOM
CLOCK
SEGMENTED SWITCHES
FOR DB9–DB1
LSB
SWITCHES
VDIFF = VOUTA – VOUTB
IOUTA
IOUTB
LATCHES
SLEEP
DIGITAL DATA INPUTS (DB9–DB0)
Figure 22. Simplified Block Diagram (SOIC/TSSOP Packages)
Rev. B | Page 12 of 32
IOUTA
IOUTB
MODE
VOUTA
VOUTB
RLOAD
50Ω
RLOAD
50Ω
02911-021
MAGNITUDE (dBm)
16
21
FREQUENCY (MHz)
Figure 20. Dual-Tone SFDR
0
–100
11
02911-019
–20
02911-017
50
–40
02911-020
–90
AD9740
FUNCTIONAL DESCRIPTION
The analog and digital sections of the AD9740 have separate
power supply inputs (that is, AVDD and DVDD) that can
operate independently over a 2.7 V to 3.6 V range. The digital
section, which is capable of operating at a clock rate of up to
210 MSPS, consists of edge-triggered latches and segment
decoding logic circuitry. The analog section includes the PMOS
current sources, the associated differential switches, a 1.2 V
band gap voltage reference, and a reference control amplifier.
The DAC full-scale output current is regulated by the reference
control amplifier and can be set from 2 mA to 20 mA via an
external resistor, RSET, connected to the full-scale adjust
(FS ADJ) pin. The external resistor, in combination with both
the reference control amplifier and voltage reference, VREFIO, sets
the reference current, IREF, which is replicated to the segmented
current sources with the proper scaling factor. The full-scale
current, IOUTFS, is 32 times IREF.
The AD9740 contains an internal 1.2 V band gap reference. The
internal reference cannot be disabled, but can be easily overridden
by an external reference with no effect on performance. Figure 23
shows an equivalent circuit of the band gap reference. REFIO
serves as either an output or an input depending on whether
the internal or an external reference is used. To use the internal
reference, simply decouple the REFIO pin to ACOM with a
0.1 μF capacitor and connect REFLO to ACOM via a resistance
less than 5 Ω. The internal reference voltage is present at
REFIO. If the voltage at REFIO is to be used anywhere else in
the circuit, then an external buffer amplifier with an input bias
current of less than 100 nA should be used. An example of the
use of the internal reference is shown in Figure 24.
AVDD
84µA
REFIO
7kΩ
REFLO
Figure 23. Equivalent Circuit of Internal Reference
3.3V
OPTIONAL
EXTERNAL
REF BUFFER
REFLO
150pF
AVDD
1.2V REF
REFIO
ADDITIONAL
LOAD
0.1μF
2kΩ
FS ADJ
CURRENT
SOURCE
ARRAY
AD9740
Figure 24. Internal Reference Configuration
An external reference can be applied to REFIO, as shown in
Figure 25. The external reference can provide either a fixed
reference voltage to enhance accuracy and drift performance
or a varying reference voltage for gain control. Note that the
0.1 μF compensation capacitor is not required because the
internal reference is overridden, and the relatively high input
impedance of REFIO minimizes any loading of the external
reference.
Rev. B | Page 13 of 32
02911-022
All of these current sources are switched to one or the other of
the two output nodes (that is, IOUTA or IOUTB) via PMOS
differential current switches. The switches are based on the
architecture that was pioneered in the AD9764 family, with
further refinements to reduce distortion contributed by the
switching transient. This switch architecture also reduces
various timing errors and provides matching complementary
drive signals to the inputs of the differential current switches.
REFERENCE OPERATION
02911-057
Figure 22 shows a simplified block diagram of the AD9740. The
AD9740 consists of a DAC, digital control logic, and full-scale
output current control. The DAC contains a PMOS current
source array capable of providing up to 20 mA of full-scale
current (IOUTFS). The array is divided into 31 equal currents that
make up the five most significant bits (MSBs). The next four
bits, or middle bits, consist of 15 equal current sources whose
value is 1/16 of an MSB current source. The remaining LSBs are
binary weighted fractions of the middle bits current sources.
Implementing the middle and lower bits with current sources,
instead of an R-2R ladder, enhances its dynamic performance
for multitone or low amplitude signals and helps maintain the
DAC’s high output impedance (that is, >100 kΩ).
AD9740
3.3V
REFLO
150pF
The two current outputs typically drive a resistive load directly
or via a transformer. If dc coupling is required, then IOUTA
and IOUTB should be directly connected to matching resistive
loads, RLOAD, that are tied to analog common, ACOM. Note that
RLOAD can represent the equivalent load resistance seen by
IOUTA or IOUTB, as would be the case in a doubly terminated
50 Ω or 75 Ω cable. The single-ended voltage output appearing
at the IOUTA and IOUTB nodes is simply
AVDD
1.2V REF
REFIO
AD9740
REFERENCE
CONTROL
AMPLIFIER
02911-023
CURRENT
SOURCE
ARRAY
FS ADJ
Figure 25. External Reference Configuration
REFERENCE CONTROL AMPLIFIER
The AD9740 contains a control amplifier that is used to regulate
the full-scale output current, IOUTFS. The control amplifier is
configured as a V-I converter, as shown in Figure 24, so that its
current output, IREF, is determined by the ratio of the VREFIO and
an external resistor, RSET, as stated in Equation 4. IREF is copied
to the segmented current sources with the proper scale factor to
set IOUTFS, as stated in Equation 3.
The control amplifier allows a wide (10:1) adjustment span of
IOUTFS over a 2 mA to 20 mA range by setting IREF between
62.5 μA and 625 μA. The wide adjustment span of IOUTFS
provides several benefits. The first relates directly to the power
dissipation of the AD9740, which is proportional to IOUTFS (see
the Power Dissipation section). The second relates to a 20 dB
adjustment, which is useful for system gain control purposes.
The small signal bandwidth of the reference control amplifier is
approximately 500 kHz and can be used for low frequency small
signal multiplying applications.
DAC TRANSFER FUNCTION
The AD9740 provides complementary current outputs, IOUTA
and IOUTB. IOUTA provides a near full-scale current output,
IOUTFS, when all bits are high (that is, DAC CODE = 1023), while
IOUTB, the complementary output, provides no current. The
current output appearing at IOUTA and IOUTB is a function of
both the input code and IOUTFS and can be expressed as:
IOUTA = (DAC CODE/1023) × IOUTFS
(1)
IOUTB = (1023 − DAC CODE)/1024 × IOUTFS
(2)
where DAC CODE = 0 to 1023 (that is, decimal representation).
As mentioned previously, IOUTFS is a function of the reference
current IREF, which is nominally set by a reference voltage,
VREFIO, and external resistor, RSET. It can be expressed as:
IOUTFS = 32 × IREF
(3)
where
IREF = VREFIO/RSET
(4)
VOUTA = IOUTA × RLOAD
(5)
VOUTB = IOUTB × RLOAD
(6)
Note that the full-scale value of VOUTA and VOUTB should not
exceed the specified output compliance range to maintain
specified distortion and linearity performance.
VDIFF = (IOUTA − IOUTB) × RLOAD
(7)
Substituting the values of IOUTA, IOUTB, IREF, and VDIFF can be
expressed as:
VDIFF = {(2 × DAC CODE − 1023)/1024}
(32 × RLOAD/RSET) × VREFIO
(8)
Equation 7 and Equation 8 highlight some of the advantages of
operating the AD9740 differentially. First, the differential
operation helps cancel common-mode error sources associated
with IOUTA and IOUTB, such as noise, distortion, and dc
offsets. Second, the differential code-dependent current and
subsequent voltage, VDIFF, is twice the value of the single-ended
voltage output (that is, VOUTA or VOUTB), thus providing twice the
signal power to the load.
Note that the gain drift temperature performance for a singleended (VOUTA and VOUTB) or differential output (VDIFF) of the
AD9740 can be enhanced by selecting temperature tracking
resistors for RLOAD and RSET due to their ratiometric relationship,
as shown in Equation 8.
B
ANALOG OUTPUTS
The complementary current outputs in each DAC, IOUTA,
and IOUTB can be configured for single-ended or differential
operation. IOUTA and IOUTB can be converted into
complementary single-ended voltage outputs, VOUTA and VOUTB,
via a load resistor, RLOAD, as described in the DAC Transfer
Function section by Equation 5 through Equation 8. The
differential voltage, VDIFF, existing between VOUTA and VOUTB, can
also be converted to a single-ended voltage via a transformer or
differential amplifier configuration. The ac performance of the
AD9740 is optimum and specified using a differential
transformer-coupled output in which the voltage swing at
IOUTA and IOUTB is limited to ±0.5 V.
The distortion and noise performance of the AD9740 can be
enhanced when it is configured for differential operation. The
common-mode error sources of both IOUTA and IOUTB can
be significantly reduced by the common-mode rejection of a
Rev. B | Page 14 of 32
AD9740
DVDD
transformer or differential amplifier. These common-mode
error sources include even-order distortion products and noise.
The enhancement in distortion performance becomes more
significant as the frequency content of the reconstructed
waveform increases and/or its amplitude decreases. This is due
to the first-order cancellation of various dynamic commonmode distortion mechanisms, digital feedthrough, and noise.
Performing a differential-to-single-ended conversion via a
transformer also provides the ability to deliver twice the
reconstructed signal power to the load (assuming no source
termination). Because the output currents of IOUTA and
IOUTB are complementary, they become additive when
processed differentially. A properly selected transformer allows
the AD9740 to provide the required power and voltage levels to
different loads.
The output impedance of IOUTA and IOUTB is determined by
the equivalent parallel combination of the PMOS switches
associated with the current sources and is typically 100 kΩ in
parallel with 5 pF. It is also slightly dependent on the output
voltage (that is, VOUTA and VOUTB) due to the nature of a PMOS
device. As a result, maintaining IOUTA and/or IOUTB at a
virtual ground via an I-V op amp configuration results in the
optimum dc linearity. Note that the INL/DNL specifications for
the AD9740 are measured with IOUTA maintained at a virtual
ground via an op amp.
IOUTA and IOUTB also have a negative and positive voltage
compliance range that must be adhered to in order to achieve
optimum performance. The negative output compliance range
of −1 V is set by the breakdown limits of the CMOS process.
Operation beyond this maximum limit can result in a
breakdown of the output stage and affect the reliability of the
AD9740.
The positive output compliance range is slightly dependent on
the full-scale output current, IOUTFS. It degrades slightly from its
nominal 1.2 V for an IOUTFS = 20 mA to 1 V for an IOUTFS = 2 mA.
The optimum distortion performance for a single-ended or
differential output is achieved when the maximum full-scale
signal at IOUTA and IOUTB does not exceed 0.5 V.
DIGITAL INPUTS
The AD9740 digital section consists of 10 input bit channels
and a clock input. The 10-bit parallel data inputs follow
standard positive binary coding, where DB9 is the most
significant bit (MSB) and DB0 is the least significant bit (LSB).
IOUTA produces a full-scale output current when all data bits
are at Logic 1. IOUTB produces a complementary output with
the full-scale current split between the two outputs as a
function of the input code.
02911-024
DIGITAL
INPUT
Figure 26. Equivalent Digital Input
The digital interface is implemented using an edge-triggered
master/slave latch. The DAC output updates on the rising edge
of the clock and is designed to support a clock rate as high as
210 MSPS. The clock can be operated at any duty cycle that
meets the specified latch pulse width. The setup and hold times
can also be varied within the clock cycle as long as the specified
minimum times are met, although the location of these transition
edges can affect digital feedthrough and distortion performance.
Best performance is typically achieved when the input data
transitions on the falling edge of a 50% duty cycle clock.
CLOCK INPUT
SOIC/TSSOP Packages
The 28-lead package options have a single-ended clock input
(CLOCK) that must be driven to rail-to-rail CMOS levels. The
quality of the DAC output is directly related to the clock quality,
and jitter is a key concern. Any noise or jitter in the clock
translates directly into the DAC output. Optimal performance is
achieved if the CLOCK input has a sharp rising edge, because
the DAC latches are positive edge triggered.
LFCSP Package
A configurable clock input is available in the LFCSP package,
which allows for one single-ended and two differential modes.
The mode selection is controlled by the CMODE input, as
summarized in Table 6. Connecting CMODE to CLKCOM
selects the single-ended clock input. In this mode, the CLK+
input is driven with rail-to-rail swings and the CLK− input is
left floating. If CMODE is connected to CLKVDD, then the
differential receiver mode is selected. In this mode, both inputs
are high impedance. The final mode is selected by floating
CMODE. This mode is also differential, but internal
terminations for positive emitter-coupled logic (PECL) are
activated. There is no significant performance difference
between any of the three clock input modes.
Table 6. Clock Mode Selection
CMODE Pin
CLKCOM
CLKVDD
Float
Clock Input Mode
Single-ended
Differential
PECL
The single-ended input mode operates in the same way as the
clock input in the 28-lead packages, as described previously.
Rev. B | Page 15 of 32
AD9740
75
70
65
55
50MHz SFDR
The final clock mode allows for a reduced external component
count when the DAC clock is distributed on the board using
PECL logic. The internal termination configuration is shown in
Figure 27. These termination resistors are untrimmed and can
vary up to ±20%. However, matching between the resistors
should generally be better than ±1%.
50
45
40
0
1
2
Sleep Mode Operation
50Ω
VTT = 1.3V NOM
3
Figure 28. SFDR vs. Clock Placement @
fOUT = 20 MHz and 50 MHz (fCLOCK = 165 MSPS)
TO DAC CORE
02911-025
50Ω
–1
ns
CLK+
CLK–
–2
02911-026
50MHz SFDR
35
–3
AD9740
CLOCK
RECEIVER
20MHz SFDR
60
dB
In the differential input mode, the clock input functions as a
high impedance differential pair. The common-mode level of
the CLK+ and CLK− inputs can vary from 0.75 V to 2.25 V, and
the differential voltage can be as low as 0.5 V p-p. This mode
can be used to drive the clock with a differential sine wave
because the high gain bandwidth of the differential inputs
converts the sine wave into a single-ended square wave internally.
Figure 27. Clock Termination in PECL Mode
DAC TIMING
Input Clock and Data Timing Relationship
Dynamic performance in a DAC is dependent on the
relationship between the position of the clock edges and the
time at which the input data changes. The AD9740 is rising
edge triggered, and so exhibits dynamic performance sensitivity
when the data transition is close to this edge. In general, the
goal when applying the AD9740 is to make the data transition
close to the falling clock edge. This becomes more important as
the sample rate increases. Figure 28 shows the relationship of
SFDR to clock placement with different sample rates. Note that
at the lower sample rates, more tolerance is allowed in clock
placement, while at higher rates, more care must be taken.
The AD9740 has a power-down function that turns off the output
current and reduces the supply current to less than 6 mA over the
specified supply range of 2.7 V to 3.6 V and the temperature range.
This mode can be activated by applying a Logic Level 1 to the
SLEEP pin. The SLEEP pin logic threshold is equal to 0.5 Ω
AVDD. This digital input also contains an active pull-down
circuit that ensures that the AD9740 remains enabled if this
input is left disconnected. The AD9740 takes less than 50 ns
to power down and approximately 5 μs to power back up.
POWER DISSIPATION
The power dissipation, PD, of the AD9740 is dependent on
several factors that include:
•
•
•
•
The power supply voltages (AVDD, CLKVDD, and
DVDD)
The full-scale current output (IOUTFS)
The update rate (fCLOCK)
The reconstructed digital input waveform
The power dissipation is directly proportional to the analog
supply current, IAVDD, and the digital supply current, IDVDD. IAVDD
is directly proportional to IOUTFS, as shown in Figure 29, and is
insensitive to fCLOCK. Conversely, IDVDD is dependent on both the
digital input waveform, fCLOCK, and digital supply DVDD. Figure 30
shows IDVDD as a function of full-scale sine wave output ratios
(fOUT/fCLOCK) for various update rates with DVDD = 3.3 V.
Rev. B | Page 16 of 32
AD9740
35
APPLYING THE AD9740
Output Configurations
30
The following sections illustrate some typical output
configurations for the AD9740. Unless otherwise noted, it is
assumed that IOUTFS is set to a nominal 20 mA. For applications
requiring the optimum dynamic performance, a differential
output configuration is suggested. A differential output
configuration can consist of either an RF transformer or a
differential op amp configuration. The transformer
configuration provides the optimum high frequency
performance and is recommended for any application that
allows ac coupling. The differential op amp configuration is
suitable for applications requiring dc coupling, bipolar output,
signal gain, and/or level shifting within the bandwidth of the
chosen op amp.
IAVDD (mA)
25
20
15
0
2
4
6
8
10
12
IOUTFS (mA)
14
16
18
02911-027
10
20
Figure 29. IAVDD vs. IOUTFS
20
A single-ended output is suitable for applications requiring a
unipolar voltage output. A positive unipolar output voltage
results if IOUTA and/or IOUTB is connected to an
appropriately sized load resistor, RLOAD, referred to ACOM.
This configuration can be more suitable for a single-supply
system requiring a dc-coupled, ground referred output voltage.
Alternatively, an amplifier could be configured as an I-V
converter, thus converting IOUTA or IOUTB into a negative
unipolar voltage. This configuration provides the best dc linearity
because IOUTA or IOUTB is maintained at a virtual ground.
18
210MSPS
16
IDVDD (mA)
14
165MSPS
12
10
125MSPS
8
6
65MSPS
4
2
1
An RF transformer can be used to perform a differential-tosingle-ended signal conversion, as shown in Figure 32. A
differentially coupled transformer output provides the
optimum distortion performance for output signals whose
spectral content lies within the transformer’s pass band. An
RF transformer, such as the Mini-Circuits® T1–1T, provides
excellent rejection of common-mode distortion (that is, evenorder harmonics) and noise over a wide frequency range. It also
provides electrical isolation and the ability to deliver twice the
power to the load. Transformers with different impedance ratios
can also be used for impedance matching purposes. Note that
the transformer provides ac coupling only.
Figure 30. IDVDD vs. Ratio @ DVDD = 3.3 V
11
10
9
DIFF
7
6
PECL
5
SE
4
3
2
IOUTA 22
1
0
0
50
100
150
200
fCLOCK (MSPS)
250
02911-056
ICLKVDD (mA)
8
MINI-CIRCUITS
T1-1T
AD9740
RLOAD
IOUTB 21
Figure 31. ICLKVDD vs. fCLOCK and Clock Mode
OPTIONAL RDIFF
Figure 32. Differential Output Using a Transformer
Rev. B | Page 17 of 32
02911-030
0.1
RATIO (fOUT/fCLOCK)
02911-055
DIFFERENTIAL COUPLING USING A TRANSFORMER
0
0.01
AD9740
500Ω
AD9740
An op amp can also be used to perform a differential-to-singleended conversion, as shown in Figure 33. The AD9740 is
configured with two equal load resistors, RLOAD, of 25 Ω. The
differential voltage developed across IOUTA and IOUTB is
converted to a single-ended signal via the differential op amp
configuration. An optional capacitor can be installed across
IOUTA and IOUTB, forming a real pole in a low-pass filter. The
addition of this capacitor also enhances the op amp’s distortion
performance by preventing the DAC’s high slewing output from
overloading the op amp’s input.
500Ω
25Ω
225Ω
IOUTB 21
25Ω
AVDD
1kΩ
SINGLE-ENDED, UNBUFFERED VOLTAGE OUTPUT
Figure 35 shows the AD9740 configured to provide a unipolar
output range of approximately 0 V to 0.5 V for a doubly
terminated 50 Ω cable because the nominal full-scale current,
IOUTFS, of 20 mA flows through the equivalent RLOAD of 25 Ω.
In this case, RLOAD represents the equivalent load resistance seen
by IOUTA or IOUTB. The unused output (IOUTA or IOUTB)
can be connected to ACOM directly or via a matching RLOAD.
Different values of IOUTFS and RLOAD can be selected as long as
the positive compliance range is adhered to. One additional
consideration in this mode is the integral nonlinearity (INL),
discussed in the Analog Outputs section. For optimum INL
performance, the single-ended, buffered voltage output
configuration is suggested.
AD9740
IOUTFS = 20mA
VOUTA = 0V TO 0.5V
IOUTA 22
50Ω
AD8047
50Ω
IOUTB 21
25Ω
COPT
25Ω
02911-031
500Ω
25Ω
1kΩ
Figure 34. Single-Supply DC Differential Coupled Circuit
225Ω
IOUTA 22
COPT
Figure 33. DC Differential Coupling Using an Op Amp
The common-mode rejection of this configuration is typically
determined by the resistor matching. In this circuit, the
differential op amp circuit using the AD8047 is configured to
provide some additional signal gain. The op amp must operate
off a dual supply because its output is approximately ±1 V. A
high speed amplifier capable of preserving the differential
performance of the AD9740 while meeting other system level
objectives (that is, cost or power) should be selected. The op
amp’s differential gain, gain setting resistor values, and full-scale
output swing capabilities should all be considered when
optimizing this circuit.
The differential circuit shown in Figure 34 provides the
necessary level shifting required in a single-supply system. In
this case, AVDD, which is the positive analog supply for both
the AD9740 and the op amp, is also used to level shift the
differential output of the AD9740 to midsupply (that is,
AVDD/2). The AD8041 is a suitable op amp for this application.
02911-033
AD9740
AD8041
225Ω
IOUTB 21
B
DIFFERENTIAL COUPLING USING AN OP AMP
225Ω
IOUTA 22
02911-032
The center tap on the primary side of the transformer must be
connected to ACOM to provide the necessary dc current path
for both IOUTA and IOUTB. The complementary voltages
appearing at IOUTA and IOUTB (that is, VOUTA and VOUTB)
swing symmetrically around ACOM and should be maintained
with the specified output compliance range of the AD9740. A
differential resistor, RDIFF, can be inserted in applications where
the output of the transformer is connected to the load, RLOAD,
via a passive reconstruction filter or cable. RDIFF is determined
by the transformer’s impedance ratio and provides the proper
source termination that results in a low VSWR. Note that
approximately half the signal power is dissipated across RDIFF.
Figure 35. 0 V to 0.5 V Unbuffered Voltage Output
SINGLE-ENDED, BUFFERED VOLTAGE OUTPUT
CONFIGURATION
Figure 36 shows a buffered single-ended output configuration
in which the op amp U1 performs an I-V conversion on the
AD9740 output current. U1 maintains IOUTA (or IOUTB) at a
virtual ground, minimizing the nonlinear output impedance
effect on the DAC’s INL performance as described in the Analog
Outputs section. Although this single-ended configuration
typically provides the best dc linearity performance, its ac
distortion performance at higher DAC update rates can be
limited by U1’s slew rate capabilities. U1 provides a negative
unipolar output voltage, and its full-scale output voltage is
simply the product of RFB and IOUTFS. The full-scale output
should be set within U1’s voltage output swing capabilities by
scaling IOUTFS and/or RFB. An improvement in ac distortion
performance can result with a reduced IOUTFS because U1 is
required to sink less signal current.
Rev. B | Page 18 of 32
AD9740
COPT
RFB
200Ω
IOUTFS = 10mA
AD9740
IOUTA 22
U1
VOUT = IOUTFS × RFB
IOUTB 21
02911-034
200Ω
Figure 36. Unipolar Buffered Voltage Output
POWER AND GROUNDING CONSIDERATIONS,
POWER SUPPLY REJECTION
Many applications seek high speed and high performance
under less than ideal operating conditions. In these application
circuits, the implementation and construction of the printed
circuit board is as important as the circuit design. Proper RF
techniques must be used for device selection, placement, and
routing as well as power supply bypassing and grounding to
ensure optimum performance. Figure 41 to Figure 44 illustrate
the recommended printed circuit board ground, power, and
signal plane layouts implemented on the AD9740 evaluation
board.
One factor that can measurably affect system performance is
the ability of the DAC output to reject dc variations or ac noise
superimposed on the analog or digital dc power distribution.
This is referred to as the power supply rejection ratio (PSRR).
For dc variations of the power supply, the resulting performance
of the DAC directly corresponds to a gain error associated with
the DAC’s full-scale current, IOUTFS. AC noise on the dc supplies
is common in applications where the power distribution is
generated by a switching power supply. Typically, switching
power supply noise occurs over the spectrum from tens of
kilohertz to several megahertz. The PSRR vs. frequency of the
AD9740 AVDD supply over this frequency range is shown in
Figure 37.
85
As a result, the PSRR measurement in Figure 37 represents a
worst-case condition in which the digital inputs remain static
and the full-scale output current of 20 mA is directed to the
DAC output being measured.
The following illustrates the effect of supply noise on the analog
supply. Suppose a switching regulator with a switching frequency
of 250 kHz produces 10 mV of noise and, for simplicity’s sake
(ignoring harmonics), all of this noise is concentrated at 250 kHz.
To calculate how much of this undesired noise appears as current
noise superimposed on the DAC’s full-scale current, IOUTFS, users
must determine the PSRR in dB using Figure 37 at 250 kHz. To
calculate the PSRR for a given RLOAD, such that the units of PSRR
are converted from A/V to V/V, adjust the curve in Figure 37 by
the scaling factor 20 Ω log (RLOAD). For instance, if RLOAD is 50 Ω,
then the PSRR is reduced by 34 dB (that is, PSRR of the DAC at
250 kHz, which is 85 dB in Figure 37, becomes 51 dB VOUT/VIN).
Proper grounding and decoupling should be a primary
objective in any high speed, high resolution system. The
AD9740 features separate analog and digital supplies and
ground pins to optimize the management of analog and digital
ground currents in a system. In general, AVDD, the analog
supply, should be decoupled to ACOM, the analog common, as
close to the chip as physically possible. Similarly, DVDD, the
digital supply, should be decoupled to DCOM as close to the
chip as physically possible.
For those applications that require a single 3.3 V supply for both
the analog and digital supplies, a clean analog supply can be
generated using the circuit shown in Figure 38. The circuit
consists of a differential LC filter with separate power supply
and return lines. Lower noise can be attained by using low ESR
type electrolytic and tantalum capacitors.
80
75
70
65
60
FERRITE
BEADS
55
TTL/CMOS
LOGIC
CIRCUITS
50
45
10μF–22μF
TANT.
0.1μF
CER.
ACOM
0
2
4
6
8
FREQUENCY (MHz)
10
12
3.3V
POWER SUPPLY
Figure 37. Power Supply Rejection Ratio (PSRR)
Figure 38. Differential LC Filter for Single 3.3 V Applications
Rev. B | Page 19 of 32
02911-036
40
AVDD
100μF
ELECT.
02911-035
PSRR (dB)
Note that the ratio in Figure 37 is calculated as amps out/volts
in. Noise on the analog power supply has the effect of modulating
the internal switches, and therefore the output current. The
voltage noise on AVDD, therefore, is added in a nonlinear
manner to the desired IOUT. Due to the relative different size of
these switches, the PSRR is very code dependent. This can produce
a mixing effect that can modulate low frequency power supply
noise to higher frequencies. Worst-case PSRR for either one of
the differential DAC outputs occur when the full-scale current
is directed toward that output.
AD9740
EVALUATION BOARD
GENERAL DESCRIPTION
The TxDAC family evaluation boards allow for easy setup and
testing of any TxDAC product in the SOIC and LFCSP packages.
Careful attention to layout and circuit design, combined with a
prototyping area, allows the user to evaluate the AD9740 easily
and effectively in any application where high resolution, high
speed conversion is required.
This board allows the user the flexibility to operate the AD9740
in various configurations. Possible output configurations
include transformer coupled, resistor terminated, and single
and differential outputs. The digital inputs are designed to be
driven from various word generators, with the on-board option
to add a resistor network for proper load termination. Provisions
are also made to operate the AD9740 with either the internal or
external reference or to exercise the power-down feature.
JP3
CKEXTX
BEAD
RED
TP2
DVDD
TB1 1
C7
0.1μF
1
2
3
4
5
6
7
8
1
2
3
4
5
6
7
DB13X
DB12X
DB11X
DB10X
DB9X
DB8X
DB7X
DB6X
DB5X
DB4X
DB3X
DB2X
DB1X
DB0X
BLK
TP4
+ C4
10μF
25V
C6
0.1μF
BLK
TP7
RP3
RP3
RP3
RP3
RP3
RP3
RP3
RP3
RP4
RP4
RP4
RP4
RP4
RP4
RP4
8 RP4
CKEXTX
RIBBON
L2
RP5
OPT
1
2
3
4
5
6
7
8
9
10
DCOM
R1
R2
R3
R4
R5
R6
R7
R8
R9
DB13X
DB12X
DB11X
DB10X
DB9X
DB8X
DB7X
DB6X
DB5X
DB4X
DB3X
DB2X
DB1X
DB0X
RP1
OPT
22Ω 16
22Ω 15
22Ω 14
22Ω 13
22Ω 12
22Ω 11
22Ω 10
22Ω 9
22Ω 16
22Ω 15
22Ω 14
22Ω 13
22Ω 12
22Ω 11
22Ω 10
DB13
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
22Ω 9
RP6
OPT
CKEXT
DCOM 1
R1 2
R2 3
R3 4
R4 5
R5 6
R6 7
R7 8
R8 9
R9 10
1
3
5
7
9
11
13
15
17
19
21
23
25
27
29
31
33
35
37
39
DCOM 1
R1 2
R2 3
R3 4
R4 5
R5 6
R6 7
R7 8
R8 9
R9 10
2
4
6
8
10
12
14
16
18
20
22
24
26
28
30
32
34
36
38
40
1 DCOM
2 R1
3 R2
4 R3
5 R4
6 R5
7 R6
8 R7
9 R8
10 R9
J1
RP2
OPT
BLK
TP8
TB1 2
L3
BEAD
RED
TP5
C9
0.1μF
BLK
TP6
+ C5
10μF
25V
C8
0.1μF
BLK
TP10
BLK
TP9
TB1 4
Figure 39. SOIC Evaluation Board—Power Supply and Digital Inputs
Rev. B | Page 20 of 32
02911-037
AVDD
TB1 3
AD9740
AVDD
+ C14
10μF
16V
C16
0.1μF
CUT
UNDER DUT
C17
0.1μF
JP6
DVDD
C18
0.1μF
DVDD
C19
0.1μF
R5
OPT
S2
IOUTA
CLOCK
CKEXT
AVDD
1
2
3
4
5
6
7
8
9
10
11
12
13
14
DB13
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
CLOCK
DVDD
DCOM
MODE
AVDD
RESERVED
IOUTA
U1
AD9740 IOUTB
ACOM
NC
FS ADJ
REFIO
REFLO
SLEEP
2
A B
3
1
JP5
INT
EXT
REF
28
27
26
25
24
23
22
21
20
19
18
17
16
15
CLOCK
TP1
WHT
JP10
A B
2
3
R11
10kΩ
S5
JP4
DB13
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
1
IX
DVDD
R4
50Ω
R2
10kΩ
DVDD
C13
OPT
JP8
JP2
IOUT
MODE
AVDD
3
T1
2
R6
OPT
5
1
REF
R1
2kΩ
4
S3
6
T1-1T
TP3
WHT
C11
0.1μF
C1
0.1μF
C2
0.1μF
C12
OPT
JP9
AVDD
SLEEP
TP11
WHT
R10
10kΩ
S1
IOUTB
R3
10kΩ
IY
Figure 40. SOIC Evaluation Board—Output Signal Conditioning
Rev. B | Page 21 of 32
1
2
A B
3
JP11
02911-038
+ C15
10μF
16V
02911-039
AD9740
02911-040
Figure 41. SOIC Evaluation Board—Primary Side
Figure 42. SOIC Evaluation Board—Secondary Side
Rev. B | Page 22 of 32
02911-041
AD9740
02911-042
Figure 43. SOIC Evaluation Board—Ground Plane
Figure 44. SOIC Evaluation Board—Power Plane
Rev. B | Page 23 of 32
02911-043
AD9740
02911-044
Figure 45. SOIC Evaluation Board Assembly—Primary Side
Figure 46. SOIC Evaluation Board Assembly—Secondary Side
Rev. B | Page 24 of 32
AD9740
RED
TP12
TB1
C3
0.1μF
TB1
CVDD
1
BLK
C2
10μF
6.3V
TP2
2
C10
0.1μF
2
4
1
3
6
5
8
7
DB10X
10
9
DB9X
11
DB8X
13
DB7X
15
DB6X
17
DB5X
19
DB4X
21
DB3X
23
DB2X
25
DB1X
27
DB0X
12
L2 BEAD
TB3
16
DVDD
1
C7
0.1μF
TB3
14
RED
TP13
18
20
BLK
C6
0.1μF
C4
10μF
6.3V
TP4
2
22
24
26
28
RED
TP5
L3 BEAD
C9
0.1μF
TB4
32
AVDD
1
BLK
36
C8
0.1μF
C5
10μF
6.3V
TP6
34
38
40
2
DB13X
DB12X
DB11X
29
31
33
35
JP3
CKEXTX
37
39
J1
R3
100Ω
R4
100Ω
R15
100Ω
R16
100Ω
R17
100Ω
R18
100Ω
R19
100Ω
DB13X
DB12X
DB11X
DB10X
DB9X
DB8X
DB7X
DB6X
DB5X
DB4X
DB3X
DB2X
DB1X
DB0X
CKEXTX
R21
100Ω
R24
100Ω
R25
100Ω
R26
100Ω
R27
100Ω
R20
100Ω
1 RP3
22Ω 16
2 RP3
22Ω 15
3 RP3
22Ω 14
4 RP3
22Ω 13
5 RP3
22Ω 12
6 RP3
7 RP3
22Ω 11
22Ω 10
8 RP3
22Ω 9
1 RP4
22Ω 16
2 RP4
22Ω 15
3 RP4
22Ω 14
4 RP4
22Ω 13
5 RP4
22Ω 12
6 RP4
7 RP4
22Ω 11
22Ω 10
8 RP4
22Ω 9
DB13
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
CKEXT
R28
100Ω
02911-045
TB4
30
HEADER STRAIGHT UP MALE NO SHROUD
L1 BEAD
Figure 47. LFCSP Evaluation Board Schematic—Power Supply and Digital Inputs
Rev. B | Page 25 of 32
AD9740
AVDD
DVDD
CVDD
C19
0.1μF
C17
0.1μF
C32
0.1μF
SLEEP
TP11
WHT
R29
10kΩ
DB7
DB6
DVDD
DB5
DB4
DB3
DB2
DB1
DB0
CVDD
CLK
CLKB
CMODE
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
DB7
DB6
DVDD
DB5
DB4
DB3
DB2
DB1
DB0
DCOM
U1
CVDD
CLK
CLKB
CCOM
CMODE
MODE
DB8
DB9
DB10
DB11
DB12
DB13
DCOM1
SLEEP
FS ADJ
REFIO
ACOM
IA
IB
ACOM1
AVDD
AVDD1
32
31
30
29
28
27
26
25
DB8
DB9
DB10
DB11
DB12
DB13
R11
50kΩ
DNP
C13
24
23
22
TP3
TP1
WHT
WHT
JP8
IOUT
3
21
20
19
18
17
TP7
R30
10kΩ
4
5
2
S3
AGND: 3, 4, 5
6
1
AVDD
T1 – 1T
C11
0.1μF
JP9
AD9740LFCSP
WHT
T1
DNP
C12
R10
50Ω
CVDD
R1
2kΩ
0.1%
JP1
02911-046
MODE
Figure 48. LFCSP Evaluation Board Schematic—Output Signal Conditioning
CVDD
1
7
U4
C20
10μF
16V
2
AGND: 5
CVDD: 8
C35
0.1μF
CVDD
R5
120Ω
3
JP2
CKEXT
CLK
4
U4
6
AGND: 5
CVDD: 8
R2
120Ω
C34
0.1μF
S5
AGND: 3, 4, 5
R6
50Ω
02911-047
CLKB
Figure 49. LFCSP Evaluation Board Schematic—Clock Input
Rev. B | Page 26 of 32
02911-048
AD9740
02911-049
Figure 50. LFCSP Evaluation Board Layout—Primary Side
Figure 51. LFCSP Evaluation Board Layout—Secondary Side
Rev. B | Page 27 of 32
02911-050
AD9740
02911-051
Figure 52. LFCSP Evaluation Board Layout—Ground Plane
Figure 53. LFCSP Evaluation Board Layout—Power Plane
Rev. B | Page 28 of 32
02911-052
AD9740
02911-053
Figure 54. LFCSP Evaluation Board Layout Assembly—Primary Side
Figure 55. LFCSP Evaluation Board Layout Assembly—Secondary Side
Rev. B | Page 29 of 32
AD9740
OUTLINE DIMENSIONS
9.80
9.70
9.60
28
15
4.50
4.40
4.30
1
6.40 BSC
14
PIN 1
0.65
BSC
0.15
0.05
COPLANARITY
0.10
0.30
0.19
1.20 MAX
0.20
0.09
SEATING
PLANE
8°
0°
0.75
0.60
0.45
COMPLIANT TO JEDEC STANDARDS MO-153-AE
Figure 56. 28-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-28)
Dimensions shown in millimeters
18.10 (0.7126)
17.70 (0.6969)
28
15
7.60 (0.2992)
7.40 (0.2913)
1
14
2.65 (0.1043)
2.35 (0.0925)
10.65 (0.4193)
10.00 (0.3937)
0.75 (0.0295)
× 45°
0.25 (0.0098)
0.30 (0.0118)
0.10 (0.0039)
COPLANARITY
0.10
8°
1.27 (0.0500) 0.51 (0.0201) SEATING 0.33 (0.0130) 0°
BSC
PLANE
0.31 (0.0122)
0.20 (0.0079)
1.27 (0.0500)
0.40 (0.0157)
COMPLIANT TO JEDEC STANDARDS MS-013-AE
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 57. 28-Lead Standard Small Outline Package [SOIC]
Wide Body (RW-28)
Dimensions shown in millimeters and (inches)
Rev. B | Page 30 of 32
AD9740
0.60 MAX
5.00
BSC SQ
0.60 MAX
PIN 1
INDICATOR
TOP
VIEW
0.50
BSC
4.75
BSC SQ
0.50
0.40
0.30
32
1
3.25
3.10 SQ
2.95
EXPOSED
PAD
(BOTTOM VIEW)
17
16
9
8
0.25 MIN
3.50 REF
0.80 MAX
0.65 TYP
12° MAX
1.00
0.85
0.80
PIN 1
INDICATOR
25
24
0.05 MAX
0.02 NOM
SEATING
PLANE
0.30
0.23
0.18
0.20 REF
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VHHD-2
Figure 58. 32-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
5 mm × 5 mm Body, Very Thin Quad
(CP-32-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD9740AR
AD9740ARRL
AD9740ARZ 1
AD9740ARZRL1
AD9740ARU
AD9740ARURL7
AD9740ARUZ1
AD9740ARUZRL71
AD9740ACP
AD9740ACPRL7
AD9740ACPZ1
AD9740ACPZRL71
AD9740-EB
AD9740ACP-PCB
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
28-Lead Wide Body SOIC
28-Lead Wide Body SOIC
28-Lead Wide Body SOIC
28-Lead Wide Body SOIC
28-Lead TSSOP
28-Lead TSSOP
28-Lead TSSOP
28-Lead TSSOP
32-Lead LFCSP
32-Lead LFCSP_VQ
32-Lead LFCSP_VQ
32-Lead LFCSP_VQ
Evaluation Board (SOIC)
Evaluation Board (LFCSP)
Z = Pb-free part.
Rev. B | Page 31 of 32
Package Option
RW-28
RW-28
RW-28
RW-28
RU-28
RU-28
RU-28
RU-28
CP-32-2
CP-32-2
CP-32-2
CP-32-2
AD9740
NOTES
© 2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C02911–0–12/05(B)
Rev. B | Page 32 of 32