LINER LTC3603 40v, 1a synchronous step-down converter 1a guaranteed output current Datasheet

LTC3646/LTC3646-1
40V, 1A Synchronous
Step-Down Converter
Features
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Description
Wide Input Voltage Range: 4.0V to 40V
1A Guaranteed Output Current
High Efficiency: Up to 95%
Wide Output Voltage Range
LTC3646: 2.0V to 30V
LTC3646-1: 0.6V to 15V
±1% Accurate Reference Voltage
Internal or External Compensation
Switching Frequency Adjustable and Synchronizable:
200kHz to 3MHz
Selectable High Efficiency Burst Mode® Operation or
Forced Continuous Mode
14 Lead 3mm × 4mm DFN or Thermally Enhanced
16-Lead MSOP Packages
The LTC®3646 is a high efficiency, step-down DC/DC
converter with internal high side and synchronous power
FETs. It draws only 140µA typical DC supply current in
Burst Mode operation at no load while maintaining output
voltage regulation.
The LTC3646 can supply up to a 1A load, and its combination of Burst Mode operation, integrated power switches
and low quiescent current provides high efficiency over a
broad range of load currents. Alternatively, it can be used
in forced continuous mode for ripple sensitive applications.
The LTC3646 has a wide 4.0V to 40V input range and its
patented controlled on-time architecture and 0.6V reference
voltage allow for large step-down ratios without the risk of
overvoltage. The frequency may be set between 200kHz
to 3.0MHz with a program resistor or synchronized to an
external clock.
Applications
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Point of Load Power Supply
Intermediate Bus Power
Automotive Applications
The internal soft-start, short-circuit protection and voltage rating make the LTC3646 a robust part ideal for high
voltage applications.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents, including 6580258, 5481178,
5994885, 6304066, 5847554, 6476589, 6774611.
Typical Application
Efficiency Curve
BOOST
PVIN
90
SVIN
SW
INTVCC
3.3µH
MODE/SYNC
EXTVCC
SWITCH
CONTROL
VON
412k
4.7µF
0.6V
SGND
+ –
PGND
15µF
VFB
3646 TA01a
VOUT = 3.3V
0.8
70
0.6
60
50
0.4
40
30
VOUT = 5V
20
VOUT = 3.3V
0.2
10
0
0.001
56.2k
1.0
VOUT = 5V
80
VOUT
5V
1A
ITH
RT
VIN = 12V
f = 1MHz
POWER LOSS (W)
10µF
0.1µF
LTC3646
RUN
EFFICIENCY (%)
VIN
40V MAX
100
0.01
0.1
LOAD CURRENT (A)
1
0
3646 TA01b
36461f
1
LTC3646/LTC3646-1
Absolute Maximum Ratings
(Notes 1, 7)
PVIN, SVIN Supply Voltage......................... –0.3V to 45V
SW Voltage (DC)............................ –0.3V to PVIN + 0.3V
BOOST – SW Voltage.................................... –0.3V to 6V
VON Voltage................................................. –0.3V to 33V
MODE/SYNC, RT, ITH,
VFB Voltage.................................–0.3V to INTVCC + 0.3V
INTVCC Voltage............................................. –0.3V to 6V
EXTVCC Voltage............................................ –0.3V to 6V
RUN Voltage............................................... –0.3V to 45V
PGOOD........................................–0.3V to INTVCC + 0.3V
Operating Junction Temperature Range
(Note 2)................................................... –40°C to 150°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec) MSE........... 300°C
Pin Configuration
TOP VIEW
SGND
1
14 SVIN
VFB
2
13 RUN
ITH
3
RT
4
15
PGND
TOP VIEW
12 EXTVCC
11 INTVCC
VON
5
10 BOOST
PGOOD
6
9 SW
MODE/SYNC
7
8 PVIN
SGND
VFB
ITH
RT
VON
PGOOD
MODE/SYNC
NC
1
2
3
4
5
6
7
8
17
PGND
16
15
14
13
12
11
10
9
SVIN
RUN
EXTVCC
INTVCC
BOOST
SW
PVIN
PVIN
MSE PACKAGE
16-LEAD PLASTIC MSOP
DE PACKAGE
14-LEAD (4mm × 3mm) PLASTIC DFN
TJMAX = 150°C, θJA = 43°C/W
EXPOSED PAD (PIN 15) IS PGND, MUST BE SOLDERED TO PCB
TJMAX = 150°C, θJA = 38°C/W
EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3646EDE#PBF
LTC3646EDE#TRPBF
3646
14-Lead (4mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3646IDE#PBF
LTC3646IDE#TRPBF
3646
14-Lead (4mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3646HDE#PBF
LTC3646HDE#TRPBF
3646
14-Lead (4mm × 3mm) Plastic DFN
–40°C to 150°C
LTC3646EMSE#PBF
LTC3646EMSE#TRPBF
3646
16-Lead Plastic MSOP
–40°C to 85°C
LTC3646IMSE#PBF
LTC3646IMSE#TRPBF
3646
16-Lead Plastic MSOP
–40°C to 125°C
LTC3646HMSE#PBF
LTC3646HMSE#TRPBF
3646
16-Lead Plastic MSOP
–40°C to 150°C
LTC3646EDE-1#PBF
LTC3646EDE-1#TRPBF
36461
14-Lead (4mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3646IDE-1#PBF
LTC3646IDE-1#TRPBF
36461
14-Lead (4mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3646HDE-1#PBF
LTC3646HDE-1#TRPBF
36461
14-Lead (4mm × 3mm) Plastic DFN
–40°C to 150°C
LTC3646EMSE-1#PBF
LTC3646EMSE-1#TRPBF
36461
16-Lead Plastic MSOP
–40°C to 85°C
LTC3646IMSE-1#PBF
LTC3646IMSE-1#TRPBF
36461
16-Lead Plastic MSOP
–40°C to 125°C
LTC3646HMSE-1#PBF
LTC3646HMSE-1#TRPBF 36461
16-Lead Plastic MSOP
–40°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
36461f
2
LTC3646/LTC3646-1
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C (Note 2), PVIN = SVIN = 12V unless otherwise noted.
SYMBOL
PARAMETER
Input Supply (PVIN, SVIN)
Input Voltage Operating Range
VIN
Input Supply Overvoltage Lockout
VIN(OV)
VIN(UV)
Input Supply Undervoltage Lockout
DC Supply Current (Note 3)
Forced Continuous
Sleep Mode
Shutdown Mode
Main Control Loop
Output Voltage Range (Note 4)
VOUT
CONDITIONS
MIN
l
VIN Rising
Hysteresis (VIN Falling)
VIN Falling
Hysteresis (VIN Rising)
l
l
4.0
43.5
2.0
3.2
IQ
VFB
Feedback Reference Voltage
VRUN = 0V
LTC3646
LTC3646-1
–40°C < TA < 85°C
gm(EA)
tON(MIN)
tOFF(MIN)
ILIM
RDS(ON)
External Clock Frequency Range
Top Switch On-Resistance (Note 5)
Bottom Switch On-Resistance (Note 5)
Switch Leakage
Internal VCC Regulator
INTVCC Voltage
VINTVCC
INTVCC Load Regulation (Note 6)
INTVCC Undervoltage Lockout
INTVCC UVLO Hysteresis
EXTVCC Switchover Voltage
EXTVCC Hysteresis
Operation
RUN Pin Threshold
VRUN
VPGOOD(UT)
RUN Pin Leakage Current
PGOOD Overvoltage Threshold
VPGOOD(LT)
PGOOD Undervoltage Threshold
tPGOOD
PGOOD Filter Time
VIN = 4.0V to 40V, ITH = 1.5V
ITH = 1.0V to 1.8V
VFB = 0.6V
External Comp
VIN = 40V, RRT = 30k, VON = 2V
2.0
0.6
0.594
0.591
l
l
VRT = VINTVCC
RRT = 450kΩ
RRT = 60kΩ
RRT = 30kΩ
0.9
1.6
0.19
1.25
2.55
0.2
VIN = 5.5V
VIN = 5.5V
VIN = VSW = 40V, VRUN = 0
VIN = 40V, VSW = 0V, VRUN = 0
MAX
40
46
2.5
3.35
250
620
140
8
l
Feedback Voltage Line Regulation
Feedback Voltage Load Regulation
Feedback Input Current
Error Amplifier Transconductance
Minimum On-Time
Minimum Off-Time
Valley Switch Current Limit
Internal Oscillator Frequency
TYP
0.6
0.6
0
300
30
80
1.2
2.25
0.2
1.5
3.0
2.9
3.5
875
190
30
15
0.606
0.609
0.05
0.12
±20
1.5
2.95
0.27
1.75
3.45
3.0
200
120
±1
±1
l
4.8
IINTVCC = 0mA to 5mA
INTVCC Falling
EXTVCC Rising
4.25
RUN Rising
RUN Falling
Hysteresis
RUN = 1.3V
FB Rising
FB Falling
FB Falling
FB Rising
RT = INTVCC
1.17
1.06
5.0
3.5
–5.0
–3.5
15
5.0
0.5
3.0
0.3
4.5
200
1.21
1.10
110
0
7.5
5
–7.5
–5
30
5.2
4.65
1.26
1.14
UNITS
V
V
V
V
mV
µA
µA
µA
V
V
V
V
%
%
nA
µS
ns
ns
A
MHz
MHz
MHz
MHz
MHz
mΩ
mΩ
µA
µA
V
%
V
V
V
mV
±1
10
V
V
mV
µA
%VFB
–10
%VFB
µs
36461f
3
LTC3646/LTC3646-1
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C (Note 2), PVIN = SVIN = 12V unless otherwise noted.
SYMBOL
RPGOOD
tSS
VMODE/SYNC
PARAMETER
PGOOD Pull-Down Resistance
PGOOD Leakage
Internal Soft-Start Time
Mode Threshold Voltage
SYNC Threshold Voltage
MODE/SYNC Input Current
Internal ITH Voltage Threshold
VON Pin Input Impedance
CONDITIONS
IPGOOD = 10mA
VPGOOD = VINTVCC
MIN
TYP
63
250
Mode VIH
Mode VIL
SYNCIH
SYNCIL
l
l
l
l
1.2
LTC3646
LTC3646-1
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3646 is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3646E is guaranteed to meet specifications from
0°C to 85°C junction temperature. Specifications over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3646I is guaranteed over the –40°C to 125°C operating junction
temperature range and the LTC3646H is guaranteed over the –40°C to
150°C operating junction temperature range. High junction temperatures
derate operating lifetimes; operating lifetime is derated for junction
temperatures greater than 125°C. Note that the maximum ambient
temperature consistent with these specifications is determined by specific
operating conditions in conjunction with board layout, the rated package
thermal impedance and other environmental factors.
1.0
500
0.3
1.2
1
l
MAX
INTVCC – 0.3V
520
600
0.3
2
UNITS
Ω
µA
µs
V
V
V
V
µA
V
kΩ
kΩ
Note 3: Dynamic supply current is higher due to gate charging being
delivered at the switch frequency.
Note 4: Limits to VOUT are a subject to PVIN, SVIN, ton(min), toff(min),
and frequency constraints. See the Applications Information section for a
further discussion. These items are tested with appropriate combinations
of VIN, VON and frequency.
Note 5: RDS(ON) is guaranteed by correlation to wafer level measurements.
Note 6: Maximum allowed current draw when used as a regulated output
is 5mA. This supply is only intended to provide additional DC load current
as needed and not to regulate large transient or AC behavior as such
waveforms may impact LTC3646 operation.
Note 7: This IC includes overtemperature protection intended to protect
the device during momentary overload conditions. The maximum junction
temperature may be exceeded when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may result in device degradation or failure.
36461f
4
LTC3646/LTC3646-1
Typical Performance Characteristics
TA = 25°C, SVIN/PVIN = 12V, fO = 1MHz unless otherwise noted.
Efficiency vs Load Current,
Burst Mode Operation
Efficiency vs Load Current,
Forced Continuous Mode Operation
100
100
VOUT = 1.8V
90 fO = 1.5MHz
80
70
70
50
40
30
EFFICIENCY (%)
80
70
60
60
50
40
0
0.001
0.01
0.1
LOAD CURRENT (A)
0
0.001
1
0.01
0.1
LOAD CURRENT (A)
95
LOAD = 500mA
LOAD = 100mA
0.35
606
0.30
602
600
0
5
10
35
0
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
36461 G06
Oscillator Internal Set Frequency
vs Temperature
2.50
VDS = 12V, MAIN SWITCH
VDS = 12V, SYNCHRONOUS SWITCH
VDS = 40V, MAIN SWITCH
VDS = 40V, SYNCHRONOUS SWITCH
2.45
8
6
2.30
2.25
2.20
2.15
2.10
2
2.05
0
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
2.00
–40 –20 0
RRT = 60k
1.7
2.35
4
36461 G07
1.8
VRT = VINTVCC
FREQUENCY (MHz)
10
Oscillator Externally Set Frequency
vs Temperature
2.40
FREQUENCY (MHz)
14
12
SYNCHRONOUS SWITCH
36461 G05
Switch Pin Leakage
vs Temperature and Supply
16
0.15
592
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
40
36461 G04
18
0.20
0.05
594
15 20 25 30
INPUT VOLTAGE (V)
MAIN SWITCH
0.25
0.10
596
L = 6.8µH
VOUT = 3.3V
1
RDS(ON) vs Temperature
608
598
70
65
0.01
0.1
LOAD CURRENT (A)
36461 G03
RESISTANCE (Ω)
VREF (V)
EFFICIENCY (%)
LOAD = 1A
75
0
0.001
1
604
80
VOUT = 5V
VOUT = 3.3V
10
Reference Voltage
vs Temperature
100
85
20
36461 G02
Efficiency vs Input Voltage,
Burst Mode Operation
90
FORCED CONTINUOUS
OPERATION
40
VIN = 5V
VIN = 8V
VIN = 12V
10
36461 G01
20
50
30
20
VIN = 5V
VIN = 8V
VIN = 12V
10
60
60
30
20
Burst Mode OPERATION
90
80
EFFICIENCY (%)
EFFICIENCY (%)
VOUT = 1.8V
90 fO = 1.5MHz
SW LEAKAGE CURRENT (µA)
Efficiency vs Load Current
100
1.6
1.5
1.4
1.3
20 40 60 80 100 120 140
TEMPERATURE (°C)
36461 G08
1.2
–40 –20 0
20 40 60 80 100 120 140
TEMPERATURE (°C)
36461 G09
36461f
5
LTC3646/LTC3646-1
Typical Performance Characteristics
TA = 25°C, SVIN/PVIN = 12V, fO = 1MHz unless otherwise noted.
No Load Quiescent Current vs
Temperature, Burst Mode Operation
Bottom Switch Current Limit
vs Temperature
180
160
140
120
100
–40 –20 0
1.6
1.4
1.2
1.3
1.2
0.4
0
–0.4
20 40 60 80 100 120 140
TEMPERATURE (°C)
Output Voltage vs Time,
Burst Mode Operation
0.2
0.8
0.4
0.6
LOAD CURRENT (A)
RUN
5V/DIV
IL
250mA/DIV
IL
250mA/DIV
IL
250mA/DIV
VOUT
20mV/DIV
AC-COUPLED
VOUT
5V/DIV
VOUT
5V/DIV
36461 G13
ILOAD = 25mA
100µs/DIV
36461 G14
ILOAD = 25mA
Load Step,
Forced Continuous Operation
Load Step, Burst Mode Operation
IL
1A/DIV
IL
1A/DIV
VOUT
AC-COUPLED
100mV/DIV
VOUT
AC-COUPLED
100mV/DIV
36461 G16
SHUTDOWN CURRENT (µA)
ILOAD
1A/DIV
10µs/DIV
VOUT = 1.2V
50mA to 1A STEP
fO = 2.25MHz
RCOMP = 200kΩ
CCOMP = 33pF
CF = 10pF
COUT = 15µF
100µs/DIV
36461 G15
IQ Shutdown vs Supply and
Temperature
60
ILOAD
1A/DIV
1
Start-Up from Shutdown,
Forced Continuous Mode
Start-Up from Shutdown,
Burst Mode Operation
RUN
5V/DIV
40µs/DIV
VOUT = 1.2V
50mA to 1A STEP
fO = 2.25MHz
RCOMP = 200kΩ
CCOMP = 33pF
CF = 10pF
COUT = 15µF
0
36461 G12
SW
10V/DIV
2µs/DIV
FORCED CONTINUOUS
Burst Mode OPERATION
36461 G11
36461 G10
VIN = 12V
VOUT = 5V
ILOAD = 20mA
VOUT = 5V
0.8
1.1
1.0
–40 –20 0
20 40 60 80 100 120 140
TEMPERATURE (°C)
Load Regulation
1.5
∆VOUT/VOUT (%)
BOTTOM SWITCH CURRENT LIMIT (A)
NO LOAD QUIESCENT CURRENT (µA)
200
36461 G17
50
VIN = 40V
VIN = 12V
40
30
20
10
0
–40 –20 0
20 40 60 80 100 120 140
TEMPERATURE (°C)
36461 G18
36461f
6
LTC3646/LTC3646-1
Pin Functions
(DFN/MSOP)
SGND (Pin 1/Pin 1): Analog Ground Pin. This pin should
have a low noise connection to the reference ground.
VFB (Pin 2/Pin 2): Output Voltage Feedback Pin. Input to
the error amplifier that compares the feedback voltage with
the internal 0.6V reference. Connect this pin to a resistor
divider network to program the desired output voltage.
ITH (Pin 3/Pin 3): Error Amplifier Output and Switching
Regulator Compensation Point. Connect this pin to appropriate external components to compensate the regulator
loop frequency response. Connect this pin to INTVCC to
use the default internal compensation.
RT (Pin 4/Pin 4): Oscillator Frequency Program Pin. Connect an external resistor between 450k and 30k from this
pin to SGND to program the switching frequency from
200kHz to 3.0MHz. When RT is connected to INTVCC, the
switching frequency will be 2.25MHz. Do not float RT.
VON (Pin 5/Pin 5): On-Time Voltage Input Pin. This pin
provides information about the output voltage (VOUT) to
the on-time control loop. Connect this pin to the regulated
output to make the on-time proportional to the output
voltage.
PGOOD (Pin 6/Pin 6): Power Good Output Pin. PGOOD
is pulled to ground when the voltage at the VFB pin is not
within 7.5% (typical) of the internal 0.6V reference. PGOOD
becomes high impedance once the voltage at the VFB pin
returns to within 5% of the internal reference.
MODE/SYNC (Pin 7/Pin 7): Mode Selection and External
Clock Input Pin. This pin forces the LTC3646 into forced
continuous operation when tied to ground, and high efficiency Burst Mode operation when tied to INTVCC. When
driven with an external clock, the LTC3646 will adjust the
top switch on-time to match the switching frequency to
the applied clock frequency and the part will operate in
forced continuous mode. During start-up or external clock
synchronization, the operating mode will be as described
in the Applications Information section.
PVIN (Pin 8/Pins 9, 10): Supply Pin for the Power Switch.
This pin connects directly to top switch. Closely decouple
this pin to PGND with a 10µF or greater, low ESR capacitor.
SW (Pin 9/Pin 11): Switch Node Output Pin. Connect this
pin to the switch side of the external inductor and boost
capacitor.
BOOST (Pin 10/Pin 12): Boosted Supply Pin. A boosted
voltage is generated at this pin by connecting a capacitor
between this pin and the SW pin. The normal operation
voltage swing of this pin ranges from INTVCC to PVIN
+ INTVCC. When necessary, connect the cathode of an
external boost diode to this pin. See the Boost Capacitor
and Diode section.
INTVCC (Pin 11/Pin 13): Internal 5.0V Regulator Output
Pin. This pin should be decoupled to PGND with a 4.7µF or
greater low ESR capacitor. When necessary, connect the
anode of an external boost diode to this pin. The internal
regulator is disabled when the RUN pin is low.
EXTVCC (Pin 12/Pin 14): Use this input pin to power the
chip’s low voltage control circuitry if a high efficiency
supply between 4.5V and 6.0V is available. Otherwise,
connect this pin to SGND. See the Applications Information section for further information.
RUN (Pin 13/Pin 15): Regulator Enable Pin. Enables chip
operation by applying a voltage greater than VRUN.
SVIN (Pin 14/Pin 16): Power Supply Input for Internal
Circuitry. Closely decouple this pin to SGND with a greater
than 1µF low ESR capacitor. SVIN pin voltage should equal
PVIN in order to correctly calculate on the on time and
maintain constant frequency operation.
PGND (Exposed Pad Pin 15/Exposed Pad Pin 17): Power
Path Ground Pin. The exposed pad must be well soldered
to the PCB to provide a low impedance electrical connection to ground and rated thermal performance.
36461f
7
LTC3646/LTC3646-1
Functional Block Diagram
CPVIN
CSVIN
SVIN
RUN
1.21V
+
–
EXTVCC
4.5V
PVIN
+
–
INTVCC
5V REG
MODE/SYNC
CVCC
OSCILLATOR
RT
RRT
BOOST
TG
VON
CBOOST
SW
ON-TIME CONTROLLER
CONTROL
LOGIC
R
V
tON = ON
VIN • fO
S
+
–
ICMP
ITH
Q
L1
VOUT
COUT
ON
+
–
BG
IREV
PGND
SENSE+
SENSE–
RCOMP
0.555V
CCOMP
+ –
EA
+ –
INTVCC – 0.3V
0.6V
REF
0.645V
SOFT-START
+
–
+
–
UV
PGOOD
OV
VFB
SGND
R1
R2
3646 BD
36461f
8
LTC3646/LTC3646-1
Operation
The LTC3646 and LTC3646-1 are current mode, monolithic,
synchronous, step-down regulators capable of 40V input
operation, and extremely high step down ratios while maintaining constant frequency. (Both will be referred to as the
LTC3646 except as specifically noted.) Part operation is
enabled by raising the voltage of the RUN pin above 1.21V.
Main Control Loop
In normal operation a switching cycle is initiated by a
signal from the inductor valley current comparator (ICMP
in the Block Diagram). The top power MOSFET is turned
on, and a timer is simultaneously started in the on-time
controller. The on-time controller computes the correct on
time (subject to tOn(min)) based on the desired switching
frequency fO, and step-down ratio VON /VIN, according to the
formula shown in the Block Diagram. In a typical application, the VON pin should be connected to the output voltage,
VOUT. When the timer expires, the top power MOSFET is
turned off and the bottom power MOSFET is turned on
until the current comparator (ICMP) trips, restarting the
timer and initiating the next cycle. The inductor current
is monitored by sensing the voltage drop across the SW
and PGND nodes of the bottom power MOSFET. The voltage at the ITH node sets the ICMP comparator threshold
corresponding to the inductor valley current. The error
amplifier (EA) adjusts the ITH voltage by comparing an
internal 0.6V reference voltage to the feedback signal,
VFB, derived from the output voltage. If, for example, the
load current increases, the output voltage will decrease
relative to the 0.6V reference. The ITH voltage will rise
until the average inductor current increases to match the
load current.
At low load currents, the inductor current can drop to
zero or become negative. If the LTC3646 is configured for
Burst Mode operation, this inductor current condition is
detected by the current reversal comparator (IREV) which
then shuts off the bottom power MOSFET and places the
part into a low quiescent current sleep state, resulting in
discontinuous operation and increased efficiency at low
load currents. Both power MOSFETs will remain off with
the part in sleep and the output capacitor supplying the
load current until the ITH voltage rises sufficiently to initiate
another cycle. Discontinuous operation is disabled by tying
the MODE/SYNC pin to ground, placing the LTC3646 into
forced continuous mode. During forced continuous mode
operation, synchronous operation occurs regardless of
the output load current, and the inductor current trough
levels are allowed to become negative.
The operating frequency is determined by the value of the
RRT resistor, which programs the current for the internal
oscillator. An internal phase-locked loop adjusts the switching regulator on-time so that the switching frequency
matches the programmed frequency, subject to tON and
tOFF time constraints shown in the Electrical Characteristics
table. Alternatively, the RT pin can be connected to the
INTVCC pin which causes the internal oscillator to run at
the default frequency of 2.25MHz.
A clock signal can be applied to the MODE/SYNC pin to
synchronize the switching frequency to an external source.
When operating in this configuration, connect a resistor to
the RT pin with a value corresponding to the applied clock
frequency. With an external clock supplied to the MODE/
SYNC pin, the part will operate in forced continuous mode.
Power Good Status Output
The PGOOD open-drain output will be pulled low if the
regulator output exits the VPGOOD window around the
regulation point. This condition is released once regulation within the specified window is achieved. To prevent
unwanted PGOOD glitches during transients or dynamic
VOUT changes, the LTC3646 PGOOD falling edge includes
a filter time of approximately 70 clock cycles.
36461f
9
LTC3646/LTC3646-1
Operation
SVIN/PVIN Overvoltage Protection
Short-Circuit Protection
In order to protect the internal power MOSFET devices
against transient voltage spikes, the LTC3646 continuously
monitors the SVIN pin for an overvoltage condition. When
SVIN rises above VIN(OV), the regulator suspends operation by shutting off both power MOSFETs and resetting
the soft-start level. Once VIN drops to below the specified
range of VIN(OV), the regulator immediately restarts normal
operation.
The LTC3646 is designed to withstand output short circuits.
In this situation the part will source ILIM (approximately
1.2A) plus half the inductor current ripple. Since VOUT is
at or near 0V, the on-time will shorten and the off-time
will lengthen considerably, resulting in a lower switching
frequency.
36461f
10
LTC3646/LTC3646-1
Applications Information
A general LTC3646 application circuit is shown on the first
page of this data sheet. External component selection is
largely driven by the load requirement and begins with the
selection of the inductor L. Once the inductor is chosen,
the input capacitor, CIN, the output capacitor, COUT, the
internal regulator capacitor, CVCC, and the boost capacitor, CBOOST, can be selected. Next, the feedback resistors
are selected to set the desired output voltage. Finally, the
remaining optional external components can be selected for
functions such as external loop compensation, externally
programmed oscillator frequency and PGOOD.
Operating Frequency
Selection of the operating frequency is a trade-off between
efficiency and component size. High frequency operation
allows the use of smaller inductor and capacitor values.
Operation at lower frequencies improves efficiency by
reducing internal gate charge losses but requires larger
inductance values and/or capacitance to maintain low
output ripple voltage. The operating frequency, fO, of the
LTC3646 is determined by an external resistor that is connected between the RT pin and ground. The value of the
resistor sets the ramp current that is used to charge and
discharge an internal timing capacitor within the oscillator
and can be calculated by using the following equation:
RRT
where RRT is in Ω and fO is in Hz.
3000
2500
FREQUENCY (kHz)
Inductor Selection
For a given input and output voltage, the inductor value and
operating frequency determine the inductor ripple current.
More specifically, the inductor ripple current decreases
with higher inductor value or higher operating frequency
according to the following equation:
⎛ V ⎞⎛ V ⎞
ΔIL = ⎜ OUT ⎟⎜1− OUT ⎟
VIN ⎠
⎝ fO • L ⎠⎝
where ΔIL = inductor ripple current (A), fO = operating
frequency (Hz) and L = inductor value (H). A trade-off between component size, efficiency and operating frequency
can be seen from this equation. Accepting larger values of
ΔIL allows the use of lower value inductors but results in
greater core loss in the inductor, greater ESR loss in the
output capacitor, and larger output ripple. Generally, the
highest efficiency operation is obtained at low operating
frequency with small ripple current.
The inductor value should be chosen to give a peak-topeak ripple current of between 30% and 40% of IOUT(MAX),
where IOUT(MAX) equals the maximum average output
current. Note that the largest ripple current occurs at
the highest VIN. To guarantee the ripple current does not
exceed a specified maximum, the inductance should be
chosen according to:
9e10
=
fO
⎛ V
⎞⎛ V ⎞
L = ⎜ OUT ⎟⎜1− OUT ⎟
VIN ⎠
⎝ fO • ΔIL ⎠⎝
2000
1500
1000
500
0
Connecting the RT pin to INTVCC will assert the internal
default fO = 2.25MHz; however, this switching frequency
will be more sensitive to process and temperature variations
than using a resistor on RT (see the Typical Performance
Characteristics section).
0
100
200
300
RRT (kΩ)
400
500
Once the value for L is known, the type of inductor must
be selected. Actual core loss is independent of core size
for a fixed inductor value, but is very dependent on the
inductance selected. As the inductance increases, core loss
decreases. Unfortunately, increased inductance requires
more turns of wire leading to increased copper loss.
3646 F01
Figure 1
36461f
11
LTC3646/LTC3646-1
Applications Information
Ferrite designs exhibit very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core materials saturate “hard,” meaning the inductance
collapses abruptly when the peak design current is
exceeded. This collapse will result in an abrupt increase
in inductor ripple current, so it is important to ensure the
core will not saturate.
Different core materials and shapes will change the size/
current and price/current relationship of an inductor.
Toroidal or shielded pot cores in ferrite or permalloy
materials are small and don’t radiate much energy but
generally cost more than powdered iron core inductors
with similar characteristics. The choice of which style
inductor to use mainly depends on the price versus size
requirements and any radiated field/EMI requirements.
New designs for surface mount inductors are available
from Toko, Vishay, NEC/Tokin, Cooper, Coilcraft, TDK and
Würth Electronik. Table 1 gives a sampling of available
surface mount inductors.
Table 1. Inductor Selection
INDUCTANCE
(µH)
DCR
(mΩ)
MAX CURRENT DIMENSIONS
(A)
(mm)
HEIGHT
(mm)
Würth Elektronik, TPC MH, L, LH Series
3.3 (MH)
3.9 (L)
6.2 (LH)
35
55
45
1.8
2.1
1.7
4.8 × 4.8
5.8 × 5.8
5.8 × 5.8
2.8
1.8
2.8
3.5
2.2
3.92 × 3.92
3.92 × 3.92
2.5
2.5
2.1
1.6
10.1 × 10.1
10.1 × 10.1
4.5
4.5
1.64
1.36
7.3 × 7.3
7.3 × 7.3
4.1
4.1
Sumida, CDRH3d23/HP Series
1.2
3.3
40
70
TDK, SLF10145 Series
22
33
59
82
Coilcraft MSS7341 Series
10
15
32
47
CIN and COUT Selection
The input capacitance, CIN, is needed to filter the trapezoidal wave current at the drain of the top power MOSFET.
To prevent large voltage transients from occurring a low
ESR input capacitor sized for the maximum RMS current
is recommended. The maximum RMS current is given by:
IRMS = IOUT(MAX)
VOUT ( VIN − VOUT )
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT /2. This simple worst-case condition is commonly
used for design because even significant deviations do
not offer much relief. Note that ripple current ratings from
capacitor manufacturers are often based on only 2000
hours of life which makes it advisable to further derate
the capacitor or choose a capacitor rated at a higher
temperature than required.
Several capacitors may be paralleled to meet the requirements of the design. For low input voltage applications
sufficient bulk input capacitance is needed to minimize
transient effects during output load changes. Even though
the LTC3646 design includes an overvoltage protection
circuit, care must always be taken to ensure input voltage
transients do not pose an overvoltage hazard to the part.
The selection of COUT is primarily determined by the effective series resistance (ESR) that is required to minimize
voltage ripple and load step transients. The output ripple,
ΔVOUT, is determined by:
⎛
⎞
1
ΔVOUT < ΔIL ⎜ESR +
⎟
8 • fO • COUT ⎠
⎝
When using low-ESR ceramic capacitors, it is more useful
to choose the output capacitor value to fulfill a charge storage requirement. During a load step, the output capacitor
must instantaneously supply the current to support the load
until the feedback loop raises the switch current enough
to support the load. The time required for the feedback
loop to respond is dependent on the compensation and the
output capacitor size. Typically, 3 to 4 cycles are required
to respond to a load step, but only in the first cycle does
the output drop linearly. The output droop, VDROOP, is
usually about 3 times the linear drop of the first cycle.
36461f
12
LTC3646/LTC3646-1
Applications Information
Thus, a good place to start is with the output capacitor
size of approximately:
COUT ≈
3 • ΔIOUT
fO • VDROOP
Though this equation provides a good approximation, more
capacitance may be required depending on the duty cycle
and load step requirements. The actual VDROOP should be
verified by applying a load step to the output.
Using Ceramic Input and Output Capacitors
Higher value, lower cost ceramic capacitors are now
available in small case sizes. Their high voltage rating
and low ESR make them ideal for switching regulator applications. However, due to the self-resonant and high-Q
characteristics of some types of ceramic capacitors, care
must be taken when these capacitors are used at the input
and output. When a ceramic capacitor is used at the input,
and the power is supplied by a wall adapter through long
wires, a load step at the output can induce ringing at the
VIN input. At best, this ringing can couple to the output and
be mistaken as loop instability. At worst, a sudden inrush
of current through the long wires can potentially cause a
voltage spike at VIN large enough to damage the part. For
a more detailed discussion, refer to Application Note 88.
When choosing the input and output ceramic capacitors
choose the X5R or X7R dielectric formulations. These
dielectrics provide the best temperature and voltage
characteristics for a given value and size.
INTVCC Regulator and EXTVCC
An internal low dropout (LDO) regulator produces a 5V
supply voltage used to power much of the internal LTC3646
circuitry including the power MOSFET gate drivers. The
INTVCC pin connects to the output of this regulator and
should have 4.7μF of decoupling capacitance to ground.
The decoupling capacitor should have low impedance
electrical connections to the INTVCC and PGND pins to
provide the transient currents required by the LTC3646.
The user may draw a maximum load current of 5mA from
this pin but must take into account the increased power
dissipation and die temperature that results. Furthermore,
this supply is intended only to supply additional DC load
currents as desired; it is not intended to regulate large
transient or AC behavior as this may impact LTC3646
operation.
Alternatively, if a suitable supply is available or can be
generated, the power required to operate the low voltage
circuitry of the LTC3646 can be supplied through the
EXTVCC pin. When the voltage on the EXTVCC pin is below 4.5V, the chip power is supplied by the internal LDO.
As shown in the Block Diagram, when EXTVCC is above
4.5V, the internal LDO is shut off, and an internal switch
is closed between the EXTVCC and INTVCC pins. Connect
EXTVCC to SGND if an external supply meeting these constraints is not available.If the voltage on the EXTVCC pin is
efficiently generated, this will result in the highest overall
system efficiency and the least amount of heat generated
by the LTC3646. This effectively decreased the no-load
quiescent current by a factor of VOUT/VIN. This topic is
further discussed in the Thermal Considerations section.
Boost Capacitor and Diode
The boost capacitor, CBOOST, is used to create a voltage rail
above the applied input voltage VIN. Specifically, the boost
capacitor is charged to a voltage equal to approximately
INTVCC each time the bottom power MOSFET is turned
on. The charge on this capacitor is then used to supply
the required transient current during the remainder of the
switching cycle. When the top MOSFET is turned on, the
BOOST pin voltage will be equal to approximately VIN +
INTVCC. For most applications a 0.1μF ceramic capacitor
will provide adequate performance.
An internal switch is used to charge the boost capacitor
when the synchronous MOSFET is turned on. An external
Schottky diode can be connected between BOOST and
INTVCC in parallel with this switch in order to improve
the capacitor refresh. For best performance and sufficient
design margin an external diode must be used in circuits
where VOUT is programmed to be above 12V or the IC
operates at a die temperature above 85°C. Forward currents through this diode are small, on the order of 10mA
to 20mA, but the diode chosen must have low reverse
leakage current at the expected voltage and temperature.
36461f
13
LTC3646/LTC3646-1
Applications Information
The design example on the back page uses a DFLS1200
based on its low reverse leakage over the voltage and
temperature ratings of the LTC3646.
Output Voltage Programming
The LTC3646 will adjust the output voltage such that VFB
equals the reference voltage of 0.6V according to:
⎛ R1 ⎞
VOUT = 0.6V ⎜1+ ⎟
⎝ R2 ⎠
Minimum Off-Time/On-Time Considerations
The minimum off-time is the smallest amount of time that
the LTC3646 can turn on the bottom power MOSFET, trip
the current comparator and turn off the power MOSFET.
This time is typically 80ns. For the controlled on-time
current mode control architecture, the minimum off-time
limit imposes a maximum duty cycle of:
DC(MAX) = 1 – (fO • tOFF(MIN))
The desired output voltage is set by the appropriate selection of resistors R1 and R2 as shown in Figure 2. Choosing
large values for R1 and R2 will result in improved efficiency
but may lead to undesired noise coupling or phase margin
reduction due to stray capacitance at the VFB node. Care
should be taken to route the FB line away from any noise
source, such as the SW line.
where fO is the switching frequency and tOFF(MIN) is the
minimum off-time. If the maximum duty cycle is surpassed,
due to a dropping input voltage for example, the output
will drop out of regulation. The minimum input voltage to
avoid this dropout condition is:
VIN(MIN) =
(
VOUT
1− fO • tOFF(MIN)
)
When programming output voltages above 12V, a Schottky
diode connected between BOOST and INTVCC is needed
(see the Boost Capacitor and Diode section.)
If there is concern about operating near the minimum
off-time limits, consider reducing the frequency to add
margin to the design.
To improve the frequency response of the main control
loop a feedforward capacitor, CF, may be used as shown
in Figure 2.
Conversely, the minimum on-time is the smallest duration of time in which the top power MOSFET can be in
its “on” state. This time is typically 30ns. In continuous
mode operation, the minimum on-time limit imposes a
minimum duty cycle of:
VOUT
R1
CF
VFB
LTC3646
R2
SGND
3646 F02
Figure 2. Optional Feedforward Capacitor
DC(MIN) = (fO • tON(MIN))
where tON(MIN) is the minimum on-time. As the equation
shows, reducing the operating frequency will alleviate the
minimum duty cycle constraint. In the rare cases where the
minimum duty cycle is surpassed, the output voltage will
still remain in regulation, but the switching frequency will
decrease from its programmed value. This is an acceptable
result in many applications, so this constraint may not be
of critical importance in some cases, and high switching
frequencies may be used in the design without any fear
of severe consequences. As the sections on Inductor and
Capacitor Selection show, high switching frequencies allow
the use of smaller board components, thus reducing the
footprint of the application circuit.
36461f
14
LTC3646/LTC3646-1
Applications Information
Minimum on-time can be affected by the output load current and the trough current level. During the transition
between the top switch turn-off and the synchronous
switch turn-on, the inductor current discharges the SW
pin capacitance. When the inductor trough current level
is low, or reversing in forced continuous operation, the
minimum on-time can increase by approximately 20nS.
ITH
RCOMP
LTC3646
CBYP
CCOMP
SGND
3634 F03
Figure 3. Compensation and Filtering Components
Output Voltage Limits
The Block Diagram shows that a sample of the output
voltage (taken through the VON pin) is used to servo the
correct on-time for a given application duty cycle and
frequency. This circuit limits the range of VOUT over which
the LTC3646 will be able to adjust the on-time in order to
match the selected frequency at a given duty cycle. The
valid output range for the LTC3646 is 2.0V to 30V. For
output voltage below 2.0V, use the LTC3646-1 which has
a valid output range of between 0.6V and 15V.
It is important to note that the LTC3646 will maintain
output voltage regulation if these limits are exceeded,
but the tON(MIN) limit may be reached resulting in the
part switching at a frequency lower than the programmed
switching frequency.
Choosing Compensation Components
Loop compensation is a complicated subject and Application Note 76 is recommended reading for a full discussion
on maximizing loop bandwidth in a current mode switching regulator. This section will provide a quick method on
choosing proper components to compensate the LTC3646
regulators.
Figure 3 shows the recommended components to be connected to the ITH pin, and Figure 4 shows an approximate
bode plot of the buck regulator loop using these components. It is assumed that the major poles in the system
(the output capacitor pole and the error amplifier output
pole) are located at a frequency lower than the crossover
frequency.
|H(s)|
–2
–1
ƒP
0dB
ƒZ
ƒC
LOG (ƒ)
3646 F04
Figure 4. Bode Plot of Regulator Loop
The first step is to choose the crossover frequency fC.
Higher crossover frequencies will result in a faster loop
transient response; however, in order to avoid higher
order loop dynamics from the switching power stage, it is
recommended that fC not exceed one-tenth the switching
frequency (fO).
Once fC is chosen, the value of RCOMP that sets this crossover frequency can be calculated by the following equation:
⎛ 2π • fC • COUT ⎞⎛ V ⎞
⎟⎜ OUT ⎟
RCOMP = ⎜⎜
⎟
⎝ gm(EA) • gm(MOD) ⎠⎝ VFB ⎠
where gm(EA) is the error amplifier transconductance
(see the Electrical Characteristics section), and gm(MOD)
is the modulator transconductance (the transfer function
from ITH voltage to current comparator threshold). For
the LTC3646, this transconductance is nominally 1Ω–1 at
room temperature.
36461f
15
LTC3646/LTC3646-1
Applications Information
Once RCOMP is determined, CCOMP can be chosen to set
the zero frequency (fZ):
1
fZ =
2π • CCOMP • RCOMP
For maximum phase margin, fZ should be chosen to be
less than one-tenth of fC.
Since the ITH node is sensitive to noise coupling, a small
bypass capacitor (CBYP) may be used to filter out board
noise. However, this cap contributes a pole at fP and may
introduce some phase loss at the crossover frequency:
1
fP =
2π • C BYP • RCOMP
For best results, fP should be set high enough such that
phase margin is not significantly affected.
If necessary, a capacitor CF (as shown in Figure 2) may
be used to add some phase lead.
Though better load transient response can generally be
achieved with external compensation, at switching frequencies above 1MHz, component count can be reduced
by connecting the ITH pin to INTVCC enabling internal
compensation. When using internal compensation, a
reasonable starting point for the minimum amount of
output capacitance necessary for stability can be found
as the greater of 15µF or COUT defined by the equation:
COUT > 3 • 10–5/VOUT
Checking Transient Response
The regulator loop response can be checked by observing
the response of the system to a load step. When configured
for external compensation, the availability of the ITH pin
not only allows optimization of the control loop behavior
but also provides a DC-coupled and AC-filtered closed-loop
response test point. The DC step, rise time, and settling
behavior at this test point reflect the system’s closedloop response. Assuming a predominantly second order
system, the phase margin and/or damping factor can be
estimated by observing the percentage of overshoot seen
at this pin. Use a high impedance, low capacitance probe
(>50MΩ, <5pF). The ITH external components shown in
Figure 3 will provide an adequate starting point for most
applications. The series R-C filter sets the pole-zero loop
compensation. The values can be modified from their
suggested values once the final PC layout is done, and the
particular switching frequency, output capacitor type and
value have been chosen. The output capacitors need to be
selected because their various types and values determine
the loop feedback factor gain and phase. An output current pulse of 20% to 100% of full load current with a rise
time of 1μs to 10μs will produce output voltage and ITH
pin waveforms that will give a sense of the overall loop
stability without breaking the feedback loop.
When observing the response of VOUT to a load step, the
initial output voltage step may not be within the bandwidth
of the feedback loop. As a result, the standard second
order overshoot/DC ratio cannot be used to estimate
phase margin. The output voltage settling behavior is
related to the stability of the closed-loop system and
will demonstrate the actual overall supply performance.
For a detailed explanation of optimizing the compensation
components, including a review of control loop theory,
refer to Application Note 76. As shown in Figure 2 a feedforward capacitor, CF, may be added across feedback
resistor R1 to improve the high frequency response of
the system. Capacitor CF provides phase lead by creating
a high frequency zero with R1.
In some applications severe transients can be caused by
switching in loads with large (>10μF) input capacitors. The
discharged input capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this output droop if the
switch connecting the load has low resistance and is driven
quickly. The solution is to limit the turn-on speed of the
load switch driver. A Hot Swap™ controller is designed
specifically for this purpose and usually incorporates current limit, short-circuit protection and soft-start functions.
MODE/SYNC Operation
The MODE/SYNC pin is a multipurpose pin allowing both
mode selection and operating frequency synchronization.
Connecting this pin to INTVCC enables Burst Mode operation
for superior efficiency at low load currents at the expense
of slightly higher output voltage ripple. When the MODE/
SYNC pin is pulled to ground, forced continuous mode
36461f
16
LTC3646/LTC3646-1
Applications Information
operation is selected creating the lowest fixed output ripple
at the expense of light load efficiency.
The LTC3646 will detect the presence of the external
clock signal on the MODE/SYNC pin and synchronize the
internal oscillator to the phase and frequency of the incoming clock. The presence of an external clock will place
the LTC3646 into forced continuous mode operation. For
proper on-time, connect a resistor corresponding to the
SYNC frequency between the RT pin and SGND (see the
Operating Frequency section). The user should be aware
that a clock with fast edges may drive this pin below the
–0.3V rating of this pin and an R-C filter may be needed
to prevent this condition.
Soft-Start
Soft-start on the LTC3646 is implemented by internally
ramping the reference signal fed to the error amplifier
over approximately a 250µs period. Figure 5 shows the
behavior of the regulator during start-up.
During the soft-start period, the inductor current is not
allowed to reverse and discontinuous operation may occur
under light load conditions.
Output Power Good
The PGOOD output of the LTC3646 is driven by a 63Ω
(typical) open-drain pull-down device. This pin will become
high impedance once the output voltage is within 5% (see
VPGOOD thresholds) of the target regulation point allowing the voltage at PGOOD to rise via an external pull-up
resistor (100k typical). If the output voltage exits a 7.5%
(see VPGOOD thresholds) regulation window around the
target regulation point the open-drain output will pull down
with 63Ω output resistance to ground, thus dropping
the PGOOD pin voltage. A filter time of 30μs (typical at
fO = 2.25MHz) acts to prevent unwanted PGOOD output
changes during VOUT transient events. This filter time
varies as a function of programmed switching period. The
output voltage must exit the 7.5% regulation window for
approximately 70 switching cycles before the PGOOD pin
pulls to ground (see Figure 6).
NOMINAL OUTPUT
PGOOD
VOLTAGE
VOUT
–7.5% –5% 0%
RUN
5V/DIV
5%
7.5%
3646 F06
Figure 6. PGOOD Pin Behavior
IL
250mA/DIV
VOUT
5V/DIV
ILOAD = 50mA
100µs/DIV
3646 F05
Figure 5. Start-Up Waveform
36461f
17
LTC3646/LTC3646-1
Applications Information
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
% Efficiency = 100% – (L1 + L2 + L3 +…)
where L1, L2, etc. are the individual loss terms as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, three main sources account for the majority of the
losses in the LTC3646: 1) I2R loss, 2) switching losses
and quiescent current loss, 3) transition losses and other
system losses.
1. I2R loss is calculated from the DC resistance of the
internal switches, RSW, and external inductor, RL. In
continuous mode, the average output current will flow
through inductor L but is chopped between the internal
top and bottom power MOSFETs. Thus, the series resistance looking into the SW pin is a function of both the
top and bottom MOSFET’s RDS(ON) and the duty cycle
(DC) as follows:
RSW = (RDS(ON)TOP)(DC) +(RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the curves in the Typical Performance
Characteristics section. Thus to obtain I2R loss:
I2R Loss = IOUT 2 • (RSW + RL)
2. The internal LDO supplies the power to the INTVCC rail.
The total power loss here is the sum of the switching
losses and quiescent current losses from the control
circuitry.
Each time a power MOSFET gate is switched from low
to high to low again, a packet of charge dQ moves
from VIN to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than
the DC control bias current. In continuous mode,
IGATECHG = fO(QT + QB), where QT and QB are the gate
charges of the internal top and bottom power MOSFETs
and fO is the switching frequency. For estimation
purposes, (QT + QB) on the LTC3646 is approximately
5nC. To calculate the total power loss from the LDO
load, simply add the gate charge current and quiescent
current and multiply by VIN:
PLDO = (IGATECHG + IQ) • SVIN
As will be discussed below, in certain cases the overall
efficiency can be improved by supplying the gate and
quiescent current through the EXTVCC pin.
3. Other hidden losses such as transition loss, copper trace
resistances, and internal load currents can account for
additional efficiency degradations in the overall power
system. Transition loss arises from the brief amount of
time the top power MOSFET spends in the saturated
region during switch node transitions. Other losses,
including diode conduction losses during dead time
and inductor core losses, generally account for less
than 2% total additional loss.
Transition loss can become significant at high VIN or high
switching frequencies. Transition loss for the LTC3646 can
be approximated by the following formula:
Loss (Watts) = IOUT • VIN2 • fO • 10–10
36461f
18
LTC3646/LTC3646-1
Applications Information
Thermal Considerations
The LTC3646 requires the exposed package backplane
metal (PGND) to be well soldered to the PC board to provide good thermal contact. This gives the QFN and MSOP
packages exceptional thermal properties, compared to
other packages of similar size, making it difficult in normal
operation to exceed the maximum junction temperature
of the part. In many applications, the LTC3646 does not
generate much heat due to its high efficiency. However,
in applications in which the LTC3646 is running at a high
ambient temperature, high input voltage, high switching
frequency, and maximum output current, the heat dissipated may cause the part to exceed the maximum allowed
junction temperature. If the junction temperature reaches
approximately 175°C, both power switches will be turned
off until temperature decreases approximately 10°C.
Thermal analysis should always be performed by the user
to ensure the LTC3646 does not exceed the maximum
junction temperature.
The temperature rise is given by:
tRISE = PD θJA
where PD is the power dissipated by the regulator and
θJA is the thermal resistance from the junction of the
die to the ambient. Consider the example in which an
LTC3646IDE-1 is operating with IOUT = 1.0A, SVIN = 12V,
f = 3MHz, VOUT = 1.8V, and a board temperature of 85°C.
From the Typical Performance Characteristics section, the
RDS(ON) of the top switch is found to be nominally 200mΩ
while that of the bottom switch is nominally 120mΩ
yielding an equivalent power MOSFET resistance RSW of:
R DS(ON)TOP • 1.8 R DS(ON)BOT • 10.2
+
= 132mΩ
12
12
From the previous section, IGATECHG + IQ is ~15mA when
f = 3MHz. Therefore, the total power dissipation due to
resistive losses and LDO losses is:
PD = IOUT 2 • RSW + SVIN • (IGATECHG + IQ)
PD = (1.0)2 • (0.132) + 12V • 15mA = 312mW
and the transition loss is:
PT = 1.0 • 122 • 10–10 • 3.0 • 106 = 43mW
The DFN 4mm × 3mm package junction-to-ambient thermal
resistance, θJA, is approximately 43°C/W. Therefore, the
junction temperature of the regulator operating in a 85°C
ambient temperature is approximately:
TJ = (0.312 + 0.043) • 43 + 85 = 100°C
which is below the specified maximum junction temperature of 125°C.
High input voltage, high frequency applications may cause
the internal LDO to generate significant heat. The INTVCC
current, which is dominated by the gate charge current,
may be supplied by either the SVIN LDO or through the
EXTVCC pin. When the voltage on the EXTVCC pin is less
than 4.5V, the VIN LDO is enabled. Power dissipation for the
IC in this case is highest and is equal to SVIN • IINTVCC. The
gate charge current is dependent on operating frequency
as discussed in the Efficiency Considerations section. For
example, the LTC3646 INTVCC current is approximately
15mA at 3MHz operation. If VIN is at the 40V maximum,
the loss in the on-chip LDO is:
40V • 0.015A = 0.60W
In these situations it will be advantageous to bias the
part through the EXTVCC pin if a suitable voltage source
is available. When the voltage applied to EXTVCC rises
above 4.5V (maximum 6.0V), the SVIN LDO is turned off
and an internal switch between the EXTVCC and INTVCC
pins is closed. This voltage is unregulated and so in this
situation, INTVCC = EXTVCC.
Using EXTVCC allows the control power to be derived from
the output if the output voltage is between 4.5V and 6.0V
during normal operation and from the SVIN LDO when the
output is out of regulation (e.g., start-up, short-circuit).
Significant efficiency and thermal gains can be realized by
powering INTVCC from the output, since the power needed
for the driver and control currents will be supplied from
the buck converter instead of the internal linear regulator.
For 4.5V to 6V regulator outputs, this means connecting
the EXTVCC pin directly to VOUT. Tying the EXTVCC pin to a
5.5V supply reduces the dissipated power in the previous
example from 0.60W to approximately:
5.5V • 0.015A = 82.5mW
36461f
19
LTC3646/LTC3646-1
Applications Information
The following list summarizes the three possible connections for EXTVCC:
1. EXTVCC grounded. This will cause INTVCC to be powered
from the internal 5.0V regulator.
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 4.5V to 6V regulated output and provides the highest efficiency.
3. EXTVCC connected to an external supply. If an external
supply is available in the 4.5V to 6V range, it may be
used to power EXTVCC. Ensure that EXTVCC < VIN.
Board Layout Considerations
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3646.
1. Does the capacitor CPVIN connect to PVIN and PGND
as close to the pins as possible? These capacitors
provide the AC current to the internal power MOSFETs
and drivers. The (–) plate of CPVIN should be closely
connected to PGND and the (–) plate of COUT.
2. The output capacitor, COUT, and inductor L1 should
be closely connected to minimize loss. The (–) plate
of COUT should be closely connected to PGND and the
(–) plate of CIN.
3. The resistive divider, R1 and R2, must be connected
between the (+) plate of COUT and a ground line terminated near SGND. The feedback signal, VFB , should be
routed away from noisy components and traces such as
the SW line, and its trace length should be minimized.
In addition, RT and the loop compensation components
should be terminated to SGND.
4. Keep sensitive components away from the SW pin. The
RRT resistor, the feedback resistors, the compensation
components, and the INTVCC bypass capacitor should
all be routed away from the SW trace and the inductor.
5. A ground plane is preferred, but if not available the
signal and power grounds should be segregated with
both connecting to a common, low noise reference
point. The point at which the ground terminals of the
VIN and VOUT bypass capacitors are connected makes a
good, low noise reference point. The connection to the
PGND pin should be made with a minimal resistance
trace from the reference point.
6. Flood all unused areas on all layers with copper in order
to reduce the temperature rise of power components.
These copper areas should be connected to the exposed
backside connection of the IC.
36461f
20
LTC3646/LTC3646-1
Applications Information
RSVIN
R2
CSVIN
R1
CF
CCOMP
CVCC
RCOMP
RRT
CBOOST
RPGOOD
SW
GND
CPVIN
PVIN
VIA TO VOUT
VIA TO INTVCC
VIA TO PVIN
VIA TO GND
COUT
L1
VOUT
3646 F07
Figure 7. Example of Power Component Layout for DFN Package.
Because of the Similar Pinout, MSE Package Layout is Similar
36461f
21
LTC3646/LTC3646-1
Applications Information
PVIN
BOOST
PVIN
RSVIN
SVIN
SW
INTVCC
RPGOOD
CIN
10µF
CBOOST
LTC3646
RUN
L1
VOUT
PGOOD
ITH
RT
CSVIN
MODE/SYNC
RCOMP
EXTVCC
CVCC
RRT
CCOMP
0.6V
SGND
COUT
SWITCH VON
CONTROL
R1
+ –
CF
VFB
PGND
R2
3646 F08
Figure 8. Board Layout Schematic
36461f
22
LTC3646/LTC3646-1
Applications Information
Design Example
As a design example, consider using the LTC3646-1 in an
application with the following specifications: VIN = 12V,
VOUT = 1.8V, IOUT(MAX) = 1A, IOUT(MIN) = 10mA.
Because efficiency is important at both high and low load
currents, Burst Mode operation and 1MHz operation is
chosen.
First the correct RRT resistor value for 1MHz switching
frequency must be chosen. Based on the equation discussed previously, the closest standard resistor value
for RRT is 90.9k.
Next, determine the inductor value for approximately 35%
ripple current using:
Next COUT is selected based on the required output transient
performance and the required ESR to satisfy the output
voltage ripple. Using a 15µF ceramic capacitor with an
ESR of 5mΩ will result in approximately 5mV of ripple.
Decoupling the PVIN pin with a 22µF capacitor and the
SVIN pin with a 1µF capacitor should be adequate for
most applications. A 0.1µF boost capacitor should also
work for most applications.
To save board space the ITH pin is connected to INTVCC
to select internal compensation. Since the switching and
Q current drawn from the 12V supply is not a significant
source of loss or heat, EXTVCC is disabled by tying the
pin to SGND.
⎛
⎞ ⎛ 1.8V ⎞
1.8V
L=⎜
⎟ ⎜1−
⎟ = 4.37µH
⎝ 1MHz • 350mA ⎠ ⎝ 12V ⎠
A standard 4.7µH inductor would work well for this application.
12V
SVIN
CSVIN
1µF
PVIN
RUN
LTC3646-1
BOOST
INTVCC
RPG
100k
CVCC
4.7µF
RRT
90.9k
MODE/SYNC
CBOOST LOUT
0.1µF
4.7µH
VOUT
SW
ITH
PGOOD
VON
RT
VFB
R1
150k
CF
1pF
COUT
15µF
R2
75k
EXTVCC
SGND
CPVIN
22µF
PGND
3646 F09
Figure 9
36461f
23
LTC3646/LTC3646-1
Typical Applications
12V to 1.8V Output with 400kHz External Sync
VIN
12V
BOOST
PVIN
2Ω
0.1µF
SVIN
LTC3646
RUN
SW
INTVCC
L1
10µH
VOUT
1.8V
22µF
ITH
RT
RCOMP
49.9k
2.2µF
MODE/SYNC
4.7µF
226k
400kHz
EXTERNAL
CLOCK
EXTVCC
0.6V
100k
CCOMP
220pF
SWITCH VON
CONTROL
SGND
274k
+ –
PGND
CF
4.7pF
100µF
VFB
137k
3646 TA02a
L1: WÜRTH WE-TPC-744-065-100
Efficiency Curve
50mA to 1A Load Step
100
90
ILOAD
1A/DIV
EFFICIENCY (%)
80
70
IL
1A/DIV
60
50
VOUT
100mV/DIV
AC-COUPLED
40
30
20
40µs/DIV
10
0
0.001
0.1
0.01
LOAD CURRENT (A)
3646 TA02c
1
3646 TA02b
36461f
24
LTC3646/LTC3646-1
Typical Applications
24V Input to 5V Output at 1MHz Frequency and EXTVCC
VIN
24V
BOOST
PVIN
2Ω
SVIN
LTC3646
RUN
SW
INTVCC
100k
22µF
0.1µF
L1
10µH
VOUT
5V
PGOOD
ITH
RT
2.2µF
RCOMP
100k
MODE/SYNC
EXTVCC
4.7µF
90.9k
CCOMP
33pF
0.6V
SGND
COUT
30µF
SWITCH VON
CONTROL
487k
+ –
20pF
VFB
PGND
66.5k
3646 TA03a
Efficiency Curve
50mA to 1A Load Step
100
90
ILOAD
1A/DIV
EFFICIENCY (%)
80
70
IL1
1A/DIV
60
50
40
VOUT
200mV/DIV
AC-COUPLED
30
20
40µs/DIV
10
0
0.001
0.1
0.01
LOAD CURRENT (A)
3646 TA03c
1
3646 TA03b
36461f
25
LTC3646/LTC3646-1
Package Description
DE Package
14-Lead Plastic DFN (4mm × 3mm)
(Reference LTC DWG # 05-08-1708 Rev B)
0.70 ±0.05
3.30 ±0.05
3.60 ±0.05
2.20 ±0.05
1.70 ± 0.05
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
3.00 REF
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
4.00 ±0.10
(2 SIDES)
R = 0.05
TYP
3.00 ±0.10
(2 SIDES)
R = 0.115
TYP
8
0.40 ± 0.10
14
3.30 ±0.10
1.70 ± 0.10
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
PIN 1
TOP MARK
(SEE NOTE 6)
(DE14) DFN 0806 REV B
7
0.200 REF
1
0.25 ± 0.05
0.50 BSC
0.75 ±0.05
3.00 REF
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WGED-3) IN JEDEC
PACKAGE OUTLINE MO-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
36461f
26
LTC3646/LTC3646-1
Package Description
MSE Package
16-Lead Plastic
, Exposed Die Pad
MSEMSOP
Package
(Reference LTC DWG # 05-08-1667 Rev E)
16-Lead
Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev E)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
5.23
(.206)
MIN
2.845 ±0.102
(.112 ±.004)
0.889 ±0.127
(.035 ±.005)
8
1
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102 3.20 – 3.45
(.065 ±.004) (.126 – .136)
0.305 ±0.038
(.0120 ±.0015)
TYP
16
0.50
(.0197)
BSC
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 ±0.076
(.011 ±.003)
REF
16151413121110 9
DETAIL “A”
0° – 6° TYP
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
4.90 ±0.152
(.193 ±.006)
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
1234567 8
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.86
(.034)
REF
0.1016 ±0.0508
(.004 ±.002)
MSOP (MSE16) 0911 REV E
36461f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC3646/LTC3646-1
Typical Application
Efficiency
D1: DFLS 1200
100
LTC3646
RUN
SW
INTVCC
100k
80
0.1µF
SVIN
22µF
90
BOOST
PVIN
2Ω
L1
47µH
VOUT
28V
PGOOD
ITH
RT
2.2µF
RCOMP
402k
MODE/SYNC
SWITCH VON
CONTROL
EXTVCC
4.7µF
180k
0.6V
CCOMP
220pF
COUT
33µF
SGND
60
50
40
30
10
0
0.001
VFB
PGND
70
20
487k
+ –
EFFICIENCY (%)
VIN
40V
10.7k
Burst Mode OPERATION
FORCED CONTINUOUS
0.1
0.01
LOAD CURRENT (A)
1
3646 TA04b
3646 TA04a
L1: WÜRTH WE-TPC-744-066-470
Related Parts
PART
NUMBER
DESCRIPTION
COMMENTS
LTC3642
45V Input Capable with 60V Transient Protection, 50mA, Synchronous
Micropower Step-Down DC/DC Converter with IQ = 12µA
VIN(MIN) = 4.5V, VIN(MAX) = 45V, 60V Transient, VOUT(MIN) = 0.8V,
IQ = 12µA, ISD < 1µA, 3mm × 3mm DFN-8, MSOP-8E Packages
LTC3631
45V Input Capable with 60V Transient Protection, 100mA, Synchronous VIN(MIN) = 4.5V, VIN(MAX) = 45V, 60V Transient, VOUT(MIN) = 0.8V,
IQ = 12µA, ISD <1µA, 3mm × 3mm DFN-8, MSOP-8E Packages
Micropower Step-Down DC/DC Converter with IQ = 12µA
LTC3632
50V Input Capable with 60V Transient Protection, 20mA, Synchronous
Micropower Step-Down DC/DC Converter with IQ = 12µA
VIN(MIN) = 4.5V, VIN(MAX) = 45V, 60V Transient, VOUT(MIN) = 0.8V,
IQ = 12µA, ISD <1µA, 3mm × 3mm DFN-8, MSOP-8E Packages
LTC3601
15V, 1.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter
VIN(MIN) = 4.5V, VIN(MAX) = 15V, VOUT(MIN) = 0.6V, IQ = 300µA,
ISD <1µA, 4mm × 4mm QFN-20, MSOP-16E Packages
LTC3603
15V, 2.5A (IOUT), 3MHz, Synchronous Step-Down DC/DC Converter
VIN(MIN) = 4.5V, VIN(MAX) = 15V, VOUT(MIN) = 0.6V, IQ = 75µA,
ISD <1µA, 4mm × 4mm QFN-20, MSOP-16E Packages
LT3991
55V, 1.2A, 2.2MHz High Efficiency Micropower Step-Down DC/DC
Converter with IQ = 2.8µA
VIN(MIN) = 4.3V, VIN(MAX) = 38V, VOUT(MIN) = 1.2V, IQ = 2.8µA,
ISD <1µA, 3mm × 3mm DFN-10, MSOP-10E Packages
LT3682
36V, 60VMAX, 1A, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter
VIN(MIN) = 3.6V, VIN(MAX) = 36V, VOUT(MIN) = 0.8V, IQ = 75µA,
ISD <1µA, 3mm × 3mm DFN-12 Package
LT3689
36V, 60V Transient Protection, 800mA, 2.2MHz High Efficiency
Micropower Step-Down DC/DC Converter with POR Reset and
Watchdog Timer
VIN(MIN) = 3.6V, VIN(MAX) = 36V, Transient to 60V, VOUT(MIN) = 0.8V,
IQ = 75µA, ISD <1µA, 3mm × 3mm QFN-16 Package
LT3480
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN(MIN) = 3.6V, VIN(MAX) = 36V, Transient to 60V, VOUT(MIN) =
0.78V,
IQ = 70µA, ISD <1µA, 3mm × 3mm DFN-10, MSOP-10E Packages
LT3980
58V with Transient Protection to 80V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN(MIN) = 3.6V, VIN(MAX) = 58V, Transient to 80V, VOUT(MIN) =
0.78V, IQ = 85µA, ISD <1µA, 3mm × 4mm DFN-16, MSOP-16E
Packages
LT8610/
LT8611
42V, 2.5A, 96% Efficiency, 2.2MHz Synchronous Micropower
Step-Down DC/DC Converter with IQ = 2.5µA
VIN = 3.4V to 42V, VOUT(MIN) = 0.97V, IQ = 2.5µA, ISD <1µA,
MSOP-16E Package
36461f
28 Linear Technology Corporation
LT 1212 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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