EUA5202 2-W Stereo Audio Power Amplifier with Mute DESCRIPTION FEATURES z The EUA5202 is a stereo audio power amplifier that delivering 2W of continuous RMS power per channel into 3-Ω loads. When driving 1W into 8-Ω speakers, the EUA5202 has less than 0.04% THD+N across its specified frequency range. Included within this device is integrated de-pop circuitry that virtually eliminates transients that cause noise in the speakers. Amplifier gain is externally configured by means of two resistors per input channel and does not require external compensation for settings of 2 to 20 in BTL mode (1 to 10 in SE mode). An internal input MUX allows two sets of stereo inputs to the amplifier. In notebook applications, where internal speakers are driven as BTL and the line (often headphone drive) outputs are required to be SE, the EUA5202 automatically switches into SE mode when the SE/BTL inputs is activated. Consume only 7mA of supply current during normal operation, and the EUA5202 also features a shutdown function for power sensitive applications, holding the supply current at 1µA. z z z z z z z z RoHS Compliant and 100% Lead (Pb)-Free APPLICATIONS z Notebook Computers z Multimedia Monitors z Digital Radios and Portable TVs Block Diagram DS5202 Ver 1.6 May. 2005 Output Power at 3Ω Load - 2W/ch at VDD=5V - 800mW/ch at 3V Low Supply Current and Shutdown Current Integrated Depop Circuit Mute and Shutdown Control Function Thermal Shutdown Protection Stereo Input MUX Bridge-Tied Load (BTL) or Single-Ended (SE) Modes. TSSOP-24 with Thermal Pad 1 EUA5202 Typical Application Circuit Figure 1. EUA5202 Minimum Configuration Application Circuit Figure 2. EUA5202 Full Configuration Application Circuit DS5202 Ver 1.6 May. 2005 2 EUA5202 Pin Configurations Package Pin Configurations TSSOP-24 with Thermal Pad, exposure on the bottom of the package Pin Description PIN PIN I/O HP/ LINE 16 LBYPASS 6 LHPIN 5 I LLINE IN LOUT+ LOUT- I O O LVDD MUTE IN MUTE OUT NC RBYPASS 4 3 10 1,12,13, 24 7 11 9 17,23 19 RHPIN 20 I RLINEIN ROUT+ ROUTRVDD SE/BTL SHUTDOWN 21 22 15 18 14 8 I O O I I I TJ 2 O GND/HS DS5202 Ver 1.6 May. 2005 I DESCRIPTION Input MUX control input, hold high to select LHP IN or RHP IN (5, 20), hold low to select LLINE IN or RLINE IN (4, 21) Tap to voltage divider for left channel internal mid-supply bias Left channel headphone input, selected when HP/LINE terminal (16) is held high Left channel line input, selected when HP/LINE terminal (16) is held low Left channel + output in BTL mode, + output in SE mode Left channel - output in BTL mode, high-impedance state in SE mode Ground connection for circuitry, directly connected to thermal pad I I O Supply voltage input for left channel and for primary bias circuits Mute all amplifiers, hold low for normal operation, hold high to mute Follows MUTE IN terminal (11), provides buffered output No internal connection Tap to voltage divider for right channel internal mid-supply bias Right channel headphone input, selected when HP/LINE terminal (16) is held high Right channel line input, selected when HP/LINE terminal (16) is held low Right channel + output in BTL mode, + output in SE mode Right channel - output in BTL mode, high-impedance state in SE mode Supply voltage input for high channel Hold low foe BTL mod, hold high for SE mode Places entire IC in shutdown mode when held high, IDD=5µA Sources a current proportional to the junction temperature. This terminal should be left unconnected during normal operation. 3 EUA5202 Ordering Information Order Number Package Type Marking Operating Temperature range EUA5202QIR1 TSSOP 24 xxxx EUA5202 -40°C to 85°C EUA5202QIR0 TSSOP 24 xxxx EUA5202 -40°C to 85°C EUA5202QIT1 TSSOP 24 xxxx EUA5202 -40°C to 85°C EUA5202QIT0 TSSOP 24 xxxx EUA5202 -40°C to 85°C EUA5202 □ □ □ □ Lead Free Code 1: Lead Free 0: Lead Packing R: Tape& Reel T: Tube Operating temperature range I: Industry Standard Package Type Q: TSSOP DS5202 Ver 1.6 May. 2005 4 EUA5202 Absolute Maximum Ratings Supply voltage , VDD------------------------------------------------------------------------------------------6V Input voltage, V1----------------------------------------------------------------------------0.3V to VDD+0.3V Continuous total power dissipation------------------------------------------------------------internally limited Operating free-air temperature range ,TA ------------------------------------------------------- –40°C to 85°C Operating junction temperature range, TJ ------------------------------------------------------- –40°C to 150°C Storage temperature range, Tstg ---------------------------------------------------------------–65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds-------------------------------------260°C Recommended Operating Conditions MIN NOM MAX UNIT Supply Voltage, VDD Operating free-air temperature, TA Common mode input voltage, VICM 3 5 5.5 VDD = 5V, 250m W/Ch average power, 4-Ω stereo BTL drive, with proper PCB design -40 VDD = 5V, 2 W/Ch average power, 3-Ω stereo BTL drive, with proper PCB design -40 and 300 CFM forced-air cooling 85 VDD = 5V 1.25 4.5 VDD = 3.3V 1.25 2.7 V 85 °C V DC Electrical Characteristics, TA=25°C Symbol Parameter Conditions Stereo BTL Stereo SE Stereo BTL Stereo SE 7.1 3.9 5.7 3.1 11 6 9 5 mA mA mA mA VDD=5V, Gain=2 5 25 mV VDD=5V 1.55 VDD=5V 1 VDD=5V IDD Supply Current VDD=3.3V VOO IDD (Mute) Output Offset Voltage (measured differentially) Supply Current in Mute Mode IDD(SD) DS5202 Ver 1.6 May. 2005 EUA5202 Unit Min. Typ Max. IDD in Shutdown 5 mA 5 µA Typical Ac Operating Characteristics, VDD=5V, TA=25°C, RL=3Ω Symbol PO THD+N BOM Parameter Output Power(each channel)*1 Total Harmonic Distortion Plus Noise Maximum Output Power Bandwidth Supply Ripple Rejection Ratio Z1 VN Mute Attenuation Channel-to- Channel Output Separation Line/HP Input Separation BTL Attenuation in SE Mode Input Impedance Signal-to-Noise Ratio Output Noise Voltage EUA5202 Conditions Typ. Unit THD=0.2%, BTL, See Figure 3 THD=1%, BTL, See Figure 3 2 2.2 W PO=2W, f=1KHz ,See Figure5 200 m% V1=1V, RL=10kΩ, AV=1V/V 100 m% AV=10V/V THD <1%, See Figure5 >20 KHz f=1KHz, See Figure37 65 f=20-20KHz, See Figure37 40 dB 85 dB 85 dB 88 dB 86 dB 2 101 22 MΩ dB µV(rms) Conditions Typ. Unit THD=0.2%, BTL, See Figure 10 THD=1%, BTL, See Figure 10 800 900 W PO=2W, f=1KHz ,See Figure11 350 m% V1=1V, RL=10kΩ, AV =1V/V 200 m% AV=10V/V THD <1%, See Figure11 >20 KHz f=1KHz, See Figure37 60 f=20-20KHz, See Figure37 40 f=1KHz, See Figure 39 PO=2W,BTL, 5V See Figure 35 *1: Output Power is measured at the output terminals of the IC at 1 KHz Typical Ac Operating Characteristics, VDD=3.3V, TA=25°C, RL=3Ω Symbol PO THD+N BOM Parameter Output Power(each channel)*1 Total Harmonic Distortion Plus Noise Maximum Output Power Bandwidth Supply Ripple Rejection Ratio Z1 VN Mute Attenuation Channel-to- Channel Output Separation Line/HP Input Separation BTL Attenuation in SE Mode Input Impedance Signal - to - Noise Ratio Output Noise Voltage f=1KHz, See Figure 40 PO=700mW,BTL, 5V See Figure 36 *1: Output Power is measured at the output terminals of the IC at 1 KHz DS5202 Ver 1.6 May. 2005 6 dB 85 dB 80 dB 88 dB 86 dB 2 96 22 MΩ dB µV(rms) EUA5202 Typical Operating Characteristics (Table of Graphs) No Item Conditions Figure Page 1 THD+N vs. Output Power VDD=5V，RL=3 & 8 ohm，BTL，f=1KHz 3 9 2 THD+N vs. Frequency VDD=5V，RL=4 ohm，BTL，Po=1.5W f=20 to 20KHz，Av= -2 & -10 & -20V/V 4 9 3 THD+N vs. Frequency VDD=5V，RL=3 & 4 ohm，BTL，Po=1.5W ，f=20 to 20KHz 5 9 4 THD+N vs. Output Power VDD=5V，RL=3 ohm，BTL，f=20 & 1K & 20KHz 6 9 5 THD+N vs. Frequency VDD=5V，RL=8 ohm，BTL，f=20 to 20KHz，Av=-2V/V 7 9 6 THD+N vs. Output Power VDD=5V，RL=8 ohm，BTL，Po=1W， Av= -2 &-10 & -20V/V，f=20 to 20KHz 8 9 7 THD+N vs. Output Power VDD=5V，RL=8 ohm，BTL，f=20 & 1K & 20KHz 9 10 8 THD+N vs. Output Power VDD=3.3V，RL=3 & 8 ohm，BTL，f=1KHz 10 10 9 THD+N vs. Frequency 11 10 12 10 10 THD+N vs. Frequency VDD=3.3V，RL=4 ohm，BTL，Po=0.75W，Av= -2 &-10 &-20V/V， f=20 to 20KHz VDD=3.3V，RL=4 ohm，BTL，Av=-2V/V， Po=0.1 & 0.35 & 0.75W & 800mW(RL=3 ohm) 11 THD+N vs. Output Power VDD=3.3V，RL=3 ohm，BTL，Av=-2V/V，f=20 & 1K & 20KHz 13 10 12 THD+N vs. Frequency VDD=3.3V，RL=8 ohm，BTL，Po=0.4W ，Av=-2 &-10 & -20V/V 14 10 13 THD+N vs. Frequency VDD=3.3V，RL=8 ohm，BTL，Av=-2V/V ，Po=0.1 & 0.25 & 0.4W 15 11 14 THD+N vs. Output Power VDD=3.3V，RL=8 ohm，BTL，Av= -2V/V，f=20 & 1K &20KHz 16 11 15 THD+N vs. Frequency VDD=5V，RL=4 ohm，SE，Po=0.5W，Av= -1&-5&-10V/V 17 11 16 THD+N vs. Frequency VDD=5V，RL=4 ohm，SE，Av= -2V/V，Po=0.1 & 0.25 & 0.5W 18 11 17 THD+N vs. Output Power VDD=5V，RL=4 ohm，SE，Av= -2V/V，f=100 & 1K & 20KHz 19 11 18 THD+N vs. Frequency VDD=5V，RL=8 ohm，SE，Po=0.25W，Av= -1 &-5 &-10V/V 20 11 19 THD+N vs. Frequency VDD=5V，RL=8 ohm，SE，Av= -2V/V，Po=0.05 & 0.1 & 0.25W 21 12 20 THD+N vs. Output Power VDD=5V，RL=8 ohm，SE，Av= -2V/V，f=100 &1K & 20KHz 22 12 21 THD+N vs. Frequency VDD=5V，RL=32 ohm，SE，Po=0.075W，Av= -1 &-5 &-10V/V 23 12 22 THD+N vs. Frequency VDD=5V，RL=32 ohm，SE，Av= -1V/V，Po=25 & 50 & 75mW 24 12 23 THD+N vs. Output Power VDD=5V，RL=32 ohm，SE，Av= -1V/V，f=20 & 1K & 20KHz 25 12 24 THD+N vs. Frequency VDD=3.3V，RL=4 ohm，SE，Po=0.2W，Av= -1 &-5 &-10V/V 26 12 25 THD+N vs. Frequency VDD=3.3V，RL=4 ohm，SE，Av= -1V/V，Po=0.05 & 0.1 & 0.2W 27 13 26 THD+N vs. Output Power VDD=3.3V，RL=4 ohm，SE，Av= -2V/V，f=100 & 1K & 20KHz 28 13 27 THD+N vs. Frequency VDD=3.3V，RL=8 ohm，SE，Po=100mW，Av= -1 &-5 &-10V/V 29 13 28 THD+N vs. Frequency VDD=3.3V，RL=8 ohm，SE，Av= -1V/V，Po=25 & 50 &100mW 30 13 29 THD+N vs. Output Power VDD=3.3V，RL=8 ohm，SE，Av= -1V/V，f=100 & 1K & 20KHz 31 13 30 THD+N vs. Frequency VDD=3.3V，RL=32 ohm，SE，Po=30mW，Av= -1 &-5 &-10V/V 32 13 31 THD+N vs. Frequency VDD=3.3V，RL=32 ohm，SE，Av= -1V/V，Po=10 & 20 & 30mW 33 14 32 THD+N vs. Output Power VDD=3.3V，RL=32 ohm，SE，Av=-1V/V，f=20 & 1K & 20KHz 34 14 35 14 33 Output Noise Voltage vs. Frequency VDD=5V，BW=22Hz to 22kHz，RL=4 DS5202 Ver 1.6 May. 2005 7 EUA5202 34 Output Noise Voltage vs. Frequency VDD=3.3V，BW=22Hz to 22kHz，RL=4 Supply Ripple Rejection Ratio vs. 35 RL=4 ohm，CB=4.7uF，BTL，VDD=3.3 & 5V Frequency Supply Ripple Rejection Ratio vs. RL=4 ohm，CB=4.7uF，SE，VDD=3.3 & 5V 36 Frequency 36 14 37 14 38 14 37 Crosstalk vs .Frequency VDD=5V，Po=1.5W，RL=4 ohm，BTL，Right to Left & Left to Right 39 15 38 Crosstalk vs .Frequency VDD=3.3V，Po=0.75W，RL=4 ohm，BTL，Right to Left & Left to Right 40 15 39 Crosstalk vs .Frequency VDD=5V，Po=75mW，RL=32 ohm，SE，Right to Left & Left to Right 41 15 40 Crosstalk vs .Frequency VDD=3.3V，Po=35mW，RL=32 ohm，SE，Right to Left & Left to Right 42 15 41 Closed Loop Response VDD=5V，Av=-2V/V，Po=1.5W，BTL，Gain & Phase 43 15 42 Closed Loop Response VDD=3.3V，Av= -2V/V，Po=0.75W，BTL，Gain &Phase 44 15 43 Closed Loop Response VDD=5V，Av=-1V/V，Po=0.5W，SE，Gain &Phase 45 16 44 Closed Loop Response VDD=3.3V，Av= -1V/V，Po=0.25W，SE，Gain &Phase 46 16 45 Supply Current vs. Supply Voltage Stereo BTL & Stereo SE 47 16 46 Output Power vs. Supply Voltage THD+N=1%，BTL，Each Channel，RL=3 & 4 & 8 ohm 48 16 47 Output Power vs. Supply Voltage THD+N=1%，SE，Each Channel，RL=3 & 4 & 8 ohm 49 16 48 Output Power vs. Load Resistance THD+N=1%，BTL，Each Channel，VDD=3.3 & 5V 50 16 49 Output Power vs. Load Resistance THD+N=1%，SE，Each Channel，VDD=3.3 & 5V 51 17 50 Power Dissipation vs. Output Power VDD=5V，BTL，Each Channel，RL=3 & 4 & 8ohm 52 17 51 Power Dissipation vs. Output Power VDD=3.3V，BTL，Each Channel，RL=3 & 4 & 8ohm 53 17 52 Power Dissipation vs. Output Power VDD=5V，SE，Each Channel，RL=4 & 8 &32 ohm 54 17 53 Power Dissipation vs. Output Power VDD=3.3V，SE，Each Channel，RL=4 & 8 &32 ohm 55 17 DS5202 Ver 1.6 May. 2005 8 EUA5202 DS5202 Ver 1.6 May. 2005 Figure 3. Figure 4. Figure5. Figure6. Figure7. Figure8. 9 EUA5202 Figure9. Figure10. Figure11. Figure12. Figure13. DS5202 Ver 1.6 May. 2005 Figure14. 10 EUA5202 Figure15. DS5202 Ver 1.6 May. 2005 Figure16. Figure17. Figure18. Figure19. Figure20. 11 EUA5202 Figure21. Figure22. Figure23. Figure24. Figure25. DS5202 Ver 1.6 May. 2005 Figure26. 12 EUA5202 Figure27. Figure28. Figure29. Figure30. Figure31. DS5202 Ver 1.6 May. 2005 Figure32. 13 EUA5202 Figure33. DS5202 Ver 1.6 May. 2005 Figure34. Figure35. Figure36. Figure37. Figure38. 14 EUA5202 Figure39. Figure40. Figure41. Figure42. Figure43. DS5202 Ver 1.6 May. 2005 Figure44. 15 EUA5202 Figure45. Figure46. Figure47. Figure48. Figure49. DS5202 Ver 1.6 May. 2005 Figure50. 16 EUA5202 Figure51. Figure52. Figure53. Figure54. Figure55. DS5202 Ver 1.6 May. 2005 17 EUA5202 Application Information Input Capacitor, CI In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency determined in equation 4. Gain Setting Resistors, RF and RI The gain for each audio input of the EUA5202 is set by resistors by resistors RF and RI according to equation 1 for BTL mode. R F BTL Gain = −2 R I -------------------------------- (1) f BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the voltage swing across the load. Given that the EUA5202 is a MOS amplifier, the input impedance is very high, value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the amplifier be set between 5kΩ and 20kΩ .The effective impedance is calculated in equation 2. F I R +R F -------------------- (2) I 1 C = I 2π R f I C As an example consider an input resistance of 10kΩ and a feedback resistor of 50kΩ. The BTL gain of the amplifier would be -10 and the effective impedance at the inverting terminal would be 8.3kΩ, which is well within the recommended range. For high performance applications metal film resistors are recommended because they tent to have lower noise levels than carbon resistors. For values of RF above 50kΩ the amplifier tends to become unstable due to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small compensation capacitor of approximately 5pF should be places in parallel with RF when RF is greater than 50kΩ. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 3. 1 f (lowpass) = c 2π R C F F 1 2π R C I I ------------------- (4) ------------------------------------ (5) In this example, CI is 0.40 µF so one would likely choose a value in the range of 0.47µF to 1µF. A further consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher that the source dc level. Please note that it is important to confirm the capacitor polarity in the application. -------------------- (3) For example, if RF is 100kΩ and CF is 5 pF then fC is 318 KHz, which is well outside of the audio range. DS5202 Ver 1.6 May. 2005 = The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit. Consider the example where RI is 10kΩ and the specification calls for a flat bass response down to 40Hz. Equation 8 is reconfigured as equation 5. R R EffectiveImpedance = c(highpass ) 18 EUA5202 Power Supply Decoupling, CS The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives the low-frequency corner higher degrading the bass response. Large values of CC are required to pass low frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 3Ω, 4Ω, 8Ω, 32Ω, 10kΩ, to 47kΩ. Table 1 summarizes the frequency response characteristics of each configuration. The EUA5202 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent – series - resistance (ESR) ceramic capacitor, typically 0.1µF placed as close as possible to the device VDD lead works best. For filtering lower – frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audio power amplifier is recommended. Bypass Capacitor, CB The bypass capacitor, CB, is the most critical capacitor and serves several important functions. During startup or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and THD+N. Bypass capacitor, CB, values of 0.1 µF to 1 µF ceramic of tantalum low-ESR capacitors are recommended for the best THD and noise performance. In Figure 2, the full feature configuration, two bypass capacitors are used. This provides the maximum separation between right and left drive circuits.When absolute minimum cost and/or component space is required, one bypass capacitor can be used as shown in Figure 1. It is critical that terminals 6 and 19 be tied together in this configuration. Output Coupling Capacitor, CC In the typical single-supply SE configuration, and output coupling capacitor (CC) is required to block the dc bias at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor and impedance of the load form a high-pass filter governed by equation 6 Table1. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode 2π R C L CC Lowest Frequency 3Ω 330 µF 161 Hz 4Ω 330 µF 120 Hz 8Ω 330 µF 60 Hz 32 Ω 330 µF 15 Hz 10000 Ω 330 µF 0.05 Hz 47000 Ω 330 µF 0.01 Hz As Table 1 indicates, most of the bass response is attenuated into 4–Ω load, an 8–Ω load is adequate, headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional. Using Low-ESR Capacitors Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more the real capacitor behaves like an ideal capacitor. Bridged-Tied Load Versus Single-Ended Mode Figure 56 show a linear audio power amplifier (APA) in a BTL configuration. The EUA 5202 BTL amplifier consists of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive configuration, but initially consider power to the load. The differential drive to the speaker means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the load as compared to a ground referenced load. Plugging 2 × VO(PP) into the power equation, where voltage is squared, yields 4 × the output power from the same supply rail and load impedance (see equation 7 ) 1 fc(high)= RL ---------------------------- (6) C V(rms)＝ DS5202 Ver 1.6 May. 2005 19 V O(PP) 2 2 Power ＝ V (rms) R L 2 ------ (7) EUA5202 Increasing power to the load does carry a penalty of increased internal power dissipation. The increased dissipation is understandable considering that the BTL configuration produces 4 × the output power of the SE configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal considerations section. Single-Ended Operation In SE mode (see Figure56 and Figure57), the load is driven from the primary amplifier output for each channel (OUT+, terminals 22 and 3). In SE mode the gain is set by the RF and RI resistors and is shown in equation 9. Since the inverting amplifier is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included. In a typical computer sound channel operating at 5V, bridging raises the power into an 8-Ω speaker from a singled -ended (SE, ground reference) limit of 250 mW to 1W. In sound power that is a 6-dB improvement— which is loudness that can be heard. In addition to increased power there are frequency response concerns. Consider the single-supply SE configuration shown in Figure 57. A coupling capacitor is required to block the dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33µF to 1000µF) so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network created with the speaker impedance and the coupling capacitance and is calculated with equation 8. fC= 1 2 π R LCC R SE Gain = − F -------------------------------------- (9) R I The output coupling capacitor required in single-supply SE mode also places additional constraints on the selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of the following relationship (see equation 10): 1 C × 25 k Ω B ------------------------------------ (8) For example, a 68µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency performance is then limited only by the input network and speaker response. Cost and PCB space are also minimized by eliminating the bulky coupling capacitor. DS5202 Ver 1.6 May. 2005 20 ≤ 1 C R I I < 1 R C L C --------------- (10) EUA5202 Input MUX Operation SE/BTL Operation The ability of the EUA5202 to easily switch between BTL and SE modes is one of its most important cost saving features. This feature eliminates the requirement for an additional headphone amplifier in applications where internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated. Internal to the EUA5202, two separate amplifiers drive LOUT- and ROUT- (terminals 10 and 15).When SE/ BTL is held high, the OUT- amplifier are in high output impedance state, which configures the EUA5202 as an SE driver from LOUT + and ROUT + (terminal 3 and 22). IDD is reduced by approximately one-half in SE mode. Control of the SE/ BTL input can be from a logic-level CMOS source, or, more typically, from a resistor divider network as shown in Figure 59. Working in concert with the SE/ BTL feature, the HP/LINE MUX feature gives the audio designer the flexibility of a multichip design in a single IC (see Figure 58). The primary function of the MUX is to allow different gain settings for BTL versus SE mode. Speakers typically require approximately a factor of 10 more gain for similar volume listening levels as compared to headphones. To achieve headphone and speaker listening parity, the resistor values would need to be set as follows: SE Gain R F(HP) = − (HP) R I (HP) -------------------------- (11) If, for example RI (HP) = 10 kΩ and RF (HP) = 10kΩ then SE Gain (HP) = -1 R F (LINE) -------------------- (12) BTL Gain = −2 (LINE) R I (LINE) If, for example RI (LINE) = 10kΩ and RF (LINE) = 50kΩ then BTL Gain (LINE) = -10 Using a readily available 1/8-in. (3.5mm) stereo headphone jack, the control switch is closed when no plug is inserted. When closed the 100-kΩ/1-kΩ divider pulls the SE/ BTL input low. When a plug is inserted, the OUT- amplifier is shutdown causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives through the output capacitor (CO) into the headphone jack. As shown n the full feature application (Figure 2), the input MUX control can be tied to the SE/ BTL input. The benefits of doing this are described in the following input MUX operation section. Another advantage of using the MUX feature is setting the gain of the headphone channel to -1. This provides the optimum distortion performance into the headphones where clear sound is more important. Refer to the SE/ BTL operation section for a description of the headphone hack control circuit. DS5202 Ver 1.6 May. 2005 Mute and Shutdown Mode The EUA5202 employs both a mute and a shutdown mode of operation designed to reduce supply current, IDD, to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal should be held low during normal operation when the amplifier is in use. Pulling SHUTDOWN high causes the outputs to mute and the amplifier to enter a low-current state, IDD = 5 µA. SHUTDOWN or MUTE IN should never be left unconnected because amplifier operation would be unpredictable. Mute mode alone reduces IDD to 1.5 mA. 21 EUA5202 Thermal Pad Considerations The thermal pad must be connected to ground. The package with thermal pad of the EUA5202 requires special attention on thermal design. If the thermal design issues are not properly addressed, the EUA5202 will go into thermal shutdown when driving a heavy load. The thermal pad on the bottom of the EUA5202 should be soldered down to a copper pad on the circuit board. Heat can be conducted away from the thermal pad through the copper plane to ambient. If the copper plane is not on the top surface of the circuit board, 8 to 10 vias of 13 mil or smaller in diameter should be used to thermally couple the thermal pad to the bottom plane. For good thermal conduction, the vias must be plated through and solder filled. The copper plane used to conduct heat away from the thermal pad should be as large as practical. If the ambient temperature is higher than 25℃,a larger copper plane or forced-air cooling will be required to keep the EUA5202 junction temperature below the thermal shutdown temperature (150℃). In higher ambient temperature, higher airflow rate and/or larger copper area will be required to keep the IC out of thermal shutdown. DS5202 Ver 1.6 May. 2005 22 EUA5202 Package Information Use as much copper area as possible Bottom view Exposed Pad NOTE 1. Package body sizes exclude mold flash protrusions or gate burrs 2. Tolerance ± 0.1mm unless otherwise specified 3. Coplanarity :0.1mm 4. Controlling dimension is millimeter. Converted inch dimensions are not necessarily exact. 5. Die pad exposure size is according to lead frame design. 6. Standard Solder Map dimension is millimeter. 7. Followed from JEDEC MO-153 SYMBOLS A A1 A2 b C D E E1 e L y θ DS5202 Ver 1.6 May. 2005 DIMENSIONS IN MILLIMETERS MIN. NOM. MAX. ----------1.15 0.00 -----0.10 0.80 1.00 1.05 0.19 -----0.30 0.09 -----0.20 7.70 7.80 7.90 -----6.40 ----4.30 4.40 4.50 -----0.65 ----0.45 0.60 0.75 ----------0.10 0 -----8 23 DIMENSIONS IN INCHES MIN. NOM. MAX. ----------0.045 0.000 -----0.004 0.031 0.039 0.041 0.007 -----0.012 0.004 -----0.008 0.303 0.307 0.311 -----0.252 -----0.169 0.173 0.177 -----0.026 -----0.018 0.024 0.030 ----------0.004 0 -----8