ON MC33262 Power factor controller Datasheet

Order this document by MC34262/D
The MC34262/MC33262 are active power factor controllers specifically
designed for use as a preconverter in electronic ballast and in off–line power
converter applications. These integrated circuits feature an internal startup
timer for stand–alone applications, a one quadrant multiplier for near unity
power factor, zero current detector to ensure critical conduction operation,
transconductance error amplifier, quickstart circuit for enhanced startup,
trimmed internal bandgap reference, current sensing comparator, and a
totem pole output ideally suited for driving a power MOSFET.
Also included are protective features consisting of an overvoltage
comparator to eliminate runaway output voltage due to load removal, input
undervoltage lockout with hysteresis, cycle–by–cycle current limiting,
multiplier output clamp that limits maximum peak switch current, an RS latch
for single pulse metering, and a drive output high state clamp for MOSFET
gate protection. These devices are available in dual–in–line and surface
mount plastic packages.
• Overvoltage Comparator Eliminates Runaway Output Voltage
•
•
•
•
•
•
•
•
POWER FACTOR
CONTROLLERS
SEMICONDUCTOR
TECHNICAL DATA
P SUFFIX
PLASTIC PACKAGE
CASE 626
8
1
Internal Startup Timer
One Quadrant Multiplier
Zero Current Detector
Trimmed 2% Internal Bandgap Reference
D SUFFIX
PLASTIC PACKAGE
CASE 751
(SO–8)
Totem Pole Output with High State Clamp
8
Undervoltage Lockout with 6.0 V of Hysteresis
1
Low Startup and Operating Current
Supersedes Functionality of SG3561 and TDA4817
PIN CONNECTIONS
Simplified Block Diagram
Zero Current Detector
5
2.5V
Reference
Undervoltage
Lockout
Zero Current
Detect Input
Voltage Feedback
Input
Compensation
Multiplier Input
Current Sense
Input
VCC
8 VCC
7 Drive Output
6 Gnd
5 Zero Current
Detect Input
1
2
3
4
(Top View)
8
Drive Output
7
Multiplier,
Latch,
PWM,
Timer,
&
Logic
Overvoltage
Comparator
+
4
Current Sense
Input
ORDERING INFORMATION
1.08 Vref
Device
Error Amp
Multiplier
Input 3
+
Multiplier
Quickstart
Gnd
6
Compensation
2
Vref
Voltage
Feedback
1 Input
MC34262D
MC34262P
MC33262D
MC33262P
Operating
Temperature Range
TA = 0° to +85°C
TA = – 40° to +105°C
 Motorola, Inc. 1996
MOTOROLA ANALOG IC DEVICE DATA
Package
SO–8
Plastic DIP
SO–8
Plastic DIP
Rev 1
1
MC34262 MC33262
MAXIMUM RATINGS
Rating
Symbol
Value
Unit
(ICC + IZ)
30
mA
Output Current, Source or Sink (Note 1)
IO
500
mA
Current Sense, Multiplier, and Voltage Feedback Inputs
Vin
–1.0 to +10
V
Zero Current Detect Input
High State Forward Current
Low State Reverse Current
Iin
Total Power Supply and Zener Current
Power Dissipation and Thermal Characteristics
P Suffix, Plastic Package, Case 626
Maximum Power Dissipation @ TA = 70°C
Thermal Resistance, Junction–to–Air
D Suffix, Plastic Package, Case 751
Maximum Power Dissipation @ TA = 70°C
Thermal Resistance, Junction–to–Air
mA
50
–10
PD
RθJA
800
100
mW
°C/W
PD
RθJA
450
178
mW
°C/W
Operating Junction Temperature
TJ
+150
°C
Operating Ambient Temperature (Note 3)
MC34262
MC33262
TA
Storage Temperature
°C
0 to + 85
– 40 to +105
Tstg
°C
– 65 to +150
ELECTRICAL CHARACTERISTICS (VCC = 12 V (Note 2), for typical values TA = 25°C, for min/max values TA is the operating ambient
temperature range that applies [Note 3], unless otherwise noted.)
Characteristic
Symbol
Min
Typ
Max
2.465
2.44
2.5
—
2.535
2.54
Unit
ERROR AMPLIFIER
Voltage Feedback Input Threshold
TA = 25°C
TA = Tlow to Thigh (VCC = 12 V to 28 V)
Line Regulation (VCC = 12 V to 28 V, TA = 25°C)
VFB
V
Regline
—
1.0
10
mV
Input Bias Current (VFB = 0 V)
IIB
—
– 0.1
– 0.5
µA
Transconductance (TA = 25°C)
gm
80
100
130
µmho
Output Current
Source (VFB = 2.3 V)
Sink (VFB = 2.7 V)
IO
—
—
10
10
—
—
VOH(ea)
VOL(ea)
5.8
—
6.4
1.7
—
2.4
VFB(OV)
1.065 VFB
1.08 VFB
1.095 VFB
V
IIB
—
– 0.1
– 0.5
µA
Input Threshold, Pin 2
Vth(M)
1.05 VOL(EA)
1.2 VOL(EA)
—
V
Dynamic Input Voltage Range
Multiplier Input (Pin 3)
Compensation (Pin 2)
VPin 3
VPin 2
0 to 2.5
Vth(M) to
(Vth(M) + 1.0)
0 to 3.5
Vth(M) to
(Vth(M) + 1.5)
—
—
K
0.43
0.65
0.87
1/V
Input Threshold Voltage (Vin Increasing)
Vth
1.33
1.6
1.87
V
Hysteresis (Vin Decreasing)
VH
100
200
300
mV
Input Clamp Voltage
High State (IDET = + 3.0 mA)
Low State (IDET = – 3.0 mA)
VIH
VIL
6.1
0.3
6.7
0.7
—
1.0
Output Voltage Swing
High State (VFB = 2.3 V)
Low State (VFB = 2.7 V)
µA
V
OVERVOLTAGE COMPARATOR
Voltage Feedback Input Threshold
MULTIPLIER
Input Bias Current, Pin 3 (VFB = 0 V)
Multiplier Gain (VPin 3 = 0.5 V, VPin 2 = Vth(M) + 1.0 V) (Note 4)
V
ZERO CURRENT DETECTOR
2
V
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
ELECTRICAL CHARACTERISTICS (VCC = 12 V (Note 2), for typical values TA = 25°C, for min/max values TA is the operating ambient
temperature range that applies (Note 3), unless otherwise noted.)
Characteristic
Symbol
Min
Typ
Max
Unit
IIB
—
– 0.15
–1.0
µA
CURRENT SENSE COMPARATOR
Input Bias Current (VPin 4 = 0 V)
Input Offset Voltage (VPin 2 = 1.1 V, VPin 3 = 0 V)
VIO
—
9.0
25
mV
Maximum Current Sense Input Threshold (Note 5)
Vth(max)
1.3
1.5
1.8
V
tPHL(in/out)
—
200
400
ns
VOL
—
—
9.8
7.8
0.3
2.4
10.3
8.4
0.8
3.3
—
—
14
16
18
Delay to Output
DRIVE OUTPUT
Output Voltage (VCC = 12 V)
Low State
(ISink = 20 mA)
Low State
(ISink = 200 mA)
High State (ISource = 20 mA)
High State (ISource = 200 mA)
V
VOH
Output Voltage (VCC = 30 V)
High State (ISource = 20 mA, CL = 15 pF)
VO(max)
V
Output Voltage Rise Time (CL = 1.0 nF)
tr
—
50
120
ns
Output Voltage Fall Time (CL = 1.0 nF)
tf
—
50
120
ns
VO(UVLO)
—
0.1
0.5
V
tDLY
200
620
—
µs
Vth(on)
11.5
13
14.5
V
VShutdown
7.0
8.0
9.0
V
VH
3.8
5.0
6.2
V
—
—
—
0.25
6.5
9.0
0.4
12
20
30
36
—
Output Voltage with UVLO Activated
(VCC = 7.0 V, ISink = 1.0 mA)
RESTART TIMER
Restart Time Delay
UNDERVOLTAGE LOCKOUT
Startup Threshold (VCC Increasing)
Minimum Operating Voltage After Turn–On (VCC Decreasing)
Hysteresis
TOTAL DEVICE
Power Supply Current
Startup (VCC = 7.0 V)
Operating
Dynamic Operating (50 kHz, CL = 1.0 nF)
ICC
Power Supply Zener Voltage (ICC = 25 mA)
VZ
mA
NOTES: 1. Maximum package power dissipation limits must be observed.
2. Adjust VCC above the startup threshold before setting to 12 V.
3. Tlow = 0°C for MC34262
3. Tlow = – 40°C for MC33262
4. K =
Thigh = + 85°C for MC34262
Thigh = +105°C for MC33262
1.0
VCC = 12 V
TA = 25°C
VPin 2 = 3.75 V
VPin 2 = 3.5 V
VPin 2 = 3.25 V
0.8
VPin 2 = 3.0 V
0.6
VPin 2 = 2.75 V
VPin 2 = 2.5 V
VPin 2 = 2.25 V
0.4
0.2
0
– 0.2
VPin 2 = 2.0 V
0.6
1.4
2.2
3.0
VM, MULTIPLIER PIN 3 INPUT VOLTAGE (V)
MOTOROLA ANALOG IC DEVICE DATA
3.8
V CS , CURRENT SENSE PIN 4 THRESHOLD (V)
V CS , CURRENT SENSE PIN 4 THRESHOLD (V)
Figure 2. Current Sense Input Threshold
versus Multiplier Input, Expanded View
1.6
1.2
Pin 4 Threshold
VPin 3 (VPin 2 – Vth(M))
5. This parameter is measured with VFB = 0 V, and VPin 3 = 3.0 V.
Figure 1. Current Sense Input Threshold
versus Multiplier Input
1.4
V
0.08
VPin 2 = 3.75 V
VPin 2 = 3.5 V
VPin 2 = 3.25 V
0.06 VPin 2 = 3.0 V
0.05 VPin 2 = 2.75 V
0.07
0.04
VCC = 12 V
TA = 25°C
VPin 2 = 2.5 V
VPin 2 = 2.25 V
0.03
0.02
0.01
0
– 0.12
VPin 2 = 2.0 V
– 0.06
0
0.06
0.12
0.18
VM, MULTIPLIER PIN 3 INPUT VOLTAGE (V)
0.24
3
Figure 3. Voltage Feedback Input Threshold
Change versus Temperature
V FB(OV) , OVERVOLTAGE INPUT THRESHOLD (%V FB )
4.0
VCC = 12 V
Pins 1 to 2
0
– 4.0
– 8.0
–12
–16
– 55
– 25
0
25
50
75
100
125
TA, AMBIENT TEMPERATURE (°C)
Figure 4. Overvoltage Comparator Input
Threshold versus Temperature
110
VCC = 12 V
109
108
107
106
– 55
Transconductance
80
VCC = 12 V
VO = 2.5 V to 3.5 V
RL = 100 k to 3.0 V
CL = 2.0 pF
TA = 25°C
30
60
60
90
40
120
20
150
0
3.0 k
75
10 k
30 k
100 k 300 k
f, FREQUENCY (Hz)
1.0 M
4.00 V
100
125
VCC = 12 V
RL = 100 k
CL = 2.0 pF
TA = 25°C
3.25 V
2.50 V
180
3.0 M
5.0 µs/DIV
Figure 8. Restart Timer Delay
versus Temperature
VCC = 12 V
1.76
800
1.72
700
Voltage
Current
600
1.68
– 25
0
25
50
75
TA, AMBIENT TEMPERATURE (°C)
100
500
125
800
VCC = 12 V
t DLY, RESTART TIME DELAY (µ s)
900
1.80
I chg , QUICKSTART CHARGE CURRENT (µ A)
Figure 7. Quickstart Charge Current
versus Temperature
Vchg , QUICKSTART CHARGE VOLTAGE (V)
50
Figure 6. Error Amp Transient Response
O , EXCESS PHASE (DEGREES)
g , TRANSCONDUCTANCE ( µ mho)
m
Phase
4
25
0
120
1.64
– 55
0
TA, AMBIENT TEMPERATURE (°C)
Figure 5. Error Amp Transconductance and
Phase versus Frequency
100
– 25
0.75 V/DIV
∆ V FB, VOLTAGE FEEDBACK THRESHOLD CHANGE (mV)
MC34262 MC33262
700
600
500
400
– 55
– 25
0
25
50
75
100
125
TA, AMBIENT TEMPERATURE (°C)
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Figure 10. Output Saturation Voltage
versus Load Current
VCC = 12 V
1.6
1.5
1.4
Lower Threshold
(Vin, Decreasing)
1.3
– 55
– 25
0
25
50
75
TA, AMBIENT TEMPERATURE (°C)
100
VCC
Source Saturation
(Load to Ground)
– 4.0
– 6.0
4.0
Sink Saturation
(Load to VCC)
2.0
0
Gnd
0
160
240
320
VO , OUTPUT VOLTAGE
Figure 12. Drive Output Cross Conduction
VCC = 12 V
CL = 15 pF
TA = 25°C
I CC , SUPPLY CURRENT
10%
100 ns/DIV
100 ns/DIV
Figure 13. Supply Current
versus Supply Voltage
Figure 14. Undervoltage Lockout Thresholds
versus Temperature
16
14
VCC , SUPPLY VOLTAGE (V)
I CC , SUPPLY CURRENT (mA)
80
IO, OUTPUT LOAD CURRENT (mA)
VCC = 12 V
CL = 1.0 nF
TA = 25°C
12
8.0
VFB = 0 V
Current Sense = 0 V
Multiplier = 0 V
CL = 1.0 nF
f = 50 kHz
TA = 25°C
4.0
0
VCC = 12 V
80 µs Pulsed Load
120 Hz Rate
– 2.0
125
Figure 11. Drive Output Waveform
90%
0
5.0 V/DIV
Upper Threshold
(Vin, Increasing)
100 mA/DIV
V th , THRESHOLD VOLTAGE (V)
1.7
Vsat , OUTPUT SATURATION VOLTAGE (V)
Figure 9. Zero Current Detector Input
Threshold Voltage versus Temperature
0
10
20
30
VCC, SUPPLY VOLTAGE (V)
MOTOROLA ANALOG IC DEVICE DATA
13
Startup Threshold
(VCC Increasing)
12
11
10
9.0
Minimum Operating Threshold
(VCC Decreasing)
8.0
40
7.0
– 55
– 25
0
25
50
75
TA, AMBIENT TEMPERATURE (°C)
100
125
5
MC34262 MC33262
FUNCTIONAL DESCRIPTION
Introduction
With the goal of exceeding the requirements of legislation
on line–current harmonic content, there is an ever increasing
demand for an economical method of obtaining a unity power
factor. This data sheet describes a monolithic control IC that
was specifically designed for power factor control with
minimal external components. It offers the designer a simple,
cost–effective solution to obtain the benefits of active power
factor correction.
Most electronic ballasts and switching power supplies use
a bridge rectifier and a bulk storage capacitor to derive raw dc
voltage from the utility ac line, Figure 15.
can be made to appear resistive to the ac line, thus
significantly reducing the harmonic current content.
Figure 16. Uncorrected Power Factor
Input Waveforms
Vpk
Rectified
DC
0
Line Sag
Figure 15. Uncorrected Power Factor Circuit
Rectifiers
AC Line
Voltage
Converter
AC
Line
+
0
Bulk
Storage
Capacitor
Load
This simple rectifying circuit draws power from the line
when the instantaneous ac voltage exceeds the capacitor
voltage. This occurs near the line voltage peak and results in
a high charge current spike, Figure 16. Since power is only
taken near the line voltage peaks, the resulting spikes of
current are extremely nonsinusoidal with a high content of
harmonics. This results in a poor power factor condition
where the apparent input power is much higher than the real
power. Power factor ratios of 0.5 to 0.7 are common.
Power factor correction can be achieved with the use of
either a passive or an active input circuit. Passive circuits
usually contain a combination of large capacitors, inductors,
and rectifiers that operate at the ac line frequency. Active
circuits incorporate some form of a high frequency switching
converter for the power processing, with the boost converter
being the most popular topology, Figure 17. Since active
input circuits operate at a frequency much higher than that of
the ac line, they are smaller, lighter in weight, and more
efficient than a passive circuit that yields similar results. With
proper control of the preconverter, almost any complex load
AC Line
Current
The MC34262, MC33262 are high performance, critical
conduction, current–mode power factor controllers
specifically designed for use in off–line active preconverters.
These devices provide the necessary features required to
significantly enhance poor power factor loads by keeping the
ac line current sinusoidal and in phase with the line voltage.
Operating Description
The MC34262, MC33262 contain many of the building
blocks and protection features that are employed in modern
high performance current mode power supply controllers.
There are, however, two areas where there is a major
difference when compared to popular devices such as the
UC3842 series. Referring to the block diagram in Figure 19,
note that a multiplier has been added to the current sense
loop and that this device does not contain an oscillator. The
reasons for these differences will become apparent in the
following discussion. A description of each of the functional
blocks is given below.
Figure 17. Active Power Factor Correction Preconverter
Rectifiers
PFC Preconverter
AC
Line
+
6
High
Frequency
Bypass
Capacitor
Converter
+
MC34362
Bulk
Storage
Capacitor
Load
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Error Amplifier
An Error Amplifier with access to the inverting input and
output is provided. The amplifier is a transconductance type,
meaning that it has high output impedance with controlled
voltage–to–current gain. The amplifier features a typical gm
of 100 µmhos (Figure 5). The noninverting input is internally
biased at 2.5 V ± 2.0% and is not pinned out. The output
voltage of the power factor converter is typically divided down
and monitored by the inverting input. The maximum input
bias current is – 0.5 µA, which can cause an output voltage
error that is equal to the product of the input bias current and
the value of the upper divider resistor R2. The Error Amp
output is internally connected to the Multiplier and is pinned
out (Pin 2) for external loop compensation. Typically, the
bandwidth is set below 20 Hz, so that the amplifier’s output
voltage is relatively constant over a given ac line cycle. In
effect, the error amp monitors the average output voltage of
the converter over several line cycles. The Error Amp output
stage was designed to have a relatively constant
transconductance over temperature. This allows the
designer to define the compensated bandwidth over the
intended operating temperature range. The output stage can
sink and source 10 µA of current and is capable of swinging
from 1.7 V to 6.4 V, assuring that the Multiplier can be driven
over its entire dynamic range.
A key feature to using a transconductance type amplifier,
is that the input is allowed to move independently with
respect to the output, since the compensation capacitor is
connected to ground. This allows dual usage of of the Voltage
Feedback Input pin by the Error Amplifier and by the
Overvoltage Comparator.
Overvoltage Comparator
An Overvoltage Comparator is incorporated to eliminate
the possibility of runaway output voltage. This condition can
occur during initial startup, sudden load removal, or during
output arcing and is the result of the low bandwidth that must
be used in the Error Amplifier control loop. The Overvoltage
Comparator monitors the peak output voltage of the
converter, and when exceeded, immediately terminates
MOSFET switching. The comparator threshold is internally
set to 1.08 Vref. In order to prevent false tripping during
normal operation, the value of the output filter capacitor C3
must be large enough to keep the peak–to–peak ripple less
than 16% of the average dc output. The Overvoltage
Comparator input to Drive Output turn–off propagation delay
is typically 400 ns. A comparison of startup overshoot without
and with the Overvoltage Comparator circuit is shown in
Figure 23.
Multiplier
A single quadrant, two input multiplier is the critical
element that enables this device to control power factor. The
ac full wave rectified haversines are monitored at Pin 3
MOTOROLA ANALOG IC DEVICE DATA
with respect to ground while the Error Amp output at Pin 2 is
monitored with respect to the Voltage Feedback Input
threshold. The Multiplier is designed to have an extremely
linear transfer curve over a wide dynamic range, 0 V to 3.2 V
for Pin 3, and 2.0 V to 3.75 V for Pin 2, Figure 1. The Multiplier
output controls the Current Sense Comparator threshold as
the ac voltage traverses sinusoidally from zero to peak line,
Figure 18. This has the effect of forcing the MOSFET on–time
to track the input line voltage, resulting in a fixed Drive Output
on–time, thus making the preconverter load appear to be
resistive to the ac line. An approximation of the Current
Sense Comparator threshold can be calculated from the
following equation. This equation is accurate only under the
given test condition stated in the electrical table.
VCS, Pin 4 Threshold ≈ 0.65 (VPin 2 – Vth(M)) VPin 3
A significant reduction in line current distortion can be
attained by forcing the preconverter to switch as the ac line
voltage crosses through zero. The forced switching is
achieved by adding a controlled amount of offset to the
Multiplier and Current Sense Comparator circuits. The
equation shown below accounts for the built–in offsets and is
accurate to within ten percent. Let Vth(M) = 1.991 V
VCS, Pin 4 Threshold = 0.544 (VPin 2 – Vth(M)) VPin 3
+ 0.0417 (VPin 2 – Vth(M))
Zero Current Detector
The MC34262 operates as a critical conduction current
mode controller, whereby output switch conduction is initiated
by the Zero Current Detector and terminated when the peak
inductor current reaches the threshold level established by
the Multiplier output. The Zero Current Detector initiates the
next on–time by setting the RS Latch at the instant the
inductor current reaches zero. This critical conduction mode
of operation has two significant benefits. First, since the
MOSFET cannot turn–on until the inductor current reaches
zero, the output rectifier reverse recovery time becomes less
critical, allowing the use of an inexpensive rectifier. Second,
since there are no deadtime gaps between cycles, the ac line
current is continuous, thus limiting the peak switch to twice
the average input current.
The Zero Current Detector indirectly senses the inductor
current by monitoring when the auxiliary winding voltage falls
below 1.4 V. To prevent false tripping, 200 mV of hysteresis is
provided. Figure 9 shows that the thresholds are
well–defined over temperature. The Zero Current Detector
input is internally protected by two clamps. The upper 6.7 V
clamp prevents input overvoltage breakdown while the lower
0.7 V clamp prevents substrate injection. Current limit
protection of the lower clamp transistor is provided in the
event that the input pin is accidentally shorted to ground. The
Zero Current Detector input to Drive Output turn–on
propagation delay is typically 320 ns.
7
MC34262 MC33262
Figure 18. Inductor Current and MOSFET
Gate Voltage Waveforms
Peak
Average
Inductor Current
0
On
MOSFET
Q1
Off
Current Sense Comparator and RS Latch
The Current Sense Comparator RS Latch configuration
used ensures that only a single pulse appears at the Drive
Output during a given cycle. The inductor current is
converted to a voltage by inserting a ground–referenced
sense resistor R7 in series with the source of output switch
Q1. This voltage is monitored by the Current Sense Input and
compared to a level derived from the Multiplier output. The
peak inductor current under normal operating conditions is
controlled by the threshold voltage of Pin 4 where:
IL(pk ) =
Pin 4 Threshold
R7
Abnormal operating conditions occur during preconverter
startup at extremely high line or if output voltage sensing is
lost. Under these conditions, the Multiplier output and Current
Sense threshold will be internally clamped to 1.5 V.
Therefore, the maximum peak switch current is limited to:
Ipk(max) =
1.5 V
R7
An internal RC filter has been included to attenuate any
high frequency noise that may be present on the current
waveform. This filter helps reduce the ac line current
distortion especially near the zero crossings. With the
component values shown in Figure 20, the Current Sense
Comparator threshold, at the peak of the haversine varies
from 1.1 V at 90 Vac to 100 mV at 268 Vac. The Current
Sense Input to Drive Output turn–off propagation delay is
typically less than 200 ns.
8
Timer
A watchdog timer function was added to the IC to eliminate
the need for an external oscillator when used in stand–alone
applications. The Timer provides a means to automatically
start or restart the preconverter if the Drive Output has been
off for more than 620 µs after the inductor current reaches
zero. The restart time delay versus temperature is shown in
Figure 8.
Undervoltage Lockout and Quickstart
An Undervoltage Lockout comparator has been
incorporated to guarantee that the IC is fully functional before
enabling the output stage. The positive power supply terminal
(VCC) is monitored by the UVLO comparator with the upper
threshold set at 13 V and the lower threshold at 8.0 V. In the
stand–by mode, with VCC at 7.0 V, the required supply current
is less than 0.4 mA. This large hysteresis and low startup
current allow the implementation of efficient bootstrap startup
techniques, making these devices ideally suited for wide
input range off–line preconverter applications. An internal
36 V clamp has been added from VCC to ground to protect
the IC and capacitor C4 from an overvoltage condition. This
feature is desirable if external circuitry is used to delay the
startup of the preconverter. The supply current, startup, and
operating voltage characteristics are shown in Figures 13
and 14.
A Quickstart circuit has been incorporated to optimize
converter startup. During initial startup, compensation
capacitor C1 will be discharged, holding the error amp output
below the Multiplier threshold. This will prevent Drive Output
switching and delay bootstrapping of capacitor C4 by diode
D6. If Pin 2 does not reach the multiplier threshold before C4
discharges below the lower UVLO threshold, the converter
will “hiccup” and experience a significant startup delay. The
Quickstart circuit is designed to precharge C1 to 1.7 V, Figure
7. This level is slightly below the Pin 2 Multiplier threshold,
allowing immediate Drive Output switching and bootstrap
operation when C4 crosses the upper UVLO threshold.
Drive Output
The MC34262/MC33262 contain a single totem–pole
output stage specifically designed for direct drive of power
MOSFETs. The Drive Output is capable of up to ± 500 mA
peak current with a typical rise and fall time of 50 ns with a
1.0 nF load. Additional internal circuitry has been added to
keep the Drive Output in a sinking mode whenever the
Undervoltage Lockout is active. This characteristic eliminates
the need for an external gate pull–down resistor. The
totem–pole output has been optimized to minimize
cross–conduction current during high speed operation. The
addition of two 10 Ω resistors, one in series with the source
output transistor and one in series with the sink output
transistor, helps to reduce the cross–conduction current and
radiated noise by limiting the output rise and fall time. A 16 V
clamp has been incorporated into the output stage to limit the
high state VOH. This prevents rupture of the MOSFET gate
when VCC exceeds 20 V.
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
APPLICATIONS INFORMATION
The application circuits shown in Figures 19, 20 and 21
reveal that few external components are required for a
complete power factor preconverter. Each circuit is a peak
detecting current–mode boost converter that operates in
critical conduction mode with a fixed on–time and variable
off–time. A major benefit of critical conduction operation is
that the current loop is inherently stable, thus eliminating the
need for ramp compensation. The application in Figure 19
operates over an input voltage range of 90 Vac to 138 Vac
and provides an output power of 80 W (230 V at 350 mA) with
an associated power factor of approximately 0.998 at
nominal line. Figures 20 and 21 are universal input
preconverter examples that operate over a continuous input
voltage range of 90 Vac to 268 Vac. Figure 20 provides an
output power of 175 W (400 V at 440 mA) while Figure 21
provides 450 W (400 V at 1.125 A). Both circuits have an
observed worst–case power factor of approximately 0.989.
The input current and voltage waveforms of Figure 20 are
shown in Figure 22 with operation at 115 Vac and 230 Vac.
The data for each of the applications was generated with the
test set–up shown in Figure 24.
Table 1. Design Equations
Notes
Calculation
Calculate the maximum required output power.
Required Converter Output Power
Calculated at the minimum required ac line voltage
for output regulation. Let the efficiency η = 0.92 for
low line operation.
Peak Inductor Current
Let the switching cycle t = 40 µs for universal input
(85 to 265 Vac) operation and 20 µs for fixed input
(92 to 138 Vac, or 184 to 276 Vac) operation.
In theory the on–time ton is constant. In practice ton
tends to increase at the ac line zero crossings due
to the charge on capacitor C5. Let Vac = Vac(LL) for initial
ton and toff calculations.
Inductance
Formula
PO = VO IO
2 2 PO
ηVac(LL)
IL(pk) =
tǒ
LP =
VO
– Vac(LL)Ǔ η Vac(LL)2
2
2 VO PO
Switch On–Time
ton =
The off–time toff is greatest at the peak of the ac line
voltage and approaches zero at the ac line zero
crossings. Theta (θ) represents the angle of the ac
line voltage.
Switch Off–Time
The minimum switching frequency occurs at the peak
of the ac line voltage. As the ac line voltage traverses
from peak to zero, toff approaches zero producing an
increase in switching frequency.
Switching Frequency
f=
Set the current sense threshold VCS to 1.0 V for
universal input (85 Vac to 265 Vac) operation and
to 0.5 V for fixed input (92 Vac to 138 Vac, or
184 Vac to 276 Vac) operation. Note that VCS must
be <1.4 V.
Peak Switch Current
R7 =
Set the multiplier input voltage VM to 3.0 V at high
line. Empirically adjust VM for the lowest distortion
over the ac line voltage range while guaranteeing
startup at minimum line.
Multiplier Input Voltage
The IIB R1 error term can be minimized with a divider
current in excess of 50 µA.
2 PO LP
η Vac2
ton
toff =
VO
2 Vac Sin θ
Converter Output Voltage
The calculated peak–to–peak ripple must be less than
16% of the average dc output voltage to prevent false
tripping of the Overvoltage Comparator. Refer to the
Overvoltage Comparator text. ESR is the equivalent
series resistance of C3
Converter Output
Peak to Peak
Ripple Voltage
The bandwidth is typically set to 20 Hz. When operating
at high ac line, the value of C1 may need to be
increased. (See Figure 25)
Error Amplifier Bandwidth
VM =
VO = Vref
ǒ
–1
1
ton + toff
ǒ
VCS
IL(pk)
Vac
2
R5
+ 1Ǔ
R3
R2
+ 1 Ǔ – IIB R2
R1
∆VO(pp) = IO
BW =
ǒ
1
2πfac C3
2
Ǔ+ ESR2
gm
2 π C1
The following converter characteristics must be chosen:
Vac — AC RMS line voltage
VO — Desired output voltage
IO — Desired output current
Vac(LL) — AC RMS low line voltage
∆VO — Converter output peak–to–peak ripple voltage
MOTOROLA ANALOG IC DEVICE DATA
9
MC34262 MC33262
Figure 19. 80 W Power Factor Controller
C5
1
D2
92 to 138 RFI
Vac
Filter
D1
100k
R6
8
D4
Zero Current
Detector
D3
1.2V
+
5
6.7V
1.6V/
1.4V
Drive
Output
RS
Latch
MUR130
D5
10
1.5V
MTP
8N50E
Q1
7
10
0.1
R7
10pF
Overvoltage
Comparator
VO
230V/0.35A
+
220
C3
1.0M
R2
4
20k
+
1.08 Vref
10µA
7.5k
R3
T
16V
Delay
0.01
C2
22k
R4
+ 13V/
8.0V
Timer R
Current Sense
Comparator
100
C4
UVLO
2.5V
Reference
2.2M
R5
+
36V
+
1N4934
D6
Multiplier
Error Amp
+
Vref
1
3
11k
R1
Quickstart
2
6
0.68
C1
Power Factor Controller Test Data
AC Line Input
DC Output
Current Harmonic Distortion (% Ifund)
Vrms
Pin
PF
Ifund
THD
2
3
5
7
VO(pp)
VO
IO
PO
η(%)
90
85.9
0.999
0.93
2.6
0.08
1.6
0.84
0.95
4.0
230.7
0.350
80.8
94.0
100
85.3
0.999
0.85
2.3
0.13
1.0
1.2
0.73
4.0
230.7
0.350
80.8
94.7
110
85.1
0.998
0.77
2.2
0.10
0.58
1.5
0.59
4.0
230.7
0.350
80.8
94.9
120
84.7
0.998
0.71
3.0
0.09
0.73
1.9
0.58
4.1
230.7
0.350
80.8
95.3
130
84.4
0.997
0.65
3.9
0.12
1.7
2.2
0.61
4.1
230.7
0.350
80.8
95.7
138
84.1
0.996
0.62
4.6
0.16
2.4
2.3
0.60
4.1
230.7
0.350
80.8
96.0
This data was taken with the test set–up shown in Figure 24.
T = Coilcraft N2881–A
Primary: 62 turns of # 22 AWG
Secondary: 5 turns of # 22 AWG
Core: Coilcraft PT2510, EE 25
Gap: 0.072″ total for a primary inductance (LP) of 320 µH
Heatsink = AAVID Engineering Inc. 590302B03600, or 593002B03400
10
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Figure 20. 175 W Universal Input Power Factor Controller
C5
1
D2
90 to 268 RFI
Vac
Filter
D1
100k
R6
8
D4
Zero Current
Detector
D3
1.2V
+
5
6.7V
1.6V/
1.4V
Drive
Output
RS
Latch
10
1.5V
MTP
14N50E
Q1
7
10
0.1
R7
10pF
Overvoltage
Comparator
VO
400V/0.44A
+
330
C3
1.6M
R2
4
20k
+
1.08 Vref
10µA
12k
R3
T
MUR460
D5
16V
Delay
0.01
C2
22k
R4
+ 13V/
8.0V
Timer R
Current Sense
Comparator
100
C4
UVLO
2.5V
Reference
1.3M
R5
+
36V
+
1N4934
D6
Multiplier
Error Amp
+
Vref
1
3
10k
R1
Quickstart
2
6
0.68
C1
Power Factor Controller Test Data
AC Line Input
DC Output
Current Harmonic Distortion (% Ifund)
Vrms
Pin
PF
Ifund
THD
2
3
5
7
VO(pp)
VO
IO
PO
η(%)
90
193.3
0.991
2.15
2.8
0.18
2.6
0.55
1.0
3.3
402.1
0.44
176.9
91.5
120
190.1
0.998
1.59
1.6
0.10
1.4
0.23
0.72
3.3
402.1
0.44
176.9
93.1
138
188.2
0.999
1.36
1.2
0.12
1.3
0.65
0.80
3.3
402.1
0.44
176.9
94.0
180
184.9
0.998
1.03
2.0
0.10
0.49
1.2
0.82
3.4
402.1
0.44
176.9
95.7
240
182.0
0.993
0.76
4.4
0.09
1.6
2.3
0.51
3.4
402.1
0.44
176.9
97.2
268
180.9
0.989
0.69
5.9
0.10
2.3
2.9
0.46
3.4
402.1
0.44
176.9
97.8
This data was taken with the test set–up shown in Figure 24.
T = Coilcraft N2880–A
Primary: 78 turns of # 16 AWG
Secondary: 6 turns of # 18 AWG
Core: Coilcraft PT4215, EE 42–15
Gap: 0.104″ total for a primary inductance (LP) of 870 µH
Heatsink = AAVID Engineering Inc. 590302B03600
MOTOROLA ANALOG IC DEVICE DATA
11
MC34262 MC33262
Figure 21. 450 W Universal Input Power Factor Controller
C5
2
D2
90 to 268 RFI
Vac
Filter
D1
100k
R6
8
D4
Zero Current
Detector
D3
1.2V
+
5
6.7V
1.6V/
1.4V
MUR460
D5
10
10
4
10pF
Overvoltage
Comparator
+
MTW
20N50E
Q1
7
20k
1.5V
VO
+
Drive
Output
RS
Latch
400V/1.125A
330
C3
1.6M
R2
330
0.05
R7
0.001
1.08 Vref
10µA
12k
R3
T
16V
Delay
0.01
C2
22k
R4
+ 13V/
8.0V
Timer R
Current Sense
Comparator
100
C4
UVLO
2.5V
Reference
1.3M
R5
+
36V
+
1N4934
D6
Multiplier
Error Amp
+
Vref
1
3
10k
R1
Quickstart
2
6
0.68
C1
Power Factor Controller Test Data
AC Line Input
DC Output
Current Harmonic Distortion (% Ifund)
Vrms
Pin
PF
Ifund
THD
2
3
5
7
VO(pp)
VO
IO
PO
η(%)
90
489.5
0.990
5.53
2.2
0.10
1.5
0.25
0.83
8.8
395.5
1.14
450.9
92.1
120
475.1
0.998
3.94
2.5
0.12
0.29
0.62
0.52
8.8
395.5
1.14
450.9
94.9
138
470.6
0.998
3.38
2.1
0.06
0.70
1.1
0.41
8.8
395.5
1.14
450.9
95.8
180
463.4
0.998
2.57
4.1
0.21
2.0
1.6
0.71
8.9
395.5
1.14
450.9
97.3
240
460.1
0.996
1.91
4.8
0.14
4.3
2.2
0.63
8.9
395.5
1.14
450.9
98.0
268
459.1
0.995
1.72
5.8
0.10
5.0
2.5
0.61
8.9
395.5
1.14
450.9
98.2
This data was taken with the test set–up shown in Figure 24.
T = Coilcraft P3657–A
Primary: 38 turns Litz wire, 1300 strands of #48 AWG, Kerrigan–Lewis, Chicago, IL
Secondary: 3 turns of # 20 AWG
Core: Coilcraft PT4220, EE 42–20
Gap: 0.180″ total for a primary inductance (LP) of 190 µH
Heatsink = AAVID Engineering Inc. 604953B04000 Extrusion
12
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Figure 22. Power Factor Corrected Input Waveforms
(Figure 20 Circuit)
Current = 1.0 A/DIV
Current = 1.0 A/DIV
Voltage = 100 V/DIV
Input = 230 VAC, Output = 175 W
Voltage = 100 V/DIV
Input = 115 VAC, Output = 175 W
2.0 ms/DIV
2.0 ms/DIV
Figure 23. Output Voltage Startup Overshoot
(Figure 20 Circuit)
With Overvoltage Comparator
Without Overvoltage Comparator
500 V
8%
432 V
400 V
26%
80 V/DIV
80 V/DIV
400 V
0V
0V
200 ms/DIV
200 ms/DIV
Figure 24. Power Factor Test Set–Up
Line
115 Vac
Input
Neutral
2X Step–Up
Isolation
Transformer
RFI Test Filter
HI
AC POWER ANALYZER
PM 1000
W
Autoformer
0
I
O
Vcf
VA
1
PF Vrms Arms
2
3
A
T
V
5
0.1
0.005
1.0
0.005
0 to 270 Vac
Output to Power
Factor
Controller Circuit
Acf Ainst FREQ HARM
LO
7
HI
9
11
13
LO
Voltech
Earth
An RFI filter is required for best performance when connecting the preconverter directly to the ac line. The filter attenuates the level of high
frequency switching that appears on the ac line current waveform. Figures 19 and 20 work well with commercially available two stage filters such
as the Delta Electronics 03DPCG5. Shown above is a single stage test filter that can easily be constructed with four ac line rated capacitors and a
common–mode transformer. Coilcraft CMT3–28–2 was used to test Figures 19 and 20. It has a minimum inductance of 28 mH and a maximum
current rating of 2.0 A. Coilcraft CMT4–17–9 was used to test Figure 21. It has a minimum inductance of 17 mH and a maximum current rating of
9.0 A. Circuit conversion efficiency η (%) was calculated without the power loss of the RFI filter.
MOTOROLA ANALOG IC DEVICE DATA
13
MC34262 MC33262
Figure 25. Error Amp Compensation
10µA
R2
Error Amp
+
1
R1
2
6
C1
The Error Amp output is a high impedance node and is susceptible to noise pickup. To minimize pickup, compensation capacitor C1 must be
connected as close to Pin 2 as possible with a short, heavy ground returning directly to Pin 6. When operating at high ac line, the voltage at Pin 2
may approach the lower threshold of the Multiplier, ≈ 2.0 V. If there is excessive ripple on Pin 2, the Multiplier will be driven into cut–off causing
circuit instability, high distortion and poor power factor. This problem can be eliminated by increasing the value of C1.
Figure 26. Current Waveform Spike Suppression
Figure 27. Negative Current Waveform
Spike Suppression
7
7
22k
10pF
4
22k
R
C
R7
10pF
4
D1
R7
Current
Sense
Comparator
Current
Sense
Comparator
A narrow turn–on spike is usually present on the leading edge of the current
waveform and can cause circuit instability. The MC34262 provides an
internal RC filter with a time constant of 220 ns. An additional external RC
filter may be required in universal input applications that are above 200 W. It
is suggested that the external filter be placed directly at the Current Sense
Input and have a time constant that approximates the spike duration.
A negative turn–off spike can be observed on the trailing edge of the current
waveform. This spike is due to the parasitic inductance of resistor R7, and if it
is excessive, it can cause circuit instability. The addition of Shottky diode D1
can effectively clamp the negative spike. The addition of the external RC filter
shown in Figure 26 may provide sufficient spike attenuation.
14
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Figure 28. Printed Circuit Board and Component Layout
(Circuits of Figures 15 and 16)
(Top View)
3.0″
4.5″
(Bottom View)
NOTE: Use 2 oz. copper laminate for optimum circuit performance.
MOTOROLA ANALOG IC DEVICE DATA
15
MC34262 MC33262
OUTLINE DIMENSIONS
8
P SUFFIX
PLASTIC PACKAGE
CASE 626–05
ISSUE K
5
NOTES:
1. DIMENSION L TO CENTER OF LEAD WHEN
FORMED PARALLEL.
2. PACKAGE CONTOUR OPTIONAL (ROUND OR
SQUARE CORNERS).
3. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
–B–
1
4
F
–A–
NOTE 2
DIM
A
B
C
D
F
G
H
J
K
L
M
N
L
C
J
–T–
N
SEATING
PLANE
D
M
K
MILLIMETERS
MIN
MAX
9.40
10.16
6.10
6.60
3.94
4.45
0.38
0.51
1.02
1.78
2.54 BSC
0.76
1.27
0.20
0.30
2.92
3.43
7.62 BSC
–––
10_
0.76
1.01
INCHES
MIN
MAX
0.370
0.400
0.240
0.260
0.155
0.175
0.015
0.020
0.040
0.070
0.100 BSC
0.030
0.050
0.008
0.012
0.115
0.135
0.300 BSC
–––
10_
0.030
0.040
G
H
0.13 (0.005)
T A
M
M
B
D SUFFIX
PLASTIC PACKAGE
CASE 751–05
(SO–8)
ISSUE N
–A–
8
M
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE
DAMBAR PROTRUSION. ALLOWABLE
DAMBAR PROTRUSION SHALL BE 0.127
(0.005) TOTAL IN EXCESS OF THE D
DIMENSION AT MAXIMUM MATERIAL
CONDITION.
5
–B–
1
4X
P
0.25 (0.010)
4
M
B
M
G
R
C
–T–
8X
K
D
0.25 (0.010)
M
T B
SEATING
PLANE
S
A
M_
S
X 45 _
F
J
DIM
A
B
C
D
F
G
J
K
M
P
R
MILLIMETERS
MIN
MAX
4.80
5.00
3.80
4.00
1.35
1.75
0.35
0.49
0.40
1.25
1.27 BSC
0.18
0.25
0.10
0.25
0_
7_
5.80
6.20
0.25
0.50
INCHES
MIN
MAX
0.189
0.196
0.150
0.157
0.054
0.068
0.014
0.019
0.016
0.049
0.050 BSC
0.007
0.009
0.004
0.009
0_
7_
0.229
0.244
0.010
0.019
Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals”
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Opportunity/Affirmative Action Employer.
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16
◊
*MC34262/D*
MOTOROLA ANALOG IC DEVICE
DATA
MC34262/D