MPS MP2560DN 2.5a, 4mhz, 42v step-down converter Datasheet

MP2560
2.5A, 4MHz, 42V
Step-Down Converter
The Future of Analog IC Technology
FEATURES
DESCRIPTION
The MP2560 is a high frequency step-down
switching regulator with an integrated internal
high-side high voltage power MOSFET. It
provides 2.5A output with current mode control for
fast loop response and easy compensation.
The wide 4.5V to 42V input range accommodates
a variety of step-down applications, including
those in an automotive input environment. A 12µA
shutdown mode quiescent current allows use in
battery-powered applications.
High power conversion efficiency over a wide
load range is achieved by scaling down the
switching frequency at light load condition to
reduce the switching and gate driving losses.
The frequency foldback helps prevent inductor
current runaway during startup and thermal
shutdown provides reliable, fault tolerant
operation.
By switching at 4MHz, the MP2560 is able to
prevent EMI (Electromagnetic Interference) noise
problems, such as those found in AM radio and
ADSL applications.
•
•
•
•
•
•
•
•
•
•
120µA Quiescent Current
Wide 4.5V to 42V Operating Input Range
220mΩ Internal Power MOSFET
Up to 4MHz Programmable Switching
Frequency
Stable with Ceramic Capacitor
Internal Soft-Start
Internally Set Current Limit without external
Current Sensing Resistor
Up to 93% Efficiency
Output Adjustable from 0.8V to 39V
Available in 3x3 QFN10 and Thermally
Enhanced SOIC8 Packages
APPLICATIONS
•
•
•
•
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High Voltage Power Conversion
Automotive Systems
Industrial Power Systems
Distributed Power Systems
Battery Powered Systems
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
The MP2560 is available in small 3mm x 3mm
QFN10 and thermally enhanced SOIC8 packages.
TYPICAL APPLICATION
95
10
VIN
8,9
VIN
BST
SW
1,2
VOUT
3.3V
7
EN
MP2560
COMP
FREQ
GND
6
MP2560 Rev. 1.2
1/25/2010
FB
85
EFFICIENCY (%)
EN
VIN=12V VIN=8V
90
D1
3
Efficiency vs.
Load Current
5
4
C6
NS
80
VIN=24V
75
VIN=36V
70
65
60
VOUT=3.3V
fs=500kHz
55
50
0
1.5
0.5
1
2
LOAD CURRENT (A)
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2.5
1
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number*
Package
Top Marking
Temperature
MP2560DQ
MP2560DN
3x3 QFN10
SOIC8E
S8YW
MP2560DN
–40°C to +85°C
* For Tape & Reel, add suffix –Z (eg. MP2560DQ–Z). For RoHS Compliant Packaging, add suffix –LF (eg.
MP2560DQ–LF–Z)
** For Tape & Reel, add suffix –Z (eg. MP2560DN–Z). For RoHS Compliant Packaging, add suffix –LF (eg.
MP2560DN–LF–Z)
PACKAGE REFERENCE
TOP VIEW
TOP VIEW
SW
1
10
BST
SW
1
8
BST
SW
2
9
VIN
EN
2
7
VIN
EN
3
8
VIN
COMP
3
6
FREQ
COMP
4
7
FREQ
FB
4
5
GND
FB
5
6
GND
EXPOSED PAD
CONNECT TO GND
EXPOSED PAD
ON BACKSIDE
CONNECT TO GND
ABSOLUTE MAXIMUM RATINGS (1)
Supply Voltage (VIN).....................–0.3V to +45V
Switch Voltage (VSW) ..........................................
............. –0.3V (-7V for < 10ns) to VIN + 0.3V
BST to SW .....................................–0.3V to +6V
All Other Pins .................................–0.3V to +6V
Continuous Power Dissipation… (TA = +25°C) (2)
3x3 QFN10 ................................................ 2.5W
SOIC8 ........................................................ 2.5W
Junction Temperature ...............................150°C
Lead Temperature ....................................260°C
Storage Temperature.............. –65°C to +150°C
Recommended Operating Conditions
(3)
Supply Voltage VIN ...........................4.5V to 42V
Output Voltage VOUT .........................0.8V to 39V
Operating Temperature............. –40°C to +85°C
MP2560 Rev. 1.2
1/25/2010
Thermal Resistance
(4)
θJA
θJC
3x3 QFN10 ............................. 50 ...... 12... °C/W
SOIC8 (Exposed Pad) ............ 50 ...... 10... °C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ (MAX), the junction-toambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD (MAX) = (TJ
(MAX)-TA)/θJA. Exceeding the maximum allowable power
dissipation will cause excessive die temperature, and the
regulator will go into thermal shutdown. Internal thermal
shutdown circuitry protects the device from permanent
damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7 4-layer board.
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2
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TA= +25°C, unless otherwise noted.
Parameter
Symbol Condition
Feedback Voltage
Upper Switch On Resistance
Upper Switch Leakage
Current Limit
COMP to Current Sense
Transconductance
Error Amp Voltage Gain
Error Amp Transconductance
Error Amp Min Source current
Error Amp Min Sink current
VIN UVLO Threshold
VIN UVLO Hysteresis
Soft-Start Time
Oscillator Frequency
Shutdown Supply Current
Quiescent Supply Current
VFB
RDS(ON)
4.5V < VIN < 42V
VBST – VSW = 5V
VEN = 0V, VSW = 0V, VIN = 42V
Duty Cycle = 50%
Min
Typ
Max
Units
0.776
0.8
220
1
3.5
0.824
V
mΩ
µA
A
2.9
8
GCS
ICOMP = ±3µA
VFB = 0.7V
VFB = 0.9V
40
2.7
10% < VOUT < 90%
RFREQ = 45kΩ
RFREQ = 18kΩ
VEN = 0V
No load, VFB = 0.9V
1.6
3.2
Thermal Shutdown
Thermal Shutdown Hysteresis
(6)
Minimum Off Time
Minimum On Time (6)
EN Up Threshold
EN Threshold Hysteresis
1.3
200
60
5
–5
3.0
0.4
1.3 (5)
2
4
12
120
A/V
80
3.3
2.4
4.8
20
145
V/V
µA/V
µA
µA
V
V
ms
MHz
MHz
µA
µA
150
°C
15
°C
ns
ns
V
mV
100
100
1.5
300
1.7
Note:
5) The Soft-Start Time is measured based on VOUT from 10% to 90% and multiplied by 1.25 to get 0% to 100% Soft-Start Time.
6) Guaranteed by design.
MP2560 Rev. 1.2
1/25/2010
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3
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
PIN FUNCTIONS
QFN
Pin #
SOIC8
Pin #
Name
1, 2
1
SW
3
2
EN
4
3
COMP
5
4
FB
6
5
7
6
8, 9
7
10
8
MP2560 Rev. 1.2
1/25/2010
Description
Switch Node. This is the output from the high-side switch. A low forward drop
Schottky diode to ground is required. The diode must be close to the SW pins to
reduce switching spikes.
Enable Input. Pulling this pin below the specified threshold shuts the chip down.
Pulling it up above the specified threshold or leaving it floating enables the chip.
Compensation. This node is the output of the error amplifier. Control loop frequency
compensation is applied to this pin.
Feedback. This is the input to the error amplifier. The output voltage is set by a
resistive divider connected between the output and GND which scales down VOUT
equal to the internal +0.8V reference.
GND,
Ground. It should be connected as close as possible to the output capacitor to
Exposed
shorten the high current switch paths. Connect exposed pad to ground plane.
Pad
Switching Frequency Program Input. Connect a resistor from this pin to ground to set
FREQ
the switching frequency.
Input Supply. This supplies power to all the internal control circuitry, both BS
VIN
regulators and the high-side switch. A decoupling capacitor to ground must be placed
close to this pin to minimize switching spikes.
Bootstrap. This is the positive power supply for the internal floating high-side
BST
MOSFET driver. Connect a bypass capacitor between this pin and SW pin.
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4
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
VIN = 12V, VOUT =5V, C1 = 10µF, C2 = 22µF, L1 = 10µH and TA = +25°C, unless otherwise noted.
EFFICIENCY (%)
90
VIN=12V
90
80
70
VIN=12V
60
50
40
OSCILLATIONG EFFICIENCY (KHz)
100
Efficiency vs
Load Current
VIN=5V
EFFICIENCY (%)
100
Efficiency vs
Load Current
VIN=24V
VOUT=2.5V
fs=500kHz
0
0.5
1.0
1.5
2.0
LOAD CURRENT (A)
2.5
VIN=24V
80
70
60
50
40
VOUT=5V
fs=500kHz
0
0.5
1.0
1.5
2.0
LOAD CURRENT (A)
Steady State
Steady State
IOUT = 0.1A
IOUT = 1A
2.5
Oscillating Frequency
vs Rfreq
4000
3500
3000
2500
2000
1500
1000
500
0
10
IOUT = 2A
VOUT
AC Coupled
20mV/div.
VOUT
AC Coupled
20mV/div.
VSW
10V/div.
VSW
10V/div.
VSW
10V/div.
IL
1A/div.
IL
2A/div.
MP2560 Rev. 1.2
1/25/2010
1000
Steady State
VOUT
AC Coupled
10mV/div.
IL
1A/div.
100
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5
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
VIN = 12V, VOUT =5V, C1 = 10µF, C2 = 22µF, L1 = 10µH and TA = +25°C, unless otherwise noted.
Startup
Shutdown
Startup
IOUT = 0.1A
IOUT = 0.1A
IOUT = 1A
VEN
5V/div.
VEN
5V/div.
VEN
5V/div.
VOUT
2V/div.
VSW
10V/div.
VOUT
2V/div.
VOUT
2V/div.
VSW
10V/div.
VSW
10V/div.
IL
1A/div.
IL
1A/div.
IL
1A/div.
1ms/div.
1ms/div.
Shutdown
Startup
IOUT = 1A
IOUT = 2A
1ms/div.
Shutdown
IOUT = 2A
VEN
5V/div.
VEN
5V/div.
VEN
5V/div.
VOUT
2V/div.
VOUT
2V/div.
VOUT
2V/div.
VSW
10V/div.
VSW
10V/div.
VSW
10V/div.
IL
1A/div.
IL
2A/div.
IL
2A/div.
1ms/div.
Short Circuit Entry
Short Circuit Recovery
IOUT = 0.1A to Short
IOUT = Short to 0.1A
VOUT
2V/div.
VOUT
2V/div.
IL
1A/div.
IL
1A/div.
MP2560 Rev. 1.2
1/25/2010
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MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
BLOCK DIAGRAM
VIN
VIN
EN
REFERENCE UVLO/
THERMAL
SHUTDOWN
5V +
-2.6V
INTERNAL
REGULATORS
+
-BST
SW
SS Time
VOUT
SS
--
ISW
+
ISW
Level
Shift
FB
SW
Gm Error Amp
SS
0V8
--
COMP
+
OSCILLATOR
CLK
VOUT
COMP
GND
FREQ
Figure 1—Functional Block Diagram
OPERATION
The MP2560 is a variable frequency,
non-synchronous,
step-down
switching
regulator with an integrated high-side high
voltage power MOSFET. It provides a single
highly efficient solution with current mode
control for fast loop response and easy
compensation. It features a wide input voltage
range, internal soft-start control and precision
current limiting. Its very low operational
quiescent current makes it suitable for battery
powered applications.
MP2560 Rev. 1.2
1/25/2010
PWM Control
At moderate to high output current, the MP2560
operates in a fixed frequency, peak current
control mode to regulate the output voltage. A
PWM cycle is initiated by the internal clock. The
power MOSFET is turned on and remains on
until its current reaches the value set by the
COMP voltage. When the power switch is off, it
remains off for at least 100ns before the next
cycle starts. If, in one PWM period, the current
in the power MOSFET does not reach the
COMP set current value, the power MOSFET
remains on, saving a turn-off operation.
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7
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
Error Amplifier
The error amplifier compares the FB pin voltage
with the internal reference (REF) and outputs a
current proportional to the difference between
the two. This output current is then used to
charge the external compensation network to
form the COMP voltage, which is used to
control the power MOSFET current.
During operation, the minimum COMP voltage
is clamped to 0.9V and its maximum is clamped
to 2.0V. COMP is internally pulled down to GND
in shutdown mode. COMP should not be pulled
up beyond 2.6V.
Internal Regulator
Most of the internal circuitries are powered from
the 2.6V internal regulator. This regulator takes
the VIN input and operates in the full VIN range.
When VIN is greater than 3.0V, the output of
the regulator is in full regulation. When VIN is
lower than 3.0V, the output decreases.
Enable Control
The MP2560 has a dedicated enable control pin
(EN). With high enough input voltage, the chip
can be enabled and disabled by EN which has
positive logic. Its falling threshold is a precision
1.2V, and its rising threshold is 1.5V (300mV
higher).
When floating, EN is pulled up to about 3.0V by
an internal 1µA current source so it is enabled.
To pull it down, 1µA current capability is needed.
When EN is pulled down below 1.2V, the chip is
put into the lowest shutdown current mode.
When EN is higher than zero but lower than its
rising threshold, the chip is still in shutdown
mode but the shutdown current increases
slightly.
Under-Voltage Lockout (UVLO)
Under-voltage lockout (UVLO) is implemented
to protect the chip from operating at insufficient
supply voltage. The UVLO rising threshold is
about 3.0V while its falling threshold is a
consistent 2.6V.
MP2560 Rev. 1.2
1/25/2010
Internal Soft-Start
The soft-start is implemented to prevent the
converter output voltage from overshooting
during startup. When the chip starts, the
internal circuitry generates a soft-start voltage
(SS) ramping up from 0V to 2.6V. When it is
lower than the internal reference (REF), SS
overrides REF so the error amplifier uses SS as
the reference. When SS is higher than REF,
REF regains control.
Thermal Shutdown
Thermal shutdown is implemented to prevent
the chip from operating at exceedingly high
temperatures. When the silicon die temperature
is higher than its upper threshold, it shuts down
the whole chip. When the temperature is lower
than its lower threshold, the chip is enabled
again.
Floating Driver and Bootstrap Charging
The floating power MOSFET driver is powered
by an external bootstrap capacitor. This floating
driver has its own UVLO protection. This
UVLO’s rising threshold is 2.2V with a
hysteresis of 150mV.
The bootstrap capacitor is charged and
regulated to about 5V by the dedicated internal
bootstrap regulator. When the voltage between
the BST and SW nodes is lower than its
regulation, a PMOS pass transistor connected
from VIN to BST is turned on. The charging
current path is from VIN, BST and then to SW.
External circuit should provide enough voltage
headroom to facilitate the charging.
As long as VIN is sufficiently higher than SW,
the bootstrap capacitor can be charged. When
the power MOSFET is ON, VIN is about equal
to SW so the bootstrap capacitor cannot be
charged. When the external diode is on, the
difference between VIN and SW is largest, thus
making it the best period to charge. When there
is no current in the inductor, SW equals the
output voltage VOUT so the difference between
VIN and VOUT can be used to charge the
bootstrap capacitor.
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8
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
At higher duty cycle operation condition, the
time period available to the bootstrap charging
is less so the bootstrap capacitor may not be
sufficiently charged.
In case the internal circuit does not have
sufficient voltage and the bootstrap capacitor is
not charged, extra external circuitry can be
used to ensure the bootstrap voltage is in the
normal operational region. Refer to External
Bootstrap Diode in Application section.
The DC quiescent current of the floating driver
is about 20µA. Make sure the bleeding current
at the SW node is higher than this value, such
that:
IO +
VO
> 20µA
(R1 + R2)
Current Comparator and Current Limit
The power MOSFET current is accurately
sensed via a current sense MOSFET. It is then
fed to the high speed current comparator for the
current mode control purpose. The current
comparator takes this sensed current as one of
its inputs. When the power MOSFET is turned
on, the comparator is first blanked till the end of
the turn-on transition to avoid noise issues. The
comparator then compares the power switch
current with the COMP voltage. When the
sensed current is higher than the COMP
voltage, the comparator output is low, turning
off the power MOSFET. The cycle-by-cycle
maximum current of the internal power
MOSFET is internally limited.
MP2560 Rev. 1.2
1/25/2010
Startup and Shutdown
If both VIN and EN are higher than their
appropriate thresholds, the chip starts. The
reference block starts first, generating stable
reference voltage and currents, and then the
internal regulator is enabled. The regulator
provides stable supply for the remaining
circuitries.
While the internal supply rail is up, an internal
timer holds the power MOSFET OFF for about
50µs to blank the startup glitches. When the
internal soft-start block is enabled, it first holds
its SS output low to ensure the remaining
circuitries are ready and then slowly ramps up.
Three events can shut down the chip: EN low,
VIN low and thermal shutdown. In the shutdown
procedure, power MOSFET is turned off first to
avoid any fault triggering. The COMP voltage
and the internal supply rail are then pulled down.
Programmable Oscillator
The MP2560 oscillating frequency is set by an
external resistor, Rfreq from the FREQ pin to
ground. The value of Rfreq can be calculated
from:
R freq (KΩ) =
180000
fs (KHz)1.1
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9
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
APPLICATION INFORMATION
COMPONENT SELECTION
Setting the Output Voltage
The output voltage is set using a resistive
voltage divider from the output voltage to FB pin.
The voltage divider divides the output voltage
down to the feedback voltage by the ratio:
VFB = VOUT
R2
R1 + R2
Thus the output voltage is:
VOUT = VFB
(R1 + R2)
R2
About 20µA current from high side BS circuitry
can be seen at the output when the MP2560is
at no load. In order to absorb this small amount
of current, keep R2 under 40KΩ. A typical
value for R2 can be 40.2kΩ. With this value, R1
can be determined by:
R1 = 50.25 × ( VOUT − 0.8)(kΩ)
For example, for a 3.3V output voltage, R2 is
40.2kΩ, and R1 is 124kΩ.
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will result in less ripple current that will
result in lower output ripple voltage. However,
the larger value inductor will have a larger
physical size, higher series resistance, and/or
lower saturation current.
MP2560 Rev. 1.2
1/25/2010
A good rule for determining the inductance to
use is to allow the peak-to-peak ripple current in
the inductor to be approximately 30% of the
maximum switch current limit. Also, make sure
that the peak inductor current is below the
maximum switch current limit. The inductance
value can be calculated by:
L1 =
⎛
⎞
VOUT
V
× ⎜1 − OUT ⎟⎟
fS × ∆IL ⎜⎝
VIN ⎠
Where VOUT is the output voltage, VIN is the input
voltage, fS is the switching frequency, and ∆IL is
the peak-to-peak inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
ILP = ILOAD +
⎛
⎞
VOUT
V
× ⎜⎜1 − OUT ⎟⎟
2 × fS × L1 ⎝
VIN ⎠
Where ILOAD is the load current.
Table 1 lists a number of suitable inductors
from various manufacturers. The choice of
which style inductor to use mainly depends on
the price vs. size requirements, the switching
frequency, and any EMI requirement.
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MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
Table 1—Inductor Selection Guide
Inductance (µH)
Max DCR (Ω)
Current Rating (A)
Dimensions
L x W x H (mm3)
7447789004
4.7
0.033
2.9
7.3x7.3x3.2
744066100
10
0.035
3.6
10x10x3.8
744771115
15
0.025
3.75
12x12x6
744771122
22
0.031
3.37
12x12x6
Part Number
Wurth Electronics
TDK
RLF7030T-4R7
4.7
0.031
3.4
7.3x6.8x3.2
SLF10145T-100
10
0.0364
3
10.1x10.1x4.5
SLF12565T-150M4R2
15
0.0237
4.2
12.5x12.5x6.5
SLF12565T-220M3R5
22
0.0316
3.5
12.5x12.5x6.5
Toko
FDV0630-4R7M
4.7
0.049
3.3
7.7x7x3
919AS-100M
10
0.0265
4.3
10.3x10.3x4.5
919AS-160M
16
0.0492
3.3
10.3x10.3x4.5
919AS-220M
22
0.0776
3
10.3x10.3x4.5
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off. To
reduce losses due to the diode forward voltage
and recovery times, use a Schottky diode.
Choose a diode whose maximum reverse
voltage rating is greater than the maximum
input voltage, and whose current rating is
greater than the maximum load current. Table 2
lists
example
Schottky
diodes
and
manufacturers.
Table 2—Diode Selection Guide
Diodes
B380-13-F
B390
CMSH3-100MA
MP2560 Rev. 1.2
1/25/2010
Voltage/
Current
Rating
80V, 3A
90V, 3A
100V, 3A
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
may also suffice.
For simplification, choose the input capacitor
with RMS current rating greater than half of the
maximum load current.
Manufacturer
Diodes Inc.
Diodes Inc.
Central Semi
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11
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
The input capacitor (C1) can be electrolytic,
tantalum or ceramic. When using electrolytic or
tantalum capacitors, a small, high quality
ceramic capacitor, i.e. 0.1µF, should be placed
as close to the IC as possible. When using
ceramic capacitors, make sure that they have
enough capacitance to provide sufficient charge
to prevent excessive voltage ripple at input. The
input voltage ripple caused by capacitance can
be estimated by:
∆VIN =
⎛
ILOAD
V
V
× OUT × ⎜1 − OUT
fS × C1 VIN ⎜⎝
VIN
⎞
⎟⎟
⎠
Output Capacitor
The output capacitor (C2) is required to
maintain the DC output voltage. Ceramic,
tantalum, or low ESR electrolytic capacitors are
recommended. Low ESR capacitors are
preferred to keep the output voltage ripple low.
The output voltage ripple can be estimated by:
∆VOUT =
VOUT ⎛
V
× ⎜⎜1 − OUT
fS × L ⎝
VIN
⎞
⎞ ⎛
1
⎟
⎟⎟ × ⎜ R ESR +
⎜
8 × f S × C2 ⎟⎠
⎠ ⎝
Where L is the inductor value and RESR is the
equivalent series resistance (ESR) value of the
output capacitor.
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance. The output
voltage ripple is mainly caused by the
capacitance. For simplification, the output
voltage ripple can be estimated by:
∆VOUT =
⎛
⎞
V
× ⎜⎜1 − OUT ⎟⎟
VIN ⎠
× L × C2 ⎝
VOUT
8 × fS
2
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the
output ripple can be approximated to:
∆VOUT =
VOUT ⎛
V
× ⎜1 − OUT
f S × L ⎜⎝
VIN
⎞
⎟⎟ × R ESR
⎠
The characteristics of the output capacitor also
affect the stability of the regulation system. The
MP2560 can be optimized for a wide range of
capacitance and ESR values.
MP2560 Rev. 1.2
1/25/2010
Compensation Components
MP2560 employs current mode control for easy
compensation and fast transient response. The
system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal error amplifier. A
series capacitor-resistor combination sets a
pole-zero
combination
to
control
the
characteristics of the control system. The DC
gain of the voltage feedback loop is given by:
A VDC = R LOAD × G CS × A VEA ×
VFB
VOUT
Where AVEA is the error amplifier voltage gain,
200V/V;
GCS
is
the
current
sense
transconductance, 8A/V; RLOAD is the load
resistor value.
The system has two poles of importance. One
is due to the compensation capacitor (C3), the
output resistor of error amplifier. The other is
due to the output capacitor and the load resistor.
These poles are located at:
fP1 =
GEA
2π × C3 × A VEA
fP2 =
1
2π × C2 × R LOAD
Where,
GEA
is
the
transconductance, 60µA/V.
error
amplifier
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). This zero is located
at:
f Z1 =
1
2π × C3 × R3
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero,
due to the ESR and capacitance of the output
capacitor, is located at:
fESR =
1
2π × C2 × R ESR
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MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
In this case (as shown in Figure 2), a third pole
set by the compensation capacitor (C6) and the
compensation resistor (R3) is used to
compensate the effect of the ESR zero on the
loop gain. This pole is located at:
f P3 =
1
2π × C6 × R3
1. Choose the compensation resistor (R3) to set
the desired crossover frequency. Determine the
R3 value by the following equation:
R3 =
2π × C2 × f C VOUT
×
G EA × G CS
VFB
Where fC is the desired crossover frequency.
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
where the feedback loop has the unity gain is
important. Lower crossover frequencies result
in slower line and load transient responses,
while higher crossover frequencies could cause
system unstable. A good rule of thumb is to set
the crossover frequency to approximately onetenth of the switching frequency. The Table 3
lists the typical values of compensation
components for some standard output voltages
with various output capacitors and inductors.
The values of the compensation components
have been optimized for fast transient
responses and good stability at given conditions.
Table 3—Compensation Values for Typical
Output Voltage/Capacitor Combinations
VOUT
(V)
L1 (µH)
C2
(µF)
R3
(kΩ)
C3
(pF)
C6
1.8
4.7
47
105
100
None
2.5
4.7 - 6.8
33
54.9
330
None
3.3
6.8 -10
22
68.1
220
None
5
15 - 22
22
100
150
None
12
22
33
147
150
None
2. Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, below one forth of
the crossover frequency provides sufficient
phase margin. Determine the C3 value by the
following equation:
C3 >
4
2π × R3 × f C
3. Determine if the second compensation
capacitor (C6) is required. It is required if the
ESR zero of the output capacitor is located at
less than half of the switching frequency, or the
following relationship is valid:
f
1
< S
2π × C2 × R ESR
2
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3
at the location of the ESR zero. Determine the
C6 value by the equation:
C6 =
C2 × R ESR
R3
To optimize the compensation components for
conditions not listed in Table 3, the following
procedure can be used.
MP2560 Rev. 1.2
1/25/2010
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MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
High Frequency Operation
The switching frequency of MP2560 can be
programmed up to 4MHz by an external resistor.
Please pay attention to the following if the
switching frequency is above 2MHz.
The minimum on time of MP2560 is about
100ns (typ). Pulse skipping operation can be
seen more easily at higher switching frequency
due to the minimum on time. Recommended
operating voltage is 12V or below, and 24V or
below at 2MHz. Refer to Figure 2 below for
detailed information.
30
Recommended VIN (max)
vs Switching Frequency
VIN (MAX) (V)
25
20
15
VOUT=3.3V
10
VOUT=2.5V
5
1500 2000 2500 3000 3500 4000
fs (KHz)
Figure 2—Recommend Max VIN vs. fS
Since the internal bootstrap circuitry has higher
impedance, which may not be adequate to
charge the bootstrap capacitor during each (1D)×Ts charging period, an external bootstrap
charging diode is strongly recommended if the
switching frequency is above 2MHz (see
External Bootstrap Diode section for detailed
implementation information).
Layout becomes more important when the
device switches at higher frequency. It is
essential to place the input decoupling
capacitor, catch diode and the MP2560 (Vin pin,
SW pin and PGND) as close as possible, with
traces that are very short and fairly wide. This
can help to greatly reduce the voltage spike on
SW node, and lower the EMI noise level as well.
Try to run the feedback trace as far from the
inductor and noisy power traces as possible. It
is often a good idea to run the feedback trace
on the side of the PCB opposite of the inductor
with a ground plane separating the two. The
compensation components should be placed
closed to the MP2560. Do not place the
compensation components close to or under
high dv/dt SW node, or inside the high di/dt
power loop. If you have to do so, the proper
ground plane must be in place to isolate those.
Switching loss is expected to be increased at
high switching frequency. To help to improve
the thermal conduction, a grid of thermal vias
can be created right under the exposed pad. It
is recommended that they be small (15mil
barrel diameter) so that the hole is essentially
filled up during the plating process, thus aiding
conduction to the other side. Too large a hole
can cause ‘solder wicking’ problems during the
reflow soldering process. The pitch (distance
between the centers) of several such thermal
vias in an area is typically 40mil. Please refer to
the layout example on EV2560 datasheet.
With higher switching frequencies, the inductive
reactance (XL) of capacitor comes to dominate,
so that the ESL of input/output capacitor
determines the input/output ripple voltage at
higher switching frequency. As a result of that,
high frequency ceramic capacitor is strongly
recommended as input decoupling capacitor
and output filtering capacitor for such high
frequency operation.
MP2560 Rev. 1.2
1/25/2010
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14
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
External Bootstrap Diode
It is recommended that an external bootstrap
diode be added when the input voltage is no
greater than 5V or the 5V rail is available in the
system. This helps improve the efficiency of the
regulator. The bootstrap diode can be a low
cost one such as IN4148 or BAT54.
5V
BS
MP2560
0.1µ F
SW
This diode is also recommended for high duty
cycle operation (when VOUT /VIN >65%) or low
VIN (<5Vin) applications.
At no load or light load, the converter may
operate in pulse skipping mode in order to
maintain the output voltage in regulation. Thus
there is less time to refresh the BS voltage. In
order to have enough gate voltage under such
operating conditions, the difference of VIN –VOUT
should be greater than 3V. For example, if the
VOUT is set to 3.3V, the VIN needs to be higher
than 3.3V+3V=6.3V to maintain enough BS
voltage at no load or light load. To meet this
requirement, EN pin can be used to program
the input UVLO voltage to Vout+3V.
Figure 3—External Bootstrap Diode
MP2560 Rev. 1.2
1/25/2010
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15
MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
TYPICAL APPLICATION CIRCUITS
C4
100nF
10
8,9
VIN
C1
10uF
50V
SW
L1
4.7uH
1,2
C2
47uF
6.3V
D1
3
EN
VIN
BST
7
EN
MP2560
FB
COMP
FREQ
VOUT
1.8V
5
4
C3
100pF
GND
6
C6
NS
Figure 4—1.8V Output Typical Application Schematic
C4
100nF
10
8,9
VIN
C1
10uF
50V
EN
VIN
BST
SW
L1
15uH
1,2
C2
22uF
6.3V
D1
3
7
EN
MP2560
FB
COMP
FREQ
GND
6
VOUT
5V
5
4
C3
150pF
C6
NS
Figure 5—5V Output Typical Application Schematic
MP2560 Rev. 1.2
1/25/2010
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MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
PCB LAYOUT GUIDE
PCB layout is very important to achieve stable
operation. It is highly recommended to duplicate
EVB layout for optimum performance.
2)
Bypass ceramic capacitors are suggested
to be put close to the VIN Pin.
3)
Ensure all feedback connections are short
and direct. Place the feedback resistors
and compensation components as close to
the chip as possible.
4)
Route SW away from sensitive analog
areas such as FB.
5)
Connect IN, SW, and especially GND
respectively to a large copper area to cool
the chip to improve thermal performance
and long-term reliability.
If change is necessary, please follow these
guidelines and take Figure 6 for reference.
1)
Keep the path of switching current short
and minimize the loop area formed by Input
cap, high-side MOSFET and external
switching diode.
4.5V to 42V
MP2560 Typical Application Circuit
TOP Layer
Bottom Layer
Figure 6―MP2560 Typical Application Circuit and PCB Layout Guide
MP2560 Rev. 1.2
1/25/2010
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MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
PACKAGE INFORMATION
3mm x 3mm QFN10
2.90
3.10
0.30
0.50
PIN 1 ID
MARKING
0.18
0.30
2.90
3.10
PIN 1 ID
INDEX AREA
1.45
1.75
PIN 1 ID
SEE DETAIL A
10
1
2.25
2.55
0.50
BSC
5
6
TOP VIEW
BOTTOM VIEW
PIN 1 ID OPTION A
R0.20 TYP.
PIN 1 ID OPTION B
R0.20 TYP.
0.80
1.00
0.20 REF
0.00
0.05
SIDE VIEW
DETAIL A
NOTE:
2.90
0.70
1) ALL DIMENSIONS ARE IN MILLIMETERS.
2) EXPOSED PADDLE SIZE DOES NOT INCLUDE MOLD FLASH.
3) LEAD COPLANARITY SHALL BE 0.10 MILLIMETER MAX.
4) DRAWING CONFORMS TO JEDEC MO-229, VARIATION VEED-5.
5) DRAWING IS NOT TO SCALE.
1.70
0.25
2.50
0.50
RECOMMENDED LAND PATTERN
MP2560 Rev. 1.2
1/25/2010
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MP2560 – 2.5A, 4MHz, 42V STEP-DOWN CONVERTER
SOIC8E (EXPOSED PAD)
0.189(4.80)
0.197(5.00)
0.124(3.15)
0.136(3.45)
8
5
0.150(3.80)
0.157(4.00)
PIN 1 ID
1
0.228(5.80)
0.244(6.20)
0.089(2.26)
0.101(2.56)
4
TOP VIEW
BOTTOM VIEW
SEE DETAIL "A"
0.051(1.30)
0.067(1.70)
SEATING PLANE
0.000(0.00)
0.006(0.15)
0.013(0.33)
0.020(0.51)
0.0075(0.19)
0.0098(0.25)
SIDE VIEW
0.050(1.27)
BSC
FRONT VIEW
0.010(0.25)
x 45o
0.020(0.50)
GAUGE PLANE
0.010(0.25) BSC
0.050(1.27)
0.024(0.61)
0o-8o
0.016(0.41)
0.050(1.27)
0.063(1.60)
DETAIL "A"
0.103(2.62)
0.138(3.51)
RECOMMENDED LAND PATTERN
0.213(5.40)
NOTE:
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN
BRACKET IS IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,
PROTRUSIONS OR GATE BURRS.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
OR PROTRUSIONS.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.004" INCHES MAX.
5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.
6) DRAWING IS NOT TO SCALE.
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
MP2560 Rev. 1.2
1/25/2010
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19