LINER LTC3448EMS8E 1.5mhz/2.25mhz, 600ma synchronous step-down regulator with ldo mode Datasheet

LTC3448
1.5MHz/2.25MHz, 600mA
Synchronous Step-Down
Regulator with LDO Mode
DESCRIPTIO
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FEATURES
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The LTC®3448 is a high efficiency, monolithic, synchronous
buck regulator using a constant frequency, current mode
architecture. Supply current during operation is only 32µA
(linear regulator mode) and drops to <1µA in shutdown. The
2.5V to 5.5V input voltage range makes the LTC3448 ideally suited for single Li-Ion battery-powered applications.
100% duty cycle provides low dropout operation, extending battery life in portable systems. At moderate output load
levels, PWM pulse skipping mode operation provides very
low output ripple voltage for noise sensitive applications.
High Efficiency: Up to 96%
Very Low Quiescent Supply Current: 32µA During
Linear Regulator Operation
600mA Output Current (Buck Converter)
Optionally Operates as Linear Regulator Below
3mA—External or Automatic ON/OFF
2.5V to 5.5V Input Voltage Range
1.5MHz or 2.25MHz Constant Frequency Operation
or External Synchronization
No Schottky Diode Required
Low Dropout Operation: 100% Duty Cycle
0.6V Reference Allows Low Output Voltages
Shutdown Mode Draws < 1µA Supply Current
Current Mode Operation for Excellent Line and
Load Transient Response
Overtemperature Protected
Low Profile (3mm × 3mm) 8-Lead DFN and 8-Lead
MSOP Packages
The LTC3448 automatically switches into linear regulator
operation at very low load currents to maintain <5mVP-P
output voltage ripple. Supply current in this mode is
typically 32µA. The switch to linear regulator mode occurs
at a threshold of 3mA. Linear regulator operation can be set
to on, off or automatic turn on/off.
Switching frequency is selectable at either 1.5MHz or
2.25MHz, allowing the use of small surface mount inductors and capacitors.
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APPLICATIO S
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Cellular Telephones
Personal Information Appliances
Wireless and DSL Modems
Digital Still Cameras
MP3 Players
Portable Instruments
The internal synchronous switch increases efficiency and
eliminates the need for an external Schottky diode. Low
output voltages are easily supported with the 0.6V feedback reference voltage. The LTC3448 is available in a low
profile 3mm × 3mm DFN package or thermally enhanced
8-lead MSOP.
, LTC and LT are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
Protected by U.S. Patents including 5481178, 6580258, 6304066, 6127815,
6498466, 6611131. Others pending.
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Efficiency and Power Loss vs Load Current
TYPICAL APPLICATIO
100
90
1.5V High Efficiency Regulator with Automatic LDO Mode
SW
VOUT
RUN
LTC3448
MODE
VFB
FREQ
SYNC
GND
3448 TA01a
22pF
474k
COUT
4.7µF
EFFICIENCY (%)
VOUT
1.5V
60
50
EFFICIENCY
POWER LOSS
0.01
40
30
316k
POWER LOSS (W)
VIN
CIN
4.7µF
0.1
70
2.2µH
VIN
2.5V TO 5.5V
80
1
VIN = 3.6V
VOUT = 1.5V
TA = 25°C
0.001
20
10
0
0.0001
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
1
23448 TA01b
3448f
1
LTC3448
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W W
W
ABSOLUTE
AXI U
RATI GS
(Note 1)
Input Supply Voltage .................................. – 0.3V to 6V
RUN, SYNC Voltages ................... – 0.3V to (VIN + 0.3V)
MODE Voltage ............................. – 0.3V to (VIN + 0.3V)
FREQ, VFB Voltages...................... – 0.3V to (VIN + 0.3V)
SW Voltage .................................. – 0.3V to (VIN + 0.3V)
VOUT Voltage ................................ – 0.3V to (VIN + 0.3V)
P-Channel Switch Source Current (DC) ............. 800mA
N-Channel Switch Sink Current (DC) ................. 800mA
VOUT (LDO) Source Current .................................. 25mA
Peak SW Sink and Source Current ........................ 1.3A
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Junction Temperature (Notes 3, 7) ...................... 125°C
Storage Temperature Range ................ – 65°C to 125°C
Lead Temperature (Soldering, 10 sec)
MSOP Only ...................................................... 300°C
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PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
TOP VIEW
VFB 1
VOUT 2
MODE 3
8
9
VIN 4
7
RUN
LTC3448EDD
SYNC
6
FREQ
5
SW
DD PACKAGE
8-LEAD (3mm × 3mm) PLASTIC DFN
TOP VIEW
VFB 1
VOUT 2
MODE 3
VIN 4
DD PART MARKING
TJMAX = 125°C, θJA = 43°C/ W
EXPOSED PAD (PIN 9) IS GND
MUST BE SOLDERED TO PCB
ORDER PART
NUMBER
LBMJ
9
8
7
6
5
LTC3448EMS8E
RUN
SYNC
FREQ
SW
MS8E PACKAGE
8-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 40°C/ W
EXPOSED PAD (PIN 9) IS GND
MUST BE SOLDERED TO PCB
MS8 PART MARKING
LTBMK
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified.
SYMBOL
PARAMETER
IVFB
Feedback Current
CONDITIONS
MIN
VFB
Regulated Feedback Voltage
(Note 4)
TA = 25°C
0°C ≤ TA ≤ 85°C
–40°C ≤ TA ≤ 85°C
●
∆VFB
Reference Voltage Line Regulation
VIN = 2.5V to 5.5V (Note 4)
●
∆VOVL
Output Overvoltage Lockout
∆VOVL = VOVL – VFB
∆VOVL = (VOVL – VOUT) • 100/VOUT
∆VOUT
Output Voltage Line Regulation
VIN = 2.5V to 5.5V (LDO)
IPK
Peak Inductor Current
VFB = 0.5V or VOUT = 90%,
Duty Cycle < 35%
VLOADREG
Output Voltage Load Regulation
LDO, 1mA to 10mA
VOUT(MAX)
Maximum Output Voltage
(Note 9)
VIN
Input Voltage Range
TYP
MAX
UNITS
±30
nA
0.6
0.6
0.6
0.6120
0.6135
0.6150
V
V
V
0.2
0.4
%/V
35
5.8
55
9.2
mV
%
0.1
0.8
%/V
1
1.3
A
●
0.5880
0.5865
0.5850
15
2.5
0.7
0.5
%/V
VIN – 0.7 VIN – 0.3
●
2.5
V
5.5
V
3448f
2
LTC3448
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
IS
Input DC Bias Current
Active Mode (Pulse Skip, No LRO)
MIN
TYP
MAX
UNITS
VIN = 3.6V (Note 5)
VFB = 0.5V or VOUT = 90%, ILOAD = 0A, 1.5MHz
VFB = 0.5V or VOUT = 90%, ILOAD = 0A, 2.25MHz
250
275
375
400
µA
µA
Linear Regulator Operation (LRO)
ILOAD ≤ ILDO(ON)
32
43
µA
Shutdown
VRUN = 0V, VIN = 5.5V
0.1
1
µA
fOSC
Oscillator Frequency
FREQ = Low, VIN = 3.6V
FREQ = High
1.5
2.25
1.8
2.7
MHz
MHz
fSYNC
Synchronization Frequency
(Note 6)
>4
MHz
VTH(SYNC)
SYNC Activation Input Threshold
RPFET
RDS(ON) of P-Channel FET
ISW = 100mA
0.4
Ω
RNFET
RDS(ON) of N-Channel FET
ISW = –150mA
0.35
Ω
ILSW
SW Leakage
VRUN = 0V, VSW = 0V or 5V, VIN = 5V
VRUNH
RUN Threshold High
●
VRUNL
RUN Threshold Low
●
IRUN
RUN Leakage Current
●
VFREQH
FREQ Threshold High
●
VFREQL
FREQ Threshold Low
●
IFREQ
FREQ Leakage Current
●
VMODEH
MODE Threshold High
●
VMODEL
MODE Threshold Low
●
IMODE
MODE Leakage Current
●
±0.1
±1
µA
ISYNC
SYNC Leakage Current
●
±0.01
±1
µA
ILDO(ON)
LRO ON Load Current Threshold
3
5
mA
ILDO(OFF)
LRO OFF Load Current Threhold
11
17
mA
●
●
1.2
1.8
1.5
1
±0.01
±1
1.5
±0.01
µA
0.3
V
±1
µA
V
±0.01
1
V
±1
µA
VIN – 0.15
V
0.12
8
V
V
VIN – 1
2.2mH Inductor (Note 8)
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC3448E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD)(43°C/W)
Note 4: The LTC3448 is tested in a proprietary test mode that connects
VFB to the output of the error amplifier.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. LRO is “linear regulator operation.”
1.3
V
Note 6: 4MHz operation is guaranteed by design but is not production
tested and is subject to duty cycle limitations.
Note 7: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.
Note 8: The load current below which the switching regulator turns off and
the LDO turns on is, to first order, inversely proportional to the value of
the inductor. This effect is covered in more detail in the Operation section.
This parameter is not production tested but is guaranteed by design.
Note 9: For 2.5V < VIN < 2.7V the output voltage is limited to VIN – 0.7V
to ensure regulation in linear regulator mode. This parameter is not
production tested but is guaranteed by design.
3448f
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LTC3448
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure1a Except for the Resistive Divider Resistor Values)
100
VOUT = 1.8V
TA = 25°C
90
IOUT = 30mA
85
80
EFFICIENCY (%)
EFFICIENCY (%)
90
IOUT = 600mA
75
70
80
70
60
50
40
65
20
55
10
3
4
5
INPUT VOLTAGE (V)
0.001
0.01
0.1
LOAD CURRENT (A)
0.615
60
50
40
30
20
1.70
VIN = 3.6V
1.60
0.605
0.600
0.595
50
25
75
0
TEMPERATURE (°C)
1.30
–50
125
OUTPUT VOLTAGE (V)
1.5
1.4
1.3
1.2
6
3448 G07
0
50
75
25
TEMPERATURE (°C)
VIN = 3.6V
1.520 TA = 25°C
0.38
1.515
0.36
1.510
0.34
1.505
1.500
1.495
125
SYNCHRONOUS
SWITCH
0.28
0.26
0.24
1.480
0.22
1.475
0.0001
0.20
3448 G08
MAIN
SWITCH
0.30
1.490
1
TA = 25°C
0.32
1.485
0.01
0.001
0.1
LOAD CURRENT (A)
100
RDS(ON) vs Input Voltage
0.40
RDS(ON) (Ω)
TA = 25°C
1.6
–25
3448 G06
Output Voltage vs Load Current
1.7
FREQUENCY (MHz)
100
1.525
3
4
5
SUPPLY VOLTAGE (V)
1.45
3448 G05
Oscillator Frequency
vs Supply Voltage
2
1.50
1.35
23448 G04
1.8
1.55
1.40
0.585
–50 –25
1
VIN = 3.6V
1.65
10
0.01
0.1
0.001
LOAD CURRENT (A)
1
Oscillator Frequency
vs Temperature
0.590
0
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
23448 G03
FREQUENCY (MHz)
REFERENCE VOLTAGE (V)
EFFICIENCY (%)
0
0.0001
1
0.610
70
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
Reference Voltage
vs Temperature
VIN = 2.7V
VOUT = 2.5V
TA = 25°C
80
40
23448 G02
Efficiency vs Load Current
(Switcher Only)
90
50
20
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
3448 G01
100
60
30
0
0.0001
6
VOUT = 1.5V
TA = 25°C
90
70
60
2
VOUT = 1.2V
TA = 25°C
80
30
50
100
EFFICIENCY (%)
IOUT = 100mA
95
Efficiency vs Load Current
Efficiency vs Load Current
Efficiency vs Input Voltage
100
2
3
4
5
INPUT VOLTAGE (V)
6
3448 G09
3448f
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LTC3448
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure1a Except for the Resistive Divider Resistor Values)
Dynamic Supply Current
vs Supply Voltage
0.6
DYNAMIC SUPPLY CURRENT (µA)
340
0.5
RDS(ON) (Ω)
0.4
0.3
0.2
0.1
MAIN SWITCH SYNCH SWITCH
2.5V
2.5V
3.6V
3.6V
4.2V
4.2V
0
–50
50
25
75
0
TEMPERATURE (°C)
–25
100
Dynamic Supply Current
vs Temperature
320
ILOAD = 0A
TA = 25°C
320
DYNAMIC SUPPLY CURRENT (µA)
RDS(ON) vs Temperature
300
280
2.25MHz
260
1.5MHz
240
220
200
125
3
2
4
SWITCH LEAKAGE (nA)
SWITCH LEAKAGE (nA)
VIN = 5.5V
RUN = 0V
250
MAIN
SWITCH
150
100
50
0
–50 –25
SYNCHRONOUS
SWITCH
1.5MHz
240
220
100
125
RUN = 0V
TA = 25°C
100
125
Start-Up from Shutdown
MAIN
SWITCH
RUN
5V/DIV
1
VOUT
1V/DIV
SYNCHRONOUS
SWITCH
0.1
IL
500mA/DIV
0.01
VIN = 3.6V
VOUT = 1.5V
ILOAD = 600mA
0
1
2
3
4
INPUT VOLTAGE (V)
5
40µs/DIV
3448 G15
6
3448 G14
3448 G13
Load Step
Load Step
VOUT
200mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
ILOAD
100mA/DIV
ILOAD
250mA/DIV
IL
500mA/DIV
IL
500mA/DIV
10µs/DIV
VIN = 3.6V
VOUT = 1.5V
ILOAD = 100µA TO 200mA
COUT = 10µF
50
25
75
0
TEMPERATURE (°C)
3448 G12
0.001
50
25
75
0
TEMPERATURE (°C)
260
Switch Leakage vs Input Voltage
10
200
2.25MHz
3448 G11
Switch Leakage vs Temperature
300
280
SUPPLY VOLTAGE (V)
3448 G10
350
300
200
–50 –25
6
5
VIN = 3.6V
ILOAD = 0A
3448 G16
10µs/DIV
VIN = 3.6V
VOUT = 1.5V
ILOAD = 50mA TO 600mA
COUT = 10µF
3448 G17
3448f
5
LTC3448
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1a Except for the Resistive Divider Resistor Values)
External Mode Control (Constant
1mA Load)
Load Step
VOUT
20mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
ILOAD
250mA/DIV
SWITCHER
SWITCHER
LDO
MODE PIN
2V/DIV
IL
500mA/DIV
VIN = 3.6V
10µs/DIV
VOUT = 1.5V
ILOAD = 100mA TO 600mA
3448 G18
VOUT = 1.5V
TA = 25°C
200µs/DIV
3448 G19
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PI FU CTIO S
VFB (Pin 1): Feedback Pin. This pin receives the feedback
voltage from an external resistive divider across the
output.
VIN (Pin 4): Main Supply Pin. This pin must be closely
decoupled to GND with a 2.2µF or greater ceramic
capacitor.
VOUT (Pin 2): Output Pin. This pin connects to an external
resistor divider and the linear regulator output. Connect
externally to the inductor and the output capacitor. The
internal linear regulator will supply current up to the
ILDO(OFF) current. Load currents above that are supplied by
the buck regulator. Internal circuitry automatically enables
the buck switching regulator at load currents higher than
the ILDO(OFF). The minimum required capacitance on this
pin is 2µF.
SW (Pin 5): Switch Node Connection to Inductor. This pin
connects to the drains of the internal main and synchronous power MOSFET switches.
MODE (Pin 3): Linear Regulator Control. Grounding this
pin turns off the linear regulator. Setting this pin to VIN
turns on the linear regulator regardless of the load current.
Tying this pin midrange (i.e., to VOUT) will place the linear
regulator in auto mode, where turn on/off is a function of
the load current. In applications where MODE is externally
driven high or low, this pin should be held low for 50µs
after the RUN pin is pulled high.
FREQ (Pin 6): Frequency Select. Switching frequency is
set to 1.5MHz when FREQ = 0V and to 2.25MHz when
FREQ = VIN. Do not float this pin.
SYNC (Pin 7): External Synchronization Pin. The oscillation frequency can be synchronized to an external oscillator applied to this pin. For external frequencies above
2.2MHz, pull FREQ high.
RUN (Pin 8): Run Control Input. Forcing this pin above
1.5V enables the part. Forcing this pin below 0.3V shuts
down the device. In shutdown, all functions are disabled
drawing <1µA supply current. Do not leave RUN floating.
Exposed Pad (Pin 9): Ground. This pin must be soldered
to PCB.
3448f
6
LTC3448
W
FU CTIO AL DIAGRA
U
U
SYNC
MODE
7
3
FREQ
LDO CONTROL
LOGIC
SLOPE
COMP
6
VIN
OSC
VOUT
LDO
DRIVE
2
4 VIN
–
+
0.6V
VFB
+
–
1
OSC
VIN
S
Q
R
Q
RS LATCH
RUN
8
0.6V REF
5Ω
+
ICOMP
– EA
–
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
ANTISHOOTTHRU
5 SW
OVDET
+
+
0.6V + ∆OVL
SHUTDOWN
IRCMP
9 GND
–
3448 F01
Figure 1
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OPERATIO (Refer to Functional Diagram)
Main Control Loop
The LTC3448 uses a constant frequency, current mode,
step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are
internal. During normal operation, the internal top power
MOSFET is turned on each cycle when the oscillator sets
the RS latch, and turned off when the current comparator,
ICOMP, resets the RS latch. The peak inductor current at
which ICOMP resets the RS latch, is controlled by the
output of error amplifier EA. When the load current
increases, it causes a slight decrease in the feedback
voltage FBINT relative to the 0.6V reference, which in turn,
causes the EA amplifier’s output voltage to increase until
the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is
turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator
IRCMP, or the beginning of the next clock cycle. The
comparator OVDET guards against transient overshoots
5.8% by turning off the main switch and keeping it off until
the fault is removed.
Pulse Skipping Mode Operation
At light loads, the inductor current may reach zero or
reverse on each pulse. The bottom MOSFET is turned off
by the current reversal comparator, IRCMP, and the switch
voltage will ring. This is discontinuous mode operation,
and is normal behavior for the switching regulator. At very
light loads, the LTC3448 will automatically skip pulses to
maintain output regulation.
Low Ripple LDO Mode Operation
At load currents below ILDO(ON), and when enabled, the
LTC3448 will switch into very low ripple, linear regulating
operation (LRO). In this mode, the current is sourced from
3448f
7
LTC3448
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OPERATIO (Refer to Functional Diagram)
the VOUT pin and both the main and synchronous switches
are turned off. The control loop is stabilized by the load
capacitor and requires a minimum value of 2µF. The
LTC3448 will change back to switching mode and turn off
the LDO when the load current exceeds approximately
11mA.
When MODE is connected to an intermediate voltage level
(i.e., VOUT), this switchover is automatic. If MODE is pulled
high to VIN, the LDO remains on and the switcher off
regardless of the load current. The LDO is capable of
providing a maximum of approximately 15mA before the
load regulation will degrade to unacceptable levels. If
MODE is pulled to GND, the switcher remains on and the
LDO off regardless of the load current.
4.5
4.0
VOUT = 1.2V
ILDO(ON) (mA)
3.5
VOUT = 1.5V
3.0
2.5
VOUT = 1.8V
2.0
1.5
1.0
TA = 25°C
L = 2.2µH
0.5
0
2
3
4
6
5
VIN (V)
3448 F02
Figure 2. ILDO(ON) vs VIN, VOUT
5.0
VIN = 3.6V
VOUT = 1.5V
TA = 25°C
4.5
4.0
ILDO(ON) (mA)
3.5
3.0
2.5
2.0
1.5
1.0
Some applications may be able to anticipate the transition
from high to low and low to high load currents. In these
cases it may be desirable to switch between modes by
controlling the MODE pin with a processor signal. In these
applications it is important that the MODE pin is pulled
high no earlier than 50µs after the RUN pin is pulled high.
This will ensure proper start-up of internal reference
circuitry.
The load current ILDO(ON) below which the switcher will
automatically turn off and the LDO turn on is independent
of the external capacitor, and to first order, independent
of supply and output voltage. There is an inverse relationship between ILDO(ON) and the value of the inductor.
These dependencies are shown in Figures 2 and 3.
Automatic operation with inductor values below 1µH is
not recommended.
At the low load currents at which the switcher to linear
regulator transition occurs, the switcher is operating in
pulse skipping mode. During each switching cycle in this
mode, while the synchronous switch (bottom MOSFET) is
on, the inductor current decays until the reverse current
comparator is triggered. At this occurrence, the bottom
MOSFET is turned off. Ideally, this occurs when the
inductor current is precisely zero. In reality, because of onchip delays, this current will be negative at higher output
voltages.
The internal algorithm which controls the LDO turn-on
load current level makes certain assumptions about the
amount of charge transferred to the output on each
switching cycle. These assumptions are no longer met
when the inductor current begins to reverse. This causes
the load current at which the transition takes place to move
to lower levels at higher output voltages. For this reason
use of the LDO auto mode is not recommended for output
levels above 2V. For output voltages above 2V, the MODE
pin should be driven externally.
Short-Circuit Protection
0.5
0
0
2
6
8
4
INDUCTOR VALUE (µH)
10
12
3448 F03
Figure 3. ILDO(ON) vs LOUT
When the output is shorted to ground, the main switch
cycle will be skipped, and the synchronous switch will
remain on for a longer duration. This allows the inductor
current more time to decay, thereby preventing runaway.
3448f
8
LTC3448
U
OPERATIO (Refer to Functional Diagram)
(see Typical Performance Characteristics). Therefore, the
user should calculate the power dissipation when the
LTC3448 is used at 100% duty cycle with low input voltage
(See Thermal Considerations in the Applications Information section).
MAXIMUM OUTPUT CURRENT (mA)
1200
1000
800
600
VOUT = 1.8V
VOUT = 2.5V
VOUT = 1.5V
Low Supply Operation
400
The LTC3448 will operate with input supply voltages as
low as 2.5V, but the maximum allowable output current is
reduced at this low voltage. Figure 4 shows the reduction
in the maximum output current as a function of input
voltage for various output voltages.
200
0
2.5
3.0
3.5
4.0
4.5
SUPPLY VOLTAGE (V)
5.0
5.5
3448 F04
Figure 4. Maximum Output Current vs Input Voltage
Dropout Operation
As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the
maximum on-time. Further reduction of the supply voltage
forces the main switch to remain on for more than one cycle
until it reaches 100% duty cycle. The output voltage will then
be determined by the input voltage minus the voltage drop
across the P-channel MOSFET and the inductor.
An important detail to remember is that at low input supply
voltages, the RDS(ON) of the P-channel switch increases
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constant frequency architectures by preventing sub-harmonic oscillations at high duty cycles. It is accomplished internally by
adding a compensating ramp to the inductor current
signal at duty cycles in excess of 40%. This normally
results in a reduction of maximum inductor peak current
for duty cycles >40%. However, the LTC3448 uses a
patent-pending scheme that counteracts this compensating ramp, which allows the maximum inductor peak
current to remain unaffected throughout all duty cycles.
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The basic LTC3448 application circuit is shown on the first
page of this data sheet. External component selection is
driven by the load requirement and begins with the selection of L followed by CIN and COUT.
Inductor Selection
For most applications, the value of the inductor will fall in
the range of 1µH to 4.7µH. Its value is chosen based on the
desired ripple current. Large value inductors lower ripple
current and small value inductors result in higher ripple
currents. Higher VIN or VOUT also increases the ripple
current as shown in equation 1. A reasonable starting point
for setting ripple current is ∆IL = 240mA (40% of 600mA).
∆IL =
⎛ V ⎞
VOUT ⎜ 1 − OUT ⎟
VIN ⎠
⎝
f L
1
( )( )
(1)
The DC current rating of the inductor should be at least
equal to the maximum load current plus half the ripple
current to prevent core saturation. Thus, a 720mA rated
inductor should be enough for most applications (600mA
+ 120mA). For better efficiency, choose a low DC-resistance inductor.
If the LTC3448 is to be used in auto LDO mode, inductor
values less than 1µH should not be used.
3448f
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Inductor Core Selection
Different core materials and shapes will change the size/
current and price/current relationship of an inductor.
Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with
similar electrical characteristics. The choice of which style
inductor to use often depends more on the price vs size
requirements and any radiated field/EMI requirements
than on what the LTC3448 requires to operate. Table 1
shows some typical surface mount inductors that work
well in LTC3448 applications.
Table 1. Representative Surface Mount Inductors
PART
NUMBER
VALUE
(µH)
DCR
(Ω MAX)
1.5
2.2
3.3
4.7
0.043
0.075
0.110
0.162
1.55
1.20
1.10
0.90
3.8 × 3.8 × 1.8
Sumida
CMD4D06
2.2
3.3
4.7
0.116
0.174
0.216
0.950
0.770
0.750
3.5 × 4.3 × 0.8
Coilcraft
ME3220
2.2
3.3
4.7
0.104
0.138
0.190
1.8
1.3
1.2
2.5 × 3.2 × 2.0
Murata
LQH3C
1.0
2.2
4.7
0.060
0.097
0.150
1.00
0.79
0.65
2.5 × 3.2 × 2.0
Sumida
CDRH3D16
MAX DC
SIZE
CURRENT (A) W × L × H (mm3)
CIN and COUT Selection
In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent
large voltage transients, a low ESR input capacitor sized
for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
CIN required IRMS ≅ IOMAX
[
(
VOUT VIN − VOUT
)]
1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design. Note that the capacitor
manufacturer’s ripple current ratings are often based on
2000 hours of life. This makes it advisable to further derate
the capacitor, or choose a capacitor rated at a higher
temperature than required. Always consult the manufacturer if there is any question.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating
generally far exceeds the IRIPPLE(P-P) requirement. In any
case, if LDO mode is enabled, the value of COUT must have
a minimum value of 2µF to ensure loop stability. The
output ripple ∆VOUT is determined by:
⎛
1 ⎞
∆VOUT ≅ ∆IL ⎜ ESR +
⎟
8fC OUT ⎠
⎝
where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since ∆IL increases with input voltage.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount configurations. In the case
of tantalum, it is critical that the capacitors are surge tested
for use in switching power supplies. An excellent choice is
the AVX TPS series of surface mount tantalum. These are
specially constructed and tested for low ESR so they give
the lowest ESR for a given volume. Other capacitor types
include Sanyo POSCAP, Kemet T510 and T495 series, and
Sprague 593D and 595D series. Consult the manufacturer
for other specific recommendations.
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. Because the
LTC3448’s control loop does not depend on the output
capacitor’s ESR for stable operation, ceramic capacitors
can be used freely to achieve very low output ripple and
small circuit size.
However, care must be taken when ceramic capacitors are
used at the input and the output. When a ceramic capacitor
is used at the input and the power is supplied by a wall
adapter through long wires, a load step at the output can
induce ringing at the input, VIN. At best, this ringing can
couple to the output and be mistaken as loop instability. At
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worst, a sudden inrush of current through the long wires
can potentially cause a voltage spike at VIN, large enough
to damage the part.
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size.
Output Voltage Programming
loss dominates the efficiency loss at low load currents,
whereas the I2R loss dominates the efficiency loss at
medium to high load currents. At very low load currents
with the part operating in LDO mode, efficiency can be
dominated by I2R losses in the pass transistor and is a
strong function of (VIN – VOUT). In a typical efficiency plot,
the efficiency curve at very low load currents can be
misleading since the actual power lost is of little consequence as illustrated in Figure 6.
The output voltage is set by tying VFB to a resistive divider
according to the following formula:
(2)
The external resistive divider is connected to the output,
allowing remote voltage sensing as shown in Figure 5.
0.6V ≤ VOUT ≤ 5.5V
R2
VFB
LTC3448
VIN = 3.6V
FREQ = 0V
LDOCNTRL = VOUT(AUTO)
0.1
⎛ R2⎞
= 0.6V ⎜ 1 + ⎟
⎝ R1⎠
POWER LOSS (W)
VOUT
1
0.01
0.001
0.0001
0.0001
1.2V
1.5V
1.8V
0.001
0.01
0.1
LOAD CURRENT (A)
1
3448 F06
R1
Figure 6. Power Loss vs Load Current
GND
3448 F05
Figure 5. Setting the LTC3448 Output Voltage
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of the
losses in LTC3448 circuits: VIN quiescent current and I2R
losses. When in switching mode, VIN quiescent current
1. The VIN quiescent current is due to two components:
the DC bias current as given in the Electrical Characteristics and the internal main switch and synchronous
switch gate charge currents. The gate charge current
results from switching the gate capacitance of the
internal power MOSFET switches. Each time the gate is
switched from high to low to high again, a packet of
charge, dQ, moves from VIN to ground. The resulting
dQ/dt is the current out of VIN that is typically larger than
the DC bias current and proportional to frequency. Both
the DC bias and gate charge losses are proportional to
VIN and thus their effects will be more pronounced at
higher supply voltages.
2. I2R losses are calculated from the resistances of the
internal switches, RSW, and external inductor RL. In
continuous mode, the average output current flowing
through inductor L is “chopped” between the main
switch and the synchronous switch. Thus, the series
resistance looking into the SW pin is a function of both
3448f
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top and bottom MOSFET RDS(ON) and the duty cycle
(DC) as follows:
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses, simply add RSW
to RL and multiply the result by the square of the
average output current.
3. At load currents below the selected threshold the
LTC3448 will switch into low ripple LDO mode if enabled. In this case the losses are due to the DC bias
currents as given in the electrical characteristics and
I2R losses due to the (VIN – VOUT) voltage drop across
the internal pass transistor.
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
The junction temperature, TJ, is given by:
TJ = TA + TR
where TA is the ambient temperature.
As an example, consider the LTC3448 in dropout at an
input voltage of 2.7V, a load current of 600mA and an
ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the
P-channel switch at 70°C is approximately 0.52Ω. Therefore, power dissipated by the part is:
PD = ILOAD2 • RDS(ON) = 187.2mW
Other losses when in switching operation, including CIN
and COUT ESR dissipative losses and inductor core losses,
generally account for less than 2% total additional loss.
For the 3mm × 3mm DFN package, the θJA is 43°C/W.
Thus, the junction temperature of the regulator is:
Thermal Considerations
which is well below the maximum junction temperature of
125°C.
The LTC3448 requires the package backplane metal (GND
pin) to be well soldered to the PC board. This gives the DFN
and MSOP packages exceptional thermal properties, making it difficult in normal operation to exceed the maximum
junction temperature of the part. In most applications the
LTC3448 does not dissipate much heat due to its high
efficiency. In applications where the LTC3448 is running at
high ambient temperature with low supply voltage and high
duty cycles, such as in dropout, the heat dissipated may
exceed the maximum junction temperature of the part if it
is not well thermally grounded. If the junction temperature
reaches approximately 150°C, both power switches will be
turned off and the SW node will become high impedance.
To avoid the LTC3448 from exceeding the maximum
junction temperature, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated exceeds the
maximum junction temperature of the part. The temperature rise is given by:
TR = PDθJA
TJ = 85°C + (0.1872)(43) = 93°C
Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance RDS(ON).
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to (∆ILOAD • ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The
regulator loop then acts to return VOUT to its steady-state
value. During this recovery time VOUT can be monitored for
overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop
theory, see Application Note 76.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
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with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25 • CLOAD).
Thus, a 10µF capacitor charging to 3.3V would require a
250µs rise time, limiting the charging current to about
130mA.
2. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground.
3. Does the (+) plate of CIN connect to VIN as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
4. Keep the switching node, SW, away from the sensitive
VFB node.
PC Board Layout Checklist
5. Keep the (–) plates of CIN and COUT as close as possible.
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3448. These items are also illustrated graphically in
Figures 7 and 8. Check the following in your layout:
Design Example
As a design example, assume the LTC3448 is used in a
single lithium-ion battery-powered cellular phone
application. The VIN will be operating from a maximum of
4.2V down to about 2.7V. The load current requirement
is a maximum of 0.6A but most of the time it will be in
standby mode, requiring only 2mA. Efficiency at both low
1. The power traces, consisting of the GND trace, the SW
trace and the VIN trace should be kept short, direct and
wide.
4
VIN
8
CIN
VIN
SW
RUN
VOUT
MODE
5
L
VOUT
2
3
COUT
RFB2
CFF
LTC3448
6
7
VFB
FREQ
SYNC
GND
9
1
RFB1
3448 F07
Figure 7. LTC3448 Layout Design
3448 F08
Figure 8. LTC3448 Layout
3448f
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LTC3448
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and high load currents is important. Output voltage is
1.8V. With this information we can calculate L using
Equation (1),
L=
⎛ V ⎞
VOUT ⎜ 1 − OUT ⎟
VIN ⎠
⎝
f ∆IL
1
( )( )
(3)
Substituting VOUT = 1.8V, VIN = 4.2V, ∆IL = 240mA and
f = 1.5MHz in Equation (3) gives:
L=
CIN will require an RMS current rating of at least 0.3A ≅
ILOAD(MAX)/2 at temperature and COUT will require an ESR
of less than 0.25Ω. In most cases, a ceramic capacitor will
satisfy this requirement.
For the feedback resistors, choose R1 = 316k. R2 can
then be calculated from Equation (2) to be:
⎛V
⎞
R2 = ⎜ OUT − 1⎟ R1 = 632k
⎝ 0.6
⎠
Figure 9 shows the complete circuit along with its efficiency curve.
1.8 V
⎛ 1.8 V ⎞
⎜1 −
⎟ = 2.86 µH
1.5MHz(240mA) ⎝ 4.2V ⎠
A 2.2µH inductor works well for this application. For best
efficiency choose a 720mA or greater inductor with less
than 0.2Ω series resistance.
4
CIN
4.7µF
CER
8
VIN
SW
5
2.2µH*
2
VOUT
RUN
LTC3448
3
MODE
6
1
VFB
FREQ
7
SYNC
GND
22pF
632k
COUT
15µF
CER
VOUT
1.8V
VIN = 3.6V
90 VOUT = 1.8V
T = 25°C
80 A
70
EFFICIENCY (%)
VIN
2.7V
TO 5.5V
100
60
50
40
30
316k
20
9
10
3448 F09a
0
0.0001
CIN: TAIYO YUDEN JMK212BJ475MG
COUT: TAIYO YUDEN JMK212BJ475MG
*MURATA LQH32CN2R2M11
0.001
0.01
0.1
LOAD CURRENT (A)
1
3448 F09b
Figure 9b
Figure 9a
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
ILOAD
250mA/DIV
ILOAD
100mA/DIV
IL
500mA/DIV
IL
500mA/DIV
VIN = 3.6V
20µs/DIV
VOUT = 1.8V
ILOAD = 100µA TO 200mA
Figure 9c
3448 F09c
VIN = 3.6V
20µs/DIV
VOUT = 1.8V
ILOAD = 50mA TO 600mA
3448 F09d
Figure 9d
3448f
14
LTC3448
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TYPICAL APPLICATIO S
Single Li-Ion 1.5V/600mA Regulator for
High Efficiency and Small Footprint
4
CIN
4.7µF
CER
8
VIN
SW
5
90
2.2µH*
VOUT
1.5V
2
VOUT
RUN
LTC3448
3
MODE
6
1
VFB
FREQ
7
SYNC
GND
22pF
474k
COUT
15µF
216k
VOUT = 1.5V
TA = 25°C
80
70
EFFICIENCY (%)
VIN
2.7V
TO 5.5V
Efficiency vs Output Current
100
60
50
40
30
9
20
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
3448 TA03
10
CIN: TAIYO YUDEN CERAMIC JMK212BJ475MG
COUT: TAIYO YUDEN CERAMIC JMK212BJ475MG
*MURATA LQH32CN2R2M33
0
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
1
23448 G03
Load Step
Load Step
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
ILOAD
250mA/DIV
ILOAD
100mA/DIV
IL
500mA/DIV
IL
500mA/DIV
VIN = 3.6V
20µs/DIV
VOUT = 1.5V
ILOAD = 100µA TO 200mA
3448 TA05
VIN = 3.6V
20µs/DIV
VOUT = 1.5V
ILOAD = 50mA TO 600mA
3448 TA06
Note: Performance data measured on the LTC3448 with external resistors
3448f
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LTC3448
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TYPICAL APPLICATIO S
Single Li-Ion 1.2V/600mA Regulator for
High Efficiency and Small Footprint
4
CIN
4.7µF
CER
8
VIN
SW
5
100
2.2µH*
2
VOUT
RUN
LTC3448
3
MODE
6
1
VFB
FREQ
7
SYNC
GND
22pF
316k
316k
9
COUT
10µF
CER
VOUT
1.2V
90
VOUT = 1.2V
TA = 25°C
80
70
EFFICIENCY (%)
VIN
2.7V
TO 5.5V
Efficiency vs Output Current
60
50
40
30
3448 TA07
20
CIN: TAIYO YUDEN JMK212BJ475MG
COUT: TAIYO YUDEN JMK212BJ475MG
*MURATA LQH32CN2R2M33
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
0
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
1
23448 G02
Load Step
Load Step
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
ILOAD
100mA/DIV
ILOAD
250mA/DIV
IL
500mA/DIV
IL
500mA/DIV
VIN = 3.6V
20µs/DIV
VOUT = 1.2V
ILOAD = 100µA TO 200mA
3448 TA09
20µs/DIV
VIN = 3.6V
VOUT = 1.2V
ILOAD = 50mA TO 600mA
3448 TA10
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LTC3448
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TYPICAL APPLICATIO S
Single Li-Ion 2.5V/600mA Regulator with 1.8MHz External
Synchronization and External MODE
VIN
2.5V TO 5.5V
4
CIN
4.7µF
CER
TO
µPROCESSOR
CONTROL
TO 0V TO 1.3V
OR GREATER 1.8MHz
EXTERNAL CLOCK
8
VIN
5
SW
2.2µH
2
VOUT
RUN
LTC3448
1.58M
3
MODE
6
1
VFB
FREQ
7
SYNC
GND
CFF
22pF
COUT
10µF
CER
VOUT
2.5V
600mA
500k
9
3448 TA12
Load Step
Load Step
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
LDOCNTRL
2V/DIV
LDOCNTRL
2V/DIV
ILOAD
250mA/DIV
ILOAD
250mA/DIV
3448 TA12b
VIN = 3.6V
40µs/DIV
VOUT = 2.5V
ILOAD = 100µA TO 300mA
40µs/DIV
VIN = 3.6V
VOUT = 2.5V
ILOAD = 100µA TO 600mA
3448 TA12c
Single Li-Ion 1.2V/600mA Regulator with 2.5MHz External Synchronization
VIN
2.5V TO 5.5V
4
CIN
4.7µF
CER
8
VIN
SW
VOUT
RUN
LTC3448
5
2.2µH
2
316k
3
TO 0V TO 1.3V OR
GREATER 2.5MHz
EXTERNAL CLOCK
MODE
6
1
VFB
FREQ
7
SYNC
GND
CFF
22pF
COUT
10µF
CER
VOUT
1.2V
600mA
316k
9
3448 TA13
3448f
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LTC3448
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PACKAGE DESCRIPTIO
DD Package
8-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698)
0.675 ±0.05
3.5 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50
BSC
2.38 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
R = 0.115
TYP
5
3.00 ±0.10
(4 SIDES)
0.38 ± 0.10
8
1.65 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(NOTE 6)
(DD8) DFN 1203
0.200 REF
0.75 ±0.05
0.00 – 0.05
4
0.25 ± 0.05
1
0.50 BSC
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
3448f
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PACKAGE DESCRIPTIO
MS8E Package
8-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1662)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.06 ± 0.102
(.081 ± .004)
1
5.23
(.206)
MIN
1.83 ± 0.102
(.072 ± .004)
0.889 ± 0.127
(.035 ± .005)
2.794 ± 0.102
(.110 ± .004)
2.083 ± 0.102 3.20 – 3.45
(.082 ± .004) (.126 – .136)
8
0.42 ± 0.038
(.0165 ± .0015)
TYP
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
0.65
(.0256)
BSC
8
7 6 5
0.52
(.0205)
REF
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
DETAIL “A”
0° – 6° TYP
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
1
2 3
4
1.10
(.043)
MAX
0.86
(.034)
REF
0.18
(.007)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
0.65
(.0256)
NOTE:
BSC
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.127 ± 0.076
(.005 ± .003)
MSOP (MS8E) 0603
3448f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC3448
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1616
500mA (IOUT), 1.4MHz, High Efficiency Step-Down
DC/DC Converter
90% Efficiency, VIN = 3.6V to 25V, VOUT ≥ 1.25V, IQ = 1.9mA,
ISD = <1µA, ThinSOT Package
LT1776
500mA (IOUT), 200kHz, High Efficiency Step-Down
DC/DC Converter
90% Efficiency, VIN = 7.4V to 40V, VOUT ≥ 1.24V, IQ = 3.2mA,
ISD = 30µA, N8, S8 Packages
LTC1877
600mA (IOUT), 550kHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, VIN = 2.7V to 10V, VOUT ≥ 0.8V, IQ = 10µA,
ISD = <1µA, MS8 Package
LTC1879
1.2A (IOUT), 550kHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, VIN = 2.7V to 10V, VOUT ≥ 0.8V, IQ = 15µA,
ISD = <1µA, TSSOP-16 Package
LTC3403
600mA (IOUT), 1.5MHz, Synchronous Step-Down
DC/DC Converter with Bypass Transistor
96% Efficiency, VIN = 2.5V to 5.5V, VOUT = Dynamically Adjustable,
IQ = 20µA, ISD = <1µA, DFN Package
LTC3405/LTC3405A
300mA (IOUT), 1.5MHz, Synchronous Step-Down
DC/DC Converter
96% Efficiency, VIN = 2.5V to 5.5V, VOUT ≥ 0.8V, IQ = 20µA,
ISD = <1µA, ThinSOT Package
LTC3406
600mA (IOUT), 1.5MHz, Synchronous Step-Down
DC/DC Converter
96% Efficiency, VIN = 2.5V to 5.5V, VOUT ≥ 0.6V, IQ = 20µA,
ISD = <1µA, ThinSOT Package
LTC3406B-2
600mA (IOUT), 2.25MHz, Synchronous Step-Down
DC/DC Converter
96% Efficiency, VIN = 2.5V to 5.5V, VOUT ≥ 0.6V, IQ = 300µA,
ISD = <1µA, ThinSOT Package
LTC3407/LTC3407-2
Dual 600mA/800mA (IOUT), 1.5MHz/2.25MHz,
Synchronous Step-Down DC/DC Converter
96% Efficiency, VIN = 2.5V to 5.5V, VOUT ≥ 0.6V, IQ = 40µA,
ISD = <1µA, MS10, DFN Packages
LTC3409
600mA Low VIN Buck Regulator
95% Efficiency, VIN = 1.6V to 5.5V, IQ = 65µA
ISD = <1µA, DFN Package
LTC3411
1.25A (IOUT), 4MHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, VIN = 2.5V to 5.5V, VOUT ≥ 0.8V, IQ = 60µA,
ISD = <1µA, MS Package
LTC3412
2.5A (IOUT), 4MHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, VIN = 2.5V to 5.5V, VOUT ≥ 0.8V, IQ = 60µA,
ISD = <1µA, TSSOP-16E Package
LTC3440
600mA (IOUT), 2MHz, Synchronous Buck-Boost
DC/DC Converter
95% Efficiency, VIN = 2.5V to 5.5V, VOUT ≥ 2.5V, IQ = 25µA,
ISD = <1µA, MS Package
LTC3441
1.2A (IOUT), 1MHz, Synchronous Buck-Boost
DC/DC Converter
95% Efficiency, VIN = 2.4V to 5.5V, VOUT ≥ 2.4V to 5.25V, IQ = 25µA,
ISD = <1µA, DFN Package
LTC3442
1.2A (IOUT), 2MHz, Synchronous Buck-Boost
DC/DC Converter
95% Efficiency, VIN = 2.4V to 5.5V, VOUT ≥ 2.4V to 5.25V, IQ = 35µA,
ISD = <1µA, DFN Package
LTC3443
1.2A (IOUT), 600kHz, Synchronous Buck-Boost
DC/DC Converter
95% Efficiency, VIN = 2.4V to 5.5V, VOUT ≥ 2.4V to 5.25V, IQ = 28µA,
ISD = <1µA, DFN Package
3448f
20
Linear Technology Corporation
LT/TP 0505 500 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2005