NSC LM2694 30v, 600 ma step down switching regulator Datasheet

LM2694
30V, 600 mA Step Down Switching Regulator
General Description
The LM2694 Step Down Switching Regulator features all of
the functions needed to implement a low cost, efficient, buck
bias regulator capable of supplying 0.6A to the load. This
buck regulator contains an N-Channel Buck Switch, and is
available in the 3 x 3 thermally enhanced LLP-10 package
and a TSSOP-14 package. The feedback regulation scheme
requires no loop compensation, results in fast load transient
response, and simplifies circuit implementation. The operating frequency remains constant with line and load variations
due to the inverse relationship between the input voltage and
the on-time. The valley current limit results in a smooth
transition from constant voltage to constant current mode
when current limit is detected, reducing the frequency and
output voltage, without the use of foldback. Additional features include: VCC under-voltage lockout, thermal shutdown, gate drive under-voltage lockout, and maximum duty
cycle limiter.
n No loop compensation required
n Ultra-Fast transient response
n Operating frequency remains constant with load current
and input voltage variations
n Maximum Duty Cycle Limited During Start-Up
n Adjustable output voltage
n Valley Current Limit At 0.6A
n Maximum switching frequency: 1 MHz
n Precision internal reference
n Low bias current
n Highly efficient operation
n Thermal shutdown
Typical Applications
n High Efficiency Point-Of-Load (POL) Regulator
n Non-Isolated Telecommunication Buck Regulator
n Secondary High Voltage Post Regulator
Features
Package
n Integrated N-Channel buck switch
n Integrated start-up regulator
n Input Voltage Range: 8V to 30V
n LLP-10 (3 mm x 3 mm) w/Exposed Pad
n TSSOP-14
Basic Step Down Regulator
20187001
© 2006 National Semiconductor Corporation
DS201870
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LM2694 30V, 600 mA Step Down Switching Regulator
May 2006
LM2694
Connection Diagrams
20187002
10-Lead LLP
20187003
14-Lead TSSOP
Ordering Information
Order Number
Package Type
NSC Package Drawing
Supplied As
LM2694SD
LLP-10 (3x3)
SDA10A
1000 Units on Tape and Reel
LM2694SDX
LLP-10 (3x3)
SDA10A
4500 Units on Tape and Reel
LM2694MT
TSSOP-14
MTC14
94 Units in Rail
LM2694MTX
TSSOP-14
MTC14
2500 Units on Tape and Reel
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2
LM2694
Pin Descriptions
PIN NUMBER
LLP-10
TSSOP-14
NAME
1
2
SW
Switching Node
Internally connected to the buck switch source.
Connect to the inductor, free-wheeling diode, and
bootstrap capacitor.
2
3
BST
Boost pin for bootstrap capacitor
Connect a 0.022 µF capacitor from SW to the BST
pin. The capacitor is charged from VCC via an
internal diode during the buck switch off-time.
3
4
ISEN
Current sense
During the buck switch off-time, the inductor current
flows through the internal sense resistor, and out of
the ISEN pin to the free-wheeling diode. The current
limit is nominally set at 0.62A.
4
5
SGND
Current Sense Ground
Re-circulating current flows into this pin to the
current sense resistor.
5
6
RTN
Circuit Ground
Ground return for all internal circuitry other than the
current sense resistor.
6
9
FB
Voltage feedback input from the
regulated output
Input to both the regulation and over-voltage
comparators. The FB pin regulation level is 2.5V.
7
10
SS
Softstart
An internal current source charges the SS pin
capacitor to 2.5V to soft-start the reference input of
the regulation comparator.
8
11
RON/SD
On-time control and shutdown
An external resistor from VIN to the RON/SD pin
sets the buck switch on-time. Grounding this pin
shuts down the regulator.
9
12
VCC
Output of the startup regulator
The voltage at VCC is nominally regulated at 7V.
Connect a 0.1 µF, or larger capacitor from VCC to
ground, as close as possible to the pins. An external
voltage can be applied to this pin to reduce internal
dissipation. MOSFET body diodes clamp VCC to
VIN if VCC > VIN.
10
13
VIN
Input supply voltage
Nominal input range is 8V to 30V. Input bypass
capacitors should be located as close as possible to
the VIN pin and RTN pins.
1,7,8,14
NC
No connection.
No internal connection. Can be connected to ground
plane to improve heat dissipation.
EP
Exposed Pad
Exposed metal pad on the underside of the LLP
package. It is recommended to connect this pad to
the PC board ground plane to aid in heat dissipation.
EP
DESCRIPTION
3
APPLICATION INFORMATION
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LM2694
Absolute Maximum Ratings (Note 1)
VCC to RTN
14V
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
SGND to RTN
-0.3V to +0.3V
VIN to RTN
BST to RTN
SW to RTN (Steady State)
-1.5V
SS to RTN
-0.3V to 4V
All Other Inputs to RTN
-0.3 to 7V
33V
Storage Temperature Range
-65˚C to +150˚C
47V
JunctionTemperature
150˚C
ESD Rating (Note 2)
Human Body Model
Operating Ratings (Note 1)
2kV
BST to VCC
33V
VIN
8.0V to 30V
VIN to SW
33V
Junction Temperature
BST to SW
14V
−40˚C to + 125˚C
Electrical Characteristics Specifications with standard type are for TJ = 25˚C only; limits in boldface type
apply over the full Operating Junction Temperature (TJ) range. Minimum and Maximum limits are guaranteed through test,
design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25˚C, and are provided for
reference purposes only. Unless otherwise stated the following conditions apply: VIN = 24V, RON = 200kΩ. See (Note 5).
Symbol
Parameter
Conditions
Min
Typ
Max
7
7.4
Units
Start-Up Regulator, VCC
VCCReg
UVLOVCC
VCC regulated output
6.6
V
VIN-VCC dropout voltage
ICC = 0 mA,
VCC = UVLOVCC + 250 mV
1.3
V
175
Ω
9
mA
VCC output impedance
0 mA ≤ ICC ≤ 5 mA, VIN = 8V
VCC current limit (Note 3)
VCC = 0V
VCC under-voltage lockout
threshold
VCC increasing
5.7
V
UVLOVCC hysteresis
VCC decreasing
150
mV
UVLOVCC filter delay
100 mV overdrive
IIN operating current
Non-switching, FB = 3V
0.5
0.8
mA
IIN shutdown current
RON/SD = 0V
90
180
µA
0.5
1.0
Ω
4.4
5.5
3
µs
Switch Characteristics
Rds(on)
Buck Switch Rds(on)
ITEST = 200 mA
UVLOGD
Gate Drive UVLO
VBST - VSW Increasing
3.0
V
UVLOGD hysteresis
490
Pull-up voltage
2.5
V
Internal current source
12
µA
mV
Softstart Pin
Current Limit
ILIM
Threshold
Current out of ISEN
0.5
0.62
0.74
A
Resistance from ISEN to
SGND
180
mΩ
Response time
150
ns
On Timer
tON - 1
On-time
VIN = 10V, RON = 200 kΩ
tON - 2
On-time
VIN = 30V, RON = 200 kΩ
Shutdown threshold
Voltage at RON/SD rising
Threshold hysteresis
Voltage at RON/SD falling
2.1
2.8
3.6
900
0.45
0.8
µs
ns
1.2
V
35
mV
265
ns
Off Timer
tOFF
Minimum Off-time
Regulation and Over-Voltage Comparators (FB Pin)
VREF
FB regulation threshold
SS pin = steady state
FB over-voltage threshold
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2.440
2.5
2.9
4
2.550
V
V
Symbol
Parameter
Conditions
Min
FB bias current
Typ
Max
Units
1
nA
Thermal shutdown
temperature
175
˚C
Thermal shutdown hysteresis
20
˚C
˚C/W
Thermal Shutdown
TSD
Thermal Resistance
θJA
θJC
Junction to Ambient
0 LFPM Air Flow
LLP Package
33
TSSOP Package
40
Junction to Case
LLP Package
8.8
TSSOP Package
5.2
˚C/W
Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device
is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.
Note 2: The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin.
Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading
Note 4: For detailed information on soldering plastic TSSOP and LLP packages, refer to the Packaging Data Book available from National Semiconductor
Corporation.
Note 5: Typical specifications represent the most likely parametric norm at 25˚C operation.
5
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LM2694
Electrical Characteristics Specifications with standard type are for TJ = 25˚C only; limits in boldface type
apply over the full Operating Junction Temperature (TJ) range. Minimum and Maximum limits are guaranteed through test,
design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25˚C, and are provided for
reference purposes only. Unless otherwise stated the following conditions apply: VIN = 24V, RON = 200kΩ. See (Note
5). (Continued)
LM2694
Typical Performance Characteristics
20187004
FIGURE 1. VCC vs VIN
20187005
FIGURE 2. VCC vs ICC
20187006
FIGURE 3. ICC vs Externally Applied VCC
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LM2694
Typical Performance Characteristics
(Continued)
20187007
FIGURE 4. ON-Time vs VIN and RON
20187008
FIGURE 5. Voltage at RON/SD Pin
20187009
FIGURE 6. IIN vs VIN
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LM2694
Typical Application Circuit and Block Diagram
20187010
FIGURE 7.
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8
LM2694
Typical Application Circuit and Block Diagram
(Continued)
20187011
FIGURE 8. Startup Sequence
9
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LM2694
reference - until then the inductor current remains zero, and
the load current is supplied by the output capacitor (C2). In
this mode the operating frequency is lower than in continuous conduction mode, and varies with load current. Conversion efficiency is maintained at light loads since the switching losses reduce with the reduction in load and frequency.
The approximate discontinuous operating frequency can be
calculated as follows:
Functional Description
The LM2694 Step Down Switching Regulator features all the
functions needed to implement a low cost, efficient buck bias
power converter capable of supplying at least 0.6A to the
load. This high voltage regulator contains a 30V N-Channel
buck switch, is easy to implement, and is available in the
TSSOP-14 and the thermally enhanced LLP-10 packages.
The regulator’s operation is based on a constant on-time
control scheme, where the on-time is determined by VIN.
This feature allows the operating frequency to remain relatively constant with load and input voltage variations. The
feedback control requires no loop compensation resulting in
very fast load transient response. The valley current limit
detection circuit, internally set at 0.62A, holds the buck
switch off until the high current level subsides. This scheme
protects against excessively high currents if the output is
short-circuited when VIN is high. The functional block diagram is shown in Figure 7.
(3)
where RL = the load resistance.
The output voltage is set by two external resistors (R1, R2).
The regulated output voltage is calculated as follows:
VOUT = 2.5 x (R1 + R2) / R2
Output voltage regulation is based on ripple voltage at the
feedback input, requiring a minimum amount of ESR for the
output capacitor C2. The LM2694 requires a minimum of 25
mV of ripple voltage at the FB pin. In cases where the
capacitor’s ESR is insufficient additional series resistance
may be required (R3 in Figure 7).
The LM2694 can be applied in numerous applications to
efficiently regulate down higher voltages. Additional features
include: Thermal shutdown, VCC under-voltage lockout, gate
drive under-voltage lockout, and maximum duty cycle limiter.
Control Circuit Overview
Start-Up Regulator, VCC
The LM2694 buck DC-DC regulator employs a control
scheme based on a comparator and a one-shot on-timer,
with the output voltage feedback (FB) compared to an internal reference (2.5V). If the FB voltage is below the reference
the buck switch is turned on for a time period determined by
the input voltage and a programming resistor (RON). Following the on-time the switch remains off for a minimum of 265
ns, and until the FB voltage falls below the reference. The
buck switch then turns on for another on-time period. Typically, during start-up, or when the load current increases
suddenly, the off-times are at the minimum of 265 ns. Once
regulation is established, the off-times are longer.
When in regulation, the LM2694 operates in continuous
conduction mode at heavy load currents and discontinuous
conduction mode at light load currents. In continuous conduction mode current always flows through the inductor,
never reaching zero during the off-time. In this mode the
operating frequency remains relatively constant with load
and line variations. The minimum load current for continuous
conduction mode is one-half the inductor’s ripple current
amplitude. The operating frequency is approximately:
The start-up regulator is integral to the LM2694. The input
pin (VIN) can be connected directly to line voltage up to 30V,
with transient capability to 33V. The VCC output regulates at
7.0V, and is current limited at 9 mA. Upon power up, the
regulator sources current into the external capacitor at VCC
(C3). When the voltage on the VCC pin reaches the undervoltage lockout threshold of 5.7V, the buck switch is enabled
and the Softstart pin is released to allow the Softstart capacitor (C6) to charge up.
The minimum input voltage is determined by the regulator’s
dropout voltage, the VCC UVLO falling threshold ()5.5V),
and the frequency. When VCC falls below the falling threshold the VCC UVLO activates to shut off the output. If VCC is
externally loaded, the minimum input voltage increases.
To reduce power dissipation in the start-up regulator, an
auxiliary voltage can be diode connected to the VCC pin.
Setting the auxiliary voltage to between 8V and 14V shuts off
the internal regulator, reducing internal power dissipation.
The sum of the auxiliary voltage and the input voltage (VCC
+ VIN) cannot exceed 47V. Internally, a diode connects VCC
to VIN. See Figure 9.
(1)
The buck switch duty cycle is equal to:
(2)
In discontinuous conduction mode current through the inductor ramps up from zero to a peak during the on-time, then
ramps back to zero before the end of the off-time. The next
on-time period starts when the voltage at FB falls below the
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LM2694
Start-Up Regulator, VCC
(Continued)
20187014
FIGURE 9. Self Biased Configuration
In high frequency applications the minimum value for tON is
limited by the maximum duty cycle required for regulation
and the minimum off-time of 265 ns, ± 15%. The minimum
off-time limits the maximum duty cycle achievable with a low
voltage at VIN. The minimum allowed on-time to regulate the
desired VOUT at the minimum VIN is determined from the
following:
Regulation Comparator
The feedback voltage at FB is compared to the voltage at the
Softstart pin (2.5V). In normal operation (the output voltage
is regulated), an on-time period is initiated when the voltage
at FB falls below 2.5V. The buck switch stays on for the
programmed on-time, causing the FB voltage to rise above
2.5V. After the on-time period, the buck switch stays off until
the FB voltage falls below 2.5V. Input bias current at the FB
pin is less than 100 nA over temperature.
(6)
The LM2694 can be remotely shut down by taking the
RON/SD pin below 0.8V. See Figure 10. In this mode the SS
pin is internally grounded, the on-timer is disabled, and bias
currents are reduced. Releasing the RON/SD pin allows
normal operation to resume. The voltage at the RON/SD pin
is normally between 1.5V and 3.0V, depending on VIN and
the RON resistor.
Over-Voltage Comparator
The voltage at FB is compared to an internal 2.9V reference.
If the voltage at FB rises above 2.9V the on-time pulse is
immediately terminated. This condition can occur if the input
voltage or the output load changes suddenly, or if the inductor (L1) saturates. The buck switch remains off until the
voltage at FB falls below 2.5V.
ON-Time Timer, and Shutdown
The on-time for the LM2694 is determined by the RON resistor and the input voltage (VIN), and is calculated from:
(4)
See Figure 4. The inverse relationship with VIN results in a
nearly constant frequency as VIN is varied. To set a specific
continuous conduction mode switching frequency (FS), the
RON resistor is determined from the following:
20187019
FIGURE 10. Shutdown Implementation
(5)
11
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LM2694
Figure 11 illustrates the inductor current waveform. During
normal operation the load current is Io, the average of the
ripple waveform. When the load resistance decreases the
current ratchets up until the lower peak reaches 0.62A.
During the Current Limited portion of Figure 11, the current
ramps down to 0.62A during each off-time, initiating the next
on-time (assuming the voltage at FB is < 2.5V). During each
on-time the current ramps up an amount equal to:
∆I = (VIN - VOUT) x tON / L1
Current Limit
Current limit detection occurs during the off-time by monitoring the recirculating current through the free-wheeling diode
(D1). Referring to Figure 7, when the buck switch is turned
off the inductor current flows through the load, into SGND,
through the sense resistor, out of ISEN and through D1. If
that current exceeds 0.62A the current limit comparator output switches to delay the start of the next on-time period if
the voltage at FB is below 2.5V. The next on-time starts
when the current out of ISEN is below 0.62A and the voltage
at FB is below 2.5V. If the overload condition persists causing the inductor current to exceed 0.62A during each ontime, that is detected at the beginning of each off-time. The
operating frequency is lower due to longer-than-normal offtimes.
During this time the LM2694 is in a constant current mode,
with an average load current (IOCL) equal to 0.62A + ∆I/2.
20187020
FIGURE 11. Inductor Current - Current Limit Operation
An internal switch grounds the SS pin if VCC is below the
under-voltage lockout threshold, if a thermal shutdown occurs, or if the RON/SD pin is grounded.
N - Channel Buck Switch and
Driver
The LM2694 integrates an N-Channel buck switch and associated floating high voltage gate driver. The peak current
allowed through the buck switch is 1.5A, and the maximum
allowed average current is 1A. The gate driver circuit works
in conjunction with an external bootstrap capacitor and an
internal high voltage diode. A 0.022 µF capacitor (C4) connected between BST and SW provides the voltage to the
driver during the on-time. During each off-time, the SW pin is
at approximately -1V, and C4 charges from VCC through the
internal diode. The minimum off-time of 265 ns ensures a
minimum time each cycle to recharge the bootstrap capacitor.
Thermal Shutdown
The LM2694 should be operated so the junction temperature
does not exceed 125˚C. If the junction temperature increases, an internal Thermal Shutdown circuit, which activates (typically) at 175˚C, takes the controller to a low power
reset state by disabling the buck switch and the on-timer,
and grounding the Softstart pin. This feature helps prevent
catastrophic failures from accidental device overheating.
When the junction temperature reduces below 155˚C (typical
hysteresis = 20˚C), the Softstart pin is released and normal
operation resumes.
Softstart
Applications Information
The softstart feature allows the converter to gradually reach
a steady state operating point, thereby reducing start-up
stresses and current surges. Upon turn-on, after VCC
reaches the under-voltage threshold, an internal 12 µA current source charges up the external capacitor at the SS pin
to 2.5V. The ramping voltage at SS (and the non-inverting
input of the regulation comparator) ramps up the output
voltage in a controlled manner.
EXTERNAL COMPONENTS
The procedure for calculating the external components is
illustrated with a design example. Referring to the Block
Diagram, the circuit is to be configured for the following
specifications:
• VOUT = 5V
•
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12
VIN = 8V to 30V
a higher efficiency, but with larger components. Generally, if
PC board space is tight, a higher frequency is better. The
resulting on-time and frequency have a ± 25% tolerance.
Using equation 5 at a VIN of 8V,
(Continued)
• FS = 250 kHz
• Minimum load current = 100 mA
• Maximum load current = 600 mA
• Softstart time = 5 ms.
R1 and R2: These resistors set the output voltage, and their
ratio is calculated from:
(7)
R1/R2 = (VOUT/2.5V) - 1
R1/R2 calculates to 1.0. The resistors should be chosen
from standard value resistors in the range of 1.0 kΩ - 10 kΩ.
A value of 2.5 kΩ will be used for R1 and for R2.
RON, FS: RON can be chosen using Equation 5 to set the
nominal frequency, or from Equation 4 if the on-time at a
particular VIN is important. A higher frequency generally
means a smaller inductor and capacitors (value, size and
cost), but higher switching losses. A lower frequency means
A value of 140 kΩ will be used for RON, yielding a nominal
frequency of 252 kHz.
L1: The guideline for choosing the inductor value in this
example is that it must keep the circuit’s operation in continuous conduction mode at minimum load current. This is
not a strict requirement since the LM2694 regulates correctly
when in discontinuous conduction mode, although at a lower
frequency. However, to provide an initial value for L1 the
above guideline will be used.
20187037
FIGURE 12. Inductor Current
To keep the circuit in continuous conduction mode, the maximum allowed ripple current is twice the minimum load current, or 200 mAp-p. Using this value of ripple current, the
inductor (L1) is calculated using the following:
IPK = ILIM + IOR(max) = 0.74A + 0.18A = 0.92A
where ILIM is the maximum guaranteed current limit threshold. At the nominal maximum load current of 0.6A, the peak
inductor current is 692 mA.
C1: This capacitor limits the ripple voltage at VIN resulting
from the source impedance of the supply feeding this circuit,
and the on/off nature of the switch current into VIN. At
maximum load current, when the buck switch turns on, the
current into VIN steps up from zero to the lower peak of the
inductor current waveform (IPK- in Figure 12), ramps up to
the peak value (IPK+), then drops to zero at turn-off. The
average current into VIN during this on-time is the load
current. For a worst case calculation, C1 must supply this
average current during the maximum on-time. The maximum
on-time is calculated at VIN = 8V using Equation 4, with a
25% tolerance added:
(8)
where FS(min) is the minimum frequency of 189 kHz (252 kHz
- 25%).
This provides a minimum value for L1 - the next higher
standard value (150 µH) will be used. To prevent saturation,
and possible destructive current levels, L1 must be rated for
the peak current which occurs if the current limit and maximum ripple current are reached simultaneously. The maximum ripple amplitude is calculated by re-arranging Equation
8 using VIN(max), FS(min), and the minimum inductor value,
based on the manufacturer’s tolerance. Assume, for this
exercise, the inductor’s tolerance is ± 20%.
The voltage at VIN should not be allowed to drop below 7.5V
in order to maintain VCC above its UVLO.
(9)
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LM2694
Applications Information
LM2694
Applications Information
should be a good quality, low ESR, ceramic capacitor, physically close to the IC pins.
(Continued)
Normally a lower value can be used for C1 since the above
calculation is a worst case calculation which assumes the
power source has a high source impedance. A quality ceramic capacitor with a low ESR should be used for C1.
C2 and R3: Since the LM2694 requires a minimum of 25
mVp-p of ripple at the FB pin for proper operation, the
required ripple at VOUT is increased by R1 and R2, and is
equal to:
VRIPPLE = 25 mVp-p x (R1 + R2)/R2 = 50 mVp-p
C4: The recommended value for C4 is 0.022 µF. A high
quality ceramic capacitor with low ESR is recommended as
C4 supplies the surge current to charge the buck switch gate
at each turn-on. A low ESR also ensures a complete recharge during each off-time.
This necessary ripple voltage is created by the inductor
ripple current acting on C2’s ESR + R3. First, the minimum
ripple current, which occurs at minimum VIN, maximum inductor value, and maximum frequency, is determined.
C6: The capacitor at the SS pin determines the soft-start
time, i.e. the time for the reference voltage at the regulation
comparator, and the output voltage, to reach their final value.
The capacitor value is determined from the following:
C5: This capacitor suppresses transients and ringing due to
lead inductance at VIN. A low ESR, 0.1 µF ceramic chip
capacitor is recommended, located physically close to the
LM2694.
For a 5 ms softstart time, C6 calculates to 0.024 µF.
D1: A Schottky diode is recommended. Ultra-fast recovery
diodes are not recommended as the high speed transitions
at the SW pin may inadvertently affect the IC’s operation
through external or internal EMI. The diode should be rated
for the maximum VIN (30V), the maximum load current
(0.6A), and the peak current which occurs when current limit
and maximum ripple current are reached simultaneously (IPK
in Figure 11), previously calculated to be 0.92A. The diode’s
forward voltage drop affects efficiency due to the power
dissipated during the off-time. The average power dissipation in D1 is calculated from:
PD1 = VF x IO x (1 - D)
where IO is the load current, and D is the duty cycle.
(10)
The minimum ESR for C2 is then equal to:
If the capacitor used for C2 does not have sufficient ESR, R3
is added in series as shown in Figure 7. The value chosen
for C2 is application dependent, and it is recommended that
it be no smaller than 3.3 µF. C2 affects the ripple at VOUT,
and transient response. Experimentation is usually necessary to determine the optimum value for C2.
C3: The capacitor at the VCC pin provides noise filtering and
stability, prevents false triggering of the VCC UVLO at the
buck switch on/off transitions, and limits the peak voltage at
VCC when a high voltage with a short rise time is initially
applied at VIN. C3 should be no smaller than 0.1 µF, and
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FINAL CIRCUIT
The final circuit is shown in Figure 13, and its performance is
shown in Figure 14 and Figure 15. Current limit measured
approximately 0.64A.
14
LM2694
Applications Information
(Continued)
20187031
FIGURE 13. Example Circuit
15
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LM2694
Applications Information
(Continued)
Item
Description
Value
C1
Ceramic Capacitor
3.3 µF, 50V
C2
Ceramic Capacitor
22 µF, 16V
C4, C6
Ceramic Capacitor
0.022 µF, 16V
C3, C5
Ceramic Capacitor
0.1 µF, 50V
D1
Schottky Diode
60V, 1A
L1
Inductor
150 µH
R1
Resistor
2.5 kΩ
R2
Resistor
2.5 kΩ
R3
Resistor
1.5 Ω
RON
Resistor
140 kΩ
U1
National Semi LM2694
MINIMUM LOAD CURRENT
The LM2694 requires a minimum load current of 500 µA. If
the load current falls below that level, the bootstrap capacitor
(C4) may discharge during the long off-time, and the circuit
will either shutdown, or cycle on and off at a low frequency.
If the load current is expected to drop below 500 µA in the
application, R1 and R2 should be chosen low enough in
value so they provide the minimum required current at nominal VOUT.
LOW OUTPUT RIPPLE CONFIGURATIONS
For applications where low output voltage ripple is required
the output can be taken directly from the low ESR output
capacitor (C2) as shown in Figure 16. However, R3 slightly
degrades the load regulation. The specific component values, and the application determine if this is suitable.
20187032
FIGURE 14. Efficiency vs Load Current and VIN
Circuit of Figure 13
20187034
FIGURE 16. Low Ripple Output
Where the circuit of Figure 16 is not suitable for reducing
output ripple, the circuits of Figure 17 or Figure 18 can be
used.
20187033
FIGURE 15. Frequency vs VIN
Circuit of Figure 13
20187035
FIGURE 17. Low Output Ripple Using a Feedforward
Capacitor
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16
close as possible to their associated pins. The two major
current loops have currents which switch very fast, and so
the loops should be as small as possible to minimize conducted and radiated EMI. The first loop is that formed by C1,
through the VIN to SW pins, L1, C2, and back to C1. The
second loop is that formed by D1, L1, C2, and the SGND and
ISEN pins. The ground connection from C2 to C1 should be
as short and direct as possible, preferably without going
through vias. Directly connect the SGND and RTN pin to
each other, and they should be connected as directly as
possible to the C1/C2 ground line without going through vias.
The power dissipation within the IC can be approximated by
determining the total conversion loss (PIN - POUT), and then
subtracting the power losses in the free-wheeling diode and
the inductor. The power loss in the diode is approximately:
PD1 = IO x VF x (1-D)
where Io is the load current, VF is the diode’s forward voltage
drop, and D is the duty cycle. The power loss in the inductor
is approximately:
PL1 = IO2 x RL x 1.1
where RL is the inductor’s DC resistance, and the 1.1 factor
is an approximation for the AC losses. If it is expected that
the internal dissipation of the LM2694 will produce high
junction temperatures during normal operation, good use of
the PC board’s ground plane can help considerably to dissipate heat. The exposed pad on the LLP package bottom
should be soldered to a ground plane, and that plane should
both extend from beneath the IC, and be connected to
exposed ground plane on the board’s other side using as
many vias as possible. The exposed pad is internally connected to the IC substrate. The use of wide PC board traces
at the pins, where possible, can help conduct heat away
from the IC. The four No Connect pins on the TSSOP
package are not electrically connected to any part of the IC,
and may be connected to ground plane to help dissipate
heat from the package. Judicious positioning of the PC
board within the end product, along with the use of any
available air flow (forced or natural convection) can help
reduce the junction temperature.
(Continued)
In Figure 17, Cff is added across R1 to AC-couple the ripple
at VOUT directly to the FB pin. This allows the ripple at VOUT
to be reduced, in some cases considerably, by reducing R3.
In the circuit of Figure 13, the ripple at VOUT ranged from 50
mVp-p at VIN = 8V to 100 mVp-p at VIN = 30V. By adding a
2700 pF capacitor at Cff and reducing R3 to 0.75Ω, the VOUT
ripple is reduced by 50%.
20187036
FIGURE 18. Minimum Output Ripple Using Ripple
Injection
To reduce VOUT ripple further, the circuit of Figure 18 can be
used. R3 has been removed, and the output ripple amplitude
is determined by C2’s ESR and the inductor ripple current.
RA and CA are chosen to generate a 40-50 mVp-p sawtooth
at their junction, and that voltage is AC-coupled to the FB pin
via CB. In selecting RA and CA, VOUT is considered a virtual
ground as the SW pin switches between VIN and -1V. Since
the on-time at SW varies inversely with VIN, the waveform
amplitude at the RA/CA junction is relatively constant. Typical values for the additional components are RA = 110k, CA
= 2700 pF, and CB = 0.01 µF.
PC BOARD LAYOUT and THERMAL CONSIDERATIONS
The LM2694 regulation, over-voltage, and current limit comparators are very fast, and will respond to short duration
noise pulses. Layout considerations are therefore critical for
optimum performance. The layout must be as neat and
compact as possible, and all the components must be as
17
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LM2694
Applications Information
LM2694
Physical Dimensions
inches (millimeters) unless otherwise noted
14-Lead TSSOP-14 Package
NS Package Number MTC14
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18
LM2694 30V, 600 mA Step Down Switching Regulator
Physical Dimensions
inches (millimeters) unless otherwise noted (Continued)
10-Lead LLP Package
NS Package Number SDA10A
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves
the right at any time without notice to change said circuitry and specifications.
For the most current product information visit us at www.national.com.
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