Maxim MAX767CAP 5v-to-3.3v, synchronous, step-down power-supply controller Datasheet

19-0224; Rev 2; 8/94
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5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
The MAX767 is a high-efficiency, synchronous buck
controller IC dedicated to converting a fixed 5V supply
into a tightly regulated 3.3V output. Two key features set
this device apart from similar, low-voltage step-down
switching regulators: high operating frequency and all
N-channel construction in the application circuit. The
300kHz operating frequency results in very small, lowcost external surface-mount components.
The inductor, at 3.3µH for 5A, is physically at least five
times smaller than inductors found in competing solutions. All N-channel construction and synchronous rectification result in reduced cost and highest efficiency.
Efficiency exceeds 90% over a wide range of loading,
eliminating the need for heatsinking. Output capacitance
requirements are low, reducing board space and cost.
The MAX767 is a monolithic BiCMOS IC available in
20-pin SSOP packages. For other fixed output voltages
and package options, please consult the factory.
____________________________Features
♦
♦
♦
♦
♦
♦
♦
♦
♦
>90% Efficiency
700µA Quiescent Supply Current
120µA Standby Supply Current
4.5V-to-5.5V Input Range
Low-Cost Application Circuit
All N-Channel Switches
Small External Components
Tiny Shrink-Small-Outline Package (SSOP)
Predesigned Applications:
Standard 5V to 3.3V DC-DC Converters up to 10A
High-Accuracy Pentium P54C VR-Spec Supply
♦ Fixed Output Voltages Available:
3.3V (Standard)
3.45V (High-Speed Pentium™)
3.6V (PowerPC™)
______________Ordering Information
REF.
TOL.
PART
TEMP. RANGE
PINPACKAGE
Local 5V-to-3.3V DC-DC Conversion
MAX767CAP
0°C to +70°C
20 SSOP
±1.8% 3.3V
Microprocessor Daughterboards
MAX767RCAP
0°C to +70°C
20 SSOP
±1.8% 3.45V
Power Supplies up to 10A or More
MAX767SCAP
0°C to +70°C
20 SSOP
±1.8% 3.6V
MAX767TCAP
0°C to +70°C
20 SSOP
±1.2% 3.3V
MAX767C/D
0°C to +70°C
Dice*
________________________Applications
________Typical Application Circuit
–
VOUT
–
Ordering Information continued at end of data sheet.
* Contact factory for dice specifications.
__________________Pin Configuration
INPUT
4.5V TO 5.5V
TOP VIEW
VCC
ON
BST
DH
MAX767
3.3µH
OUTPUT
3.3V
AT 5A
CS 1
20 FB
SS 2
19 DH
ON 3
LX
GND 4
DL
17 BST
GND 5
16 DL
PGND
GND 6
15 VCC
CS
GND 7
14 VCC
REF 8
13 PGND
FB
REF
18 LX
MAX767
GND
SYNC 9
12 N.C.
VCC 10
11 GND
SSOP
™ Pentium is a trademark of Intel.
PowerPC is a trademark of IBM.
________________________________________________________________ Maxim Integrated Products
Call toll free 1-800-998-8800 for free samples or literature.
1
MAX767
_______________General Description
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
ABSOLUTE MAXIMUM RATINGS
VCC to GND .................................................................-0.3V, +7V
PGND to GND ........................................................................±2V
BST to GND ...............................................................-0.3V, +15V
LX to BST.....................................................................-7V, +0.3V
Inputs/Outputs to GND
(ON, REF, SYNC, CS, FB, SS) .....................-0.3V, VCC + 0.3V
DL to PGND .....................................................-0.3V, VCC + 0.3V
DH to LX...........................................................-0.3V, BST + 0.3V
REF Short to GND.......................................................Momentary
REF Current.........................................................................20mA
Continuous Power Dissipation (TA = +70°C)
20-Pin SSOP (derate 8.00mW/°C above +70°C) ..........640mW
Operating Temperature Ranges:
MAX767CAP/MAX767_CAP.................................0°C to +70°C
MAX767EAP/MAX767_EAP ..............................-40°C to +85°C
Lead Temperature (soldering, 10sec) .............................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VCC = ON = 5V, GND = PGND = SYNC = 0V, IREF = 0mA, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
VCC Input Supply Range
CONDITIONS
Output Voltage (FB)
0mV < (CS - FB) < 80mV, MAX767, MAX767T
4.5V < VCC < 5.5V
MAX767R
(includes load and
MAX767S
line regulation)
Load Regulation
(CS - FB) = 0mV to 80mV
Line Regulation
VCC = 4.5V to 5.5V
VCC Fault Lockout Voltage
Falling edge, hysteresis = 1%
Current-Limit Voltage
CS - FB
SS Source Current
SS Fault Sink Current
Reference Voltage (REF)
MIN
4.5
TYP
MAX
5.5
3.17
3.35
3.46
3.32
3.50
3.60
3.46
3.65
3.75
2.5
V
%
0.1
3.80
UNITS
V
%
4.20
V
mV
80
100
120
2.50
4
6.5
2
µA
mA
MAX767, MAX767R, MAX767S
3.24
3.30
3.36
MAX767T
3.26
3.30
3.34
V
VCC Standby Current
ON = 0V, VCC = 5.5V
120
200
µA
VCC Quiescent Current
FB = CS = 3.5V
0.7
1.0
mA
Oscillator Frequency
SYNC = 3.3V
SYNC = 0V or 5V
300
200
340
260
Oscillator SYNC Range
240
SYNC High Pulse Width
200
SYNC Low Pulse Width
350
kHz
kHz
ns
200
ns
SYNC Rise/Fall Time
Not tested
Oscillator Maximum Duty Cycle
SYNC = 3.3V
SYNC = 0V
Input Low Voltage
SYNC, ON
Input High Voltage
ON
SYNC
Input Current
SYNC, ON = 0V or 5V
DL Sink/Source Current
DL = 2V
1
DH Sink/Source Current
(BST - LX) = 4.5V, DH = 2V
1
DL On Resistance
High or low
7
Ω
DH On Resistance
High or low, (BST - LX) = 4.5V
7
Ω
2
200
89
92
95
ns
%
0.8
2.40
VCC - 0.5
V
V
±1
_______________________________________________________________________________________
µA
A
A
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
70
80
70
60
50
0.001
0.01
0.1
1
10
0.01
0.1
1
OUTPUT CURRENT (A)
EFFICIENCY vs. OUTPUT CURRENT
(7A CIRCUIT)
EFFICIENCY vs. OUTPUT CURRENT
(10A CIRCUIT)
10
EFFICIENCY (%)
70
80
70
60
60
0.001
0.01
0.1
1
10
OUTPUT CURRENT (A)
0.001
0.01
0.1
1
1000
10
SYNC = REF (300kHz)
100
10
1
0.1
10
0.001
0.01
0.1
1
10
100
LOAD CURRENT (% FULL LOAD)
OUTPUT CURRENT (A)
IDLE-MODE WAVEFORMS
PWM-MODE WAVEFORMS
3.3V OUTPUT
50mV/div, AC COUPLED
3.3V OUTPUT
50mV/div, AC COUPLED
LX
5V/div
LX
5V/div
5µs/div
ILOAD = 300mA
1
0.01
50
50
0.1
0.01
SWITCHING FREQUENCY vs.
PERCENT OF FULL LOAD
90
80
0.001
OUTPUT CURRENT (A)
MAX767-05
MAX767-04
100
90
70
50
0.001
OUTPUT CURRENT (A)
100
80
60
SWITCHING FREQUENCY (kHz)
50
MAX767-03
90
EFFICIENCY (%)
80
60
EFFICIENCY (%)
100
90
EFFICIENCY (%)
EFFICIENCY (%)
90
EFFICIENCY vs. OUTPUT CURRENT
(5A CIRCUIT)
MAX767-02
100
MAX767-01
100
EFFICIENCY vs. OUTPUT CURRENT
(3A CIRCUIT)
MAX767-06
EFFICIENCY vs. OUTPUT CURRENT
(1.5A CIRCUIT)
MAX767
__________________________________________Typical Operating Characteristics
(Circuit of Figure 1 (5A configuration), VIN = 5V, oscillator frequency = 300kHz, TA = +25°C, unless otherwise noted.)
1µs/div
ILOAD = 5A
_______________________________________________________________________________________
3
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
____________________________Typical Operating Characteristics (continued)
(Circuit of Figure 1 (5A configuration), VIN = 5V, oscillator frequency = 300kHz, TA = +25°C, unless otherwise noted.)
1.5A CIRCUIT LOAD-TRANSIENT RESPONSE
3A CIRCUIT LOAD-TRANSIENT RESPONSE
1.5A
3A
LOAD CURRENT
LOAD CURRENT
0A
0A
3.3V OUTPUT
50mV/div
AC-COUPLED
3.3V OUTPUT
50mV/div
AC-COUPLED
200µs/div
200µs/div
5A CIRCUIT LOAD-TRANSIENT RESPONSE
7A CIRCUIT LOAD-TRANSIENT RESPONSE
7A
5A
LOAD CURRENT
LOAD CURRENT
0A
0A
3.3V OUTPUT
50mV/div
AC-COUPLED
3.3V OUTPUT
50mV/div
AC-COUPLED
200µs/div
200µs/div
10A CIRCUIT LOAD-TRANSIENT RESPONSE
10A
LOAD CURRENT
0A
3.3V OUTPUT
50mV/div
AC-COUPLED
200µs/div
4
_______________________________________________________________________________________
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
PIN
NAME
FUNCTION
1
CS
Current-sense input: +100mV = nominal current-limit level referred to FB.
2
SS
3
ON
Soft-start input. Ramp time to full current limit is 1ms/nF of capacitance to GND.
——–
ON/OFF control input to disable the PWM. Tie directly to VCC for automatic start-up.
4–7, 11
GND
Low-current analog ground. Feedback reference point for the output.
8
REF
3.3V internal reference output. Bypass to GND with 0.22µF minimum capacitor.
9
SYNC
10, 14, 15
VCC
Supply voltage input: 4.5V to 5.5V
12
N.C.
No internal connection
13
PGND
16
DL
Gate-drive output for the low-side synchronous rectifier MOSFET
17
BST
Boost capacitor connection (0.1µF)
18
LX
Inductor connection. Can swing 2V below GND without latchup.
19
DH
Gate-drive output for the high-side MOSFET
20
FB
Feedback and current-sense input for the PWM
Oscillator control/synchronization input. Connect to VCC or GND for 200kHz; connect to REF for
300kHz. For external clock synchronization in the 240kHz to 350kHz range, a high-to-low transition
causes a new cycle to start.
Power ground
INPUT
4.5V TO 5.5V
R2
C4
4.7µF
SHUTDOWN
ON/OFF
ON
10Ω
D1
SMALLSIGNAL
SCHOTTKY
VCC
BST
DH
N1
C3
0.1µF
L1
C5
(OPTIONAL)
R1
OUTPUT
3.3V
LX
MAX767
0.01µF
C1
D2
SS
DL
N2
C2
PGND
SYNC
CS
REF
FB
GND
C6
0.22µF
Figure 1. Standard Application Circuit
_______________________________________________________________________________________
5
MAX767
______________________________________________________________Pin Description
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
_____Standard Application Circuits
This data sheet shows five predesigned circuits with
output current capabilities from 1.5A to 10A. Many
users will find one of these standard circuits appropriate for their needs. If a standard circuit is used, the
remainder of this data sheet (Detailed Description and
Applications Information and Design Procedure ) can
be bypassed.
Figure 1 shows the Standard Application Circuit. Table 1
gives component values and part numbers for five different implementations of this circuit: 1.5A, 3A, 5A, 7A,
and 10A output currents.
Each of these circuits is designed to deliver the full
rated output load current over the temperature range
listed. In addition, each will withstand a short circuit of
several seconds duration from the output to ground. If
the circuit must withstand a continuous short circuit,
refer to the Short-Circuit Duration section for the
required changes.
Layout and Grounding
Good layout is necessary to achieve the designed output power, high efficiency, and low noise. Good layout
includes the use of a ground plane, appropriate component placement, and correct routing of traces using
appropriate trace widths. The following points are in
order of decreasing importance.
1. A ground plane is essential for optimum performance. In most applications, the circuit will be
located on a multilayer board and full use of the four
or more copper layers is recommended. Use the
top and bottom layers for interconnections and the
inner layers for an uninterrupted ground plane.
2. Because the sense resistance values are similar to
a few centimeters of narrow traces on a printed circuit board, trace resistance can contribute significant errors. To prevent this, Kelvin connect CS and
FB to the sense resistor; i.e., use separate traces
not carrying any of the inductor or load current, as
shown in Figure 2. These signals must be carefully
shielded from DH, DL, BST, and the LX node.
Important: place the sense resistor as close as possible to and no further than 10mm from the MAX767.
3. Place the LX node components N1, N2, L1, and D2
as close together as possible. This reduces resistive and switching losses and confines noise due to
ground inductance.
4. The input filter capacitor C1 should be less than
10mm away from N1’s drain. The connecting copper trace carries large currents and must be at least
2mm wide, preferably 5mm.
6
5. Keep the gate connections to the MOSFETs short
for low inductance (less than 20mm long and more
than 0.5mm wide) to ensure clean switching.
6. To achieve good shielding, it is best to keep all
switching signals (MOSFET gate drives DH and DL,
BST, and the LX node) on one side of the board
and all sensitive nodes (CS, FB, and REF) on the
other side.
7. Connect the GND and PGND pins directly to the
ground plane, which should ideally be an inner
layer of a multilayer board.
_______________Detailed Description
Note: The remainder of this document contains the
detailed information necessary to design a circuit that
differs substantially from the five standard application
circuits. If you are using one of the predesigned standard circuits, the following sections are provided only
for your reading pleasure.
The MAX767 converts a 4.5V to 5.5V input to a 3.3V
output. Its load capability depends on external components and can exceed 10A. The 3.3V output is generated by a current-mode, pulse-width-modulation (PWM)
step-down regulator. The PWM regulator operates at
either 200kHz or 300kHz, with a corresponding tradeoff between somewhat higher efficiency (200kHz) and
smaller external component size (300kHz). The
MAX767 also has a 3.3V, 5mA reference voltage. Faultprotection circuitry shuts off the output should the reference lose regulation or the input voltage go below 4V
(nominally).
External components for the MAX767 include two Nchannel MOSFETs, a rectifier, and an LC output filter.
The gate-drive signal for the high-side MOSFET, which
must exceed the input voltage, is provided by a boost
circuit that uses a 0.1µF capacitor. The synchronous
rectifier keeps efficiency high by clamping the voltage
across the rectifier diode. An external low-value current-sense resistor sets the maximum current limit, preventing excessive inductor current during start-up or
under short-circuit conditions. An optional external
capacitor sets the programmable soft-start, reducing
in-rush surge currents upon start-up and providing
adjustable power-up time.
The PWM regulator is a direct-summing type, lacking a
traditional integrator-type error amplifier and the phase
shift associated with it. It therefore does not require
external feedback-compensation components, as long
as you follow the ESR guidelines in the Applications
Information and Design Procedure sections.
_______________________________________________________________________________________
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
MAX767
Table 1. Component Values
Part
1.5A Circuit
L1
10µH
Sumida CDR74B-100
R1
N1,
N2
3A Circuit
5A Circuit
5µH
Sumida CDR125
DRG# 4722-JPS-001
3.3µH
CoilCraft
DO3316-332
0.012Ω
0.04Ω
0.02Ω
DD WSL-2512-R012
IRC LR2010-01-R040 IRC LR2010-01-R020 or 2 x 0.025Ω
or DD WSL-2512-R040 or DD WSL-2512-R020 IRC LR2010-01-R025
(in parallel)
International
Siliconix Si9410DY,
Rectifier IRF7101,
International
Siliconix Si9936DY
Rectifier IRF7101
Motorola
or Motorola
or Motorola
MTD20N03HDL
MMDF3N03HD
MMDF3N03HD
(dual N-channel)
(both FETs in parallel)
7A Circuit
2.1µH, 5mΩ
Coiltronics
CTX03-12338-1
10A Circuit
1.5µH, 3.5mΩ
Coiltronics
CTX03-12357-1
3 x 0.020Ω
3 x 0.025Ω
IRC LR2010-01-R020
IRC LR2010-01-R025
or 2 x 0.012Ω
or DD WSL-2512-R025
DD WSL-2512-R012
(in parallel)
(in parallel)
Motorola
MTD75N03HDL
(N1)
MTD20N03HDL
(N2)
Motorola
MTD75N03HDL
C1
47µF, 20V
AVX TPSD476K020R
2 x 47µF, 20V
AVX TPSD476K020R
220µF, 10V
Sanyo
OS-CON 10SA220M
2 x 100µF, 10V
Sanyo
OS-CON 10SA100M
2 x 220µF, 10V
Sanyo
OS-CON 10SA220M
C2
220µF, 6.3V
Sprague
595D227X06R3D2B
2 x 150µF, 10V
Sprague
595D157X0010D7T
2 x 220µF, 10V
Sanyo
OS-CON 10SA220M
2 x 220µF, 10V
Sanyo
OS-CON 10SA220M
4 x 220µF, 10V
Sanyo
OS-CON 10SA220M
D2
1N5817
Nihon EC10QS02,
or Motorola
MBRS120T3
1N5817
Nihon EC10QS02,
or Motorola
MBRS120T3
1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3
1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3
1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3
to +85°C
to +85°C
to +85°C
to +85°C
to +85°C
Temp.
Range
Table 2. Component Suppliers
Company
Factory Fax [Country Code] USA Telephone
AVX
[1] (207) 283-1941
(800) 282-4975
CoilCraft
[1] (708) 639-1469
(708) 639-6400
Coiltronics
[1] (407) 241-9339
(407) 241-7876
DD
[1] (402) 563-6418
(402) 563-6582
IRC
[1] (512) 992-3377
(512) 992-7900
International
Rectifier
[1] (310) 322-3332
(310) 322-3331
Motorola
[1] (602) 244 4015
(602) 244- 3576
Nihon
[81] 3-3494-7414
(805) 867-2555
Sanyo
[81] 7-2070-1174
(619) 661-6835
Siliconix
[1] (408) 970-3950
(408) 988-8000
Sprague
[1] (603) 224-1430
(603) 224-1961
Sumida
[81] 3-3607-5144
(708) 956-0666
_______________________________________________________________________________________
7
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
FAT, HIGH-CURRENT TRACES
because the minimum-current comparator immediately
resets the high-side latch at the beginning of each
cycle, unless the FB signal falls below the reference
voltage level.
MAIN CURRENT PATH
Soft-Start
SENSE RESISTOR
MAX767
Connecting a capacitor from the soft-start pin (SS) to
ground allows a gradual build-up of the 3.3V output
after power is applied or ON is driven high. When ON is
low, the soft-start capacitor is discharged to GND.
When ON is driven high, a 4µA constant current source
charges the capacitor up to 4V. The resulting ramp voltage on SS linearly increases the current-limit comparator set-point, increasing the duty cycle to the external
power MOSFETs. With no soft-start capacitor, the full
output current is available within 10µs (see Applications
Information and Design Procedure section).
Synchronous Rectifier
Figure 2. Kelvin Connections for the Current-Sense Resistor
The main gain block is an open-loop comparator that
sums four signals: output voltage error signal, currentsense signal, slope-compensation ramp, and the 3.3V
reference. This direct-summing method approaches
the ideal of cycle-by-cycle control of the output voltage.
Under heavy loads, the controller operates in full PWM
mode. Every pulse from the oscillator sets the output
latch and turns on the high-side switch for a period
determined by the duty factor (approximately
VOUT / VIN).
As the high-side switch turns off, the synchronous rectifier latch is set; 60ns later, the low-side switch turns on.
The low-side switch stays on until the beginning of the
next clock cycle (in continuous-conduction mode) or
until the inductor current reaches zero (in discontinuous-conduction mode). Under fault conditions where
the inductor current exceeds the 100mV current-limit
threshold, the high-side latch resets and the high-side
switch turns off.
At light loads, the inductor current fails to exceed the
25mV threshold set by the minimum-current comparator. When this occurs, the PWM goes into Idle-Mode™,
skipping most of the oscillator pulses to reduce the
switching frequency and cut back switching losses.
The oscillator is effectively gated off at light loads
Synchronous rectification allows for high efficiency by
reducing the losses associated with the Schottky rectifier. Also, the synchronous-rectifier MOSFET is necessary for correct operation of the MAX767’s boost gatedrive supply.
When the external power MOSFET (N1) turns off, energy stored in the inductor causes its terminal voltage to
reverse instantly. Current flows in the loop formed by
the inductor (L1), Schottky diode (D2), and the load—
an action that charges up the output filter capacitor
(C2). The Schottky diode has a forward voltage of
about 0.5V which, although small, represents a significant power loss and degrades efficiency. The synchronous-rectifier MOSFET parallels the diode and is turned
on by DL shortly after the diode conducts. Since the
synchronous rectifier’s on resistance (rDS(ON)) is very
low, the losses are reduced. The synchronous-rectifier
MOSFET is turned off when the inductor current falls to
zero.
The MAX767’s internal break-before-make timing
ensures that shoot-through (both external switches
turned on at the same time) does not occur. The
Schottky rectifier conducts during the time that neither
MOSFET is on, which improves efficiency by preventing
the synchronous-rectifier MOSFET’s lossy body diode
from conducting.
The synchronous rectifier works under all operating
conditions, including discontinuous-conduction mode
and idle-mode.
™ Idle-Mode is a trademark of Maxim Integrated Products.
8
_______________________________________________________________________________________
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
MAX767
CS
1X
VCC
60kHz
LPF
REF
FB
MAIN PWM
COMPARATOR
+3.3V
REFERENCE
BST
R
LEVEL
SHIFT
Q
S
DH
LX
SLOPE COMP
4V
MINIMUM
CURRENT
(IDLE-MODE)
25mV
VCC
CURRENT
LIMIT
4µA
FAULT
SHOOTTHROUGH
CONTROL
2.8V
0mV TO 100mV
30R
SS
3.3V
N
200kHz/300kHz
OSCILLATOR
ON
1R
ON
SYNCHRONOUS
RECTIFIER CONTROL
R
VCC
Q
LEVEL
SHIFT
S
MAX767
DL
PGND
SYNC
Figure 3. MAX767 Block Diagram
_______________________________________________________________________________________
9
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
MAX767
Modes of Operation
VIN
C1
VCC
MAX767
D1
BST
VOUT
%ON = ________
VIN
C3
DH
LEVEL
TRANSLATOR
N1
LX
PWM
L1
VCC
DL
N2
Figure 4. Boost Supply for High-Side Gate Driver
Gate-Driver Boost Supply
Gate-drive voltage for the high-side N-channel switch is
generated with the flying-capacitor boost circuit shown
in Figure 4. The capacitor (C3) is alternately charged
from the 5V input via the diode (D1) and placed in parallel with the high-side MOSFET’s gate-source terminals. On start-up, the synchronous rectifier (low-side)
MOSFET (N2) forces LX to 0V and charges the BST
capacitor to 5V. On the second half-cycle, the PWM
turns on the high-side MOSFET (N1); it does this by
closing an internal switch between BST and DH, which
connects the capacitor to the MOSFET gate. This provides the necessary enhancement voltage to turn on
the high-side switch, an action that “boosts” the 5V
gate-drive signal above the input voltage.
Ringing seen at the high-side MOSFET gates (DH) in
discontinuous-conduction mode (light loads) is a natural operating condition. It is caused by the residual
energy in the tank circuit, formed by the inductor and
stray capacitance at the LX node. The gate-driver negative rail is referred to LX, so any ringing there is directly coupled to the gate-drive supply.
10
PWM Mode
Under heavy loads—over approximately 25% of full
load—the supply operates as a continuous-current
PWM supply (see Typical Operating Characteristics).
The duty cycle, %ON, is approximately:
Current flows continuously in the inductor: first, it ramps
up when the power MOSFET conducts; second, it
ramps down during the flyback portion of each cycle as
energy is put into the inductor and then discharged into
the load. Note that the current flowing into the inductor
when it is being charged is also flowing into the load,
so the load is continuously receiving current from the
inductor. This minimizes output ripple and maximizes
inductor use, allowing very small physical and electrical
sizes. Output ripple is primarily a function of the filter
capacitor’s effective series resistance (ESR), and is
typically under 50mV (see Design Procedure section).
Idle-Mode
Under light loads (<25% of full load), the MAX767
enhances efficiency by turning the drive voltage on and
off for only a single clock period, skipping most of the
clock pulses entirely. Asynchronous switching, seen as
“ghosting” on an oscilloscope, is thus a normal operating condition whenever the load current is less than
approximately 25% of full load.
At certain input voltage and load conditions, a transition
region exists where the controller can pass back and
forth from idle-mode to PWM mode. In this situation,
short pulse bursts occur, which make the current waveform look erratic but do not materially affect the output
ripple. Efficiency remains high.
Current Limiting
The voltage between CS and FB is continuously monitored. An external, low-value shunt resistor is connected between these pins, in series with the inductor,
allowing the inductor current to be continuously measured throughout the switching cycle. Whenever this
voltage exceeds 100mV, the drive voltage to the external high-side MOSFET is cut off. This protects the MOSFET, the load, and the input supply in case of short circuits or temporary load surges. The current-limiting
resistance is typically 20mΩ for 3A.
______________________________________________________________________________________
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
Internal Reference
The internal 3.3V bandgap reference (REF) remains
active, even when the switching regulator is turned off.
It can furnish up to 5mA, and can be used to supply
memory keep-alive power or for other purposes.
Bypass REF to GND with 0.22µF, plus 1µF/mA of load
current.
Applications Information and
__________________Design Procedure
Most users will be able to work with one of the standard
application circuits; others may want to implement a
circuit with an output current rating that lies between or
beyond the standard values.
If you want an output current level that lies between two
of the standard application circuits, you can interpolate
many of the component values from the values given
for the two circuits. These components include the
input and output filter capacitors, the inductor, and the
sense resistor. The capacitors must meet ESR and ripple current requirements (see Input Filter Capacitor and
Output Filter Capacitor sections). The inductor must
meet the required current rating (see Inductor section).
You may use the rectifier and MOSFETs specified for
the circuit with the greater output current capability, or
choose a new rectifier and MOSFETs according to the
requirements detailed in the Rectifier and MOSFET
Switches sections. For more complete information, or
for output currents in excess of 10A, refer to the design
information in the following sections.
Inductor, L1
Three inductor parameters are required: the inductance
value (L), the peak inductor current (ILPEAK), and the
coil resistance (RL). The inductance is:
1.32
L1 = ______________
f x IOUT x LIR
where:
f = switching frequency, normally 300kHz
IOUT = maximum 3.3V DC load current (A)
LIR = ratio of inductor peak-to-peak AC
current to average DC load current,
typically 0.3.
A higher LIR value allows smaller inductance, but
results in higher losses and ripple.
The highest peak inductor current (ILPEAK) equals the
DC load current (IOUT) plus half the peak-to-peak AC
inductor current (ILPP). The peak-to-peak AC inductor
current is typically chosen as 30% of the maximum DC
load current, so the peak inductor current is 1.15 x IOUT.
The peak inductor current at any load is given by:
1.32
ILPEAK = IOUT + __________
2 x f x L1
The coil resistance should be as low as possible,
preferably in the low milliohms. The coil is effectively in
series with the load at all times, so the wire losses alone
are approximately:
Power Loss = IOUT2 x RL
In general, select a standard inductor that meets the L,
ILPEAK, and RL requirements. If a standard inductor is
unavailable, choose a core with an LI2 parameter
greater than L x ILPEAK2, and use the largest wire that
will fit the core.
Current-Sense Resistor, R1
The current-sense resistor must carry the peak current
in the inductor, which exceeds the full DC load current.
The internal current limiting starts when the voltage
across the sense resistors exceeds 100mV nominally,
80mV minimum. Use the minimum value to ensure adequate output current capability: R1 = 80mV / ILPEAK.
The low VIN/VOUT ratio creates a potential problem with
start-up under full load or with load transients from noload to full load. If the supply is subjected to these conditions, reduce the sense resistor:
70mV
R1 = ———
ILPEAK
Since the sense-resistance values are similar to a few
centimeters of narrow traces on a printed circuit board,
trace resistance can contribute significant errors. To
prevent this, Kelvin connect the CS and FB pins to the
sense resistors; i.e., use separate traces not carrying
any of the inductor or load current, as shown in Figure 2.
______________________________________________________________________________________
11
MAX767
Oscillator Frequency
The SYNC input controls the oscillator frequency.
Connecting SYNC to GND or to VCC selects 200kHz
operation; connecting it to REF selects 300kHz operation. SYNC can also be driven with an external 240kHz
to 350kHz CMOS/TTL source to synchronize the internal oscillator. Normally, 300kHz operation is chosen to
minimize the inductor and output filter capacitor sizes,
but 200kHz operation may be chosen for a small (about
1%) increase in efficiency at heavy loads.
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
Place R1 as close as possible to the MAX767, preferably less than 10mm. Run the traces at minimum spacing from one another. If they are longer than 20mm,
bypass CS to FB with a 1nF capacitor placed as close
as possible to these pins. The wiring layout for these
traces is critical for stable, low-ripple outputs (see
Layout and Grounding section).
In idle-mode, the ripple has a capacitive and a resistive
component:
.
0.0004 x L
VOUT(RPL) (C) = _____________
x 0.89 Volts
R12 x C2
Input Filter Capacitor, C1
Use at least 6µF per watt of output power for C1. If the
5V input is some distance away or comes through a PC
bus, greater capacitance may be desirable to improve
the load-transient response. Use a low-ESR capacitor
located no further than 10mm from the MOSFET switch
(N1) to prevent ringing. The ripple current rating must
be at least IRMS = 0.5 x IOUT. For high-current applications, two or more capacitors in parallel may be needed
to meet these requirements.
The ESR of C1 is effectively in series with the input. The
resistive dissipation of C1, IRMS2 x ESRC1, can significantly impact the circuit’s efficiency.
Output Filter Capacitor, C2
The output filter capacitor determines the loop stability,
output voltage ripple, and output load-transient
response.
Stability
To ensure stability, stay above the minimum capacitance value and below the maximum ESR value. These
values are:
3Ω
C2 > —— µF
R1
and
ESRC2 < R1
Be sure to satisfy both these requirements. To achieve
the low ESR required, it may be appropriate to parallel
two or more capacitors and/or use a total capacitance
2 or 3 times larger than the calculated minimum.
Output Ripple
The output ripple in continuous-conduction mode is:
VOUT(RPL) = IOUT(max) x LIR x
1
(ESRC2 + ———————
)
2 x π x f x C2
where f is the switching frequency (200kHz or 300kHz).
12
x ESRC2
VOUT(RPL) (R) = 0.02
_____________
R1
The total ripple, VOUT(RPL), can be approximated as
follows:
if
VOUT(RPL) (R) < 0.5 VOUT(RPL) (C)
then
VOUT(RPL) = VOUT(RPL)(C)
otherwise
VOUT(RPL) = 0.5 VOUT(RPL) (C) +
VOUT(RPL) (R)
Load-Transient Performance
In response to a large step increase in load current, the
output voltage will sag for several microseconds unless
C2 is increased beyond the values that satisfy the
above requirements. Note that an increase in capacitance is all that’s required to improve the transient
response, and that the ESR requirements don’t change.
Therefore, the added capacitance can be supplied by
an additional low-cost bulk capacitor in parallel with the
normal low-ESR switching-regulator capacitor. The
equation for voltage sag under a step load change is:
ISTEP2 x L
VSAG = ________________________________
2 x C2 x (VIN(min) x DMAX - 3.3V)
where DMAX is the maximum duty cycle. Higher duty
cycles are possible when the oscillator frequency is
reduced to 200kHz, since fixed propagation delays
through the PWM comparator become a lesser part of
the whole period. The tested worst-case limit for DMAX
is 92% at 200kHz or 89% at 300kHz. Lower inductance
values can reduce the filter capacitance requirement,
but only at the expense of increased output ripple (due
to higher peak currents).
______________________________________________________________________________________
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
Rectifier, D2
Use a 1N5817 or similar Schottky diode for applications
up to 3A, or a 1N5820 for up to 10A. Surface-mount
equivalents are available from N.I.E.C. with part numbers EC10QS02 and NSQ03A02, or from Motorola with
part numbers MBRS120T3 and MBRS320T3. D2 must
be a Schottky diode to prevent the lossy MOSFET body
diode from turning on.
Soft-Start
A capacitor connected from GND to SS causes the
supply’s current-limit level to ramp up slowly. The ramp
time to full current limit is approximately 1ms for every
nF of capacitance on SS, with a minimum value of
10µs. Typical values for the soft-start capacitor are in
the 10nF to 100nF range; a 5V rating is sufficient.
The time required for the output voltage to ramp up to
its rated value depends upon the output load, and is
not necessarily the same as the time it takes for the current limit to reach full capacity.
Duty Cycle
The duty cycle for the high-side MOSFET (N1) in continuous-conduction mode is:
100%
x ( VOUT + VN2)
___________________
VIN - VN1
where:
VOUT = 3.3V
VIN = 5V
VN1 and VN2 = ILOAD x rDS(ON) for each MOSFET.
It is apparent that, in continuous-conduction mode, N1
will conduct for about twice the time as N2. Under shortcircuit conditions, however, N2 can conduct as much
90% of the time. If there is a significant chance of short
circuiting the output, select N2 to handle the resulting
duty cycle (see Short-Circuit Duration section).
MOSFET Switches, N1 and N2
The two N-channel MOSFETs must be “logic-level”
FETs; that is, they must be fully on (have low
rDS(ON)) with only 4V gate-source drive voltage. For
high-current applications, FETs with low gatethreshold voltage specifications (i.e., maximum
VGS(TH) = 2V rather than 3V) are preferred. In addition, they should have low total gate charge (<70nC)
to minimize switching losses.
For output currents in excess of the five standard application circuits, placing MOSFETs with very low gate
charge in parallel increases output current and lowers
resistive losses. N2 does not normally require the same
current capacity as N1 because it conducts only about
33% of the time, while N1 conducts about 66% of the
time.
Short-Circuit Duration
At their highest rated temperatures (+70°C or +85°C),
each of the five standard application circuits will withstand a short circuit of several seconds duration. In
most cases, the MAX767 will be used in applications
where long-term short circuiting of the output is unlikely.
If it is desirable for the circuit to withstand a continuous
short circuit, the MOSFETs must be able to dissipate
the required power. This depends on physical factors
such as the mounting of the transistor, any heatsinking used, and ventilation provided, as well as the
actual current the transistor must deliver. The shortcircuit current is approximately 100mV / R1, but may
vary by ±20%.
Cautious design requires that the transistors
withstand the maximum possible current, which is
I SC = 120mV / R1. N1 and N2 must withstand this
current scaled by their maximum duty factors. The
maximum duty factor for N1 occurs under highload (but not short-circuit) conditions, and is approximately V OUT / V IN (min) or about 0.7. The maximum duty factor for N2 occurs during short-circuit
conditions and is:
ISC x rDS(ON)N2
1 - —————————————
VIN(max) - ISC x rDS(ON)N1
which can exceed 0.9. The total power dissipated in
both MOSFETs together is ISC2 x rDS(ON).
______________________________________________________________________________________
13
MAX767
RC Filter for VCC
R2 and C4 form a lowpass filter to remove switching
noise from the VCC input to the MAX767. C4 must have
fairly low ESR (<5Ω). Switching noise can interfere with
proper output voltage regulation, resulting in an excessive output voltage decrease (>100mV) at full load.
Overheating during soldering can damage the surfacemount capacitors specified for C4, causing the regulation problems described above. Take care to heat the
capacitor for as short a time as possible, especially if it
is soldered by hand.
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
Proper circuit operation requires that the short-circuit
current be at least ILOAD x (1 + LIR / 2). However, the
standard application circuits are designed for a shortcircuit current slightly in excess of this amount. This
excess design current guarantees proper start-up
under constant full-load conditions and proper full-load
transient response, and is particularly necessary with
low input voltages. If the circuit will not be subjected to
full-load transients or to loads approaching the full-load
at start-up, you can decrease the short-circuit current
by increasing R1, as described in the Current-Sense
Resistor section. This may allow use of MOSFETs with a
lower current-handling capability.
Heavy-Load Efficiency
Losses due to parasitic resistances in the switches,
coil, and sense resistor dominate at high load-current
levels. Under heavy loads, the MAX767 operates deep
in the continuous-conduction mode, where there is a
large DC offset to the inductor current (plus a small
sawtooth AC component) (see Inductor section). This
DC current is exactly equal to the load current, a fact
which makes it easy to estimate resistive losses via the
simplifying assumption that the total inductor current is
equal to this DC offset current. The major loss mechanisms under heavy loads, in usual order of importance,
are:
♦ I2R losses
♦ gate-charge losses
♦ diode-conduction losses
♦ transition losses
♦ capacitor-ESR losses
♦ losses due to the operating supply current of the IC.
Inductor-core losses, which are fairly low at heavy
loads because the AC component of the inductor current is small, are not accounted for in this analysis.
POUT x 100% =
Efficiency = ______
PIN
POUT
_______________
x 100%
POUT + PDTOTAL
PDTOTAL = PD(I2R) + PDGATE + PDDIODE +
PDTRAN + PDCAP + PDIC
14
I2R Losses
PD(I2R) = resistive loss = (ILOAD2) x
(RCOIL + rDS(ON) + R1)
where R COIL is the DC resistance of the coil and
rDS(ON) is the drain-source on resistance of the MOSFET. Note that the rDS(ON) term assumes that identical
MOSFETs are employed for both the synchronous rectifier and high-side switch, because they time-share the
inductor current. If the MOSFETs are not identical, estimate losses by averaging the two individual rDS(ON)
terms according to their duty factors: 0.66 for N1 and
0.34 for N2.
Gate-Charge Losses
PDGATE = gate driver loss = qG x f x 5V
where qG is the sum of the gate charge for low- and
high-side switches. Note that gate-charge losses are
dissipated in the IC, not the MOSFETs, and therefore
contribute to package temperature rise. For a pair of
matched MOSFETs, qG is simply twice the gate capacitance of a single MOSFET (a data sheet specification).
Diode Conduction Losses
PDDIODE = diode conduction losses =
ILOAD x VD x tD x f
where VD is the forward voltage of the Schottky diode
at the output current, tD is the diode’s conduction time
(typically 110ns), and f is the switching frequency.
Transition Losses
PDTRAN = transition loss =
VIN2 x CRSS x ILOAD x f
______________________
IDRIVE
where CRSS is the reverse transfer capacitance of the
high-side MOSFET (a data sheet parameter), f is the
switching frequency, and IDRIVE is the peak current
available from the high-side gate driver output (approximately 1A).
Additional switching losses are introduced by other
sources of stray capacitance at the switching node,
including the catch-diode capacitance, coil interwinding capacitance, and low-side switch drain capacitance, and are given as PDSW = VIN2 x CSTRAY x f, but
these are usually negligible compared to CRSS losses.
The low-side switch introduces only tiny switching losses, since its drain-source voltage is already low when it
turns on.
______________________________________________________________________________________
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
IC Supply-Current Losses
PDIC is the quiescent power dissipation of the IC and is
5V times the quiescent supply current (a data sheet
parameter), or about 5mW.
Light-Load Efficiency
Under light loads, the PWM will operate in discontinuous-conduction mode, where the inductor current discharges to zero at some point during each switching
cycle. New loss mechanisms, insignificant at heavy
loads, begin to become important. The basic difference
is that in discontinuous mode, the AC component of the
inductor current is large compared to the load current.
This increases losses in the core and in the output filter
capacitors. Ferrite cores are recommended over powdered-material types for best light-load efficiency.
At light loads, the inductor delivers triangular current
pulses rather than the nearly square waves found in
continuous-conduction mode. These pulses ramp up to
a point set by the idle-mode current comparator, which
is internally fixed at approximately 25% of the full-scale
current-limit level. This 25% threshold provides an optimum balance between low-current efficiency and output voltage noise (the efficiency curve would actually
look better with this threshold set at about 45%, but the
output noise would be too high).
____Additional Application Circuits
High-Accuracy Power Supplies
The standard application circuit’s accuracy is dominated by reference voltage error (±1.8%) and load regulation error (-2.5%). Both of these parameters can be
improved as shown in Figures 5 and 6. Both circuits
rely on an external integrator amplifier to increase the
DC loop gain in order to reduce the load regulation
error to 0.1%. Reference error is improved in the first
circuit by employing a version of the MAX767 (“T”
grade) which has a ±1.2% reference voltage tolerance.
Reference error of the second circuit is further
improved by substituting a highly accurate external reference chip (MAX872), which contributes ±0.38% total
error over temperature.
These two circuits were designed with the latest generation of dynamic-clock µPs in mind, which place great
demands on the transient-response performance of the
power supply. As the µP clock starts and stops, the
load current can change by several amps in less than
100ns. This tremendous ∆i/∆t can cause output voltage
overshoot or sag that results in the CPU VCC going out
of tolerance unless the power supply is carefully
designed and located close to the CPU. These circuits
have excellent dynamic response and low ripple, with
transient excursions of less than 40mV under zero to
full-load step change. In particular, these two circuits
support the “VR” (voltage regulator) version of the Intel
P54C Pentium™ CPU, which requires that its supply
voltage, including noise and transient errors, be within
the 3.30V to 3.45V range.
To configure these circuits for a given load current
requirement, substitute standard components from
Table 1 for the power switching elements (N1, N2, L1,
C1, C2) or use the Design Procedure. R1 can also be
taken from Table 1, but must be adjusted approximately 10% higher in order to maintain the correct currentlimit threshold. This increased value is due to the 0.9
gain factor introduced by the H-bridge resistor divider
(R3–R6).
If the remote sense line must sense the output voltage
on the far side of a connector or jumper that has the
possibility of becoming disconnected while the power
supply is operating, an additional 10kΩ resistor should
connect the sense line to the output voltage in the connector’s power-supply side in order to prevent accidental overvoltage at the CPU.
For applications that are powered from a fixed +12V or
battery input rather than from +5V, use a MAX797 IC
instead of the MAX767. The MAX797 is capable of
accepting inputs up to 30V. See the MAX796–MAX799
data sheet for a high-accuracy circuit schematic.
______________________________________________________________________________________
15
MAX767
Capacitor ESR Losses
PDCAP = capacitor ESR loss = IRMS2 x ESR
where IRMS = RMS AC input current, approximately
ILOAD / 2.
Note that losses in the output filter capacitors are small
when the circuit is heavily loaded, because the current
into the capacitor is not chopped. The output capacitor
sees only the small AC sawtooth ripple current. Ensure
that the input bypass capacitor has a ripple current rating that exceeds the value of IRMS.
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
INPUT
4.75V TO 5.5V
VOUT = VREF
C1
D1
+ 1)
( R10
R9
R2
10Ω
N1
DH
VCC
BST
C4
4.7µF
L1
R1
C2
MAX767T
N2
C5
0.01µF
D2
C8
620pF
REF
R8
10k
C10
0.01µF
(OPTIONAL)
R4
1k,
1%
SYNC
GND
C6
0.01µF
R7
330k
CS
SS
R3
1k,
1%
PGND
ON
FB
TO MAX767
VCC
R5
10k,
1%
C7
R11
5.1k 10µF CERAMIC
(LOCATE AT
MIN
µP PINS)
LOAD
R6
10k,
1%
C9
0.22µF
MAX495
R9
332k, 1%
R10
8.06k, 1%
REMOTE SENSE LINE
Figure 5. High-Accuracy CPU Power Supply with Internal Reference
16
3.38V OUTPUT
3.427V MAX
3.330V MIN
LX
DL
SHUTDOWN
ON/OFF
C3
0.1µF
______________________________________________________________________________________
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
MAX767
INPUT
4.75V TO 5.5V
C1
D1
VOUT = VREF
+ 1)
( R10
R9
R2
20Ω
N1
DH
VCC
BST
C4
22µF
C3
0.1µF
MAX767
ON
C2
3.38V OUTPUT
3.408V MAX
3.369V MIN
N2
D2
C5
0.01µF
R3
1k,
1%
R4
1k,
1%
PGND
REF
CS
SYNC
FB
SS
C9
0.22µF
R1
LX
DL
SHUTDOWN
ON/OFF
L1
TO MAX767
VCC
GND
C6
0.01µF
R7
330k
C8
1000pF
R5
10k,
1%
C7
R11
5.1k 10µF CERAMIC
(LOCATE AT
MIN
µP PINS)
LOAD
R6
10k,
1%
C10
0.01µF
(OPTIONAL)
VIN
VOUT
R8
10k
MAX495
MAX872
GND
R9
332k, 0.1%
R10
118k, 0.1%
REMOTE SENSE LINE
Figure 6. High-Accuracy CPU Power Supply with External Reference
______________________________________________________________________________________
17
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
_Ordering Information (continued)
REF.
TOL.
PART
TEMP. RANGE
PINPACKAGE
MAX767EAP
0°C to +70°C
20 SSOP
±1.8% 3.3V
MAX767REAP
0°C to +70°C
20 SSOP
±1.8% 3.45V
___________________Chip Topography
VOUT
SS CS
ON
FB
DH
LX
MAX767SEAP
0°C to +70°C
20 SSOP
±1.8% 3.6V
GND
MAX767TEAP
0°C to +70°C
20 SSOP
±1.2% 3.3V
GND
BST
DL
0.181"
V CC (4.597mm)
GND
V CC
GND
V CC
GND
PGND
REF
SYNC
V CC
GND
0.109"
(2.769mm)
TRANSISTOR COUNT: 1294
SUBSTRATE CONNECTED TO GND
18
______________________________________________________________________________________
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
DIM
A
A1
B
C
D
E
e
H
L
α
e
E
H
INCHES
MAX
MIN
0.078
0.068
0.008
0.002
0.015
0.010
0.009
0.005
0.289
0.278
0.212
0.205
0.0256 BSC
0.311
0.301
0.037
0.022
8˚
0˚
MILLIMETERS
MIN
MAX
1.73
1.99
0.05
0.21
0.25
0.38
0.13
0.22
7.07
7.33
5.20
5.38
0.65 BSC
7.65
7.90
0.55
0.95
0˚
8˚
21-0003A
D
α
A
0.127mm
0.004in.
B
A1
C
L
20-PIN PLASTIC
SHRINK
SMALL-OUTLINE
PACKAGE
______________________________________________________________________________________
19
MAX767
________________________________________________________Package Information
MAX767
5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller
20
______________________________________________________________________________________